WO2018073869A1

WO2018073869A1 - Motor drive device, electric fan, electric vacuum cleaner, and hand dryer

Info

Publication number: WO2018073869A1
Application number: PCT/JP2016/080730
Authority: WO
Inventors: 裕次 ▲高▼山; 啓介植村; 有澤　浩一; 酒井　顕
Original assignee: 三菱電機株式会社
Priority date: 2016-10-17
Filing date: 2016-10-17
Publication date: 2018-04-26
Also published as: JP6644159B2; JPWO2018073869A1

Abstract

A motor drive device 2 according to the present invention drives a single-phase motor 12 and is provided with: a single-phase inverter 11 for supplying an AC voltage to the single-phase motor 12; and a control unit 25 for generating a PWM signal for performing PWM control on a plurality of switching elements in the single-phase inverter 11 on the basis of a signal for detecting a rotational position of the single-phase motor 12 and a voltage command. The control unit 25 makes the amplitude of the voltage command constant and increases an advance angle of the voltage command in accordance with an increase in the rotational speed of the single-phase motor 12.

Description

Motor drive device, electric blower, vacuum cleaner and hand dryer

The present invention relates to a motor driving device that drives a single-phase motor, and an electric blower, a vacuum cleaner, and a hand dryer equipped with a single-phase motor that is driven by the motor driving device.

A single-phase motor has the following advantages compared to a three-phase motor having a different number of phases.
(1) While a three-phase motor requires a three-phase inverter, a single-phase motor may be a single-phase inverter.
(2) When a full-bridge inverter that is generally used as a three-phase inverter is used, six switching elements are required, whereas in the case of a single-phase motor, four switching elements are used even if a full-bridge inverter is used. Can be configured.
(3) Due to the features of (1) and (2), the single-phase motor can be downsized as compared with the three-phase motor.

When driving a single-phase motor with a single-phase inverter, it is required to reduce the harmonic component of the current flowing through the single-phase motor. Patent Document 1 listed below describes a PWM for making a current (hereinafter referred to as “motor current”) flowing through a single-phase motor into a sine wave by controlling a supply voltage supplied to the single-phase motor in order to reduce harmonic components. (Pulse Width Modulation: pulse width modulation) control technology is disclosed. Patent Document 2 below discloses a method of switching output voltage pulses in accordance with switching of position sensor signals.

JP 2012-257457 A Japanese Patent No. 5524925

However, when PWM control is applied to a product with a wide motor rotational speed drive range such as a vacuum cleaner, the motor rotational speed drive range is narrowed only by controlling the supply voltage to the single-phase motor. There was a problem that it could not be driven until rotation.

The present invention has been made in view of the above, and is a motor drive device that does not interfere with application to a product that requires high-speed rotation even when PWM control is used for control of a single-phase motor. An object is to obtain an electric blower, a vacuum cleaner and a hand dryer.

In order to solve the above-described problems and achieve the object, a motor driving device according to the present invention includes a plurality of switching elements, a single-phase inverter that applies an AC voltage to a single-phase motor, and rotation of the single-phase motor. A control unit is provided that generates a PWM signal for PWM control of a plurality of switching elements based on a signal for detecting a position and a voltage command. The control unit performs control to make the amplitude of the voltage command constant and increase the advance angle of the voltage command in accordance with an increase in the rotation speed of the single-phase motor.

According to the present invention, even when PWM control is used for controlling a single-phase motor, there is an effect that application to a product that requires high-speed rotation is not hindered.

The block diagram which shows the structure of the motor drive system containing the motor drive device in this Embodiment Circuit diagram of the inverter shown in FIG. Block diagram showing a functional configuration for generating a PWM signal The block diagram which showed the structure shown in FIG. 3 in detail Time chart showing waveform examples of voltage command, PWM signal and motor applied voltage Waveform diagram showing change in inverter output voltage according to modulation factor FIG. 3 is a block diagram showing a functional configuration for calculating an advance phase that is an input signal to the carrier generation unit and the carrier comparison unit shown in FIGS. The figure which shows an example of the calculation method of advance angle phase Time chart showing the relationship between rotor magnetic pole position, position sensor signal and voltage command Time chart showing time change of voltage amplitude command The figure which shows the path of the motor current when the polarity of the inverter output voltage is positive and negative The figure which shows the path of the motor current when the return current flows through the body diode of the MOSFET The figure which shows the route of the motor current when flowing the reflux current through the channel of MOSFET The figure which shows an example of a structure of a vacuum cleaner as an application example of the motor drive device in this Embodiment The figure which shows an example of a structure of a hand dryer as another application example of the motor drive device in this Embodiment. The figure which uses for description of the modulation control suitable for the motor drive device in this Embodiment

Hereinafter, a motor drive device, an electric blower, and a vacuum cleaner according to an embodiment of the present invention will be described in detail based on the drawings. The present invention is not limited to the following embodiments. Moreover, although the following description demonstrates centering on the application example to a vacuum cleaner and a hand dryer, it is not the meaning which excludes the application to another use.

Embodiment.
FIG. 1 is a block diagram showing a configuration of a motor drive system including a motor drive device according to the present embodiment. As shown in FIG. 1, the motor drive system 1 includes a single-phase motor 12, a motor drive device 2 that drives the single-phase motor 12 by supplying AC power to the single-phase motor 12, and DC power to the motor drive device 2. , A voltage sensor 20 that detects a DC applied voltage applied to the motor drive device 2 by the power supply 10, and a rotor rotational position that is a rotational position of the rotor 12a in the single-phase motor 12. Position sensor 21. Examples of the load including the single-phase motor 12 include a vacuum cleaner and a hand dryer provided with an electric blower. In the above description, the voltage sensor 20 detects a DC applied voltage, but may detect a power supply voltage that is an output voltage of the power supply 10.

In addition, the motor drive device 2 is connected to the single-phase motor 12 and applies single-phase inverter 11 that applies an AC voltage to the single-phase motor 12 and analog data that is a detected value of the DC applied voltage detected by the voltage sensor 20. An analog / digital converter 30 for converting into digital data, a DC applied voltage read from the analog / digital converter 30, a position sensor signal from the position sensor 21, and a rotational speed command, that is, a rotational speed command, which is a command value not shown. A control unit 25 that generates a PWM signal based on the value, and a drive signal generation unit 32 that generates a drive signal for driving the switching element in the single-phase inverter 11 based on the PWM signal output from the control unit 25. .

The control unit 25 includes a processor 31, a carrier generation unit 33, and a memory 34. The processor 31 generates a PWM signal described later by PWM control. The drive signal generation unit 32 generates a drive signal for driving the single-phase inverter 11 based on the PWM signal from the processor 31 and outputs the drive signal to the single-phase inverter 11.

The single-phase motor 12 is preferably a brushless motor. In the brushless motor, a plurality of permanent magnets (not shown) arranged in the circumferential direction are arranged on the rotor 12a. The plurality of permanent magnets are arranged so that the magnetization direction is alternately reversed in the circumferential direction, and form a plurality of magnetic poles of the rotor 12a. A winding (not shown) is wound around the stator. The motor current is an alternating current flowing through the winding. In this embodiment, the number of magnetic poles is four, but the number of magnetic poles may be other than four.

The position sensor 21 outputs a position sensor signal that is a digital signal to the control unit 25. The position sensor signal is a signal for detecting the rotor rotational position, and shows a binary value of high and low according to the direction of the magnetic flux from the rotor 12a.

FIG. 2 is a circuit configuration diagram of the single-phase inverter 11. The single-phase inverter 11 includes

switching elements

51, 52, 53, and 54 that are bridge-connected. The

switching elements

51 and 53 located on the high potential side are referred to as upper arm switching elements. The

switching elements

52 and 54 located on the low potential side are referred to as lower arm switching elements. The connection end of the switching element 51 and the switching element 52 and the connection end of the switching element 53 and the switching element 54 constitute an AC end in the bridge circuit, and the AC phase is connected to the single-phase motor 12. .

MOSFETs (Metal-Oxide-Semiconductor Field Effect Transistor) are used for the

switching elements

51, 52, 53, and 54. A MOSFET is an example of an FET. A MOSFET is a switching element that allows a current to flow in both directions between a drain and a source. In each of the

switching elements

51, 52, 53, and 54, a diode connected in parallel between the drain and the source is referred to as a freewheeling diode. However, in the present embodiment, a body diode, which is a parasitic diode formed inside the MOSFET, is used as the freewheeling diode.

Further, at least one of the

switching elements

51, 52, 53, and 54 can be formed using a wide band gap semiconductor. The wide gap semiconductor is, for example, GaN (gallium nitride), SiC (silicon carbide), or diamond. By using a wide bandgap semiconductor for at least one of the

switching elements

51, 52, 53, and 54, the voltage resistance and allowable current density of the switching element are increased. Miniaturization is possible. In addition, since the wide band gap semiconductor has high heat resistance, it is possible to reduce the size of the heat radiating portion and simplify the heat radiating structure.

FIG. 3 is a block diagram showing a functional configuration for generating a PWM signal. 4 is a block diagram showing in detail the configuration shown in FIG. The function of generating the PWM signal can be realized by the carrier generation unit 33 and the carrier comparison unit 38 as shown in FIG.

In FIG. 3, the advance phase θ _v used when generating voltage commands V _m1 and V _m2 to be described later is input to the carrier generation unit 33. Here, “advance angle phase” represents “advance angle”, which is the “advance angle” of the voltage command, in terms of phase. The “advance angle” here is a phase difference between a motor applied voltage applied to the stator winding by the single-phase inverter 11 and a motor induced voltage induced in a stator winding (not shown). The “advance angle” takes a positive value when the motor applied voltage is ahead of the motor induced voltage.

Returning to FIG. 3, the advance angle controlled θ _v , the DC applied voltage V _dc , and the voltage amplitude command V * that is the amplitude value of the voltage command V _m are input to the carrier comparison unit 38. Carrier generating unit 33 generates a carrier based on the advanced angle phase theta _v. The carrier comparison unit 38 generates the PWM signals Q1 to Q4 based on the carrier, the DC applied voltage _Vdc, and the voltage amplitude command V *.

As illustrated in FIG. 4, the carrier generation unit 33 includes a carrier frequency setting unit 33 a. In the carrier frequency setting unit 33a, a carrier frequency f _C [Hz], which is a carrier frequency, is set, and a carrier synchronized with the advance phase θ _v is generated and output to the carrier comparison unit 38. As an example of the carrier waveform, a triangular wave carrier that goes up and down between “0” and “1” is shown at the tip of the arrow of the carrier frequency setting unit 33a. Note that the PWM control for the single-phase inverter 11, synchronous PWM control and it is asynchronous PWM control, there is no need to synchronize the carrier to advance the phase theta _v in the case of the asynchronous PWM control.

Further, as shown in FIG. 4, the carrier comparison unit 38 includes an absolute value calculation unit 38a, a division unit 38b,

multiplication units

38c, 38d, and 38f, an addition unit 38e,

comparison units

38g and 38h, and

output inversion units

38i and 38j. Has functional blocks.

The absolute value calculator 38a calculates the absolute value | V * | of the voltage amplitude command V *. In the division unit 38b, the absolute value | V * | is divided by the DC applied voltage _Vdc detected by the voltage sensor 20. When the power supply 10 is a battery, the battery voltage fluctuates, but by dividing by the detection value of the voltage sensor 20, the modulation rate can be increased so that the motor applied voltage does not decrease due to the decrease in the battery voltage. In FIG. 1, the DC applied voltage V _dc is detected by the voltage sensor 20. However, when the output voltage of the power source 10 is stable, such as when the power source 10 is connected to a commercial power source. Instead of using the detection value of the voltage sensor 20, a value generated internally may be used.

The multiplier unit 38c, the sine of advanced angle phase theta _v is calculated and multiplied by the output from the divider 38b. The multiplier 38d multiplies the output of the multiplier 38c by 1/2. The adder 38e adds 1/2 to the output result of the multiplier 38d. The multiplication unit 38f multiplies the output of the division unit 38b by -1. Here, the output of the adder 38e is input to the comparator 38g as a voltage command V _m1 used for driving the

switching elements

51, 52 among the switching

elements

51, 52, 53, 54, and the output of the multiplier 38f is The voltage command V _m2 used for driving the

switching elements

53 and 54 is input to the comparison unit 38h.

The output of the comparison unit 38g becomes a PWM signal to the switching element 51, and the output of the output inversion unit 38i obtained by inverting the output of the comparison unit 38g becomes a PWM signal to the switching element 52. Similarly, the output of the comparison unit 38h becomes a PWM signal to the switching element 53, and the output of the output inversion unit 38j obtained by inverting the output of the comparison unit 38h becomes a PWM signal to the switching element 54. The switching element 51 and the switching element 52 are not simultaneously turned on due to the presence of the output inverting part 38i, and the switching element 53 and the switching element 54 are not simultaneously turned on due to the presence of the output inverting part 38j.

FIG. 5 is a time chart showing waveform examples of the voltage commands V _m1 and V _m2 , the PWM signal, and the motor applied voltage. The upper part of FIG. 5 shows the waveform of the voltage command V _m1 output from the adder 38e, and the middle upper part of FIG. 5 shows the waveform of the voltage command V _m2 output from the multiplier 38f. ing. By using the voltage command V _m1 , PWM signals Q1 and Q2 as shown in the middle and lower part of FIG. 5 can be generated. Further, by using the voltage command V _m2 , PWM signals Q3 and Q4 as shown in the middle lower part of FIG. 5 can be generated. Further, by using such PWM signals Q1, Q2, Q3, and Q4 to control the switching

elements

51, 52, 53, and 54 in the single-phase inverter 11, the PWM as shown in the lower part of FIG. A controlled voltage pulse waveform can be applied to the single phase motor 12.

By the way, as a modulation method used when generating a PWM signal, bipolar modulation that outputs a voltage pulse that changes at a positive or negative potential, a positive or zero potential, or a negative or zero potential every half cycle of the power source, That is, unipolar modulation that outputs voltage pulses that change at three potentials of positive, zero, or negative is known. The waveform shown in FIG. 5 is based on unipolar modulation. Any modulation method may be used as the motor driving device in the present embodiment. In applications where it is necessary to control the motor current waveform to a sine wave, it is preferable to employ unipolar modulation with a lower harmonic content than bipolar modulation.

As described above, the motor applied voltage is determined by comparing the carrier and the voltage command value. Voltage pulse included in the motor applied voltage, during high-speed rotation, and also increased the frequency of the voltage command V _m, because the number of voltage pulses outputted to the electric angle in one cycle is decreased, the number of voltage pulse current The effect on waveform distortion is also increased. In general, when an even number of voltage pulses are applied, even-order harmonics are superimposed, and the symmetry of the positive and negative waveforms is lost. Therefore, it is preferable to control the number of voltage pulses in one electrical angle cycle to be an odd number so that the motor current waveform approaches a sine wave with a reduced harmonic content. By controlling the number of voltage pulses in one electrical angle cycle to be an odd number, the motor current waveform can be made closer to a sine wave.

Next, a method for setting the number of voltage pulses in one electrical angle cycle to an odd number will be described. In order to control the output voltage pulse number to be an odd number during high-speed rotation, there is a method of synchronizing the triangular wave carrier with the motor rotation number. In addition, when performing control using the rotational speed as a command value in the product specification, there is a method of determining a carrier frequency in advance with respect to the rotational speed command. Since either method is a known technique, a detailed description thereof is omitted here. In the present embodiment, the control is not limited to this as long as the output voltage is an odd number of times.

FIG. 6 is a waveform diagram showing changes in the inverter output voltage in accordance with the modulation rate. From the upper side, three patterns in the case of modulation rate = 1.0, modulation rate = 1.2, and modulation rate = 2.0 are shown. The “inverter output voltage” has the same meaning as the “motor applied voltage” shown in the lower part of FIG.

As described with reference to FIGS. 4 and 5, the voltage commands V _m1 and V _m2 are compared with the carrier, and PWM signals Q1, Q2, Q3, and Q4 corresponding to the comparison result are generated. Further, the switching

elements

51, 52, 53, and 54 in the single-phase inverter 11 are controlled by the generated PWM signals Q1, Q2, Q3, and Q4, and according to the modulation rate as shown in each stage of FIG. An inverter output voltage is generated and applied to the single phase motor 12.

There are various modulation rate definitions. Here, the ratio between the voltage amplitude command V * and the amplitude of the triangular wave carrier, that is, “voltage amplitude command V * / triangular wave carrier amplitude” is defined as the modulation rate. The upper part of FIG. 6 shows the case where the modulation rate is 1.0, but the same waveform is obtained when the modulation rate is less than 1.0. When the modulation factor is less than 1.0, the inverter output voltage is generated according to the frequency of the triangular wave carrier, and therefore, the inverter output voltage also outputs a voltage pulse corresponding to the carrier frequency.

On the other hand, when the modulation factor exceeds 1.0, the waveforms are as shown in the middle and lower parts of FIG. When the modulation rate exceeds 1.0, it is called “overmodulation”, and the region where the modulation rate exceeds 1.0 is called “overmodulation region”. In the overmodulation region, a portion where the voltage commands V _m1 and V _m2 exceed the carrier amplitude occurs, and therefore an interval in which an inverter drive signal cannot be generated occurs according to the carrier frequency. In this section, since the inverter output voltage is fixed to a positive power supply voltage or a negative power supply voltage, it is possible to obtain a larger output voltage than the inverter output voltage when the modulation factor is 1.

Next, the advance angle control in the present embodiment will be described. Figure 7 is a block diagram showing a functional configuration for calculating the input signal is advanced angle phase theta _v of the carrier generating unit 33 and the carrier comparison section 38 shown in FIGS. Functions for calculating the advance phase theta _v, as shown in FIG. 7, the rotational speed of the single-phase motor 12 based on the position sensor signals omega, and the rotor mechanical angle theta is the angle from a reference position of the rotor 12a A rotation speed calculation unit 42 that calculates a reference phase θ _{e in} which _m is converted to an electrical angle, and an advance angle phase θ _v are calculated based on information about the rotation speed ω and the reference phase θ _e calculated by the rotation speed calculation unit 42. This can be realized by the advance angle calculation unit 44.

Figure 8 is a diagram showing an example of a calculation method of the advanced angle phase theta _v. Advanced angle phase theta _v, as shown in FIG. 8, may be determined using a function advanced phase theta _v increases with increasing rotational speed N. In the example of FIG. 8, the advance phase θ _v is determined by a linear function, but not limited to this, as long as the advance phase θ _v increases with an increase in the number of rotations. Any function may be used.

9, the magnetic pole position of the rotor is a time chart showing the relationship between the position sensor signal and the voltage command V _m. In addition, as shown in the lowermost stage of FIG. 9, the rotor is provided with a four-pole magnet, and four stators are provided on the outer periphery of the rotor. When the rotor rotates in the clockwise direction, a position sensor signal corresponding to the rotor mechanical angle θ _m is detected, and a reference phase θ _e converted to an electrical angle is calculated according to the detected position sensor signal.

In the middle part of FIG. 9, the case where the advance angle phase θ _v = 0 is shown as Example 1. When the advance phase θ _v = 0, the voltage command V _m outputs a sine wave voltage in phase with the reference phase θ _e . The amplitude of the voltage command V _m at this time is determined based on the voltage amplitude command V * as described above. A case where the advance phase θ _v = π / 4 is shown as Example 2. When the advance phase θ _v = π / 4, the voltage command V _m has a waveform that is advanced from the reference phase θ _e by the advance angle phase θ _v , that is, π / 4.

The functions of the carrier comparison unit 38 shown in FIG. 4 and the functions of the rotation speed calculation unit 42 and the advance angle calculation unit 44 shown in FIG. 7 can be realized by the processor 31 shown in FIG. The processor 31 is a processing unit that performs various calculations related to PWM control and advance angle control. The memory 34 stores a program read by the processor 31. The memory 34 is used as a work area when the processor 31 performs arithmetic processing. The processor 31 may be a CPU (Central Processing Unit), a microprocessor, a microcomputer, or a DSP (Digital Signal Processor). The memory 34 is typically a nonvolatile or volatile semiconductor memory such as a RAM (Random Access Memory), a flash memory, an EPROM (Erasable Programmable ROM), or an EEPROM (Electrically EPROM).

Here, the reason why the PWM control is not used so much in the control of the single phase motor in the prior art will be described. As a method of driving a single-phase motor, a method of switching output voltage pulses in accordance with switching of position sensor signals as shown in Patent Document 2 described above is common. Here, as an example of a product that drives a single-phase motor, if a vacuum cleaner or a hand dryer is taken as an example, it is necessary to rotate the single-phase motor at a high speed of 100,000 rotations or more in these products. High-speed arithmetic processing was necessary to control the sine wave. In the conventional environment, it has been difficult to obtain a microcomputer capable of high-speed arithmetic processing at low cost, and thus PWM control has not been performed.

Next, how to give the voltage amplitude command V * will be described. FIG. 10 is a time chart showing a time change of the voltage amplitude command V *. In the present embodiment, the voltage amplitude command V * is an operation mode that changes stepwise according to time t, as shown. Specifically, first, a constant first voltage V ₁ set in advance is applied at the time of startup, and a constant second voltage V ₂ greater than the first voltage V1 is applied during steady operation after acceleration. Further, at the time of acceleration is changed from the first voltages V ₁ to the second voltage V _2, to raise the voltage amplitude command V * as acceleration rate set in advance is obtained. That is, in the present embodiment, the voltage amplitude command V * is controlled to be constant during startup and during steady operation. At the time of start-up, the time τ1 for applying the first voltage V1 can be set to an arbitrary time in consideration of the stabilization time of the control system.

Next, the effect of the voltage amplitude command V * being constant will be described. The following effects can be obtained by controlling the voltage amplitude command V * to be constant during steady operation.

(1) Even when the load suddenly changes, a constant voltage command can be output based on the phase detected from the position sensor signal.
(2) Since the voltage command is not affected even when the rotation speed fluctuates, the output voltage can be kept stable.

The above effect is effective for an application where the load varies depending on the contact area between the suction port of the vacuum cleaner and the floor, such as a vacuum cleaner.

When performing constant rotation speed control that is implemented in a general electric blower, overcurrent may flow through the motor. The reason for the overcurrent flowing is that the current fluctuates abruptly because it tries to keep the rotation speed constant when the load fluctuates. More specifically, when the rotational speed constant control is performed when the state is changed from the “light load” state, that is, the “load torque is small state” to the “heavy load state”, that is, the “load torque is large state”, This is because the motor output torque must be increased in order to maintain the same rotation speed, and the amount of change in motor current increases.

On the other hand, in the control of the present embodiment, as described above, the control is performed to keep the voltage amplitude command V * constant during steady operation. Here, when the voltage amplitude command V * is constant, the voltage amplitude command V * is not changed when the load becomes heavy, and therefore the motor rotation speed decreases as the load torque increases. By this control, it is possible to prevent a steep change in motor current and an overcurrent, and thus it is possible to realize an electric blower and an electric vacuum cleaner that rotate stably.

In the case of an electric blower, the load torque increases with an increase in the number of rotations of the blades, which is the load of the motor, and also increases with an increase in the diameter of the air passage. The diameter of the air passage represents the size of the suction port when an electric vacuum cleaner is taken as an example. When the diameter of the air passage is wide, if nothing is in contact with the suction port, a force for sucking in the wind is required. Therefore, when the blades are rotating at the same rotation speed, the load torque is increased. On the other hand, when the diameter of the air passage is narrow, when the suction port is in contact with something and is blocked, the force to suck in the wind is no longer necessary, so when the blades are rotating at the same speed, the load torque is Get smaller.

Next, the effect of the advance angle control will be described. First, if to increase the advance phase theta _v in accordance with an increase in rotational speed, it is possible to extend the rotational speed range. The advance phase theta _v when 0, the rotational speed is saturated at which the voltage applied to the motor and the motor induced voltage is balanced. To further increase the rotational speed advances the advanced angle phase theta _v, suppresses motor induced voltage by weakening the magnetic flux to be generated in the stator by an armature reaction, increasing the rotational speed. Thus, by selecting the advanced angle phase theta _v according to the rotation speed, it is possible to obtain a wide speed range.

When the advance angle control according to this embodiment is applied to a vacuum cleaner, the voltage command is made constant regardless of the change in the suction port closing state, that is, regardless of the load torque, and according to the increase in the rotational speed. it suffices to increase the advance phase theta _v is a lead angle of the voltage command Te. By controlling in this way, stable driving is possible in a wide rotational speed range.

Next, the loss reduction technique in this embodiment will be described. 11 to 13 are diagrams showing motor current paths according to the polarity of the inverter output voltage. When the polarity of the inverter output voltage is positive, it flows into the single-phase motor 12 through the channel portion of the MOSFET 51, which is the upper arm of the first phase, as shown by the thick solid line (a) in FIG. It flows out of the single-phase motor 12 through the channel portion of the MOSFET 54 that is an arm. Further, when the polarity of the inverter output voltage is negative, as shown by the thick broken line (b) in FIG. 11, the inverter flows into the single-phase motor 12 through the channel portion of the MOSFET 53 that is the upper arm of the second phase. The single-phase motor 12 flows out through the channel portion of the MOSFET 52 which is the lower arm.

Next, the current path when the inverter output voltage is zero, that is, when zero voltage is output from the single-phase inverter 11, will be described. When the inverter output voltage becomes zero after the positive inverter output voltage is generated, no current flows from the power source side as shown by the thick solid line (c) in FIG. 12, and the single-phase inverter 11 and the single-phase motor 12 It becomes the recirculation | reflux mode in which an electric current goes back and forth between. At this time, since the direction of the current that flows immediately before the single-phase motor 12 does not change, the current that flows from the single-phase motor 12 is the channel portion of the MOSFET 54, which is the lower arm of the second phase, and the lower phase of the first phase. The arm 52 returns to the single-phase motor 12 through the body diode portion of the MOSFET 52. In addition, when the inverter output voltage becomes zero after the negative inverter output voltage is generated, the direction of the current that has flowed immediately before is reversed, and as shown by the thick broken line (d) in FIG. The direction of the current is reversed. More specifically, the current that flows out of the single-phase motor 12 passes through the body diode portion of the MOSFET 51 that is the upper arm of the first phase and the channel portion of the MOSFET 53 that is the upper arm of the second phase. Return to.

As described above, in the reflux mode in which current flows back between the single-phase motor 12 and the single-phase inverter 11, a current flows through the body diode in one of the first phase and the second phase. Generally, it is known that the conduction loss is smaller when a current is passed through a MOSFET channel than when a current is passed in the forward direction of a diode. Therefore, in the present embodiment, in the return mode in which the return current flows, the MOSFET on the side having the body diode is controlled to be turned on in order to reduce the current flowing through the body diode.

In the return mode, the MOSFET 52 is controlled to be turned on at the timing when the return current indicated by the thick solid line (c) in FIG. 12 flows. If controlled in this way, as shown by the thick solid line (e) in FIG. 13, most of the reflux current flows through the channel side of the MOSFET 52 having a small resistance value. Thereby, the semiconductor loss in MOSFET52 is reduced. Further, the MOSFET 51 is controlled to be turned on at the timing when the return current indicated by the thick broken line (d) in FIG. 12 flows. By controlling in this way, as shown by the thick broken line (f) in FIG. 13, most of the reflux current flows through the channel side of the MOSFET 51 having a small resistance value. Thereby, the semiconductor loss in MOSFET51 is reduced.

As described above, the loss of the switching element can be reduced by turning on the MOSFET on the side having the body diode at the timing when the return current flows through the body diode. For this reason, the structure of the MOSFET is made a surface mount type so that heat can be radiated on the substrate, and if necessary, part or all of the switching element is formed of a wide band gap semiconductor, so that the MOSFET generates heat only on the substrate. The structure which suppresses is realized. Note that if heat can be radiated only by the substrate, a heat sink is unnecessary, which contributes to the miniaturization of the inverter and can lead to the miniaturization of the product.

In addition to the heat dissipation method described above, a further heat dissipation effect can be obtained by installing the substrate in the air path. Here, the air passage is a portion that generates an air flow, such as an electric blower, or an air flow passage generated by the electric blower. By installing the substrate in the air path, the semiconductor element on the substrate can be radiated by the wind generated by the electric blower, so that the heat generation of the semiconductor element can be significantly suppressed.

Next, an application example of the motor drive device in the present embodiment will be described. FIG. 14 is a diagram illustrating an example of a configuration of a vacuum cleaner as an application example of the motor driving device in the present embodiment. In FIG. 14, the vacuum cleaner 61 includes a battery 67 that is a DC power source and an electric blower 64 that is driven by the single-phase motor 12 described above, and further includes a dust collection chamber 65, a sensor 68, a suction port body 63, and an extension pipe. 62 and an operation unit 66.

The electric vacuum cleaner 61 drives the electric blower 64 using the battery 67 as a power source, performs suction from the suction port body 63, and sucks dust into the dust collection chamber 65 through the extension pipe 62. In use, the operation unit 66 is held and the electric vacuum cleaner 61 is operated.

The electric vacuum cleaner 61 is a product in which the drive range of the motor rotation speed varies from 0 to 100,000 rpm or more. When such a motor drives a product that rotates at high speed, the control method according to the above-described embodiment is suitable. And constant voltage amplitude command V *, by changing the advanced angle phase theta _v according to the rotational speed, while widening the rotational speed drive range from a low speed to a high speed rotation region, it is possible to correspond to abrupt load change. In addition, since the motor current can be controlled to a sine wave by PWM control, high-efficiency driving can be achieved, so that the operation time can be extended.

Moreover, it can contribute to size reduction and weight reduction by the reduction of the heat dissipation component mentioned above. Furthermore, since a current sensor for detecting current is not required and a high-speed analog-digital converter is not required, cost can be suppressed.

FIG. 15 is a diagram illustrating an example of a configuration of a hand dryer as another application example of the motor drive device according to the present embodiment. The hand dryer 90 includes a casing 91, a hand detection sensor 92, a water receiver 93, a drain container 94, a cover 96, a sensor 97, and an intake port 98. Here, the sensor 97 is either a gyro sensor or a human sensor. The hand dryer 90 has an electric blower (not shown) in the casing 91. The hand dryer 90 has a structure in which water is blown off by blowing with an electric blower by inserting a hand into the hand insertion portion 99 at the top of the water receiving portion 93 and water is stored from the water receiving portion 93 into the drain container 94. ing.

The hand dryer 90 is also a product in which the drive range of the motor rotation speed varies from 0 to 100,000 rpm or more. For this reason, also in the hand dryer 90, the control method which concerns on this Embodiment mentioned above is suitable, and the effect similar to the vacuum cleaner 61 can be acquired.

FIG. 16 is a diagram for explaining modulation control suitable for the motor drive device according to the present embodiment. The left side of the figure shows the relationship between the rotational speed and the modulation rate. The right side of the figure shows the waveform of the inverter output voltage when the modulation rate is 1 or less and exceeds 1. In general, the load torque of the rotating body increases as the number of rotations increases. For this reason, it is necessary to increase the motor output torque as the rotational speed increases. In general, the motor output torque increases in proportion to the motor current, and the inverter output voltage needs to be increased to increase the motor current. Therefore, it is possible to increase the rotational speed by increasing the modulation rate and increasing the inverter output voltage.

Next, the rotational speed control in the present embodiment will be described. In the following description, an electric blower is assumed as a load, and an operation range of the electric blower is divided as follows.
(1) Low speed range (low speed range): 0 to 70,000 rpm
(2) High speed rotation range (high rotation speed range): 100,000 rpm or more

In addition, the area | region between said (1) and (2) is a gray zone, and depending on a use, it may be included in a low-speed rotation area, and may be included in a high-speed rotation area.

First, the control in the low speed rotation region will be described. In the low-speed rotation range, PWM control is performed with a modulation rate of 1 or less. Note that, by setting the modulation rate to 1 or less, the motor current can be controlled to a sine wave, and the efficiency of the motor can be increased. When operating at the same carrier frequency in the low-speed rotation region and the high-speed rotation region, the carrier frequency becomes a carrier frequency that matches the high-speed rotation region, and therefore the PWM pulse tends to increase more than necessary in the low-speed rotation region. . For this reason, in the low-speed rotation region, a method of reducing the carrier frequency and reducing the switching loss may be used. Further, by changing the carrier frequency in synchronization with the rotation speed, control may be performed so that the pulse number does not change according to the rotation speed.

Next, control in the high speed rotation range will be described. In the high speed rotation range, the modulation rate is set to a value larger than 1. By increasing the modulation factor above 1, it is possible to suppress an increase in switching loss by increasing the inverter output voltage and reducing the number of switching operations performed by the switching elements in the inverter. Here, when the modulation rate exceeds 1, the motor output voltage increases, but since the number of switching times decreases, there is a concern about current distortion. However, during high speed rotation, the reactance component of the motor increases and di / dt, which is a change component of the motor current, decreases. Therefore, current distortion is smaller than in the low speed rotation range, and the influence on waveform distortion is small. Become. Therefore, in the high-speed rotation region, the modulation rate is set to a value larger than 1 and control is performed to reduce the number of switching pulses. By this control, an increase in switching loss can be suppressed and high efficiency can be achieved.

As mentioned above, the boundary between the low-speed rotation range and the high-speed rotation range is gray. For this reason, the first rotation speed that determines the boundary between the low-speed rotation range and the high-speed rotation range is set. When the rotation speed of the first rotation speed exceeds the first rotation speed, the modulation rate may be controlled to be greater than 1.

As described above, in the present embodiment, the vacuum cleaner 61 and the hand dryer 90 have been described as application examples of the motor driving device, but the present invention can be applied to general electric devices equipped with a motor. Electrical equipment equipped with motors are incinerators, crushers, dryers, dust collectors, printing machines, cleaning machines, confectionery machines, tea making machines, woodworking machines, plastic extruders, cardboard machines, packaging machines, hot air generators, objects It is a device equipped with an electric blower such as transport, dust absorption, general air supply / exhaust, or OA device.

Further, the configuration shown in the above embodiment shows an example of the content of the present invention, and can be combined with another known technique, and can be combined without departing from the gist of the present invention. It is also possible to omit or change a part of.

1 motor drive system, 2 motor drive device, 10 power supply, 11 single-phase inverter, 12a rotor, 12 single-phase motor, 20 voltage sensor, 21 position sensor, 25 control unit, 30 analog-digital converter, 31 processor, 32 drive signal Generation unit, 33 carrier generation unit, 33a carrier frequency setting unit, 34 memory, 38 carrier comparison unit, 38a absolute value calculation unit, 38b division unit, 38c, 38d, 38f multiplication unit, 38e addition unit, 38g, 38h comparison unit, 38i, 38j, output reversal unit, 42 rotation speed calculation unit, 44 advance phase calculation unit, 51, 52, 53, 54 switching element, 61 vacuum cleaner, 62 extension pipe, 63 suction port, 64 electric blower, 65 collection Dust chamber, 66 operation unit, 67 battery, 68 sec Sa, 90 hand dryers, 91 casing, 92 hand detecting sensor, 93 water receiving unit, 94 drain container 96 cover, 97 sensors, 98 inlet, 99 hand insertion portion.

Claims

A motor driving device for driving a single-phase motor,
Comprising a plurality of switching elements, comprising a single-phase inverter for applying an AC voltage to the single-phase motor,
PWM control of the plurality of switching elements based on a voltage command and a signal for detecting the rotational position of the single-phase motor,
A motor drive device that makes the amplitude of the voltage command constant and increases the advance angle of the voltage command in accordance with an increase in the rotational speed of the single-phase motor.
The plurality of switching elements are MOSFETs,
The motor driving device according to claim 1, wherein at a timing when a return current flows through the single-phase inverter, a MOSFET in which the return current flows through a body diode is controlled to be on.
The motor drive device according to claim 1 or 2, wherein the PWM signal for performing the PWM control is generated by unipolar modulation.
At startup, a predetermined first voltage is applied to the voltage amplitude command, which is the amplitude of the voltage command,
The motor drive device according to any one of claims 1 to 3, wherein a constant second voltage larger than the first voltage is applied during steady operation after acceleration.
Set the first rotation speed that determines the boundary between the low-speed rotation range and the high-speed rotation range,
When the rotational speed of the single-phase motor is equal to or lower than the first rotational speed, the modulation rate of the PWM control is set to 1 or lower, and when the rotational speed exceeds the first rotational speed, the modulation rate The motor driving device according to any one of claims 1 to 4, wherein the motor driving device is set to a value exceeding 1.
6. The motor driving device according to claim 1, wherein at least one of the plurality of switching elements is formed of a wide band gap semiconductor.
The motor driving device according to claim 6, wherein the wide band gap semiconductor is silicon carbide, gallium nitride, or diamond.
An electric blower comprising the motor driving device according to any one of claims 1 to 7.
A vacuum cleaner comprising the electric blower according to claim 8.
The electric vacuum cleaner according to claim 9, wherein a substrate on which a plurality of the switching elements are mounted is installed in a passage of air flow generated by the electric blower.
A hand dryer equipped with the electric blower according to claim 8.
The hand dryer according to claim 11, wherein a substrate on which a plurality of the switching elements are mounted is installed in a passage of air flow generated by the electric blower.