JP6671516B2 - Motor drive, electric blower, vacuum cleaner and hand dryer - Google Patents

Description

  The present invention relates to a motor drive device for driving a single-phase motor, an electric blower equipped with a single-phase motor driven by the motor drive device, a vacuum cleaner, and a hand dryer.

A single-phase motor has the following advantages over a three-phase motor having three phases.
(1) While a three-phase inverter must be used for a three-phase motor, a single-phase inverter whose configuration is simpler than that of a three-phase inverter may be used for a single-phase motor.
(2) While a three-phase inverter using a full-bridge inverter requires six switching elements, a single-phase motor can be configured with four switching elements even if a full-bridge inverter is used.
(3) Due to the features of (1) and (2), a device using a single-phase motor can be downsized compared to a device using a three-phase motor.

  When a single-phase motor is driven by a single-phase inverter, the single-phase inverter is required to reduce harmonic components of a current flowing through the single-phase motor.

  Patent Literature 1 discloses a pulse width modulation (PWM) control for controlling a voltage supplied to a single-phase motor to control a current supplied to the single-phase motor to a sine wave in order to reduce harmonic components. A technique for performing this is disclosed. The current flowing through the single-phase motor may be simply referred to as “motor current” below.

  Patent Literature 2 discloses a method of switching output voltage pulses according to switching of a position sensor signal.

  Patent Document 3 discloses that in a control drive device of a three-phase sensorless DC brushless motor, a delay angle of a conduction phase is changed according to a value of power consumption obtained by detecting a DC current.

JP 2012-257457 A Patent No. 5524925 Patent No.3183071

  However, in a motor drive device that drives a small-diameter motor that requires high-speed rotation, such as a motor mounted on a vacuum cleaner, if the position of the position sensor is shifted, the position sensor signal and the motor induced A phase difference occurs between the voltage and the voltage, and the driving speed varies. Patent Literatures 1, 2, and 3 have a problem in that a method for suppressing a variation in the driving rotation speed due to such a phase difference is not disclosed.

  SUMMARY OF THE INVENTION The present invention has been made in view of the above, and has been made in consideration of the above-described problems, and has been made in consideration of the above-described problems, and has been made in consideration of the following circumstances. The aim is to obtain a device.

  In order to solve the above-described problems and achieve the object, a motor driving device according to the present invention is a motor driving device that drives a single-phase motor using a battery as a power source, the motor driving device including a plurality of switching elements, and A single-phase inverter that applies an AC voltage to the single-phase motor, a position sensor that is attached to the single-phase motor and detects a rotor rotation position of the single-phase motor and outputs a rotation position detection signal, and a voltage command and a rotation position detection signal. A control unit for performing pulse width modulation control of the plurality of switching elements, the control unit including a phase adjustment angle for adjusting a phase difference between the rotational position detection signal and the output voltage of the single-phase inverter, and a single-phase A lead angle adjustment width corresponding to a variation range of the position where the position sensor is attached to the motor is set, and the control unit is used to calculate a voltage command using the phase adjustment angle and the lead angle adjustment width. Setting the advanced angle phase relative to the constant rotation speed.

  ADVANTAGE OF THE INVENTION The motor drive device which concerns on this invention has the effect that even when using the single phase motor driven by the electric power supplied from a battery, the fluctuation | variation of the drive rotation speed resulting from the position displacement of a position sensor can be suppressed.

1 is a block diagram illustrating a configuration of a motor drive system including a motor drive device according to an embodiment. Circuit configuration diagram of the single-phase inverter shown in FIG. The figure which shows the functional structure for producing | generating a PWM signal. FIG. 3 is a diagram showing details of a carrier comparison unit and a carrier generation unit shown in FIG. 4 is a time chart showing respective waveform examples of the positive voltage command, the negative voltage command, the PWM signal, and the motor applied voltage shown in FIG. Diagram showing change in inverter output voltage according to modulation factor The figure which shows the functional structure for calculating the advance phase input into the carrier generation part and the carrier comparison part shown in FIG.3 and FIG.4. The figure which shows an example of the calculation method of an advance angle phase 1st diagram showing the positional relationship between a position sensor, a stator and a rotor 2nd diagram showing the positional relationship between the position sensor, the stator and the rotor Diagram showing position sensor signal and motor induced voltage Flow chart for explaining the operation for determining the advance angle phase The figure which shows the relationship between a position sensor signal, a rotor mechanical angle, a reference phase, and a voltage command. Diagram showing time change of voltage amplitude command FIG. 1 is a first diagram illustrating a path of a motor current according to a polarity of an inverter output voltage. FIG. 2 is a second diagram illustrating a path of the motor current according to the polarity of the inverter output voltage. Third diagram showing the path of the motor current according to the polarity of the inverter output voltage Configuration diagram of a vacuum cleaner including a motor drive device according to an embodiment Configuration diagram of a hand dryer including a motor drive device according to an embodiment Diagram for explaining modulation control in a motor drive device according to an embodiment

  Hereinafter, a motor drive device, an electric blower, a vacuum cleaner, and a hand dryer according to an embodiment of the present invention will be described in detail with reference to the drawings. The present invention is not limited by the embodiment.

Embodiment.
FIG. 1 is a block diagram showing a configuration of a motor drive system including a motor drive device according to an embodiment. The motor drive system 1 shown in FIG. 1 supplies a single-phase motor 12, a motor drive device 2 that supplies AC power to the single-phase motor 12 to drive the single-phase motor 12, and supplies DC power to the motor drive device 2. A power supply 10 that is a DC power supply, a voltage sensor 20 that detects a DC voltage _Vdc output from the power supply 10 to the motor driving device 2, and a rotor rotation position that is a rotation position of a rotor 12 a built in the single-phase motor 12 are determined. And a position sensor 21 for detecting.

  The single-phase motor 12 is used as a rotating electric machine that rotates an electric blower (not shown), and the electric blower and the single-phase motor 12 are mounted on devices such as a vacuum cleaner and a hand dryer.

In the present embodiment, the voltage sensor 20 detects the DC voltage V _dc , but the detection target of the voltage sensor 20 is not limited to the DC voltage V _dc output from the power supply 10, and the output of the motor driving device 2 is not limited. The voltage may be an inverter output voltage. “Inverter output voltage” is synonymous with “motor applied voltage” described later.

The motor drive device 2 is connected to the single-phase motor 12, and converts a single-phase inverter 11 that applies an AC voltage to the single-phase motor 12 and analog data that is a DC voltage _Vdc detected by the voltage sensor 20 to digital data. And an analog-to-digital converter 30.

  The motor drive device 2 includes a control unit 25 that generates PWM signals Q1, Q2, Q3, and Q4, and a switching unit in the single-phase inverter 11 based on the PWM signals Q1, Q2, Q3, and Q4 output from the control unit 25. A drive signal generation unit 32 that generates a drive signal for driving the element. The control unit 25 uses the DC voltage converted by the analog-to-digital converter 30, the position sensor signal 21 a that is a rotation position detection signal output from the position sensor 21, and a rotation speed command value output from a device (not shown). PWM signals Q1, Q2, Q3, and Q4 are generated based on a certain rotational speed command. The position sensor signal 21a is a binary digital signal that changes according to the direction of the magnetic flux generated by the rotor 12a, and is input to the control unit 25.

  The control unit 25 has a processor 31, a carrier generation unit 33, and a memory. The processor 31 generates PWM signals Q1, Q2, Q3, and Q4 by PWM control.

  The processor 31 is a processing unit that performs various calculations related to the PWM control and the advance angle control. The function of the carrier comparison unit 38, the function of the rotation speed calculation unit 42, and the function of the advance phase calculation unit 44, which will be described later, are realized by the processor 31. The processor 31 may be referred to as a CPU (Central Processing Unit), a microprocessor, a microcomputer, or a DSP (Digital Signal Processor).

  The memory 34 stores a program read by the processor 31. The memory 34 is used as a work area when the processor 31 performs arithmetic processing. The memory 34 is generally a nonvolatile or volatile semiconductor memory such as a RAM (Random Access Memory), a flash memory, an EPROM (Erasable Programmable ROM), and an EEPROM (Electrically EPROM). Details of the configuration of the carrier generation unit 33 will be described later.

  The drive signal generation unit 32 converts the PWM signals Q1, Q2, Q3, and Q4 output from the processor 31 into drive signals for driving the single-phase inverter 11 and outputs the drive signals to the single-phase inverter 11.

  The single-phase motor 12 is a brushless motor. On the rotor 12a of the single-phase motor 12, a plurality of permanent magnets (not shown) are arranged in the circumferential direction. The plurality of permanent magnets are arranged so that the magnetization directions are alternately reversed in the circumferential direction, and form a plurality of magnetic poles of the rotor 12a. A winding (not shown) is wound around the stator 12 b of the single-phase motor 12. The alternating current flowing through the winding corresponds to the “motor current” described above. In the present embodiment, the number of magnetic poles of the rotor 12a is four, but the number of magnetic poles of the rotor 12a may be other than four.

  FIG. 2 is a circuit configuration diagram of the single-phase inverter shown in FIG. The single-phase inverter 11 has a plurality of switching elements 51, 52, 53, 54 connected in a bridge. Each of the two switching elements 51 and 53 located on the high potential side is called an upper arm switching element. Each of the two switching elements 52 and 54 located on the low potential side is called a lower arm switching element. A connection end between the switching element 51 and the switching element 52 and a connection end between the switching element 53 and the switching element 54 constitute AC terminals in a bridge circuit, and the single-phase motor 12 is connected to these AC terminals.

  As each of the plurality of switching elements 51, 52, 53, and 54, a MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor), which is a metal oxide semiconductor field-effect transistor, is used. A MOSFET is an example of a FET (Field-Effect Transistor).

  In the switching element 51, a body diode 51a connected in parallel between the drain and the source of the switching element 51 is formed. In the switching element 52, a body diode 52a connected in parallel between the drain and the source of the switching element 52 is formed. In the switching element 53, a body diode 53a connected in parallel between the drain and the source of the switching element 53 is formed. In the switching element 54, a body diode 54a connected in parallel between the drain and the source of the switching element 54 is formed. Each of the plurality of body diodes 51a, 52a, 53a, 54a is a parasitic diode formed inside the MOSFET, and is used as a freewheeling diode.

  At least one of the plurality of switching elements 51, 52, 53, 54 is not limited to a MOSFET formed of a silicon-based material, but is formed of a wide band gap semiconductor such as silicon carbide, a gallium nitride-based material, or diamond. A MOSFET may be used.

  Generally, a wide band gap semiconductor has higher withstand voltage and heat resistance than a silicon semiconductor. Therefore, by using a wide band gap semiconductor for at least one of the plurality of switching elements 51, 52, 53, and 54, the withstand voltage and the allowable current density of the switching elements are increased, and the semiconductor module incorporating the switching elements is provided. Can be reduced in size. Wide bandgap semiconductors also have high heat resistance, so the heat radiating part for radiating the heat generated by the semiconductor module can be downsized, and the heat radiating structure for radiating the heat generated by the semiconductor module can be simplified. It is possible.

  FIG. 3 is a diagram showing a functional configuration for generating a PWM signal. FIG. 4 is a diagram showing details of the carrier comparison unit and the carrier generation unit shown in FIG. As described above, the function of generating the PWM signals Q1, Q2, Q3, and Q4 can be realized by the carrier generation unit 33 and the carrier comparison unit 38 shown in FIG.

In FIG. 3, the advance phase θ _v and the reference phase θ _e that are advanced and controlled to be used when generating a voltage command V _m described later are input to the carrier comparison unit 38. Reference phase theta _e is a phase obtained by converting the rotor mechanical angle theta _m in electrical angle which is the angle from the reference position of the rotor 12a. Here, the “advance angle phase” represents the “advance angle”, which is the “advance angle” of the voltage command, as a phase. The “lead angle” here is a phase difference between a motor applied voltage applied to the stator winding by the single-phase inverter 11 and a motor induced voltage induced in the stator winding (not shown). Note that when the motor applied voltage is ahead of the motor induced voltage, the “lead angle” takes a positive value.

In addition to the advance phase θ _v and the reference phase θ _e , the carrier comparison unit 38 also includes a carrier generated by the carrier generation unit 33, a DC voltage V _dc, and a voltage amplitude that is an amplitude value of the voltage command V _m. Command V * is input. The carrier comparing unit 38 generates the PWM signals Q1, Q2, Q3, and Q4 based on the carrier, the advance phase θ _v , the reference phase θ _e , the DC voltage _Vdc, and the voltage amplitude command V *.

A carrier frequency f _C [Hz], which is a carrier frequency, is set in the carrier generation unit 33. The arrowheads of the carrier frequency f _C, as an example of a carrier wave, a triangular wave carrier up and down between "0" and "1" is shown. Note that the PWM control of the single-phase inverter 11, there are a synchronous PWM control and the asynchronous PWM control, when the asynchronous PWM control, it is not necessary to synchronize the carrier to advance the phase theta _v.

  As shown in FIG. 4, the carrier comparison unit 38 includes an absolute value calculation unit 38a, a division unit 38b, a multiplication unit 38c, a multiplication unit 38d, a multiplication unit 38f, an addition unit 38e, a comparison unit 38g, a comparison unit 38h, and an output inversion unit. 38i and an output inversion unit 38j.

The absolute value calculator 38a calculates the absolute value | V * | of the voltage amplitude command V *. In the divider 38b, the absolute value | V * | is divided by the DC voltage _Vdc detected by the voltage sensor 20. When the power supply 10 is a battery, the battery voltage fluctuates. However, by dividing the absolute value | V * | by the DC voltage _Vdc , the modulation rate is increased so that the motor applied voltage does not decrease due to the decrease in the battery voltage. Can be done.

The multiplier unit 38c, the sine of the reference phase theta _e and advanced phase theta _v is calculated, the sine value of the computed advanced phase theta _v is multiplied by the output from the divider 38b. The multiplier unit 38d, 1/2 is multiplied by the voltage command _{V m} is the output of the multiplication unit 38c. The adder 38e adds １ ／ to the output of the multiplier 38d. The multiplier 38f multiplies the output of the adder 38e by -1. Here, the output of the adder 38e, of the plurality of switching elements 51, 52, 53, 54, comparing section 38g as a voltage command _{V m} of positive side for driving the two switching elements 51, 53 of the upper arm It is input, the output of the multiplication unit 38f is input to the comparison unit 38h as a voltage command V _m2 of the negative side for driving the two switching elements 52, 54 of the lower arm.

  The output of the comparing section 38g becomes a PWM signal Q1 to the switching element 51, and the output of the output inverting section 38i obtained by inverting the output of the comparing section 38g becomes a PWM signal Q2 to the switching element 52. Similarly, the output of the comparing section 38h becomes a PWM signal Q3 to the switching element 53, and the output of the output inverting section 38j obtained by inverting the output of the comparing section 38h becomes a PWM signal Q4 to the switching element 54. The switching element 51 and the switching element 52 are not simultaneously turned on by the output inversion section 38i, and the switching element 53 and the switching element 54 are not simultaneously turned on by the output inversion section 38j.

FIG. 5 is a time chart showing respective waveform examples of the positive voltage command, the negative voltage command, the PWM signal, and the motor applied voltage shown in FIG. FIG. 5 shows, in order from the top, the waveform of the voltage command V _m1 output from the adder 38e, the waveform of the voltage command V _m2 output from the multiplier 38f, and the waveforms of the PWM signals Q1, Q2, Q3, and Q4. , And the waveform of the motor applied voltage. The PWM signals Q1, Q2, Q3, and Q4 are generated by using the voltage commands _Vm1 and _Vm2 . The motor driving device 2 shown in FIG. 1 controls a plurality of switching elements 51, 52, 53, and 54 in the single-phase inverter 11 using the PWM signals Q1, Q2, Q3, and Q4, thereby obtaining the configuration shown in FIG. A motor application voltage as shown, that is, a PWM-controlled voltage pulse is applied to the single-phase motor 12.

  By the way, as a modulation method used when generating the PWM signals Q1, Q2, Q3, and Q4, bipolar modulation that outputs a voltage pulse that changes at a positive or negative potential and change at three potentials every power supply half cycle are available. 2. Description of the Related Art Unipolar modulation that outputs a voltage pulse, that is, a voltage pulse that changes between a positive potential, a negative potential, and zero potential is known. The waveform shown in FIG. 5 is based on unipolar modulation. Any modulation method may be used for the motor drive device 2 according to the embodiment. In applications where it is necessary to control the motor current waveform to a sine wave, it is preferable to employ unipolar modulation having a lower harmonic content than bipolar modulation.

FIG. 6 is a diagram showing a change in the inverter output voltage according to the modulation factor. The upper part of FIG. 6 shows the voltage command _Vm , the carrier, and the inverter output voltage when the modulation factor is 1.0. The middle part of FIG. 6 shows the voltage command _Vm , the carrier, and the inverter output voltage when the modulation factor is 1.2. The lower part of FIG. 6 shows the voltage command _Vm , the carrier, and the inverter output voltage when the modulation factor is 2.0.

As described in FIG. 4, the voltage command V _m1 is compared with the carrier in the comparing unit 38g, and the voltage command V _m2 is compared with the carrier in the comparing unit 38h. When the voltage command V _m1,2 is larger than the carrier, the switching element of the single-phase inverter 11 is turned on, and when the voltage command V _m1,2 is smaller than the carrier, the switching element of the single-phase inverter 11 is turned off. . Therefore, as shown in FIG. 4, the inverter output voltage subjected to the PWM control is applied to the single-phase motor 12.

  Although there are various definitions of the modulation rate, here, the ratio between the voltage amplitude command V * and the amplitude of the triangular wave carrier, that is, “voltage amplitude command V * / triangle wave carrier amplitude” is defined as the modulation rate. The upper part of FIG. 6 shows a waveform when the modulation rate is 1.0, but the same waveform is obtained when the modulation rate is less than 1.0. When the modulation rate is less than 1.0, the inverter output voltage is generated according to the frequency of the triangular wave carrier, and therefore, the inverter output voltage also outputs a voltage pulse corresponding to the carrier frequency.

On the other hand, when the modulation factor exceeds 1.0, the waveform is as shown in the middle part and the lower part of FIG. When the modulation rate exceeds 1.0, it is called “overmodulation”, and the area where the modulation rate exceeds 1.0 is called “overmodulation area”. In the overmodulation region, the voltage command V _m exceeds the amplitude of the carrier, sections that can not be generated inverter drive signal is generated in accordance with the carrier frequency. In this section, since the inverter output voltage is fixed to the positive power supply voltage or the negative power supply voltage, a larger output voltage can be obtained for the inverter output voltage than when the modulation factor is 1.

  Here, a problem in the case where a battery is used for the power supply 10 shown in FIG. 1 will be described. A battery has an internal impedance as a structure, and the battery output voltage greatly changes according to the current output from the battery. Specifically, it is known that when a current of 20 [A] flows in a battery of 20 [V], the battery output voltage drops to about 19.5 [V]. In the case where the above-mentioned modulation rate is 1 or more, the output voltage pulse is reduced, so that there is a problem that the output voltage cannot be accurately obtained with respect to the voltage command. Further, it is known that the battery current becomes a pulsating current due to the influence of the switching by the inverter, so that the voltage output from the battery also pulsates. In order to solve these problems, it is possible to suppress both the variation in the voltage supplied from the battery to the inverter and the variation in the voltage output from the inverter by successively changing the advance angle without making it constant.

Next, the advance angle control in the present embodiment will be described. FIG. 7 is a diagram illustrating a functional configuration for calculating an advance phase input to the carrier generation unit and the carrier comparison unit illustrated in FIGS. 3 and 4. The function for calculating the advance angle phase θ _v can be realized by the rotation speed calculation section 42 and the advance angle phase calculation section 44 as shown in FIG.

Rotation speed calculation unit 42, the position sensor signal 21a calculates the rotation speed ω of the single-phase motor 12 based on the further is the angle rotor mechanical angle theta _m the reference phase in terms of electrical angle from the reference position of the rotor 12a to calculate the θ _e. Advanced angle phase calculator 44, the rotation speed calculation unit 42 based on the calculated information of the rotation speed ω and the reference phase theta _e, calculates the advance phase theta _v.

FIG. 8 is a diagram illustrating an example of a method of calculating the advance angle phase. The horizontal axis in FIG. 8 is the rotation speed, and the vertical axis in FIG. 8 is the advance phase. The advance phase θ _v can be determined using a function in which the advance phase θ _v increases with an increase in the rotation speed N, as shown in FIG. In the example of FIG. 8, the advance angle phase θ _v is determined by a first-order linear function. However, the present invention is not limited to this. If the advance angle phase θ _v increases as the rotational speed increases, 1 A function other than the following linear function may be used. The advance angle adjustment width Δθdel indicates a range of variation in the mounting position of the position sensor 21.

  FIG. 9 is a first diagram illustrating a positional relationship among a position sensor, a stator, and a rotor. FIG. 10 is a second diagram illustrating a positional relationship among the position sensor, the stator, and the rotor.

  9 and 10 show a plurality of teeth 12b1, a rotor 12a arranged at the center of the plurality of teeth 12b1, and a position sensor 21. The center line CL is a line passing through the center between two teeth 12b1 adjacent to each other in the rotation direction D1 of the rotor 12a and the center axis AX of the rotor 12a.

  The position sensor 21 shown in FIG. 9 is arranged between the two teeth 12b1, and the center of the position sensor 21 in the rotation direction D1 coincides with the center line CL. The position sensor 21 shown in FIG. 10 is disposed between the two teeth 12b1, and the center of the position sensor 21 in the rotation direction D1 is shifted from the center line CL.

  In general, in motor control, phase calculation is performed on the assumption that the position sensor 21 is arranged such that the center between the adjacent teeth 12b1 and the center of the position sensor 21 match. However, in the actual assembling process of the single-phase motor 12, the position sensor 21 is fixed at a position where the center of the position sensor 21 is shifted from the center between the adjacent teeth 12b1. Therefore, a phase shift, that is, a phase difference occurs between the position sensor signal 21a and the induced voltage due to the position shift amount. If it is known in advance that the center of the position sensor 21 is shifted from the center between the adjacent teeth 12b1, the phase is calculated in consideration of the shift amount.

  FIG. 11 is a diagram showing the position sensor signal and the motor induced voltage. As shown in FIG. 9, the waveform of the motor induced voltage indicated by the dotted line indicates that the position sensor 21 is arranged so that the center between the adjacent teeth 12 b 1 and the center of the position sensor 21 coincide with each other. This is a waveform in the case where there is no occurrence. As shown in FIG. 10, the waveform of the motor induced voltage indicated by the solid line indicates that the position sensor 21 is disposed so that the center between the adjacent teeth 12b1 and the center of the position sensor 21 are displaced. FIG. In FIG. 11, a case where no position shift has occurred is described as “no position shift”, and a case where a position shift has occurred is described as “position shift”.

  When no displacement occurs, the edge of the position sensor signal 21a matches the zero cross point of the motor induced voltage, and the phase difference becomes zero. Note that an edge of the position sensor signal 21a indicates a rising point of the signal or a falling point of the signal. However, when the position sensor 21 is displaced, the edge of the position sensor signal 21a does not coincide with the zero cross point of the motor induced voltage, and a phase difference occurs between the position sensor signal and the motor induced voltage.

  The degree of the influence on the motor control due to the displacement of the position sensor 21 varies depending on the rotor diameter of the rotor 12a provided in the single-phase motor 12 to be controlled and the number of magnetic poles of the rotor 12a. I do. The smaller the rotor diameter, the greater the change in phase according to the distance of the arc on the outer peripheral surface of the rotor. Further, as the number of magnetic poles increases, the rate of change in the electrical angle during one rotation of the rotor 12a increases. Therefore, the phase difference increases as the rotor diameter decreases and the number of magnetic poles of the magnet increases.

  From the above, the phase difference between the induced voltage and the position sensor signal 21a is generated due to the position shift, so that even if an advance angle phase with respect to the rotation speed set in motor control is given, the phase difference is different from the actual phase. Voltage is applied to the motor. When voltages are applied in different phases, there arises a problem that the rotation speed does not increase to a specific value.

  Here, when the rotation speed does not reach a specific rotation speed, the phase shift of the position sensor 21 is changed by using a method of changing the phase within a range of the advance adjustment width Δθdel and fixing the phase at a predetermined rotation speed. Even when this occurs, a specific rotational speed can be obtained.

  In addition, by using a plurality of motors in advance and grasping the phase difference between the induced voltage of each of the plurality of motors and the position sensor signal 21a, the maximum value of the position shift of the position sensor 21 is set to the advance adjustment width Δθdel. can do. If the maximum value of the advance angle adjustment width Δθdel is not set in this way, the motor control attempts to increase the rotation speed beyond the maximum value of the advance angle adjustment width Δθdel, and thus the control may fail. Therefore, stable control can be realized by determining the advance adjustment width Δθdel in advance.

  FIG. 12 is a flowchart illustrating an operation for determining the advance phase. The control unit 25 calculates the number of rotations determined according to the time between edges of the position sensor 21 (S1). The control unit 25 compares the calculated rotation speed with the set target rotation speed (S2).

  When the current rotational speed is equal to or less than the target rotational speed (Step S2, Yes), the controller 25 updates the phase adjustment angle θadj by adding a predetermined phase adjustment angle Δθadj to the phase adjustment angle θadj ( S3).

  When the current rotation speed exceeds the target rotation speed (step S2, No), the control unit 25 updates the phase adjustment angle θadj by subtracting the phase adjustment angle Δθadj from the phase adjustment angle θadj (S4).

Next, the controller 25 compares the phase adjustment angle θadj with the advance angle adjustment width Δθdel (S5). If the phase adjustment angle θadj is equal to or larger than the advance angle adjustment width Δθdel (step S5, Yes), the control unit 25 sets the phase adjustment angle θadj to an advance angle so that the phase adjustment angle θadj does not become larger than the advance angle adjustment width Δθdel. fixed (S6) the adjusted width Derutashitadel, adds the phase adjustment angle θadj obtained in S3 or S4 in the advance phase θ _v (S7). Thus, the control unit 25 derives the final voltage command _{V m} (S8).

If the phase adjustment angle θadj is less than the advance adjustment range Derutashitadel (step S5, No), the control unit 25 executes the processing of S7, the deriving the final voltage command _{V m} (S8).

By setting these advance angle phases θ _v , when the current rotation speed is lower than the target rotation speed, the rotation speed is increased by advancing the phase, and when the current rotation speed is higher than the target rotation speed, By delaying the phase, the number of rotations can be reduced.

In the present embodiment, an example in which control is performed so that the rotation speed follows the target rotation speed has been described. However, the control example of the present embodiment is limited to control in which the rotation speed follows the target rotation speed. Sarezu may be a control for adjusting the advance phase theta _v so as to obtain the maximum speed.

  In a three-phase sensorless DC brushless motor, the position of the rotor cannot be directly detected, and a method of estimating the rotor rotational position from the motor current is generally used. In this case, since an error occurs between the estimated rotor rotation position and the actual rotor rotation position, it is known to adjust the advance angle by motor control.

  On the other hand, in the DC brushless motor with a position sensor, the position sensor signal can be directly read, so that the position between the magnetic poles can be grasped. Since it is easy to grasp the rotor rotation position, in the DC brushless motor with the position sensor, there are few cases where the correction caused by the variation in the mounting position of the position sensor 21 is performed by the motor control. By performing the lead angle correction with the DC brushless motor with the position sensor, the influence on the motor control due to the variation in the mounting position of the position sensor 21 is suppressed, and highly accurate control according to the rotor position can be realized. .

  Since the power that can be output from the battery is large when the battery voltage is close to full charge, if the inverter tries to extract power from the battery in a state close to the maximum, the modulation rate exceeds 1 and the output voltage error Becomes larger. In this state, the control unit 25 reduces the phase adjustment angle θadj because the influence of the rotational speed variation due to the advance phase becomes large. When the battery voltage decreases, the maximum output power decreases due to the limitation of the discharge current. Therefore, when the battery voltage decreases, the control unit 25 can increase the phase adjustment angle θadj, thereby shortening the time required to reach the maximum rotation speed.

FIG. 13 is a diagram illustrating a relationship among a position sensor signal, a rotor mechanical angle, a reference phase, and a voltage command. The lowermost portion of FIG. 13, the rotor mechanical angle theta _m when the rotor 12a is rotated in the clockwise direction is 0 °, 45 °, 90 ° , the single-phase motor 12 is 135 ° and 180 ° are shown. The rotor 12a of the single-phase motor 12 is provided with four magnets. Four teeth 12b1 are provided on the outer periphery of the rotor 12a. If the rotor 12a is rotated clockwise, the control unit 25, the position sensor signal 21a corresponding to the rotor mechanical angle theta _m is detected, it converted to an electrical angle in accordance with the detected position sensor signals 21a the reference phase theta _e is calculated.

Voltage command V _m shown as "Example 1" in the middle portion of FIG. 13 is a voltage command in a case of advanced angle phase θ _{v =} 0. When the advance phase θ _v = 0, a sinusoidal voltage command V _{m in} phase with the reference phase θ _e is output. The amplitude of the voltage command V _m at this time is determined based on the voltage amplitude command V * as described above.

Voltage command V _m shown as "Example 2" in the middle portion of FIG. 13 is a voltage command in a case of advanced angle phase θ _{v =} π / 4. For advanced angle phase θ _{v =} π / 4, components of the reference phase theta _e from advanced angle phase theta _v, i.e. [pi / 4 sinusoidal voltage command V _m has proceeded is output.

Next, how to give the voltage amplitude command V * will be described. FIG. 14 is a diagram showing a time change of the voltage amplitude command. In the present embodiment, an operation mode in which the voltage amplitude command V * changes stepwise according to the time t as shown in the figure. Specifically, first, the first voltage V ₁ of the constant set in advance is given at startup, during steady operation after acceleration, the second voltage V ₂ of greater constant than the first voltages V ₁ Ataeraru . Further, at the time of acceleration is changed from the first voltages V ₁ to the second voltage V _2, to raise the voltage amplitude command V * as acceleration rate set in advance is obtained. That is, in the present embodiment, the control is performed so that the voltage amplitude command V * is constant at the time of startup and during steady operation. At the time of startup, the time τ1 for applying the first voltage V1 can be set to an arbitrary time in consideration of the stabilization time of the control system.

  Next, the effect of the constant voltage amplitude command V * will be described. The following effects can be obtained by controlling the voltage amplitude command V * to be constant during the steady operation.

(1) Even when the load suddenly changes, a constant voltage command can be output based on the phase detected from the position sensor signal 21a.
(2) Even if the rotational speed changes, the voltage command is not affected, so that the output voltage can be kept stable.

  The above-described effect is effective in an application such as a vacuum cleaner in which the load varies according to the contact area between the suction port of the vacuum cleaner and the floor surface.

  In the constant rotation speed control performed by a general electric blower, an overcurrent may flow through the motor. The reason why the overcurrent flows is that the current fluctuates rapidly in order to keep the rotation speed constant during a load change. To explain in more detail, when performing the rotation speed constant control when transitioning from the “light load” state, that is, the “load torque small state”, to the “heavy load state”, that is, the “high load torque state”, This is because the motor output torque must be increased while maintaining the same rotational speed, and the amount of change in the motor current increases.

  On the other hand, in the control according to the present embodiment, as described above, control is performed to keep the voltage amplitude command V * constant during steady operation. Here, when the voltage amplitude command V * is constant, when the load becomes heavy, the voltage amplitude command V * is not changed, so that the motor torque is reduced by an increase in the load torque. With this control, a steep change in motor current and overcurrent can be prevented, so that an electric blower and an electric vacuum cleaner that rotate stably can be realized.

  In the case of an electric blower, the load torque increases with an increase in the number of rotations of the blade, which is a load of the motor, and also increases with an increase in the diameter of the air path. The diameter of the air path represents the size of the suction port when an electric vacuum cleaner is used as an example. When the diameter of the air passage is large, when nothing is in contact with the suction port, a force for sucking the wind is required, so that the load torque when the blade is rotating at the same rotation speed increases. On the other hand, when the diameter of the wind path is small, the suction torque is not required when the blades are rotating at the same rotation speed because the suction force is not required when the suction port is in contact with something and is closed. Become.

Next, the effect of the advance angle control will be described. First, if the advance angle phase θ _v is increased in accordance with an increase in the number of revolutions, the range of the number of revolutions can be expanded. The advance phase theta _v when 0, the rotational speed is saturated at which the voltage applied to the motor and the motor induced voltage is balanced. In order to further increase the rotation speed, the advance angle θ _v is advanced and the magnetic flux generated in the stator due to the armature reaction is weakened to suppress the motor induced voltage and increase the rotation speed. Thus, by selecting the advanced angle phase theta _v according to the rotation speed, it is possible to obtain a wide speed range.

  Next, the effect of providing the advance angle adjustment width Δθdel in the advance angle control will be described. First, by providing the advance adjustment width Δθdel, a specific number of rotations can be stably obtained even when the position sensor 21 is displaced during manufacturing. In addition, a specific rotation speed can be obtained even in the case where a characteristic deviation of the sensitivity unique to the position sensor occurs. Therefore, it is possible to suppress the generation of the cost for adjusting the mounting position of the position sensor 21 to eliminate the variation in the manufacturing process.

When the advance angle control according to the present embodiment is applied to a vacuum cleaner, the voltage command is kept constant regardless of the change in the closed state of the suction port, that is, regardless of the load torque, and the voltage command is adjusted according to the increase in the rotation speed. it suffices to increase the advance phase theta _v is a lead angle of the voltage command Te. With such control, stable driving can be performed in a wide rotation speed range. Further, by providing the advance angle adjustment width, it is possible to suppress the influence on the drive rotation speed even when the position sensor 21 is displaced.

  Next, a loss reducing method according to the present embodiment will be described. FIG. 15 is a first diagram illustrating a path of a motor current according to the polarity of the inverter output voltage. FIG. 16 is a second diagram showing the path of the motor current depending on the polarity of the inverter output voltage. FIG. 17 is a third diagram showing the path of the motor current depending on the polarity of the inverter output voltage.

  When the polarity of the inverter output voltage is positive, the current flows into the single-phase motor 12 through the channel of the switching element 51, which is the upper arm of the first phase, as shown by the thick solid line (a) in FIG. It flows out of the single-phase motor 12 through the channel of the switching element 54 which is the lower arm of the two-phase. When the polarity of the inverter output voltage is negative, the current flows into the single-phase motor 12 through the channel of the switching element 53, which is the upper arm of the second phase, as shown by the thick broken line (b) in FIG. , From the single-phase motor 12 through the channel of the switching element 52 which is the lower arm of the first phase.

  Next, a current path when the inverter output voltage is zero, that is, when the single-phase inverter 11 outputs a zero voltage will be described. When the inverter output voltage becomes zero after the positive inverter output voltage is generated, no current flows from the power supply side as shown by the thick solid line (c) in FIG. 16, and the single-phase inverter 11 and the single-phase motor 12 A reflux mode in which current flows back and forth between. At this time, since the direction of the current flowing immediately before to the single-phase motor 12 does not change, the current flowing out of the single-phase motor 12 is applied to the channel of the switching element 54 which is the lower arm of the second phase and the first phase. The light returns to the single-phase motor 12 through the body diode 52a of the switching element 52 serving as the lower arm. Note that when the inverter output voltage becomes zero after the negative inverter output voltage is generated, the direction of the current flowing immediately before is opposite, and therefore, as shown by the thick broken line (d) in FIG. The direction of the current is reversed. More specifically, the current flowing from the single-phase motor 12 passes through the body diode 51a of the switching element 51, which is the upper arm of the first phase, and the channel of the switching element 53, which is the upper arm of the second phase. Return to the single-phase motor 12.

  As described above, in the reflux mode in which the current flows between the single-phase motor 12 and the single-phase inverter 11, the current flows through the body diode in one of the first phase and the second phase. In general, it is known that conduction loss is smaller when a current flows through a channel of a MOSFET than when a current flows in a forward direction of a diode. Therefore, in the present embodiment, in the return mode in which the return current flows, the MOSFET on the side having the body diode is controlled to be turned on in order to reduce the conduction current flowing in the body diode.

  In the reflux mode, the switching element 52 is controlled to be turned on at the timing when the reflux current shown by the thick solid line (c) in FIG. 16 flows. With this control, most of the return current flows on the channel side of the switching element 52 having a small resistance value, as shown by the thick solid line (e) in FIG. Thereby, the semiconductor loss in the switching element 52 is reduced. At the timing when the return current shown by the thick broken line (d) in FIG. 16 flows, the switching element 51 is controlled to be turned on. With this control, most of the return current flows on the channel side of the switching element 51 having a small resistance value, as shown by the thick broken line (f) in FIG. Thus, semiconductor loss in the switching element 51 is reduced.

  As described above, at the timing when the return current flows through the body diode, the MOSFET having the body diode is controlled to be turned on, so that the loss of the switching element can be reduced. For this reason, the shape of the MOSFET is made to be a surface mount type so that heat can be dissipated on the substrate, and if necessary, part or all of the switching element is formed of a wide band gap semiconductor, so that the heat generated by the MOSFET is generated only on the substrate. To realize a structure for suppressing noise. If heat radiation is possible only with the substrate, no heat sink is required, which contributes to downsizing of the inverter and downsizing of the product.

  In addition to the above-described heat dissipation method, a further heat dissipation effect can be obtained by installing the substrate in the air passage. Here, the air path is a portion that generates an air flow like an electric blower or a passage of an air flow generated by the electric blower. By arranging the substrate in the air passage, heat generated by the electric blower can radiate heat from the semiconductor element on the substrate, so that heat generation of the semiconductor element can be significantly suppressed.

  Next, an application example of the motor drive device according to the embodiment will be described. FIG. 18 is a configuration diagram of a vacuum cleaner including the motor driving device according to the embodiment. The vacuum cleaner 61 includes a battery 67 that is a DC power supply, the motor driving device 2 illustrated in FIG. 1, an electric blower 64 driven by the single-phase motor 12 illustrated in FIG. 1, a dust collection chamber 65, 68, a suction port body 63, an extension tube 62, and an operation unit 66. Battery 67 corresponds to power supply 10 shown in FIG.

  A user who uses the vacuum cleaner 61 has the operation unit 66 and operates the vacuum cleaner 61. The motor drive device 2 of the vacuum cleaner 61 drives the electric blower 64 using the battery 67 as a power supply. When the electric blower 64 is driven, dust is sucked from the suction port 63, and the sucked dust is collected in the dust collecting chamber 65 via the extension pipe 62.

The vacuum cleaner 61 is a product whose motor rotation speed varies from 0 [rpm] to 100,000 [rpm]. As described above, when the single-phase motor 12 drives a product that rotates at a high speed, the control method according to the above-described embodiment is preferable. And constant voltage amplitude command V *, by changing the advanced angle phase theta _v in accordance with the rotational speed, it is possible while expanding the rotational speed drive range from a low speed to a high speed rotation region, corresponding to the sudden load change. In addition, since the motor current is controlled to a sine wave by the PWM control, high-efficiency driving can be performed, so that a longer operation time can be expected.

  Further, in a product such as the vacuum cleaner 61 on which a small motor is mounted, the phase difference due to the variation in the mounting position of the position sensor 21 has a large influence, which greatly affects the control. Therefore, generally, the amount of displacement of the position sensor 21 is measured in advance in manufacturing, and the vacuum cleaner 61 performs control in consideration of the amount of displacement of the position sensor 21. However, in this case, since a step of measuring the amount of displacement of the position sensor 21 occurs in the manufacturing process, there is a problem that the manufacturing cost increases. Therefore, by controlling the motor that suppresses the influence of the displacement of the position sensor without measuring the displacement amount of the position sensor 21, it is possible to improve the product quality at low cost.

  Further, the vacuum cleaner 61 according to the embodiment can be reduced in size and weight by reducing the above-described heat radiating components. Further, the vacuum cleaner 61 does not need a current sensor for detecting a current and does not need a high-speed analog-to-digital converter, so that the cost can be reduced.

  FIG. 19 is a configuration diagram of a hand dryer including the motor drive device according to the embodiment. The hand dryer 90 includes the motor driving device 2, the casing 91, the hand detection sensor 92, the water receiving portion 93, the drain container 94, the cover 96, the sensor 97, the intake port 98, and the electric blower 95. Prepare. Here, the sensor 97 is either a gyro sensor or a human sensor. In the hand dryer 90, when a hand is inserted into the hand insertion portion 99 above the water receiving portion 93, water is blown off by the air blower 95, and the blown water is collected in the water receiving portion 93. After that, it is stored in the drain container 94.

  The hand dryer 90 is a product whose motor rotation speed varies from 0 [rpm] to 100,000 [rpm], like the vacuum cleaner 61 shown in FIG. Therefore, also in the hand dryer 90, the control method according to the above-described embodiment is suitable, and the same effect as that of the electric vacuum cleaner 61 can be obtained.

  FIG. 20 is a diagram for explaining modulation control in the motor drive device according to the embodiment. The relationship between the number of rotations and the modulation rate is shown on the left side of FIG. On the right side of the figure, a waveform of the inverter output voltage when the modulation factor is 1 or less and a waveform of the inverter output voltage when the modulation factor exceeds 1 are shown. Generally, the load torque of the rotating body increases as the number of rotations increases. For this reason, it is necessary to increase the motor output torque as the number of rotations increases. In general, the motor output torque increases in proportion to the motor current, and an increase in the motor current requires an increase in the inverter output voltage. Therefore, by increasing the modulation rate and increasing the inverter output voltage, the number of rotations can be increased.

Next, rotation speed control in the present embodiment will be described. In the following description, an electric blower is assumed as a load, and the operating range of the electric blower is divided as follows.
(A) Low-speed rotation range (low-speed range): 0 [rpm] to 100,000 [rpm]
(B) High-speed rotation region (high rotation speed region): 100,000 [rpm] or more

  The region between (A) and (B) is a gray zone, and may be included in the low-speed rotation region or the high-speed rotation region depending on the application.

  First, control in the low-speed rotation range will be described. In the low-speed rotation range, the modulation rate is set to 1 or less and PWM control is performed. By setting the modulation factor to 1 or less, the motor current can be controlled to a sine wave, and the efficiency of the motor can be improved. When operating at the same carrier frequency in the low-speed rotation region and the high-speed rotation region, the carrier frequency becomes the carrier frequency adjusted to the high-speed rotation region, so that the PWM pulse tends to increase more than necessary in the low-speed rotation region. . For this reason, a method of reducing the carrier frequency in the low-speed rotation range to reduce the switching loss may be used. Further, by varying the carrier frequency in synchronization with the rotation speed, control may be performed so that the pulse number does not change in accordance with the rotation speed.

  Next, control in the high-speed rotation range will be described. In the high-speed rotation range, the modulation factor is set to a value larger than 1. By setting the modulation factor to be greater than 1, the number of switching operations performed by the switching elements in the inverter is reduced while increasing the inverter output voltage, thereby suppressing an increase in switching loss. Here, when the modulation factor exceeds 1, the motor output voltage increases, but the number of times of switching decreases, so that current distortion may be a concern. However, during high-speed rotation, the reactance component of the motor increases, and di / dt, which is the change component of the motor current, decreases. Therefore, the current distortion is smaller than in the low-speed rotation region, and the influence on the waveform distortion is small. Become. Therefore, in the high-speed rotation range, the modulation factor is set to a value larger than 1, and control is performed to reduce the number of switching pulses. With this control, an increase in switching loss can be suppressed, and higher efficiency can be achieved.

  As described above, the boundary between the low-speed rotation region and the high-speed rotation region is ambiguous. For this reason, the first rotation speed that determines the boundary between the low-speed rotation region and the high-speed rotation region is set in the control unit 25, and the control unit 25 determines that the rotation speed of the motor or the load is equal to or less than the first rotation speed. The modulation rate may be set to 1 or less, and if the rotation speed of the motor or the load exceeds the first rotation speed, control may be performed to set the modulation rate to more than 1.

  As described above, in the present embodiment, the configuration example in which the motor driving device 2 is applied to the vacuum cleaner 61 and the hand dryer 90 has been described. However, the motor driving device 2 is generally applied to electric equipment having a motor. can do. Electric equipment equipped with a motor includes incinerators, crushers, dryers, dust collectors, printing machines, cleaning machines, confectionery machines, tea making machines, woodworking machines, plastic extruders, cardboard machines, packaging machines, hot air generators, objects It is a device equipped with an electric blower, such as a device for transportation, dust suction, general ventilation, or OA equipment.

  The configurations described in the above embodiments are merely examples of the contents of the present invention, and can be combined with other known technologies, and can be combined with other known technologies without departing from the gist of the present invention. Parts can be omitted or changed.

  Reference Signs List 1 motor drive system, 2 motor drive device, 10 power supply, 11 single-phase inverter, 12 single-phase motor, 12a rotor, 12b stator, 12b1 teeth, 20 voltage sensor, 21 position sensor, 21a position sensor signal, 25 control unit, 30 Analog-to-digital converter, 31 processor, 32 drive signal generator, 33 carrier generator, 34 memory, 38 carrier comparator, 38a absolute value calculator, 38b divider, 38c, 38d, 38f multiplier, 38e adder, 38g , 38h comparison unit, 38i, 38j output inversion unit, 42 rotation speed calculation unit, 44 advance angle phase calculation unit, 51, 52, 53, 54 switching element, 51a, 52a, 53a, 54a body diode, 61 vacuum cleaner, 62 Extension tube, 63 Suction port, 64 Electric feed Blower, 65 dust collection chamber, 66 operation unit, 67 battery, 68 sensor, 90 hand dryer, 91 casing, 92 hand detection sensor, 93 water receiver, 94 drain container, 95 electric blower, 96 cover, 97 sensor, 98 intake Mouth, 99 hand insertion part.

Claims (13)

A motor drive device for driving a single-phase motor using a battery as a power supply, comprising a plurality of switching elements, a single-phase inverter for applying an AC voltage to the single-phase motor, and a single-phase inverter attached to the single-phase motor, A position sensor that detects a rotor rotation position of the motor and outputs a rotation position detection signal, and a control unit that performs pulse width modulation control on the plurality of switching elements based on a voltage command and the rotation position detection signal,
Wherein the control unit, the voltage command of the advance angle a is advanced angle phase relative to the motor induced voltage, and against the specific rotational speed, the variation range of the mounting position of the position sensor to the single-phase motor A motor drive device that changes within the corresponding advance adjustment width.   In the control unit, a phase adjustment angle for adjusting a phase difference between the rotation position detection signal and the output voltage of the single-phase inverter is set, and the phase adjustment angle and the advance angle adjustment width are set to the voltage. The motor drive device according to claim 1, which is used for calculating a command. The plurality of switching elements are metal oxide semiconductor field effect transistors,
2. The motor drive device according to claim 1, wherein the control unit turns on the metal oxide semiconductor field effect transistor in which the return current flows through the body diode at a timing when the return current flows through the single-phase inverter. 3.   4. The motor drive device according to claim 1, wherein the control unit generates a pulse width modulation signal for performing the pulse width modulation control by unipolar modulation. 5.   2. A method according to claim 1, wherein a predetermined first voltage is applied to a voltage amplitude command that is an amplitude of the voltage command at a start time, and a constant second voltage larger than the first voltage is applied during a steady operation after acceleration. The motor drive device according to any one of claims 1 to 4. The control unit sets a first rotation speed that determines a boundary between a low-speed rotation region and a high-speed rotation region,
When the rotation speed of the single-phase motor is equal to or less than the first rotation speed, the modulation rate of the pulse width modulation control is set to 1 or less, and when the rotation speed exceeds the first rotation speed, The motor drive device according to any one of claims 1 to 4, wherein the modulation factor is set to a value exceeding 1.   The motor drive device according to claim 1, wherein at least one of the plurality of switching elements is formed of a wide band gap semiconductor.   The motor driving device according to claim 7, wherein the wide band gap semiconductor is silicon carbide, gallium nitride, or diamond.   An electric blower comprising the motor drive device according to claim 1.   An electric vacuum cleaner comprising the electric blower according to claim 9.   The vacuum cleaner according to claim 10, wherein the board on which the plurality of switching elements are mounted is installed in a passage of an airflow generated by the electric blower.   A hand dryer comprising the electric blower according to claim 9.   The hand dryer according to claim 12, wherein a substrate on which the plurality of switching elements are mounted is installed in a passage of an airflow generated by the electric blower.

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