WO2019180967A1 - Motor drive device, electric vacuum cleaner, and hand dryer - Google Patents

Abstract

This motor drive device (2) is provided with an inverter (11) for outputting AC voltage to a single-phase motor (12) and a control unit (25) for controlling the AC voltage output by the inverter (11). The control unit (25) causes a square wave voltage to be output to the inverter (11) both when the single-phase motor (12) is starting up and when in a low-speed rotation range. The control unit (25) causes a sine wave voltage to be output to the inverter when the single-phase motor (12) is in a medium-speed rotation range. The control unit (25) causes a trapezoidal wave voltage to be output to the inverter (11) when the single-phase motor (12) is in a high-speed range.

Description

Motor drive device, vacuum cleaner and hand dryer

The present invention relates to a motor driving device for driving a single-phase motor, and a vacuum cleaner and a hand dryer provided with the motor driving device.

There is the following Patent Document 1 as a related art relating to a motor driving device for driving a single-phase motor. Patent Document 1 discloses a technique for applying a rectangular wave voltage of a plurality of pulses synchronized with a detection signal of a Hall sensor and flowing a rectangular wave current synchronized with a motor induced voltage during an acceleration mode period in which the motor is accelerated. Has been.

Japanese Patent Laying-Open No. 2015-2673

In addition to the fundamental wave component current required for rotational driving, the high frequency component current is superimposed on the rectangular wave current. In the technique of Patent Document 1, since a rectangular wave current flows when the motor is accelerated, there is a problem that vibration and noise are deteriorated by a high-frequency component current superimposed on a fundamental wave component current.

The present invention has been made in view of the above, and an object of the present invention is to obtain a motor drive device capable of suppressing vibration and noise during motor acceleration.

In order to solve the above-described problems and achieve the object, a motor drive device according to the present invention includes an inverter that outputs an AC voltage to a single-phase motor, and a control unit that controls the AC voltage output by the inverter. . The control unit causes the inverter to output a rectangular wave voltage at the start of the single-phase motor and in the low-speed rotation range, and causes the inverter to output a sine wave voltage in the medium-speed rotation range of the single-phase motor. In the high-speed rotation range, the inverter outputs a trapezoidal wave voltage.

According to the motor drive device of the present invention, there is an effect that vibration and noise can be suppressed during acceleration of the motor.

Configuration diagram of a motor drive system including a motor drive device according to an embodiment Circuit diagram of the inverter shown in FIG. 1 is a block diagram showing a functional part that generates a pulse width modulation (PWM) signal among the functional parts of the control unit shown in FIG. The block diagram which shows an example of the carrier comparison part shown by FIG. FIG. 4 is a time chart showing an example of the waveform of the main part in the carrier comparison unit shown in FIG. The block diagram which shows the other example of the carrier comparison part shown by FIG. FIG. 6 is a time chart showing a waveform example of a main part in the carrier comparison unit shown in FIG. The block diagram which shows the function structure for calculating the advance angle phase input into the carrier comparison part shown by FIG.4 and FIG.6 The figure which shows an example of the calculation method of the advance angle phase in embodiment Time chart used for explaining the relationship between the voltage command and the advance phase shown in FIGS. The figure which shows an example of the waveform of the carrier in the rotation area | region from the starting at the time of rotating the single phase motor shown by FIG. 1 to a low speed rotation area, and an inverter output voltage. The figure which shows an example of the waveform of the carrier in the middle speed rotation area at the time of rotating the single phase motor shown by FIG. 1, a voltage command, and an inverter output voltage The figure which shows an example of the waveform of the carrier in the high speed rotation area at the time of rotating the single phase motor shown by FIG. 1, a voltage command, and an inverter output voltage The figure which shows the equivalent circuit of the single phase motor shown by FIG. The figure used for description of the phase difference between the counter electromotive voltage and the inverter output voltage in the rectangular wave PWM, the sine wave PWM, and the trapezoidal wave PWM in the embodiment. Time chart used for explanation of operation at the time of switching from rectangular wave PWM to sine wave PWM in the embodiment The figure which shows the relationship between the brake torque which generate | occur | produces in the single phase motor shown by FIG. 1, and rotational speed. The figure which shows the time change of the brake current which flows into the single phase motor shown by FIG. The figure which shows the relationship between the amplitude of the voltage command before voltage limitation and the fundamental wave component of the inverter output voltage The block diagram of the vacuum cleaner provided with the motor drive device which concerns on embodiment Configuration diagram of a hand dryer provided with a motor drive device according to an embodiment

Hereinafter, a motor drive device, a vacuum cleaner, and a hand dryer according to an embodiment of the present invention will be described in detail with reference to the accompanying drawings. The present invention is not limited to the following embodiments. In the following description, electric connection and physical connection are not distinguished from each other and simply referred to as “connection”.

Embodiment.
FIG. 1 is a configuration diagram of a motor drive system 1 including a motor drive device 2 according to an embodiment. A motor drive system 1 shown in FIG. 1 includes a single-phase motor 12, a motor drive device 2, a battery 10, a voltage sensor 20, and a position sensor 21.

The motor drive device 2 drives the single-phase motor 12 by supplying AC power to the single-phase motor 12. The battery 10 is a DC power source that supplies DC power to the motor driving device 2. The voltage sensor 20 detects a DC voltage V _dc output from the battery 10 to the motor driving device 2. The position sensor 21 detects a rotor rotational position that is a rotational position of the rotor 12 a built in the single-phase motor 12.

The single phase motor 12 is used as a rotating electric machine that rotates an electric blower (not shown). The single-phase motor 12 and the electric blower are mounted on devices such as a vacuum cleaner and a hand dryer.

In the present embodiment, voltage sensor 20 detects DC voltage V _dc , but the detection target of voltage sensor 20 is not limited to DC voltage V _dc output from battery 10. The detection target of the voltage sensor 20 may be an inverter output voltage that is an output voltage of the motor drive device 2. “Inverter output voltage” has the same meaning as “motor applied voltage” described later.

The motor drive device 2 includes an inverter 11, a control unit 25, and a drive signal generation unit 32. The inverter 11 is connected to the single phase motor 12 and outputs an AC voltage to the single phase motor 12. The control unit 25 controls the AC voltage output from the inverter 11. The inverter 11 is assumed to be a single-phase inverter, but any inverter that can drive a single-phase motor may be used.

The control unit 25 receives a DC voltage V _dc detected by the voltage sensor 20, a position sensor signal 21 a that is a rotational position detection signal output from the position sensor 21, and a voltage amplitude command V *. * The voltage amplitude command V is the amplitude of the voltage command V _m to be described later. The control unit 25 generates PWM signals Q1, Q2, Q3, and Q4 based on the DC voltage _Vdc , the position sensor signal 21a, and the voltage amplitude command V *.

The drive signal generation unit 32 generates a drive signal for driving the switching element of the inverter 11 based on the PWM signals Q1, Q2, Q3, Q4 output from the control unit 25. The position sensor signal 21a is a binary digital signal that changes according to the direction of the magnetic flux generated in the rotor 12a.

The control unit 25 includes a processor 31, a carrier generation unit 33, and a memory 34. The processor 31 generates the PWM signals Q1, Q2, Q3, and Q4 described above. The processor 31 performs arithmetic processing related to advance angle control in addition to arithmetic processing related to PWM control. The functions of a carrier comparison unit 38, a rotation speed calculation unit 42, and an advance angle phase calculation unit 44 described later are realized by the processor 31. The processor 31 may be called a CPU (Central Processing Unit), a microprocessor, a microcomputer, or a DSP (Digital Signal Processor).

The memory 34 stores a program read by the processor 31. The memory 34 is used as a work area when the processor 31 performs arithmetic processing. The memory 34 is generally a nonvolatile or volatile semiconductor memory such as a ROM (Read Only Memory), a RAM (Random Access Memory), a flash memory, an EPROM (Erasable Programmable ROM), or an EEPROM (registered trademark) (Electrically EPROM). is there. Details of the configuration of the carrier generation unit 33 will be described later.

The drive signal generation unit 32 converts the PWM signals Q1, Q2, Q3, and Q4 output from the processor 31 into drive signals S1, S2, S3, and S4 for driving the inverter 11, and outputs the drive signals to the inverter 11. .

An example of the single-phase motor 12 is a brushless motor. When the single-phase motor 12 is a brushless motor, a plurality of permanent magnets (not shown) are arranged in the circumferential direction on the rotor 12 a of the single-phase motor 12. The plurality of permanent magnets are arranged so that the magnetization direction is alternately reversed in the circumferential direction, and form a plurality of magnetic poles of the rotor 12a. A winding (not shown) is wound around the stator 12 b of the single-phase motor 12. An alternating current flows through the winding. The current flowing through the winding of the single-phase motor 12 is appropriately referred to as “motor current”. In the present embodiment, the number of magnetic poles of the rotor 12a is assumed to be four, but the number of magnetic poles of the rotor 12a may be other than four.

FIG. 2 is a circuit configuration diagram of the inverter 11 shown in FIG. The inverter 11 includes a plurality of switching elements 51, 52, 53, and 54 that are bridge-connected. The switching elements 51 and 52 constitute the first leg 5A. In the first leg 5A, the switching element 51 and the switching element 52 are connected in series. The switching elements 53 and 54 constitute the second leg 5B. In the second leg 5B, the switching element 53 and the switching element 54 are connected in series.

The switching elements 51 and 53 are located on the high potential side, and the switching elements 52 and 54 are located on the low potential side. In the inverter circuit, the high potential side is generally called “upper arm” and the low potential side is called “lower arm”. In the following description, the switching element 51 of the first leg 5A may be referred to as “upper arm first element”, and the switching element 53 of the second leg 5B may be referred to as “upper arm second element”. The switching element 52 of the first leg 5A may be referred to as a “lower arm first element”, and the switching element 54 of the second leg 5B may be referred to as a “lower arm second element”.

The connection point 6A between the switching element 51 and the switching element 52 and the connection point 6B between the switching element 53 and the switching element 54 constitute an AC terminal in the bridge circuit. A single-phase motor 12 is connected between the connection point 6A and the connection point 6B.

A MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor), which is a metal oxide semiconductor field effect transistor, is used for each of the plurality of switching elements 51, 52, 53, 54. The MOSFET is an example of an FET (Field-Effect Transistor).

In the switching element 51, a body diode 51a connected in parallel between the drain and source of the switching element 51 is formed. In the switching element 52, a body diode 52a connected in parallel between the drain and source of the switching element 52 is formed. In the switching element 53, a body diode 53a connected in parallel between the drain and source of the switching element 53 is formed. The switching element 54 is formed with a body diode 54 a connected in parallel between the drain and source of the switching element 54. Each of the plurality of body diodes 51a, 52a, 53a, 54a is a parasitic diode formed inside the MOSFET, and is used as a freewheeling diode.

The plurality of switching elements 51, 52, 53, and 54 are not limited to MOSFETs formed of silicon-based materials, but may be MOSFETs formed of wide band gap semiconductors such as silicon carbide, gallium nitride-based materials, or diamond.

Generally, wide band gap semiconductors have higher withstand voltage and heat resistance than silicon semiconductors. Therefore, by using a wide band gap semiconductor for the plurality of switching elements 51, 52, 53, and 54, the voltage resistance and allowable current density of the switching elements are increased, and the semiconductor module incorporating the switching elements can be downsized. In addition, wide bandgap semiconductors have high heat resistance, so it is possible to reduce the size of the heat dissipation part to dissipate the heat generated in the semiconductor module, and simplify the heat dissipation structure that dissipates the heat generated in the semiconductor module. Is possible.

FIG. 3 is a block diagram showing a functional part that generates a PWM signal among the functional parts of the control unit 25 shown in FIG.

In FIG. 3, the advance phase θ _v and the reference phase θ _e, which are advance angle controlled and used when generating a voltage command V _m described later, are input to the carrier comparison unit 38. The reference phase θ _e is a phase obtained by converting the rotor mechanical angle θ _m that is an angle from the reference position of the rotor 12a into an electrical angle. Here, “advance angle phase” represents “advance angle”, which is the “advance angle” of the voltage command, in terms of phase. The “advance angle” here is a phase difference between a motor applied voltage applied to the winding of the stator 12b and a motor induced voltage induced in the winding of the stator 12b. The “advance angle” takes a positive value when the motor applied voltage is ahead of the motor induced voltage.

In addition to the advance phase θ _v and the reference phase θ _e , the carrier comparison unit 38 includes a carrier generated by the carrier generation unit 33, a DC voltage V _dc, and a voltage that is an amplitude value of the voltage command V _m. An amplitude command V * is input. The carrier comparison unit 38 generates PWM signals Q1, Q2, Q3, and Q4 based on the carrier, the advance angle phase θ _v , the reference phase θ _e , the DC voltage V _dc, and the voltage amplitude command V *.

FIG. 4 is a block diagram showing an example of the carrier comparison unit 38 shown in FIG. FIG. 4 shows detailed configurations of the carrier comparison unit 38A and the carrier generation unit 33.

In FIG. 4, a carrier frequency f _C [Hz], which is a carrier frequency, is set in the carrier generation unit 33. As an example of the carrier waveform, a triangular wave carrier that goes up and down between “0” and “1” is shown at the tip of the arrow of the carrier frequency f _C. The PWM control of the inverter 11 includes synchronous PWM control and asynchronous PWM control. For synchronous PWM control, it is necessary to synchronize the carrier to advance the phase theta _v. On the other hand, when the asynchronous PWM control, it is not necessary to synchronize the carrier to advance the phase theta _v.

As shown in FIG. 4, the carrier comparison unit 38A includes an absolute value calculation unit 38a, a division unit 38b, a multiplication unit 38c, a multiplication unit 38d, a multiplication unit 38f, an addition unit 38e, a comparison unit 38g, a comparison unit 38h, and output inversion. Part 38i and output inverting part 38j.

The absolute value calculator 38a calculates the absolute value | V * | of the voltage amplitude command V *. In the dividing unit 38b, the absolute value | V * | is divided by the DC voltage V _dc detected by the voltage sensor 20. In the configuration of FIG. 4, the output of the division unit 38b is the modulation rate. The battery voltage that is the output voltage of the battery 10 varies as the current continues to flow. On the other hand, by dividing the absolute value | V * | by the DC voltage V _dc , the value of the modulation factor can be adjusted so that the motor applied voltage does not decrease due to a decrease in battery voltage.

In the multiplication unit 38c, a sine value of “θ _e + θ _v ” obtained by adding the advance phase θ _v to the reference phase θ _e is calculated. The calculated sine value of “θ _e + θ _v ” is multiplied by the modulation factor that is the output of the division unit 38b. In the multiplication unit 38d, the voltage command V _m that is the output of the multiplication unit 38c is multiplied by “½”. In the adder 38e, “½” is added to the output of the multiplier 38d. The multiplication unit 38f multiplies the output of the addition unit 38e by “−1”. The output of the adder 38e is input to the comparator 38g as a positive voltage command V _m1 for driving the two switching elements 51, 53 of the upper arm among the plurality of switching elements 51, 52, 53, 54. The output of the multiplication unit 38f is input to the comparison unit 38h as a negative voltage command V _m2 for driving the two switching elements 52 and 54 of the lower arm.

The comparison unit 38g compares the positive voltage command V _m1 with the carrier amplitude. The output of the output inverting unit 38i obtained by inverting the output of the comparing unit 38g becomes the PWM signal Q1 to the switching element 51, and the output of the comparing unit 38g becomes the PWM signal Q2 to the switching element 52. Similarly, the comparison unit 38h compares the negative side voltage command V _m2 with the carrier amplitude. The output of the output inverting unit 38j obtained by inverting the output of the comparing unit 38h becomes the PWM signal Q3 to the switching element 53, and the output of the comparing unit 38h becomes the PWM signal Q4 to the switching element 54. The switching element 51 and the switching element 52 are not simultaneously turned on by the output inverting part 38i, and the switching element 53 and the switching element 54 are not simultaneously turned on by the output inverting part 38j.

FIG. 5 is a time chart showing a waveform example of a main part in the carrier comparison unit 38A shown in FIG. FIG. 5 shows the waveform of the positive voltage command V _m1 output from the adder 38e, the waveform of the negative voltage command V _m2 output from the multiplier 38f, and the waveform of the PWM signals Q1, Q2, Q3, and Q4. And the waveform of the inverter output voltage.

PWM signal Q1 is "high (High)" when "low (Low)" next when the positive voltage command _{V m1} is greater than the carrier, the positive voltage command _{V m1} is smaller than the carrier. The PWM signal Q2 is an inverted signal of the PWM signal Q1. PWM signal Q3 is "high (High)" when "low (Low)" becomes when negative voltage instruction _{V m2} is larger than the carrier, the negative-side voltage instruction _{V m2} smaller than the carrier. The PWM signal Q4 is an inverted signal of the PWM signal Q3. As described above, the circuit shown in FIG. 4 is configured by “low active”, but may be configured by “high active” in which each signal has an opposite value. Good.

As shown in FIG. 5, the waveform of the inverter output voltage shows a voltage pulse due to the difference voltage between the PWM signal Q1 and the PWM signal Q4 and a voltage pulse due to the difference voltage between the PWM signal Q3 and the PWM signal Q2. These voltage pulses are applied to the single-phase motor 12 as a motor applied voltage.

Bipolar modulation and unipolar modulation are known as modulation schemes used when generating the PWM signals Q1, Q2, Q3, and Q4. Bipolar modulation is positive or for each cycle of the voltage command V _m is the modulation scheme for outputting a voltage pulse which varies in a negative potential. Unipolar modulation is a modulation system that outputs voltage pulses that change at three potentials for each period of the voltage command V _m , that is, voltage pulses that change between a positive potential, a negative potential, and a zero potential. The waveform shown in FIG. 5 is due to unipolar modulation. In the motor drive device 2 of the present embodiment, any modulation method may be used. In applications where it is necessary to control the motor current waveform to a sine wave, it is preferable to employ unipolar modulation with a lower harmonic content than bipolar modulation.

Moreover, the waveform shown in FIG. 5, the half period T / 2 of the interval of the voltage command _{V m,} a switching element 51, 52 constituting the first leg 5A, the switching elements 53 and 54 constituting the second leg 5B These four switching elements can be obtained by a switching operation. This method is called “both sides PWM” because the switching operation is performed by both the positive side voltage command V _m1 and the negative side voltage command V _m2 . In contrast, in the one half cycle of one cycle T of the voltage command V _m, to suspend the switching operation of the switching elements 51 and 52, the other half cycle of one cycle T of the voltage command V _m is There is also a method of stopping the switching operation of the switching elements 53 and 54. This method is called “one-side PWM”. Hereinafter, “one-side PWM” will be described.

FIG. 6 is a block diagram showing another example of the carrier comparison unit 38 shown in FIG. FIG. 6 shows an example of a PWM signal generation circuit based on the above-described “one-side PWM”, and specifically shows detailed configurations of the carrier comparison unit 38B and the carrier generation unit 33. The configuration of the carrier generating unit 33 shown in FIG. 6 is the same as or equivalent to that shown in FIG. Further, in the configuration of the carrier comparison unit 38B shown in FIG. 6, the same or equivalent components as those of the carrier comparison unit 38A shown in FIG.

As shown in FIG. 6, the carrier comparison unit 38B includes an absolute value calculation unit 38a, a division unit 38b, a multiplication unit 38c, a multiplication unit 38k, an addition unit 38m, an addition unit 38n, a comparison unit 38g, a comparison unit 38h, and output inversion. Part 38i and output inverting part 38j.

The absolute value calculator 38a calculates the absolute value | V * | of the voltage amplitude command V *. In the dividing unit 38b, the absolute value | V * | is divided by the DC voltage V _dc detected by the voltage sensor 20. Also in the configuration of FIG. 6, the output of the division unit 38 b becomes the modulation rate.

In the multiplication unit 38c, a sine value of “θ _e + θ _v ” obtained by adding the advance phase θ _v to the reference phase θ _e is calculated. The calculated sine value of “θ _e + θ _v ” is multiplied by the modulation factor that is the output of the division unit 38b. In the multiplication unit 38k, the voltage command V _m that is the output of the multiplication unit 38c is multiplied by “−1”. In the addition unit 38m, “1” is added to the voltage command V _m that is the output of the multiplication unit 38c. The adder unit 38n, the output of the multiplying unit 38k, that is, "1" to the inverted output of the voltage command _{V m} is added. The output of the adder 38m is input to the comparator 38g as a first voltage command V _m3 for driving the two switching elements 51, 53 of the upper arm among the plurality of switching elements 51, 52, 53, 54. . The output of the adder 38n is input to the comparator 38h as a second voltage command V _m4 for driving the two switching elements 52 and 54 of the lower arm.

The comparison unit 38g compares the first voltage command V _m3 with the carrier amplitude. The output of the output inverting unit 38i obtained by inverting the output of the comparing unit 38g becomes the PWM signal Q1 to the switching element 51, and the output of the comparing unit 38g becomes the PWM signal Q2 to the switching element 52. Similarly, the comparison unit 38h compares the second voltage command V _m4 with the carrier amplitude. The output of the output inverting unit 38j obtained by inverting the output of the comparing unit 38h becomes the PWM signal Q3 to the switching element 53, and the output of the comparing unit 38h becomes the PWM signal Q4 to the switching element 54. The switching element 51 and the switching element 52 are not simultaneously turned on by the output inverting part 38i, and the switching element 53 and the switching element 54 are not simultaneously turned on by the output inverting part 38j.

FIG. 7 is a time chart showing a waveform example of a main part in the carrier comparison unit 38B shown in FIG. FIG. 7 shows the waveform of the first voltage command V _m3 output from the adder 38m, the waveform of the second voltage command V _m4 output from the adder 38n, and the waveforms of the PWM signals Q1, Q2, Q3, and Q4. And a waveform of the motor applied voltage. In FIG. 7, for convenience, the waveform portion of the first voltage command V _m3 whose amplitude value is larger than the peak value of the carrier and the second voltage command V _m4 whose amplitude value is larger than the peak value of the carrier. The waveform portion is represented by a flat straight line.

PWM signal Q1 is "low (Low)" next when the first voltage command _{V m3} is greater than the carrier, the first voltage command _{V m3} is "high (High)" when less than the carrier. The PWM signal Q2 is an inverted signal of the PWM signal Q1. PWM signal Q3 is "low (Low)" next when a second voltage command _{V m4} is greater than the carrier, the second voltage command _{V m4} is "high (High)" when less than the carrier. The PWM signal Q4 is an inverted signal of the PWM signal Q3. In this manner, the circuit shown in FIG. 6 is configured with “Low Active”, but may be configured with “High Active” in which each signal has an opposite value. Good.

As shown in FIG. 7, the waveform of the inverter output voltage shows a voltage pulse due to the difference voltage between the PWM signal Q1 and the PWM signal Q4 and a voltage pulse due to the difference voltage between the PWM signal Q3 and the PWM signal Q2. These voltage pulses are applied to the single-phase motor 12 as a motor applied voltage.

In the waveforms shown in FIG. 7, in one half cycle of one cycle T of the voltage command V _m, the switching operation of the switching elements 51 and 52 are at rest, the other of the one period T of the voltage command V _m In the half cycle, the switching operations of the switching elements 53 and 54 are suspended.

Further, as shown in FIG. 7, the waveform of the inverter output voltage is a unipolar modulation which changes at three potentials per cycle of the voltage command V _m. As described above, bipolar modulation may be used instead of unipolar modulation, but it is preferable to use unipolar modulation in applications where the motor current waveform needs to be controlled to a more sine wave.

Next, the advance angle control in the present embodiment will be described with reference to the drawings of FIGS. Figure 8 is a block diagram showing the carrier comparison section 38A shown in FIG. 4, and the functional configuration for calculating the advance phase theta _v inputted to the carrier comparison section 38B shown in FIG. Figure 9 is a diagram showing an example of a method of calculating the advanced angle phase theta _v in the embodiment. FIG. 10 is a time chart used for explaining the relationship between the voltage command V _m and the advance angle phase θ _v shown in FIGS. 4 and 6.

As shown in FIG. 8, the calculation function of the advance angle phase θ _v can be realized by a rotation speed calculation unit 42 and an advance angle phase calculation unit 44. The rotation speed calculation unit 42 calculates the rotation speed ω of the single-phase motor 12 based on the position sensor signal 21 a detected by the position sensor 21. In addition, the rotation speed calculation unit 42 calculates a reference phase θ _{e obtained} by converting a rotor mechanical angle θ _m that is an angle from the reference position of the rotor 12a into an electrical angle. In the example of FIG. 10, the edge portion where the position sensor signal 21a falls is the reference position of the rotor 12a. The advance phase calculation unit 44 calculates the advance phase θ _v based on the rotation speed ω and the reference phase θ _e calculated by the rotation speed calculation unit 42.

The horizontal axis of FIG. 9 rotational speed N is shown, have been shown advanced angle phase theta _v is the vertical axis in FIG. As shown in FIG. 9, the advance angle phase θ _v can be determined using a function in which the advance angle phase θ _v increases as the rotational speed N increases. In the example of FIG. 9, the advance phase θ _v is determined by a linear function, but is not limited to a linear function. If either advanced angle phase theta _v according to an increase of the rotational speed N is equal or greater relationship may be used functions other than first-order linear function.

In the upper portion of FIG. 10, a position sensor signal 21a output from the position sensor 21 shown in FIG. 2, the rotor mechanical angle theta _m is the angle from a reference position of the rotor 12a illustrated in FIG. 2, the rotor mechanical angle theta _A reference phase θ _e that is a phase obtained by converting _m into an electrical angle is shown.

In the middle part of FIG. 10, waveform examples of two voltage commands V _m are shown as “Example 1” and “Example 2”.

The lowermost portion of FIG. 10, the rotor mechanical angle theta _m when the rotor 12a is rotated in the clockwise direction is 0 °, 45 °, 90 ° , shows a state is 135 ° and 180 °. Four magnets are provided on the rotor 12a of the single phase motor 12, and four teeth 12b1 are provided on the outer periphery of the rotor 12a. If the rotor 12a is rotated clockwise, the position sensor signal 21a corresponding to the rotor machine angle theta _m is detected. The rotation speed calculation unit 42 calculates a reference phase θ _e converted into an electrical angle based on the detected position sensor signal 21a.

In the middle part of FIG. 10, the voltage command V _m shown as “Example 1” is a voltage command in the case of the advance angle phase θ _v = 0. When the advance angle phase θ _v = 0, the voltage command V _m having the same phase as the reference phase θ _e is output. The amplitude of the voltage command V _m at this time is determined based on the voltage amplitude command V * as described above.

In the middle part of FIG. 10, the voltage command V _m shown as “Example 2” is a voltage command in the case of the advance angle phase θ _v = π / 4. When the advance angle phase θ _v = π / 4, a voltage command V _m advanced by π / 4, which is a component of the advance angle phase θ _v , is output from the reference phase θ _e .

Next, a driving method in the motor driving apparatus 2 according to the embodiment will be described. In the present embodiment, control is performed to change the voltage waveform applied to the single-phase motor 12 according to the rotational speed. In the following description, it is assumed that the single-phase motor 12 is applied to an electric blower of a vacuum cleaner. In addition, one Hall sensor is used as the position sensor 21 that detects the rotor rotation position of the single-phase motor 12.

In this Embodiment, the rotation area of an electric blower is divided as follows.
(A) At startup: 0 [rpm] to 10,000 [rpm]
(B) Low speed range (low speed range): 10,000 [rpm] to 20,000 [rpm]
(C) Medium speed rotation range (medium rotation speed range): 50,000 [rpm] to 70,000 [rpm]
(D) High speed rotation range (high rotation speed range): 100,000 [rpm] or more

Note that the region sandwiched between (B) and (C) and the region sandwiched between (C) and (D) are gray zones. The rotational speed from 20,000 [rpm] to 50,000 [rpm] may be included in the low-speed rotation region or in the medium-speed rotation region. In addition, the rotational speed from 70,000 [rpm] to 100,000 [rpm] may be included in the medium-speed rotation region or in the high-speed rotation region.

FIG. 11 is a diagram illustrating an example of waveforms of the carrier, the voltage command _Vm, and the inverter output voltage in the rotation range from the startup to the low-speed rotation range. Voltage pulse sequence shown in the lower portion of FIG. 11 is generated by the voltage command V _m shown in the upper portion of FIG. 11. The voltage command V _m shown in the left half cycle is obtained by setting the value of “θ _e + θ _v ” obtained by adding the advance angle phase θ _v to the reference phase θ _e to “π / 2” in FIG. It is done. Further, the voltage command V _m shown in the right half cycle is obtained by setting the value of “θ _e + θ _v ” to “3π / 2”. As shown in the upper portion of FIG. 11, every half cycle of the voltage command V _m, the magnitude of the voltage command V _m is compared with the carrier is constant. Therefore, every half cycle of the voltage command V _m, a voltage pulse train of positive and negative to remain at a constant amplitude is generated.

If the switching frequency component of the PWM component is removed from the frequency component included in the waveform of the voltage pulse train in the lower part of FIG. 11, the waveform substantially matches the waveform of the rectangular wave indicated by the broken line in the lower part of FIG. Therefore, applying the voltage pulse train shown in the lower part of FIG. 11 to the single-phase motor 12 is equivalent to applying a rectangular wave to the single-phase motor 12. Hereinafter, generating a PWM signal using a voltage command V _m having a constant amplitude every half cycle as shown in the upper part of FIG. 11 is referred to as “rectangular wave PWM”, and the generated PWM signal is “ This is called “rectangular wave PWM signal”.

As described above, the control unit 25 in the present embodiment generates and drives a rectangular wave PWM signal so that the inverter 11 outputs a rectangular wave voltage at the time of startup and after the startup in the low speed rotation range. Output to the signal generator 32. Note that the inverter 11 outputting a rectangular wave voltage means that the voltage waveform obtained by removing the switching frequency component of the PWM control from the frequency component of the inverter output voltage is substantially a rectangular wave.

FIG. 12 is a diagram illustrating an example of waveforms of the carrier, the voltage command _Vm, and the inverter output voltage in the medium speed rotation region. Voltage pulse sequence shown in the lower part of FIG. 12 is generated by the voltage command V _m is a sine wave as shown in the upper portion of FIG. 12. In the lower portion of Figure 12, as it approaches the peak of the voltage command V _m is a sine wave, a pulse width becomes gradually wider, as the distance from the peak of the voltage command V _m, the pulse width is gradually narrower voltage pulse train It is shown.

If the switching frequency component of the PWM control is removed from the frequency component included in the waveform of the voltage pulse train in the lower part of FIG. 12, the waveform substantially coincides with the waveform of the sine wave indicated by the broken line in the lower part of FIG. Therefore, applying the voltage pulse train shown in the lower part of FIG. 12 to the single-phase motor 12 is equivalent to applying a sine wave to the single-phase motor 12. Hereinafter, generating a PWM signal using a sine wave voltage command V _m as shown in the upper part of FIG. 12 is referred to as “sine wave PWM”, and the generated PWM signal is referred to as “sine wave PWM signal”. Call it.

As described above, in the present embodiment, a sine wave PWM signal is generated and output to the drive signal generation unit 32 so that the inverter 11 outputs a sine wave voltage in the medium speed rotation region. The fact that the inverter 11 outputs a sine wave voltage means that the voltage waveform obtained by removing the switching frequency component of the PWM control from the frequency component of the inverter output voltage is substantially a sine wave.

FIG. 13 is a diagram illustrating an example of waveforms of the carrier, the voltage command _Vm, and the inverter output voltage in the high-speed rotation range. Voltage pulse train shown in the lower portion of FIG. 13 is generated by the voltage command V _m shown in the upper portion of FIG. 13. The difference from FIG. 12 is that the peak value of the voltage command V _m is greater than the carrier amplitude. More specifically, FIG. 12 shows a waveform with a modulation factor = 1.0, whereas FIG. 13 shows a waveform with a modulation factor = 1.2.

In FIG. 13, in the region A where the voltage command V _m is larger than the carrier peak, a wide and single voltage pulse is generated as shown in the lower part. In the regions B1 and B2 located on both sides of the region A, a voltage pulse train whose pulse width gradually increases as the region A is approached is generated.

If the switching frequency component of PWM control is removed from the frequency components included in the waveform of the voltage pulse train in the lower part of FIG. 13, the waveform substantially matches the waveform of the trapezoidal wave indicated by the broken line in the lower part of FIG. Therefore, applying the voltage pulse train shown in the lower part of FIG. 13 to the single-phase motor 12 is equivalent to applying a trapezoidal wave to the single-phase motor 12. Hereinafter, as shown in the upper part of FIG. 13, generating a PWM signal using a sine wave voltage command V _m having a modulation rate exceeding 1.0 is referred to as “trapezoidal wave PWM” and is generated. The PWM signal is referred to as a “trapezoidal wave PWM signal”.

As described above, in the present embodiment, a trapezoidal wave PWM signal is generated and output to the drive signal generation unit 32 so that the inverter 11 outputs a trapezoidal wave voltage in the high-speed rotation range. Note that that the inverter 11 outputs a trapezoidal voltage means that the voltage waveform obtained by removing the switching frequency component of the PWM control from the frequency component of the inverter output voltage is substantially a trapezoidal wave.

Next, the significance of controlling the voltage waveform applied to the single-phase motor 12 according to the rotation speed as described above will be described with reference to the drawings of FIGS.

FIG. 14 is a diagram showing an equivalent circuit of the single-phase motor 12 shown in FIG. FIG. 15 is a diagram used for explaining the phase difference between the counter electromotive voltage and the inverter output voltage in the rectangular wave PWM, the sine wave PWM, and the trapezoidal wave PWM in the embodiment. FIG. 16 is a time chart used for explaining the operation at the time of switching from the rectangular wave PWM to the sine wave PWM in the embodiment. FIG. 17 is a diagram showing the relationship between the brake torque generated in the single-phase motor 12 shown in FIG. 1 and the rotational speed. FIG. 18 is a diagram showing a change over time of the brake current flowing through the single-phase motor 12 shown in FIG. FIG. 19 is a diagram illustrating the relationship between the amplitude of the voltage command before voltage limitation and the fundamental wave component of the inverter output voltage.

When magnetic pole position detection is performed using a single hall sensor for the single-phase motor 12 using a permanent magnet, the hall sensor signal is switched only once for one cycle of the counter electromotive voltage as shown in FIG. . For this reason, the resolution of the magnetic pole position is 180 degrees. When the resolution is 180 degrees, it is only known that the magnetic pole position exists in either the range of 0 to 180 degrees or the range of 180 to 360 degrees, for example.

On the other hand, when the motor starts to rotate, the accurate rotation speed can be easily obtained by measuring the switching interval of the edge of the Hall sensor signal. Further, if the phase addition amount of the magnetic pole position from the current edge to the next edge is obtained based on the rotation speed, the magnetic pole position can be estimated.

FIG. 15 shows a waveform when there is a phase difference between the counter electromotive voltage and the motor applied voltage. In the upper part of FIG. 15, in the case of control by the rectangular wave PWM, a case where there is no phase difference is indicated by a broken line, and a case where there is a phase difference is indicated by a one-dot chain line. The counter electromotive voltage is indicated by a broken line and the motor applied voltage is indicated by a solid line in the middle part and the lower part of FIG.

In the case of the rectangular wave PWM, even if there is a slight phase difference, as shown in the upper part of FIG. 15, the application timing of the rectangular wave is merely shifted, and it can be seen that the influence of the phase difference is small. On the other hand, when controlling with sine wave PWM and trapezoidal wave PWM, if there is a phase difference, the difference between the back electromotive voltage and the motor applied voltage is one cycle as shown in the middle and lower parts of FIG. Therefore, the influence of the phase difference becomes large.

Therefore, in the present embodiment, as described above, the inverter output voltage is generated by the rectangular wave PWM in the low-speed rotation range at the start and after the start. As a result, even when the magnetic pole position cannot be estimated accurately, a stable voltage can be supplied.

The effect of making the inverter output voltage constant will be described with reference to FIG. In the equivalent circuit of FIG. 14, V is the inverter output voltage, E is the counter electromotive voltage generated in the stator winding of the single phase motor 12, R is the resistance of the stator winding, L is the inductance of the stator winding, and I is the single phase motor. This is a motor current flowing through 12 stator windings. At this time, the motor current I is expressed by the following equation.

I = (VE) / (R + sL) ... (1)

In the above equation (1), s is a Laplace operator.

In the low-speed rotation range at the start and after the start-up, since the rotation speed is low, the impedance of the single-phase motor 12 is low and the back electromotive voltage is zero or extremely low. Therefore, the value of E in the above equation (1) is small and the value of sL is also small, so the motor current tends to be excessive. Further, since the rotational speed is not stable at the time of starting and in the low speed rotating region after starting, it is difficult to control the current to be constant. On the other hand, when the inverter output voltage is constant, the maximum current can be obtained theoretically. For this reason, it is preferable to control with the rectangular wave PWM in which the amplitude of the applied voltage is constant at the time of starting and in the low speed rotation region after starting. By adopting the rectangular wave PWM, it is possible to reduce the possibility of start-up failure due to overcurrent, uneven rotation in the low-speed rotation region, and the like.

Next, let us consider the case where the control is performed by the rectangular wave PWM up to the medium speed rotation range and the high speed rotation range. Regarding the rectangular wave PWM control, in the upper part of FIG. 15, the waveform of the motor applied voltage is indicated by a solid line, and the waveform of the motor induced voltage is indicated by a broken line and an alternate long and short dash line. As can be understood from the equivalent circuit of FIG. 14, a current corresponding to the difference voltage between the rectangular wave voltage and the sinusoidal motor induced voltage flows through the single-phase motor 12. For this reason, not only the fundamental wave current component synchronized with the rotation speed but also the high frequency current component is superimposed on the single phase motor 12.

In the single-phase motor 12, the fundamental torque current component, which is a current component synchronized with the rotation speed, is closely related to the driving torque that gives the single-phase motor 12 driving force. On the other hand, current components other than the fundamental wave current component serve as brake torque that inhibits rotation. Since the brake torque gives a braking force to the rotation of the single-phase motor 12, it causes an instantaneous decrease in the number of rotations, vibration and noise. A high frequency current having a frequency higher than the fundamental current causes high frequency iron loss in the single-phase motor 12. An increase in high-frequency iron loss reduces the efficiency of the single-phase motor 12. For this reason, it is preferable to switch to a sine wave PWM that can efficiently output a fundamental wave current component synchronized with the rotation speed as the rotation speed increases.

When switching from the rectangular wave PWM to the sine wave PMW, as described above, information on the magnetic pole position estimated based on the Hall sensor signal is used. In the estimation of the magnetic pole position, the rotational speed is calculated by measuring the edge switching interval of the Hall sensor signal. Further, based on the calculated rotational speed, a continuous magnetic pole position of 0 to 360 degrees is calculated as shown in FIG.

When the rotation speed reaches the middle speed rotation range, the rectangular wave PWM is switched to the sine wave PWM based on the magnetic pole position. When switching from rectangular wave PWM to sine wave PWM, a phenomenon called motor current pulsation occurs due to a change in output voltage at the time of switching, and there is a possibility that vibration and noise increase due to a sudden change in motor torque. Therefore, it is preferable to reflect the fundamental wave component of the rectangular wave voltage in the rectangular wave PWM in the sine wave voltage in the sine wave PWM. Specifically, the following control is performed.

As is well known, if a rectangular wave having an amplitude of “1” is Fourier-transformed, the coefficient added to the fundamental wave component of the rectangular wave is “4 / π”. Therefore, when switching from the rectangular wave PWM sinusoidal PWM, the amplitude of the voltage command V _m immediately before switching may output (4 / π) times to. When the amplitude of the voltage command V _m in the rectangular wave PWM is “K”, the voltage command V _m in the sine wave PWM may be a sine wave having an amplitude of “(4 / π) × K”. When the voltage command V _m is controlled based on the modulation rate, the modulation rate immediately after switching may be obtained by multiplying the modulation rate immediately before switching by (4 / π). By doing so, it is possible to keep the torque generated at the time of switching constant. As a result, it is possible to perform switching while suppressing generation of vibration and noise.

Next, considerations regarding the switching frequency, which is the on / off switching frequency of the switching elements 51, 52, 53, and 54, will be described.

If the switching frequency is set to, for example, 20 kHz or more outside the audible frequency, it is possible to reduce annoying noise and to realize a motor driving device with low noise. Further, increasing the switching frequency has the effect of improving the resolution of the output voltage during high-speed operation and improving the accuracy of the rotational speed. However, when the switching frequency is increased, heat generation and switching loss of the switching element increase. For this reason, it is preferable to determine the switching frequency in consideration of the heat generation amount and the efficiency.

To reduce the number of switching times, performed once or twice a voltage output of the half cycle of the voltage command V _m, that of the rest section and freewheel period, there is a method of the motor current substantially sinusoidal. The free wheel period is a period during which current flows back between the stator winding of the single-phase motor 12 and the switching elements 52 and 54. In the freewheel period, the switching element 52 that is the lower arm first element and the switching element 54 that is the lower arm second element are controlled to be on, and the switching element 51 that is the upper arm first element and the upper arm second element The switching element 53 which is an element is controlled to be turned off.

FIG. 17 shows the relationship between the brake torque and the rotational speed during the freewheel period. When the free wheel period continues for a long time, as shown in FIG. 17, brake torque is generated according to the rotational speed of the single-phase motor 12. For this reason, current is consumed by the resistance R of the stator winding, and brake torque is generated. When brake torque is generated, speed fluctuation occurs. Since the frequency at which the brake torque is generated is within the audible frequency, deterioration of noise is inevitable.

FIG. 18 shows a time-varying waveform of the brake current during the freewheel period. The brake current is a current that flows due to the inductance L of the stator winding and the resistance R of the stator winding. For this reason, when the free wheel period becomes longer, the period during which the brake current flows becomes longer as shown in FIG. When the brake current is relatively large and the flow period is long, demagnetization of the permanent magnet becomes a problem. Further, the loss due to the resistance R of the stator winding also increases due to the brake current.

On the other hand, as shown in the lower part of FIG. 12, the sine wave PWM has a large number of voltage outputs and a short free wheel period in which no voltage pulse is generated. For this reason, the loss in the stator winding is reduced, and the influence of demagnetization of the permanent magnet can be reduced.

Also, if the switching frequency is set to 20 kHz or higher, the speed pulsation due to the brake torque during the freewheel period can be set to 20 kHz or higher. Therefore, if the switching frequency is set to 20 kHz or more, the generation of noise due to speed pulsation can be suppressed.

However, in the case of sine wave PWM, the number of switching of the switching element increases as can be understood from the waveform in the lower part of FIG. For this reason, the switching loss increases as the number of times of switching increases. The increase in switching loss and the increase in high-frequency iron loss are in a toled-off relationship, but the method of the present embodiment has a great effect of reducing high-frequency iron loss. For this reason, if the method of this Embodiment is employ | adopted, efficiency improvement can be aimed at even if the increase in switching loss is considered.

In order to further improve the switching loss, it is better to use the one-sided PWM shown in FIG. 7 than the two-sided PWM shown in FIG. If one-sided PWM is used, the number of times of switching can be halved compared to using both-sided PWM, so that switching loss can be improved.

In the case of double-sided PWM, the frequency twice the switching frequency is dominant, and in the case of single-sided PWM, the same frequency as the switching frequency is dominant. Therefore, as described above, if the switching frequency is set to 20 kHz or higher, noise deterioration can be suppressed even if one-side PWM is used.

In addition, it is preferable that the phase of the motor induced voltage and the phase of the sine wave current flowing by the sine wave PWM coincide with each other. However, in an inductive load such as the single-phase motor 12, the sine wave current has a delayed phase corresponding to the rotational speed with respect to the motor-induced voltage. For this reason, it is preferable to control the phase of the fundamental wave component included in the motor applied voltage in accordance with the rotational speed. Thereby, efficient driving | operation becomes possible.

Further, when the single-phase motor 12 is accelerated, the load torque of the single-phase motor 12 increases as the rotational speed increases. Here, if the motor torque is T _M , the load torque is T _L , the inertia moment of the rotor 12 a is J _M , and the rotation speed is ω, the change rate of the rotation speed can be expressed by the following equation.

dω / dt = (T _M −T _L ) / J _M (2)

For example, when the load is a propeller fan, the load torque is generally said to be proportional to the second to third power of the rotation speed. When accelerating the load, it is necessary to output a motor torque that is equal to or greater than the load torque, as shown in the above equation (2). If acceleration is performed without considering the characteristics of the load torque, the current when the target rotational speed is reached becomes excessive, which may cause demagnetization of the motor. For this reason, as the target rotational speed is approached, it is preferable to operate with a reduced acceleration rate. Thereby, the motor drive device which suppressed the electric current increase at the time of acceleration is realizable. In an application example such as a vacuum cleaner or a hand dryer, if the rotational speed of about 80% of the target rotational speed can be reached quickly, even if the subsequent acceleration rate decreases, the performance is sufficient.

Next, the trapezoidal wave PWM will be described. As described above, a trapezoidal wave PWM signal is generated and applied to the single-phase motor 12 in the high-speed rotation range. In the sine wave PWM, when the rotation speed increases, the amplitude of the voltage command V _m increases. However, when the amplitude of the voltage command V _m exceeds the DC voltage V _dc , the inverter 11 cannot output a voltage exceeding the DC voltage V _dc , so the voltage is limited. As a result, in the lower part of FIG. 15, the voltage waveform becomes a trapezoidal wave as shown by the solid line.

In the case of trapezoidal wave PWM, switching operation is not performed near the peak of the trapezoidal wave. For this reason, the switching loss is improved as compared with the sine wave PWM. In the case of trapezoidal wave PWM, the freewheel period occurs only before and after the zero crossing. For this reason, the speed fluctuation resulting from the brake torque is alleviated, and stable operation is possible.

FIG. 19 shows the relationship between the amplitude of the voltage command before voltage limitation and the fundamental wave component of the inverter output voltage. In FIG. 19, the region on the right side of the broken line drawn in parallel with the vertical axis is a region where the voltage is limited and the inverter output voltage becomes a trapezoidal wave. Even if it becomes a trapezoidal wave and the inverter output voltage is limited, the fundamental wave component of the inverter output voltage continues to rise as shown in FIG. Theoretically, a time when the amplitude of the voltage command V _m becomes infinite is a rectangular wave. Assuming that the voltage limit value is “1”, the value of the fundamental component of the inverter output voltage is “4 / π (= 1.27)”, which is a coefficient of the fundamental component when the square wave is expanded by Fourier series. Asymptotically. That is, if the trapezoidal wave PWM is used, a voltage up to 1.27 times the voltage limit value can be output.

On the other hand, when the rectangular wave PWM is used, it is necessary to consider the timing at which the rectangular wave voltage is switched from positive to negative or from negative to positive. When an error occurs in the timing at which the rectangular wave voltage is switched from positive to negative or from negative to positive, an imbalance occurs between the positive voltage application period and the negative voltage application period. This imbalance is not a problem in low-speed operation because the voltage cycle is long, but in high-speed operation, the voltage cycle is short, and therefore the influence of timing errors becomes large. When an imbalance occurs, a direct current component is superimposed on the motor applied voltage, so that a braking force may be generated or a loss may be deteriorated.

Therefore, when the trapezoidal wave PWM, 1 times or more voltage limit the amplitude of the voltage command V _m, and is preferably 2 times or less. Thereby, the switching of the inverter output voltage before and after the zero crossing becomes gradual, and the influence of the timing error is mitigated.

In addition, as described above, the trapezoidal wave PWM can output a voltage that is 1.27 times the voltage limit value. However, if the voltage is insufficient, the phase of the trapezoidal wave PWM signal may be controlled to a lead phase. . By this control, the magnetic force of the single-phase motor 12 is weakened and the back electromotive voltage is suppressed, so that operation up to higher speed rotation is possible.

As described above, the control unit according to the embodiment causes the inverter to output a rectangular wave voltage at the time of starting the single phase motor and in the low speed rotation range, and the sine wave voltage to the inverter in the medium speed rotation range of the single phase motor. And output a trapezoidal wave voltage to the inverter in the high-speed rotation range of the single-phase motor. As a result, vibration and noise can be suppressed when the motor is accelerated.

Next, application examples of the motor drive device according to the embodiment will be described. FIG. 20 is a configuration diagram of the electric vacuum cleaner 61 including the motor driving device 2 according to the embodiment. The vacuum cleaner 61 includes a battery 10 shown in FIG. 1, a motor drive device 2 shown in FIG. 1, an electric blower 64 driven by the single-phase motor 12 shown in FIG. 1, a dust collection chamber 65, The sensor 68, the suction inlet 63, the extension pipe 62, and the operation part 66 are provided.

The user who uses the vacuum cleaner 61 has the operation unit 66 and operates the vacuum cleaner 61. The motor drive device 2 of the electric vacuum cleaner 61 drives the electric blower 64 using the battery 10 as a power source. When the electric blower 64 is driven, dust is sucked from the suction port body 63. The sucked dust is collected in the dust collection chamber 65 through the extension pipe 62.

The vacuum cleaner 61 is a product in which the rotation speed of the single-phase motor 12 varies from 0 [rpm] to over 100,000 [rpm]. When such a single-phase motor 12 drives a product that rotates at a high speed, the control method according to the above-described embodiment is suitable.

For example, the inverter 11 outputs a rectangular wave voltage to the single-phase motor 12 at the time of startup and in a low-speed rotation range. In the middle speed range, a sine wave voltage is output. In the high-speed rotation range, the inverter 11 outputs a trapezoidal wave voltage. By controlling in this way, vibration and noise can be suppressed when the single-phase motor 12 is accelerated.

For example, when the output is switched from a rectangular wave voltage to a sine wave voltage, the control unit 25 switches the fundamental wave component of the rectangular wave voltage and the fundamental wave component of the sine wave voltage in a matched state. By doing in this way, the torque generated at the time of switching can be kept constant, and switching can be performed while suppressing the generation of vibration and noise.

For example, when the rotational speed of the single-phase motor 12 is controlled to the target rotational speed, the control unit 25 reduces the acceleration rate as the rotational speed approaches the target rotational speed. Thereby, it can suppress that the electric current at the time of reaching | attaining target rotational speed becomes excessive, and it becomes possible to suppress the demagnetization of the single phase motor 12. FIG.

In addition, when outputting a voltage based on the voltage command to the single-phase motor 12, the control unit 25 performs a switching operation between the upper arm first element and the lower arm first element in one half cycle of the voltage command cycle. And the switching operation of the upper arm second element and the lower arm second element is suspended in the other half cycle of the voltage command cycle. Thereby, the increase in switching loss is suppressed and the efficient vacuum cleaner 61 is realizable.

Moreover, the vacuum cleaner 61 according to the embodiment can be reduced in size and weight by simplifying the heat dissipation component described above. Furthermore, since the vacuum cleaner 61 does not require a current sensor for detecting a current and does not require a high-speed analog-digital converter, the vacuum cleaner 61 in which an increase in design cost and manufacturing cost is suppressed can be realized. .

FIG. 21 is a configuration diagram of a hand dryer provided with the motor drive device according to the embodiment. The hand dryer 90 includes a motor driving device 2, a casing 91, a hand detection sensor 92, a water receiver 93, a drain container 94, a cover 96, a sensor 97, an air inlet 98, and an electric blower 95. Is provided. Here, the sensor 97 is either a gyro sensor or a human sensor. In the hand dryer 90, when a hand is inserted into the hand insertion part 99 at the upper part of the water receiver 93, water is blown off by the air blow by the electric blower 95, and the blown water is collected by the water receiver 93. After that, it is stored in the drain container 94.

The hand dryer 90 is a product in which the motor rotation speed fluctuates from 0 [rpm] to over 100,000 [rpm], similarly to the electric vacuum cleaner 61 shown in FIG. For this reason, also in the hand dryer 90, the control method which concerns on embodiment mentioned above is suitable, and the effect similar to the vacuum cleaner 61 can be acquired.

As described above, in the present embodiment, the configuration example in which the motor driving device 2 is applied to the electric vacuum cleaner 61 and the hand dryer 90 has been described. However, the motor driving device 2 is applied to an electric device in which the motor is mounted. can do. Electric equipment equipped with motors is incinerator, crusher, dryer, dust collector, printing machine, cleaning machine, confectionery machine, tea making machine, woodworking machine, plastic extruder, cardboard machine, packaging machine, hot air generator, OA Equipment, electric blower, etc. The electric blower is a blowing means for transporting objects, for sucking dust, or for general air supply / discharge.

Note that the configurations shown in the above embodiments are examples of the contents of the present invention, and can be combined with other known techniques, and can be combined without departing from the gist of the present invention. It is also possible to omit or change a part of.

1 motor drive system, 2 motor drive device, 5A 1st leg, 5B 2nd leg, 6A, 6B connection point, 10 battery, 11 inverter, 12 single phase motor, 12a rotor, 12b stator, 12b1 teeth, 20 voltage sensor, 21 position sensor, 21a position sensor signal, 25 control unit, 31 processor, 32 drive signal generation unit, 33 carrier generation unit, 34 memory, 38, 38A, 38B carrier comparison unit, 38a absolute value calculation unit, 38b division unit, 38c , 38d, 38f, 38k multiplication unit, 38e, 38m, 38n addition unit, 38g, 38h comparison unit, 38i, 38j output inversion unit, 42 rotation speed calculation unit, 44 advance phase calculation unit, 51, 52, 53, 54 Switching element 51a, 52a, 53a, 5 a body diode, 61 vacuum cleaner, 62 extension pipe, 63 suction port, 64 electric blower, 65 dust collection chamber, 66 operation unit, 68 sensor, 90 hand dryer, 91 casing, 92 hand detection sensor, 93 water receiver Part, 94 drain container, 95 electric blower, 96 cover, 97 sensor, 98 air inlet, 99 manual insertion part.

Claims (8)

1. An inverter that outputs an AC voltage to a single-phase motor;
   A control unit for controlling the AC voltage output by the inverter;
   With
   The controller is
   In the start-up and low-speed rotation range of the single-phase motor, let the inverter output a rectangular wave voltage,
   In the medium speed rotation region of the single-phase motor, the inverter outputs a sine wave voltage,
   A motor driving device that causes the inverter to output a trapezoidal wave voltage in a high-speed rotation range of the single-phase motor.
2. When the control unit switches the output from the rectangular wave voltage to the sine wave voltage,
   The motor drive device according to claim 1, wherein switching is performed in a state in which a fundamental wave component of the rectangular wave voltage and a fundamental wave component of the sine wave voltage are matched.
3. 3. The motor driving device according to claim 1, wherein the control unit reduces the acceleration rate as the rotational speed approaches the target rotational speed when controlling the rotational speed of the single-phase motor to the target rotational speed. 4. .
4. The inverter includes a first leg in which an upper arm first element and a lower arm first element are connected in series, a parallel connection to the first leg, and an upper arm second element and a lower arm second element. A second leg connected in series,
   The single-phase motor is connected between a connection point between the upper arm first element and the lower arm first element and a connection point between the upper arm second element and the lower arm second element,
   When outputting a voltage based on a voltage command to the single-phase motor,
   In one half cycle of the voltage command cycle, the upper arm first element and the lower arm first element pause the switching operation,
   The motor drive device according to any one of claims 1 to 3, wherein the upper arm second element and the lower arm second element pause switching operation in the other half period of the voltage command period. .
5. The motor driving device according to claim 4, wherein the upper arm first element, the lower arm first element, the upper arm second element, and the lower arm second element are formed of a wide band gap semiconductor.
6. The motor driving device according to claim 5, wherein the wide band gap semiconductor is silicon carbide, gallium nitride, or diamond.
7. A vacuum cleaner comprising the motor drive device according to any one of claims 1 to 6.
8. A hand dryer provided with the motor drive device according to any one of claims 1 to 6.

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