JP6644159B2 - Motor drive, electric blower, vacuum cleaner and hand dryer - Google Patents

Description

  The present invention relates to a motor driving device for driving a single-phase motor, and an electric blower, a vacuum cleaner and a hand dryer equipped with a single-phase motor driven by the motor driving device.

A single-phase motor has the following advantages as compared with a three-phase motor having a different number of phases.
(1) While a three-phase motor requires a three-phase inverter, a single-phase motor may be a single-phase inverter.
(2) When a full-bridge inverter generally used as a three-phase inverter is used, six switching elements are required. On the other hand, in the case of a single-phase motor, even if a full-bridge inverter is used, four switching elements are required. Can be configured.
(3) Due to the features of (1) and (2), the size of the single-phase motor can be reduced as compared with the three-phase motor.

  When a single-phase motor is driven by a single-phase inverter, it is required to reduce harmonic components of a current flowing through the single-phase motor. Patent Document 1 below discloses a PWM for controlling a supply voltage to be supplied to a single-phase motor to reduce a harmonic component so that a current (hereinafter, referred to as “motor current”) flowing through the single-phase motor is made into a sine wave. (Pulse Width Modulation) control technology is disclosed. Also, Patent Document 2 below discloses a method of switching output voltage pulses in accordance with switching of a position sensor signal.

JP 2012-257457 A Japanese Patent No. 5524925

  However, when PWM control is applied to a product such as a vacuum cleaner having a wide driving range of the motor rotation speed, simply controlling the supply voltage to the single-phase motor narrows the driving range of the motor rotation speed, thereby increasing the speed of the motor. There was a problem that it was not possible to drive until rotation.

  The present invention has been made in view of the above, and even when PWM control is used for controlling a single-phase motor, a motor driving device that does not hinder application to products that require high-speed rotation It is an object to obtain an electric blower, a vacuum cleaner and a hand dryer.

  In order to solve the above-described problems and achieve the object, a motor drive device according to the present invention includes a plurality of switching elements, a single-phase inverter that applies an AC voltage to a single-phase motor, and a rotation of the single-phase motor. A control unit is provided for generating a PWM signal for performing PWM control of the plurality of switching elements based on a signal for detecting a position and a voltage command. The control unit controls the amplitude of the voltage command to be constant, and increases the lead angle of the voltage command according to the increase in the rotation speed of the single-phase motor.

  Advantageous Effects of Invention According to the present invention, even when PWM control is used for controlling a single-phase motor, there is an effect that application to a product requiring high-speed rotation is not hindered.

FIG. 2 is a block diagram illustrating a configuration of a motor drive system including a motor drive device according to the present embodiment. Circuit configuration diagram of the inverter shown in FIG. FIG. 2 is a block diagram showing a functional configuration for generating a PWM signal. FIG. 3 is a block diagram showing the configuration shown in FIG. 3 in detail. Time chart showing waveform examples of voltage command, PWM signal and motor applied voltage Waveform diagram showing change in inverter output voltage according to modulation rate FIG. 3 is a block diagram showing a functional configuration for calculating an advance phase which is an input signal to the carrier generation unit and the carrier comparison unit shown in FIGS. 3 and 4; The figure which shows an example of the calculation method of an advance angle phase Time chart showing the relationship between rotor magnetic pole position, position sensor signal and voltage command Time chart showing time change of voltage amplitude command The figure which shows the path | route of the motor current when the polarity of an inverter output voltage is positive and negative. The figure which shows the path | route of the motor current when the return current flows into the body diode of MOSFET The figure which shows the path | route of the motor current when flowing a return current to the channel of MOSFET. The figure which shows an example of a structure of a vacuum cleaner as an example of application of the motor drive device in this Embodiment. The figure which shows an example of a structure of a hand dryer as another application example of the motor drive device in this Embodiment. FIG. 4 is a diagram for describing modulation control suitable for a motor drive device according to the present embodiment.

  Hereinafter, a motor drive device, an electric blower, and a vacuum cleaner according to an embodiment of the present invention will be described in detail with reference to the drawings. The present invention is not limited by the following embodiments. Further, in the following description, an example of application to a vacuum cleaner and a hand dryer will be mainly described, but this is not intended to exclude application to other uses.

Embodiment.
FIG. 1 is a block diagram illustrating a configuration of a motor drive system including a motor drive device according to the present embodiment. As shown in FIG. 1, a motor drive system 1 includes a single-phase motor 12, a motor drive device 2 that supplies AC power to the single-phase motor 12 to drive the single-phase motor 12, and a DC power supply to the motor drive device 2. , A voltage sensor 20 for detecting a DC applied voltage applied to the motor drive device 2 by the power supply 10, and a rotor rotational position of the single-phase motor 12, which is a rotational position of the rotor 12 a. A position sensor 21. Examples of the load including the single-phase motor 12 include a vacuum cleaner having an electric blower and a hand dryer. In the above description, the voltage sensor 20 detects a DC applied voltage. However, the voltage sensor 20 may detect a power supply voltage that is an output voltage of the power supply 10.

  Further, the motor driving device 2 is connected to the single-phase motor 12 and applies a single-phase inverter 11 that applies an AC voltage to the single-phase motor 12 and analog data that is a detection value of a DC applied voltage detected by the voltage sensor 20. An analog-to-digital converter 30 for converting into digital data; a DC applied voltage read from the analog-to-digital converter 30; a position sensor signal from the position sensor 21; A control unit 25 that generates a PWM signal based on the value; a drive signal generation unit 32 that generates a drive signal for driving a switching element in the single-phase inverter 11 based on the PWM signal output from the control unit 25; , Is provided.

  The control unit 25 has a processor 31, a carrier generation unit 33, and a memory. The processor 31 generates a PWM signal described later by the PWM control. The drive signal generation unit 32 generates a drive signal for driving the single-phase inverter 11 based on the PWM signal from the processor 31 and outputs the drive signal to the single-phase inverter 11.

  Single-phase motor 12 is preferably a brushless motor. In the brushless motor, a plurality of permanent magnets (not shown) arranged in the circumferential direction are arranged on the rotor 12a. The plurality of permanent magnets are arranged so that the magnetization directions are alternately reversed in the circumferential direction, and form a plurality of magnetic poles of the rotor 12a. A winding (not shown) is wound around the stator. The motor current is an alternating current flowing through the winding. In this embodiment, the number of magnetic poles is four. However, the number of magnetic poles may be other than four.

  The position sensor 21 outputs a position sensor signal, which is a digital signal, to the control unit 25. The position sensor signal is a signal for detecting the rotor rotational position, and indicates a high or low value according to the direction of the magnetic flux from the rotor 12a.

  FIG. 2 is a circuit configuration diagram of the single-phase inverter 11. The single-phase inverter 11 has switching elements 51, 52, 53, 54 connected in a bridge. The switching elements 51 and 53 located on the high potential side are called upper-arm switching elements. The switching elements 52 and 54 located on the low potential side are called lower arm switching elements. The connection end of the switching element 51 and the switching element 52 and the connection end of the switching element 53 and the switching element 54 form AC terminals in a bridge circuit, and these AC terminals have a circuit configuration in which the single-phase motor 12 is connected. .

  As the switching elements 51, 52, 53, 54, use is made of a MOSFET (Metal-Oxide-Semiconductor Field Effect Transistor). A MOSFET is an example of an FET. A MOSFET is a switching element capable of flowing current bidirectionally between a drain and a source. In each of the switching elements 51, 52, 53, and 54, a diode connected in parallel between the drain and the source is called a free wheel diode. However, in the present embodiment, a body diode which is a parasitic diode formed inside the MOSFET is used as a freewheeling diode.

  Further, at least one of the switching elements 51, 52, 53, 54 can be formed using a wide band gap semiconductor. The wide gap semiconductor is, for example, GaN (gallium nitride), SiC (silicon carbide), or diamond. By using a wide bandgap semiconductor for at least one of the switching elements 51, 52, 53, and 54, the withstand voltage and the allowable current density of the switching element are increased. The size can be reduced. Further, since the wide band gap semiconductor has high heat resistance, it is possible to reduce the size of the heat radiating portion and simplify the heat radiating structure.

  FIG. 3 is a block diagram showing a functional configuration for generating a PWM signal. FIG. 4 is a block diagram showing the configuration shown in FIG. 3 in detail. The function of generating a PWM signal can be realized by a carrier generation unit 33 and a carrier comparison unit 38, as shown in FIG.

In FIG. 3, an advance phase θ _v used when generating voltage commands V _m1 and V _m2 described later is input to the carrier generation unit 33. Here, the “advance angle phase” represents the “advance angle”, which is the “advance angle” of the voltage command, as a phase. The “lead angle” here is a phase difference between the motor applied voltage applied to the stator winding by the single-phase inverter 11 and the motor induced voltage induced in the stator winding (not shown). Note that when the motor applied voltage is ahead of the motor induced voltage, the “lead angle” takes a positive value.

Returning to FIG. 3, the advance phase θ _v , the DC applied voltage V _dc , and the voltage amplitude command V * which is the amplitude value of the voltage command V _m are input to the carrier comparison unit 38. Carrier generating unit 33 generates a carrier based on the advanced angle phase theta _v. The carrier comparing unit 38 generates the PWM signals Q1 to Q4 based on the carrier, the DC applied voltage _Vdc, and the voltage amplitude command V *.

The carrier generation unit 33 has a carrier frequency setting unit 33a, as shown in FIG. In the carrier frequency setting unit 33a, a carrier frequency f _C [Hz], which is the frequency of the carrier, is set, a carrier synchronized with the advance phase θ _v is generated, and output to the carrier comparing unit 38. At the tip of the arrow of the carrier frequency setting unit 33a, a triangular wave carrier that moves up and down between “0” and “1” is shown as an example of the carrier waveform. Note that the PWM control for the single-phase inverter 11, synchronous PWM control and it is asynchronous PWM control, there is no need to synchronize the carrier to advance the phase theta _v in the case of the asynchronous PWM control.

  As shown in FIG. 4, the carrier comparing section 38 includes an absolute value calculating section 38a, a dividing section 38b, multiplying sections 38c, 38d, 38f, an adding section 38e, comparing sections 38g, 38h, and output inverting sections 38i, 38j. Has functional blocks.

The absolute value calculator 38a calculates the absolute value | V * | of the voltage amplitude command V *. In the divider 38b, the absolute value | V * | is divided by the DC applied voltage _Vdc detected by the voltage sensor 20. When the power supply 10 is a battery, the battery voltage fluctuates, but by dividing by the detection value of the voltage sensor 20, the modulation rate can be increased so that the motor applied voltage does not decrease due to the decrease in the battery voltage. In FIG. 1, the DC applied voltage _Vdc is detected by the voltage sensor 20. However, when the output voltage of the power supply 10 is stable, such as when the power supply 10 is connected to a commercial power supply, Instead of using the detection value of the voltage sensor 20, an internally generated value may be used.

The multiplier unit 38c, the sine of advanced angle phase theta _v is calculated and multiplied by the output from the divider 38b. The multiplier 38d multiplies the output of the multiplier 38c by １ ／. The adder 38e adds １ ／ to the output result of the multiplier 38d. In the multiplier 38f, the output of the divider 38b is multiplied by -1. Here, the output of the addition unit 38e is input to the comparison unit 38g as the voltage command _Vm1 used for driving the switching elements 51, 52 among the switching elements 51, 52, 53, 54, and the output of the multiplication unit 38f is It is input to the comparison unit 38h as a voltage command _{V m2} to be used to drive the switching elements 53 and 54.

  The output of the comparing section 38g becomes a PWM signal to the switching element 51, and the output of the output inverting section 38i obtained by inverting the output of the comparing section 38g becomes a PWM signal to the switching element 52. Similarly, the output of the comparing section 38h becomes a PWM signal to the switching element 53, and the output of the output inverting section 38j obtained by inverting the output of the comparing section 38h becomes a PWM signal to the switching element 54. Due to the presence of the output inversion unit 38i, the switching element 51 and the switching element 52 are not simultaneously turned on, and due to the presence of the output inversion unit 38j, the switching element 53 and the switching element 54 are not simultaneously turned on.

FIG. 5 is a time chart illustrating waveform examples of the voltage commands V _m1 and V _m2 , the PWM signal, and the motor applied voltage. The upper part of FIG. 5 shows the waveform of voltage command V _m1 output from adder 38e, and the upper middle part of FIG. 5 shows the waveform of voltage command V _m2 output from multiplier 38f. ing. By using the voltage command _Vm1 , the PWM signals Q1 and Q2 as shown in the lower middle part of FIG. 5 can be generated. Further, by using the voltage command V _{m @ 2,} it is possible to generate a PWM signal Q3, Q4 as shown in the lower part in Figure 5. Further, by controlling the switching elements 51, 52, 53, 54 in the single-phase inverter 11 using such PWM signals Q1, Q2, Q3, Q4, the PWM as shown in the lower part of FIG. The controlled voltage pulse waveform can be applied to the single-phase motor 12.

  By the way, as a modulation method used when generating a PWM signal, a bipolar modulation that outputs a voltage pulse that changes at a positive or negative potential, a positive or zero potential every power supply half cycle, or a negative or zero potential, That is, a unipolar modulation that outputs a voltage pulse that changes at three positive, zero, or negative potentials is known. The waveform shown in FIG. 5 is based on unipolar modulation. Any modulation method may be used as the motor drive device in the present embodiment. In applications where it is necessary to control the motor current waveform to a sine wave, it is preferable to employ unipolar modulation having a higher harmonic content than bipolar modulation.

As described above, the motor applied voltage is determined by comparing the carrier with the voltage command value. Voltage pulse included in the motor applied voltage, during high-speed rotation, and also increased the frequency of the voltage command V _m, because the number of voltage pulses outputted to the electric angle in one cycle is decreased, the number of voltage pulse current The effect on waveform distortion also increases. Generally, when an even-numbered voltage pulse is applied, even-order harmonics are superimposed and the symmetry between the positive and negative waveforms is lost. Therefore, in order to make the motor current waveform approach a sine wave in which the content of harmonics is suppressed, it is preferable to control the number of voltage pulses in one cycle of the electrical angle to be an odd number. By controlling the number of voltage pulses in one cycle of the electrical angle to be an odd number, the motor current waveform can be approximated to a sine wave.

  Next, a method for setting the number of voltage pulses in one cycle of the electrical angle to an odd number will be described. In order to control the number of output voltage pulses to be an odd number during high-speed rotation, there is a method of synchronizing the triangular wave carrier with the number of motor rotations. Further, in the case of performing control using a rotational speed as a command value in product specifications, there is a method of determining a carrier frequency in advance for a rotational speed command. Since each method is a well-known technique, a detailed description is omitted here. In the present embodiment, the control is not limited to this as long as the control is such that the output voltage is odd.

  FIG. 6 is a waveform diagram showing a change in the inverter output voltage according to the modulation factor. From the upper side, three patterns when the modulation rate is 1.0, the modulation rate is 1.2, and the modulation rate is 2.0 are shown. The “inverter output voltage” has the same meaning as the “motor applied voltage” shown in the lower part of FIG.

As described with reference to FIGS. 4 and 5, the voltage commands V _m1 and V _m2 are compared with the carrier, and PWM signals Q1, Q2, Q3 and Q4 corresponding to the comparison result are generated. The switching elements 51, 52, 53, and 54 in the single-phase inverter 11 are controlled by the generated PWM signals Q1, Q2, Q3, and Q4, and correspond to the modulation rates shown in each stage of FIG. An inverter output voltage is generated and applied to single-phase motor 12.

  Although there are various definitions of the modulation rate, here, the ratio between the voltage amplitude command V * and the amplitude of the triangular wave carrier, that is, “voltage amplitude command V * / triangle wave carrier amplitude” is defined as the modulation rate. Although the case where the modulation rate is 1.0 is shown in the upper part of FIG. 6, a similar waveform is obtained when the modulation rate is less than 1.0. When the modulation rate is less than 1.0, the inverter output voltage is generated according to the frequency of the triangular wave carrier, and therefore, the inverter output voltage also outputs a voltage pulse corresponding to the carrier frequency.

On the other hand, when the modulation factor exceeds 1.0, the waveform is as shown in the middle and lower parts of FIG. When the modulation rate exceeds 1.0, it is called “overmodulation”, and the area where the modulation rate exceeds 1.0 is called “overmodulation area”. In the overmodulation region, there are portions where the voltage commands V _m1 and V _m2 exceed the amplitude of the carrier, so that a section occurs in which an inverter drive signal cannot be generated according to the carrier frequency. In this section, since the inverter output voltage is fixed to the positive power supply voltage or the negative power supply voltage, it is possible to obtain a large output voltage as compared with the case where the modulation factor is 1.

Next, the advance angle control in the present embodiment will be described. Figure 7 is a block diagram showing a functional configuration for calculating the input signal is advanced angle phase theta _v of the carrier generating unit 33 and the carrier comparison section 38 shown in FIGS. As shown in FIG. 7, the function for calculating the advance phase θ _v is a rotation speed ω of the single-phase motor 12 and a rotor mechanical angle θ which is an angle from the reference position of the rotor 12a based on the position sensor signal. a rotation speed calculation unit 42 for calculating a reference phase θ _{e in} which _m is converted to an electrical angle, and an advance phase θ _v based on information on the rotation speed ω and the reference phase θ _e calculated by the rotation speed calculation unit 42. This can be realized by the advance angle phase calculation unit 44.

Figure 8 is a diagram showing an example of a calculation method of the advanced angle phase theta _v. The advance phase θ _v can be determined using a function in which the advance phase θ _v increases with an increase in the rotation speed N, as shown in FIG. In the example of FIG. 8, but determines the advanced angle phase theta _v by first-order linear function, and as long advanced angle phase theta _v a larger relationship with the increase of the rotational speed , Any function may be used.

9, the magnetic pole position of the rotor is a time chart showing the relationship between the position sensor signal and the voltage command V _m. As shown in the lowermost part of FIG. 9, this is an example in which the rotor is provided with four-pole magnets and four stators are provided on the outer periphery of the rotor. When the rotor is rotated in the clockwise direction, the position sensor signals corresponding to the rotor mechanical angle theta _m is detected, the reference phase theta _e which is converted to an electrical angle in accordance with the detected position sensor signal is calculated.

In the middle part of FIG. 9, the case where the advance angle phase θ _v = 0 is shown as Example 1. For advanced angle phase θ _{v =} 0, the voltage command V _m is a sine wave voltage of the reference phase theta _e in phase are outputted. The amplitude of the voltage command V _m at this time is determined based on the voltage amplitude command V * as described above. Further, a case where the advance phase θ _v = π / 4 is shown as Example 2. When the advance phase θ _v = π / 4, the voltage command _Vm has a component advanced by the advance phase θ _v from the reference phase θ _e , that is, a waveform advanced by π / 4.

  Note that the function of the carrier comparison unit 38 shown in FIG. 4 and the functions of the rotation speed calculation unit 42 and the advance phase calculation unit 44 shown in FIG. 7 can be realized by the processor 31 shown in FIG. The processor 31 is a processing unit that performs various calculations related to the PWM control and the advance angle control. The memory 34 stores a program read by the processor 31. The memory 34 is used as a work area when the processor 31 performs arithmetic processing. In addition, the processor 31 may be what is called a CPU (Central Processing Unit), a microprocessor, a microcomputer, a DSP (Digital Signal Processor), or the like. The memory 34 is generally a non-volatile or volatile semiconductor memory such as a RAM (Random Access Memory), a flash memory, an EPROM (Erasable Programmable ROM), and an EEPROM (Electrically EPROM).

  Here, the reason why PWM control is not often used for controlling the single-phase motor in the related art will be described. As a method of driving a single-phase motor, a method of switching an output voltage pulse in accordance with switching of a position sensor signal, as shown in Patent Document 2 described above, has generally been used. Here, as an example of a product for driving a single-phase motor, a vacuum cleaner or a hand dryer is taken as an example. In these products, it is necessary to rotate the single-phase motor at a high speed of 100,000 rotations or more. Requires a high-speed operation to control the sine wave. In a conventional environment, it was difficult to obtain a microcomputer capable of high-speed arithmetic processing at low cost, so that PWM control was not performed.

Next, how to give the voltage amplitude command V * will be described. FIG. 10 is a time chart showing a time change of the voltage amplitude command V *. In the present embodiment, an operation mode in which the voltage amplitude command V * changes stepwise according to the time t as shown in the figure. When specifically described, first, at the time of start giving first voltages V ₁ constant set in advance, during steady operation after acceleration, it gives a second voltage V ₂ of greater constant than the first voltage V1. Further, at the time of acceleration is changed from the first voltages V ₁ to the second voltage V _2, to raise the voltage amplitude command V * as acceleration rate set in advance is obtained. That is, in the present embodiment, the control is performed so that the voltage amplitude command V * is constant at the time of startup and during steady operation. At the time of startup, the time τ1 for applying the first voltage V1 can be set to an arbitrary time in consideration of the stabilization time of the control system.

  Next, the effect of the constant voltage amplitude command V * will be described. The following effects can be obtained by controlling the voltage amplitude command V * to be constant during the steady operation.

(1) Even when the load suddenly changes, a constant voltage command can be output based on the phase detected from the position sensor signal.
(2) Even if the rotational speed fluctuates, the voltage command is not affected, so that the output voltage can be kept stable.

  The above-described effect is effective in an application such as a vacuum cleaner in which the load varies according to the contact area between the suction port of the vacuum cleaner and the floor surface.

  When the rotational speed constant control performed by a general electric blower is performed, an overcurrent may flow through the motor. The reason why the overcurrent flows is that the current fluctuates rapidly because the rotational speed is kept constant during a load change. To explain in more detail, when performing the rotation speed constant control when transitioning from the “light load” state, that is, the “load torque small state”, to the “heavy load state”, that is, the “high load torque state”, This is because the motor output torque must be increased while maintaining the same rotational speed, and the amount of change in the motor current increases.

  On the other hand, in the control according to the present embodiment, as described above, control is performed to keep the voltage amplitude command V * constant during steady operation. Here, when the voltage amplitude command V * is constant, when the load becomes heavy, the voltage amplitude command V * is not changed, so that the motor torque is reduced by an increase in the load torque. With this control, a steep change in motor current and overcurrent can be prevented, so that an electric blower and an electric vacuum cleaner that rotate stably can be realized.

  In the case of an electric blower, the load torque increases with an increase in the number of rotations of the blade, which is a load of the motor, and also increases with an increase in the diameter of the air path. The diameter of the air path represents the size of the suction port when an electric vacuum cleaner is used as an example. When the diameter of the air path is large, when nothing is in contact with the suction port, a force for sucking the wind is required. Therefore, when the blades are rotating at the same rotation speed, the load torque is large. On the other hand, when the diameter of the air passage is small, the suction port contacts with something, and when the blade is rotating at the same rotation speed, the load torque is not required when the blade is rotating at the same rotation speed in a state where the suction port is closed. Become smaller.

Next, effects of the advance angle control will be described. First, if to increase the advance phase theta _v in accordance with an increase in rotational speed, it is possible to extend the rotational speed range. The advance phase theta _v when 0, the rotational speed is saturated at which the voltage applied to the motor and the motor induced voltage is balanced. In order to further increase the rotation speed, the advance angle θ _v is advanced and the magnetic flux generated in the stator due to the armature reaction is weakened to suppress the motor induced voltage and increase the rotation speed. Thus, by selecting the advanced angle phase theta _v according to the rotation speed, it is possible to obtain a wide speed range.

When the advance angle control according to the present embodiment is applied to a vacuum cleaner, the voltage command is kept constant regardless of the change in the closed state of the suction port, that is, regardless of the load torque, and the voltage command is adjusted according to the increase in the rotation speed. it suffices to increase the advance phase theta _v is a lead angle of the voltage command Te. With such control, stable driving can be performed in a wide rotation speed range.

  Next, a loss reducing method according to the present embodiment will be described. FIGS. 11 to 13 are diagrams showing paths of the motor current depending on the polarity of the inverter output voltage. When the polarity of the inverter output voltage is positive, as shown by the bold solid line (a) in FIG. 11, the current flows into the single-phase motor 12 through the channel portion of the MOSFET 51 that is the upper arm of the first phase, and flows downward to the second phase. The current flows out of the single-phase motor 12 through the channel of the MOSFET 54 serving as an arm. When the polarity of the inverter output voltage is negative, as shown by the bold broken line (b) in FIG. 11, the current flows into the single-phase motor 12 through the channel portion of the MOSFET 53 which is the upper arm of the second phase, and And flows out of the single-phase motor 12 through the channel portion of the MOSFET 52 which is the lower arm.

  Next, a current path when the inverter output voltage is zero, that is, when the single-phase inverter 11 outputs a zero voltage will be described. When the inverter output voltage becomes zero after the positive inverter output voltage is generated, no current flows from the power supply side as shown by the thick solid line (c) in FIG. A reflux mode in which current flows back and forth between. At this time, since the direction of the current flowing immediately before the single-phase motor 12 does not change, the current flowing out of the single-phase motor 12 is reduced by the channel portion of the MOSFET 54 which is the lower arm of the second phase and the lower portion of the first phase. The control returns to the single-phase motor 12 through the body diode portion of the MOSFET 52 as an arm. When the inverter output voltage becomes zero after the negative inverter output voltage is generated, the direction of the current flowing immediately before is reversed, and therefore, as shown by the thick broken line (d) in FIG. The direction of the current is reversed. More specifically, the current flowing from the single-phase motor 12 passes through the body diode portion of the MOSFET 51 that is the upper arm of the first phase and the channel portion of the MOSFET 53 that is the upper arm of the second phase. Return to

  As described above, in the freewheel mode in which the current flows between the single-phase motor 12 and the single-phase inverter 11, the current flows through the body diode in one of the first phase and the second phase. In general, it is known that conduction loss is smaller when a current flows through a channel of a MOSFET than when a current flows in a forward direction of a diode. Therefore, in the present embodiment, in the return mode in which the return current flows, the MOSFET on the side having the body diode is turned on in order to reduce the conduction current flowing through the body diode.

  In the reflux mode, the MOSFET 52 is controlled to be turned on at the timing when the reflux current shown by the thick solid line (c) in FIG. 12 flows. With this control, most of the return current flows on the channel side of the MOSFET 52 having a small resistance value, as shown by the thick solid line (e) in FIG. Thereby, semiconductor loss in MOSFET 52 is reduced. At the timing when the return current shown by the thick broken line (d) in FIG. 12 flows, the MOSFET 51 is controlled to be turned on. With this control, most of the return current flows on the channel side of the MOSFET 51 having a small resistance value, as shown by the thick broken line (f) in FIG. Thus, semiconductor loss in the MOSFET 51 is reduced.

  As described above, the loss of the switching element can be reduced by controlling the MOSFET having the body diode to be ON at the timing when the return current flows through the body diode. For this reason, the shape of the MOSFET is made to be a surface mount type so that heat can be dissipated on the substrate, and if necessary, part or all of the switching element is formed of a wide band gap semiconductor, so that the heat generated by the MOSFET is generated only on the substrate. To realize a structure for suppressing noise. If heat radiation is possible only with the substrate, no heat sink is required, which contributes to downsizing of the inverter and downsizing of the product.

  In addition to the above-described heat dissipation method, a further heat dissipation effect can be obtained by installing the substrate in the air passage. Here, the air path is a portion that generates an air flow like an electric blower or a passage of an air flow generated by the electric blower. By arranging the substrate in the air passage, heat generated by the electric blower can radiate heat from the semiconductor element on the substrate, so that heat generation of the semiconductor element can be significantly suppressed.

  Next, an application example of the motor driving device according to the present embodiment will be described. FIG. 14 is a diagram illustrating an example of a configuration of a vacuum cleaner as an application example of the motor drive device according to the present embodiment. 14, the vacuum cleaner 61 includes a battery 67 that is a DC power supply, an electric blower 64 driven by the above-described single-phase motor 12, and further includes a dust collecting chamber 65, a sensor 68, a suction port 63, and an extension pipe. 62 and an operation unit 66.

  The vacuum cleaner 61 drives the electric blower 64 using the battery 67 as a power supply, sucks in air from the suction opening 63, and sucks dust into the dust collecting chamber 65 via the extension pipe 62. In use, the user holds the operation unit 66 and operates the vacuum cleaner 61.

The vacuum cleaner 61 is a product in which the driving range of the motor rotation speed varies from 0 to 100,000 rpm or more. When such a motor drives a product that rotates at a high speed, the control method according to the present embodiment described above is suitable. And constant voltage amplitude command V *, by changing the advanced angle phase theta _v according to the rotational speed, while widening the rotational speed drive range from a low speed to a high speed rotation region, it is possible to correspond to abrupt load change. In addition, since the motor current is controlled to a sine wave by the PWM control, high-efficiency driving can be performed, so that a longer operation time can be expected.

  In addition, the reduction in the number of heat radiating components can contribute to downsizing and weight reduction. Further, since a current sensor for detecting a current is not required and a high-speed analog-to-digital converter is not required, it is possible to suppress the cost.

  FIG. 15 is a diagram illustrating an example of a configuration of a hand dryer as another application example of the motor drive device according to the present embodiment. The hand dryer 90 includes a casing 91, a hand detection sensor 92, a water receiving section 93, a drain container 94, a cover 96, a sensor 97, and an air inlet 98. Here, the sensor 97 is either a gyro sensor or a human sensor. The hand dryer 90 has an electric blower (not shown) in a casing 91. The hand dryer 90 has a structure in which water is blown off by blowing with an electric blower by inserting a hand into a hand insertion portion 99 above the water receiving portion 93, and water is stored from the water receiving portion 93 to the drain container 94. ing.

  The hand dryer 90 is also a product in which the driving range of the motor rotation speed varies from 0 to 100,000 rpm or more. Therefore, also in the hand dryer 90, the control method according to the present embodiment described above is suitable, and the same effect as that of the vacuum cleaner 61 can be obtained.

  FIG. 16 is a diagram provided for describing modulation control suitable for the motor drive device according to the present embodiment. The relationship between the number of rotations and the modulation rate is shown on the left side of FIG. Also, the right side of the figure shows the waveform of the inverter output voltage when the modulation factor is 1 or less and exceeds 1. Generally, the load torque of the rotating body increases as the number of rotations increases. For this reason, it is necessary to increase the motor output torque as the number of rotations increases. Generally, the motor output torque increases in proportion to the motor current, and an increase in the motor current requires an increase in the inverter output voltage. Therefore, it is possible to increase the number of rotations by increasing the modulation rate and increasing the inverter output voltage.

Next, rotation speed control in the present embodiment will be described. In the following description, an electric blower is assumed as a load, and the operating range of the electric blower is divided as follows.
(1) Low-speed rotation range (low-speed range): 0 to 70,000 rpm
(2) High speed rotation range (high rotation speed range): 100,000 rpm or more

  The area sandwiched between (1) and (2) is a gray zone, and may be included in the low-speed rotation range or the high-speed rotation range depending on the application.

  First, control in the low-speed rotation range will be described. In the low-speed rotation region, the PWM control is performed with the modulation factor set to 1 or less. By setting the modulation factor to 1 or less, the motor current can be controlled to a sine wave, and the efficiency of the motor can be improved. When operating at the same carrier frequency in the low-speed rotation range and the high-speed rotation range, the carrier frequency becomes the carrier frequency in accordance with the high-speed rotation range, so that the PWM pulse tends to increase more than necessary in the low-speed rotation range. . For this reason, a method of reducing the carrier frequency in the low-speed rotation range to reduce the switching loss may be used. Further, by changing the carrier frequency in synchronization with the rotation speed, control may be performed so that the pulse number does not change in accordance with the rotation speed.

  Next, control in the high-speed rotation range will be described. In the high-speed rotation range, the modulation factor is set to a value larger than 1. By making the modulation factor larger than 1, it is possible to increase the inverter output voltage and reduce the number of switching operations performed by the switching elements in the inverter, thereby suppressing an increase in switching loss. Here, when the modulation factor exceeds 1, the motor output voltage increases, but since the number of switching times decreases, there is a concern about current distortion. However, during high-speed rotation, the reactance component of the motor increases, and di / dt, which is the change component of the motor current, decreases. Therefore, the current distortion is smaller than in the low-speed rotation range, and the influence on the waveform distortion is small. Become. Therefore, in the high-speed rotation range, the modulation factor is set to a value larger than 1, and control is performed to reduce the number of switching pulses. With this control, an increase in switching loss can be suppressed, and high efficiency can be achieved.

  As described above, the boundary between the low-speed rotation region and the high-speed rotation region may be gray. For this reason, the first rotation speed that determines the boundary between the low-speed rotation region and the high-speed rotation region is set, and when the rotation speed of the motor or the load is equal to or less than the first rotation speed, the modulation factor is set to 1 or less. If the rotation speed exceeds the first rotation speed, control may be performed so as to set the modulation rate to exceed 1.

  As described above, in the present embodiment, the vacuum cleaner 61 and the hand dryer 90 have been described as application examples of the motor driving device. However, the present invention can be applied to general electric devices equipped with a motor. Electrical equipment equipped with motors include incinerators, crushers, dryers, dust collectors, printing machines, cleaning machines, confectionery machines, tea machines, woodworking machines, plastic extruders, cardboard machines, packaging machines, hot air generators, and objects. It is a device provided with an electric blower, such as a device for transportation, dust suction, general ventilation, or OA equipment.

  Further, the configuration shown in the above embodiment is an example of the content of the present invention, and can be combined with another known technology, and the configuration is not deviated from the gist of the present invention. Can be omitted or changed.

  Reference Signs List 1 motor drive system, 2 motor drive device, 10 power supply, 11 single-phase inverter, 12a rotor, 12 single-phase motor, 20 voltage sensor, 21 position sensor, 25 control unit, 30 analog / digital converter, 31 processor, 32 drive signal Generating section, 33 carrier generating section, 33a carrier frequency setting section, 34 memory, 38 carrier comparing section, 38a absolute value calculating section, 38b dividing section, 38c, 38d, 38f multiplying section, 38e adding section, 38g, 38h comparing section, 38i, 38j output reversing unit, 42 rotation speed calculation unit, 44 advance angle phase calculation unit, 51, 52, 53, 54 switching element, 61 vacuum cleaner, 62 extension pipe, 63 suction port body, 64 electric blower, 65 collection Dust chamber, 66 operation unit, 67 battery, 68 sensor, 90 hand dry Chromatography, 91 casing, 92 hand detecting sensor, 93 water receiving unit, 94 drain container 96 cover, 97 sensors, 98 inlet, 99 hand insertion portion.

Claims (12)

A motor driving device for driving a single-phase motor,
A single-phase inverter that includes a plurality of switching elements and applies an AC voltage to the single-phase motor,
By performing a PWM control on the plurality of switching elements based on a voltage command and a signal for detecting a rotational position of the single-phase motor, a current waveform flowing through the single-phase motor approaches a sine wave,
A motor drive device , wherein a modulation rate of the PWM control is constant when a load change of the motor occurs, and a lead angle of the voltage command is increased in accordance with an increase in a rotation speed of the single-phase motor. The plurality of switching elements are MOSFETs,
The motor drive device according to claim 1, wherein at a timing at which the return current flows through the single-phase inverter, a MOSFET through which the return current flows through the body diode is controlled to be on.   The motor drive device according to claim 1, wherein the PWM signal for performing the PWM control is generated by unipolar modulation. At the time of activation, a predetermined first voltage is applied to a voltage amplitude command that is the amplitude of the voltage command,
The motor drive device according to any one of claims 1 to 3, wherein a constant second voltage that is higher than the first voltage is applied during a steady operation after acceleration. Set the first rotation speed that determines the boundary between the low speed rotation region and the high speed rotation region,
When the rotation speed of the single-phase motor is equal to or less than the first rotation speed, the modulation rate of the PWM control is set to 1 or less, and when the rotation speed exceeds the first rotation speed, the modulation rate is set. Is set to a value exceeding 1. The motor drive device according to claim 1.   The motor drive device according to claim 1, wherein at least one of the plurality of switching elements is formed of a wide band gap semiconductor.   The motor driving device according to claim 6, wherein the wide band gap semiconductor is silicon carbide, gallium nitride, or diamond.   An electric blower comprising the motor drive device according to claim 1.   An electric vacuum cleaner comprising the electric blower according to claim 8.   The vacuum cleaner according to claim 9, wherein the board on which the plurality of switching elements are mounted is installed in a passage of an airflow generated by the electric blower.   A hand dryer comprising the electric blower according to claim 8.   The hand dryer according to claim 11, wherein the substrate on which the plurality of switching elements are mounted is installed in a passage of an airflow generated by the electric blower.

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