WO2023181181A1

WO2023181181A1 - Motor drive device, electric blower, electric vacuum cleaner, and hand dryer

Info

Publication number: WO2023181181A1
Application number: PCT/JP2022/013539
Authority: WO
Inventors: 裕次 ▲高▼山; 和徳畠山
Original assignee: 三菱電機株式会社
Priority date: 2022-03-23
Filing date: 2022-03-23
Publication date: 2023-09-28

Abstract

A motor drive device (2) comprises: an inverter (11) to which a battery voltage output from a battery (10) is applied; and a voltage detector (21) which detects a motor induced voltage induced in a single-phase motor (12). The inverter (11) converts the battery voltage into an AC voltage and applies the converted AC voltage to the single-phase motor (12). The inverter (11) is gated on during a first period to apply a first voltage to the single-phase motor (12) and is gated off during a second period after the application of the first voltage. The motor drive device (2) increases a conduction width, which represents the application period of the first voltage by the phase of an electrical angle, according to a decrease in the battery voltage, and when the lower limit value of the battery voltage is reached after the battery voltage further decreases, stops increasing the conduction width.

Description

Motor drive devices, electric blowers, vacuum cleaners and hand dryers

The present disclosure relates to a motor drive device that drives a single-phase motor, and an electric blower, a vacuum cleaner, and a hand dryer equipped with a single-phase motor driven by the motor drive device.

Conventionally, when starting a multiphase brushless motor without a position sensor, there is a method of applying a high-frequency voltage so that the motor rotates following the rotating magnetic field generated by an inverter. Patent Document 1 below describes a method for starting a three-phase brushless motor without a position sensor, in which the initial position of the rotor is set with one energization, and the rotational speed of the motor is increased based on information about the set initial position. , discloses a method of detecting the position of the rotor after the rotational speed has increased.

Japanese Unexamined Patent Publication No. 1-308192

As mentioned above, various starting methods have been proposed for polyphase motors. On the other hand, in the case of a single-phase motor, a rotating magnetic field cannot be generated by an inverter. For this reason, it is difficult to apply the starting method of a polyphase motor as is to a single-phase motor. Further, even if part of the method for starting a polyphase motor is applied to a single-phase motor, it is difficult to start the motor in the correct direction.

Furthermore, in some applied products of motor drive devices that drive single-phase motors, the input power source is a battery. When the input power source is a battery, when the battery voltage, which is the power supply voltage, decreases, the inverter input voltage also decreases, making it difficult to drive the single-phase motor at a desired rotation speed. For this reason, this type of applied product has a problem in that the lower limit of the power supply voltage at which operation is permitted is large and the operating time per power supply capacity is short.

The present disclosure has been made in view of the above, and an object of the present disclosure is to obtain a motor drive device that can lengthen the operating time per power supply capacity.

In order to solve the above-mentioned problems and achieve the objectives, a motor drive device according to the present disclosure is a motor drive device that drives a single-phase motor without a position sensor. The motor drive device includes an inverter to which a power supply voltage output from a DC power supply is applied, and a detector that detects a physical quantity correlated with a motor induced voltage induced in a single-phase motor. The inverter converts the power supply voltage into AC voltage and applies the converted AC voltage to the single-phase motor. Further, the inverter is gated on in a first period to apply a first voltage to the single-phase motor, and gated off in a second period after application of the first voltage. The motor drive device increases the energization width, which represents the application period of the first voltage in electrical angle phase, in response to a decrease in the power supply voltage, and when the power supply voltage further decreases and reaches the lower limit value of the power supply voltage. stops increasing the energization width.

According to the motor drive device according to the present disclosure, there is an effect that the operating time per power supply capacity can be made longer.

A block diagram showing the configuration of a motor drive system including a motor drive device according to Embodiment 1. Circuit diagram of the inverter shown in Figure 1 A circuit diagram showing a modification of the inverter shown in Figure 2 A block diagram showing a functional part that generates a pulse width modulation (PWM) signal among the functional parts of the control unit shown in FIG. 1. A block diagram showing an example of the carrier comparison section shown in FIG. 4 A diagram showing an example of waveforms of main parts when operated using the carrier comparison section shown in FIG. A block diagram showing another example of the carrier comparison section shown in FIG. 4 A diagram showing an example of waveforms of main parts when operated using the carrier comparison section shown in FIG. A block diagram showing the functional configuration for calculating the advance phase input to the carrier comparison section shown in FIG. 4 A diagram showing an example of an operation waveform used to explain the first control, which is the drive control at low speed in the first embodiment. A diagram showing an example of an operation waveform used to explain the second control, which is the drive control at high speed in the first embodiment. Diagram used to explain third control common to both low-speed and high-speed drive control in Embodiment 1 Configuration diagram of a vacuum cleaner according to Embodiment 2 Configuration diagram of a hand dryer according to Embodiment 2

Below, with reference to the accompanying drawings, a motor drive device, an electric blower, a vacuum cleaner, and a hand dryer according to embodiments of the present disclosure will be described in detail based on the drawings.

Embodiment 1.
FIG. 1 is a block diagram showing the configuration of a motor drive system 1 including a motor drive device 2 according to the first embodiment. The motor drive system 1 shown in FIG. 1 includes a single-phase motor 12, a motor drive device 2, a battery 10,

voltage detectors

20 and 21, and

current detectors

22 and 24. The battery 10 is a DC power source that supplies DC power to the motor drive device 2. The motor drive device 2 includes an inverter 11 , an analog-to-digital converter 30 , a control section 25 , and a drive signal generation section 32 . Inverter 11 and single-phase motor 12 are connected by two

connection wires

18a and 18b.

The motor drive system 1 configured as described above is a so-called position sensorless control drive system that does not use a position sensor signal to detect the rotational position of the rotor 12a. Further, the motor drive device 2 is a drive device that supplies AC power to the single-phase motor 12 and drives the single-phase motor 12 without a position sensor.

The voltage detector 20 is a detector that detects a power supply voltage V _dc , which is a DC voltage output from the battery 10 to the motor drive device 2 . Power supply voltage V _dc is the output voltage of battery 10 and is applied to inverter 11 .

The voltage detector 21 is a detector that detects the alternating current voltage _Vac generated between the connecting

lines

18a and 18b. The AC voltage V _ac is a voltage in which the motor applied voltage applied by the inverter 11 to the single-phase motor 12 and the motor induced voltage induced in the single-phase motor 12 are superimposed. When the inverter 11 stops operating and the single-phase motor 12 is rotating, a motor induced voltage is observed. Therefore, the detected value of the voltage detector 21 is a physical quantity correlated with the motor induced voltage. Therefore, in this paper, the detected value of the voltage detector 21 is sometimes described as "a first physical quantity correlated with the motor induced voltage." Furthermore, in this paper, a state in which the inverter 11 stops operating and does not output voltage is referred to as "gate off." Further, the voltage output by the inverter 11 is appropriately referred to as "inverter output voltage."

Current detector 22 is a detector that detects motor current I _m . Motor current I _m is an alternating current that flows in and out between inverter 11 and single-phase motor 12 . The motor current I _m is equal to an alternating current flowing through a winding (not shown in FIG. 1 ) wound around the stator 12 b of the single-phase motor 12 . Examples of the current detector 22 include a current transformer (CT) or a current detector that detects current using a shunt resistor.

The current detector 24 is a detector that detects the power supply current I _dc . Power supply current I _dc is a direct current flowing between battery 10 and inverter 11 . The current detector 24 generally has a configuration using a shunt resistor as shown in the figure. The detected value of the power supply current I _dc flowing through the current detector 24 is converted into a voltage value and input to the analog-to-digital converter 30 . Note that in this paper, the detected value of the current detector 24 is appropriately referred to as a "shunt voltage." Further, the shunt voltage, which is the detected value of the power supply current I _dc , has a correlation with the motor current I _m . That is, as the motor current I _m increases, the shunt voltage also increases, and as the motor current I _m decreases, the shunt voltage also decreases. Therefore, in this paper, the shunt voltage is sometimes described as "a second physical quantity correlated with the motor current I _m ."

The single-phase motor 12 is used as a rotating electrical machine that rotates an electric blower (not shown). Electric blowers are installed in devices such as vacuum cleaners and hand dryers.

The inverter 11 is a power converter that converts the power supply voltage V _dc output from the battery 10 into an alternating current voltage. The inverter 11 supplies AC power to the single-phase motor 12 by applying the converted AC voltage to the single-phase motor 12 .

The analog-to-digital converter 30 is a signal converter that converts analog data into digital data. The analog-to-digital converter 30 converts the detected value of the power supply voltage V _dc detected by the voltage detector 20 and the detected value of the AC voltage V _ac detected by the voltage detector 21 into digital data, and sends the digital data to the control unit 25 . Output. Further, the analog-to-digital converter 30 converts the detected value of the motor current I _m detected by the current detector 22 and the detected value of the power supply current I _dc detected by the current detector 24 into digital data, and converts the detected value of the motor current I m detected by the current detector 24 into digital data. Output to.

The control unit 25 generates PWM signals Q1, Q2, Q3, Q4 (hereinafter appropriately referred to as "Q1 to Q4") based on the digital output value 30a converted by the analog-to-digital converter 30 and the voltage amplitude command V*. ) is generated. The voltage amplitude command V* will be described later.

The drive signal generation unit 32 generates drive signals S1, S2, S3, and S4 (hereinafter referred to as “S1 to S4") is generated.

The control section 25 includes a processor 31, a carrier generation section 33, and a memory 34. Processor 31 generates PWM signals Q1 to Q4 for performing PWM control. The processor 31 is a processing unit that performs various calculations related to PWM control and advance angle control. Examples of the processor 31 include a CPU (Central Processing Unit), a microprocessor, a microcomputer, a DSP (Digital Signal Processor), or a system LSI (Large Scale Integration).

A program read by the processor 31 is stored in the memory 34. The memory 34 is also used as a work area when the processor 31 performs arithmetic processing. The memory 34 is generally a nonvolatile or volatile semiconductor memory such as RAM (Random Access Memory), flash memory, EPROM (Erasable Programmable ROM), or EEPROM (registered trademark) (Electrically EPROM). It is true. The details of the configuration of the carrier generation section 33 will be described later.

FIG. 2 is a circuit diagram of the inverter 11 shown in FIG. 1. The inverter 11 includes a plurality of switching

elements

51, 52, 53, and 54 (hereinafter appropriately referred to as "51 to 54") connected in a bridge.

The switching

elements

51 and 52 constitute a leg 5A which is a first leg. The leg 5A is a series circuit in which a switching element 51, which is a first switching element, and a switching element 52, which is a second switching element, are connected in series.

The switching

elements

53 and 54 constitute leg 5B, which is the second leg. The leg 5B is a series circuit in which a switching element 53, which is a third switching element, and a switching element 54, which is a fourth switching element, are connected in series.

The

legs

5A and 5B are connected in parallel to each other between the DC bus 16a on the high potential side and the DC bus 16b on the low potential side. Thereby,

legs

5A and 5B are connected to both ends of battery 10 in parallel.

The switching

elements

51 and 53 are located on the high potential side, and the

switching elements

52 and 54 are located on the low potential side. Generally, in an inverter circuit, the high potential side is called the "upper arm" and the low potential side is called the "lower arm." Therefore, the switching element 51 of the leg 5A may be referred to as the "first switching element of the upper arm", and the switching element 52 of the leg 5A may be referred to as the "second switching element of the lower arm". Similarly, the switching element 53 of the leg 5B may be referred to as the "third switching element of the upper arm", and the switching element 54 of the leg 5B may be referred to as the "fourth switching element of the lower arm".

A connection end 6A between the switching element 51 and the switching element 52 and a connection end 6B between the switching element 53 and the switching element 54 constitute an AC end in the bridge circuit. A single-phase motor 12 is connected between the connection end 6A and the connection end 6B.

A MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor), which is a metal-oxide-semiconductor field-effect transistor, is used for each of the switching elements 51 to 54. MOSFET is an example of a FET (Field-Effect Transistor).

A body diode 51a connected in parallel between the drain and source of the switching element 51 is formed in the switching element 51. A body diode 52a connected in parallel between the drain and source of the switching element 52 is formed in the switching element 52. A body diode 53a connected in parallel between the drain and source of the switching element 53 is formed in the switching element 53. A body diode 54a connected in parallel between the drain and source of the switching element 54 is formed in the switching element 54. Each of the plurality of

body diodes

51a, 52a, 53a, and 54a is a parasitic diode formed inside the MOSFET, and is used as a freewheeling diode. Note that a separate free wheel diode may be connected. Furthermore, an insulated gate bipolar transistor (IGBT) may be used instead of the MOSFET.

The switching elements 51 to 54 are not limited to MOSFETs formed of silicon-based materials, but may be MOSFETs formed of wide band gap (WBG) semiconductors such as silicon carbide, gallium nitride, gallium oxide, or diamond.

In general, WBG semiconductors have higher voltage resistance and heat resistance than silicon semiconductors. Therefore, by using a WBG semiconductor for at least one of the plurality of switching elements 51 to 54, the voltage resistance and allowable current density of the switching element are increased, and the semiconductor module incorporating the switching element can be miniaturized. Furthermore, WBG semiconductors also have high heat resistance. Therefore, it is possible to downsize the heat dissipation section for dissipating the heat generated in the semiconductor module. Furthermore, it is possible to simplify the heat dissipation structure for dissipating heat generated in the semiconductor module.

Further, FIG. 3 is a circuit diagram showing a modification of the inverter 11 shown in FIG. 2. The inverter 11A shown in FIG. 3 has the configuration of the inverter 11 shown in FIG. 2, but further includes

shunt resistors

55a and 55b. Shunt resistor 55a is a detector for detecting the current flowing through leg 5A, and shunt resistor 55b is a detector for detecting the current flowing through leg 5B. As shown in FIG. 3, the shunt resistor 55a is connected between the low potential side terminal of the switching element 52 and the DC bus 16b, and the shunt resistor 55b is connected between the low potential side terminal of the switching element 54 and the DC bus 16b. When the inverter 11A including the

shunt resistors

55a and 55b is used, the current detector 22 shown in FIG. 1 can be omitted. In this configuration, the detected values of the

shunt resistors

55a and 55b are sent to the processor 31 via the analog-to-digital converter 30. The processor 31 implements activation control, which will be described later, based on the detected values of the

shunt resistors

55a and 55b.

Note that the shunt resistor 55a is not limited to the one shown in FIG. 3 as long as it can detect the current flowing through the leg 5A. The shunt resistor 55a is connected between the DC bus 16a and the high potential terminal of the switching element 51, between the low potential terminal of the switching element 51 and the connection end 6A, or between the connection end 6A and the high potential of the switching element 52. It may also be placed between the side terminals. Similarly, the shunt resistor 55b is connected between the DC bus 16a and the high potential side terminal of the switching element 53, between the low potential side terminal of the switching element 53 and the connection end 6B, or between the connection end 6B and the switching element 54. It may also be placed between the terminal on the high potential side of the terminal. Further, instead of the

shunt resistors

55a and 55b, the on-resistance of a MOFFET may be used, and the current may be detected by the voltage generated across the on-resistance.

FIG. 4 is a block diagram showing a functional part of the control unit 25 shown in FIG. 1 that generates a PWM signal. In FIG. 4, the carrier comparison section 38 is illustrated together with the carrier generation section 33 shown in FIG. 1.

In FIG. 4, the carrier comparator 38 receives an advanced phase θ _v that is subjected to advance angle control and a reference phase θ _e that are used when generating a voltage command V _m to be described later. The reference phase θ _e is a phase obtained by converting a rotor mechanical angle, which is an angle from the reference position of the rotor 12a, into an electrical angle. Note that, as described above, the motor drive device 2 according to the first embodiment has a so-called position sensorless configuration that does not use a position sensor signal from a position sensor. Therefore, the rotor mechanical angle and the reference phase θ _e are estimated by calculation. Moreover, the "advanced angle phase" referred to here is the "advanced angle" which is the angle of advance of the voltage command V _m expressed in phase. Furthermore, the "advance angle" referred to here is the phase difference between the motor applied voltage applied to the windings of the stator 12b and the motor induced voltage induced in the windings of the stator 12b. Note that when the motor applied voltage leads the motor induced voltage, the "advance angle" takes a positive value.

In addition to the advance phase θ _v and the reference phase θ _e , the carrier comparator 38 also contains the carrier generated by the carrier generator 33, the power supply voltage V _dc , and a voltage that is the amplitude value of the voltage command V _m . An amplitude command V* is input. The carrier comparator 38 generates PWM signals Q1 to Q4 based on the carrier, advance phase θ _v , reference phase θ _e , power supply voltage V _dc , and voltage amplitude command V*.

FIG. 5 is a block diagram showing an example of the carrier comparison unit 38 shown in FIG. 4. FIG. 5 shows detailed configurations of the carrier comparison section 38A and the carrier generation section 33.

In FIG. 5, the carrier generation unit 33 is set with a carrier frequency f _C [Hz] that is the frequency of the carrier. At the tip of the arrow of carrier frequency _fC , a triangular wave carrier that fluctuates between "0" and "1" is shown as an example of a carrier waveform. PWM control of the inverter 11 includes synchronous PWM control and asynchronous PWM control. In the case of synchronous PWM control, it is necessary to synchronize the carrier with the advance phase θ _v . On the other hand, in the case of asynchronous PWM control, there is no need to synchronize the carrier with the advance phase θ _v .

As shown in FIG. 5, the carrier comparison section 38A includes an absolute value calculation section 38a, a division section 38b, a multiplication section 38c, a multiplication section 38d, a multiplication section 38f, an addition section 38e, a comparison section 38g, a comparison section 38h, and an output inversion section. 38i and an output inverting section 38j.

The absolute value calculation unit 38a calculates the absolute value |V*| of the voltage amplitude command V*. The dividing unit 38b divides the absolute value |V*| by the power supply voltage V _dc detected by the voltage detector 20. In the configuration of FIG. 5, the output of the divider 38b becomes the modulation rate. The battery voltage, which is the output voltage of the battery 10, fluctuates as current continues to flow. On the other hand, by dividing the absolute value |V*| by the power supply voltage V _dc , the value of the modulation factor can be adjusted to prevent the voltage applied to the motor from decreasing due to a decrease in battery voltage.

The multiplier 38c calculates the sine value of "θ _e +θ _v ", which is the addition of the advance phase θ _v to the reference phase θ _e . The calculated sine value of “θ _e +θ _v ” is multiplied by the modulation factor that is the output of the divider 38b. In the multiplier 38d, the voltage command _Vm , which is the output of the multiplier 38c, is multiplied by "1/2". The adder 38e adds "1/2" to the output of the multiplier 38d. The multiplier 38f multiplies the output of the adder 38e by "-1". The output of the adder 38e is input to the comparator 38g as a positive voltage command V _m1 for driving the two switching

elements

51 and 53 of the upper arm among the plurality of switching elements 51 to 54, and is input to the comparator 38g as a positive voltage command V m1 for driving the two switching

elements

51 and 53 of the upper arm among the plurality of switching elements 51 to 54. The output is input to the comparator 38h as a negative side voltage command V _m2 for driving the two switching

elements

52 and 54 of the lower arm.

The comparison unit 38g compares the positive side voltage command V _m1 and the amplitude of the carrier. The output of the output inverter 38i, which inverts the output of the comparator 38g, becomes the PWM signal Q1 to the switching element 51, and the output of the comparator 38g becomes the PWM signal Q2 to the switching element 52. Similarly, the comparison unit 38h compares the negative side voltage command V _m2 and the amplitude of the carrier. The output of the output inverting section 38j, which inverts the output of the comparing section 38h, becomes the PWM signal Q3 to the switching element 53, and the output of the comparing section 38h becomes the PWM signal Q4 to the switching element 54. The output inversion section 38i prevents the switching element 51 and the switching element 52 from being turned on at the same time, and the output inversion section 38j prevents the switching element 53 and the switching element 54 from being turned on at the same time.

FIG. 6 is a diagram showing an example of waveforms of main parts when operating using the carrier comparator 38A shown in FIG. 5. FIG. 6 shows the waveform of the positive voltage command V _m1 output from the adder 38e, the waveform of the negative voltage command V _m2 output from the multiplier 38f, the waveforms of the PWM signals Q1 to Q4, and the inverter output. A voltage waveform is shown.

The PWM signal Q1 becomes "Low" when the positive side voltage command V _m1 is larger than the carrier, and becomes "High" when the positive side voltage command V _m1 is smaller than the carrier. PWM signal Q2 is an inverted signal of PWM signal Q1. The PWM signal Q3 becomes "Low" when the negative side voltage command V _m2 is larger than the carrier, and becomes "High" when the negative side voltage command V _m2 is smaller than the carrier. PWM signal Q4 is an inverted signal of PWM signal Q3. In this way, the circuit shown in FIG. 5 is configured with "Low Active", but even if it is configured with "High Active" where each signal has the opposite value. good.

As shown in FIG. 6, the waveform of the inverter output voltage shows a voltage pulse due to the voltage difference between the PWM signal Q1 and the PWM signal Q4, and a voltage pulse due to the voltage difference between the PWM signal Q3 and the PWM signal Q2. These voltage pulses are applied to the single-phase motor 12 as motor applied voltage.

Bipolar modulation and unipolar modulation are known as modulation methods used to generate the PWM signals Q1 to Q4. Bipolar modulation is a modulation method that outputs a voltage pulse that changes in positive or negative potential every cycle T of the voltage command _Vm . Unipolar modulation is a modulation method that outputs a voltage pulse that changes in three potentials every cycle T of the voltage command _Vm , that is, a voltage pulse that changes in positive potential, negative potential, and zero potential. The waveform shown in FIG. 6 is due to unipolar modulation. In the motor drive device 2 according to the first embodiment, any modulation method may be used. Note that in applications where it is necessary to control the motor current waveform to a more sinusoidal waveform, it is preferable to employ unipolar modulation, which has a lower harmonic content, than bipolar modulation.

Moreover, the waveform shown in FIG. 6 shows the switching of four switching

elements

51 and 52 forming leg 5A and switching

elements

53 and 54 forming leg 5B during a period of half cycle T/2 of voltage command _Vm . This is achieved by a method of switching elements. This method is called "both-side PWM" because the switching operation is performed using both the positive side voltage command V _m1 and the negative side voltage command V _m2 . On the other hand, in one half period T/2 of one period T of the voltage command V _m , the switching operations of the switching

elements

51 and 52 are stopped, and in the other half period T/2 of one period T of the voltage command V _m . There is also a method in which the switching operations of the switching

elements

53 and 54 are stopped during the period T/2. This method is called "one-sided PWM." Hereinafter, "one-sided PWM" will be explained. In the following description, the operation mode in which the device operates with PWM on both sides is referred to as the “both-side PWM mode”, and the operation mode in which the device operates with PWM on one side is referred to as the “one-side PWM mode”. Further, a PWM signal based on "both-side PWM" may be referred to as a "both-side PWM signal", and a PWM signal based on "one-sided PWM" may be called a "one-sided PWM signal".

FIG. 7 is a block diagram showing another example of the carrier comparison section 38 shown in FIG. 4. FIG. 7 shows an example of a one-sided PWM signal generation circuit, and specifically shows detailed configurations of the carrier comparison section 38B and the carrier generation section 33. Note that the configuration of the carrier generation section 33 shown in FIG. 7 is the same or equivalent to that shown in FIG. 5. Furthermore, in the configuration of the carrier comparison section 38B shown in FIG. 7, the same or equivalent components as the carrier comparison section 38A shown in FIG. 5 are denoted by the same reference numerals.

As shown in FIG. 7, the carrier comparison section 38B includes an absolute value calculation section 38a, a division section 38b, a multiplication section 38c, a multiplication section 38k, an addition section 38m, an addition section 38n, a comparison section 38g, a comparison section 38h, and an output inversion section. It has a section 38i and an output inverting section 38j.

The absolute value calculation unit 38a calculates the absolute value |V*| of the voltage amplitude command V*. The dividing unit 38b divides the absolute value |V*| by the power supply voltage V _dc detected by the voltage detector 20. In the configuration of FIG. 7 as well, the output of the divider 38b becomes the modulation rate.

The multiplier 38c calculates the sine value of "θ _e +θ _v ", which is the addition of the advance phase θ _v to the reference phase θ _e . The calculated sine value of “θ _e +θ _v ” is multiplied by the modulation factor that is the output of the divider 38b. The multiplier 38k multiplies the voltage command V _m , which is the output of the multiplier 38c, by "-1". The adder 38m adds "1" to the voltage command _Vm , which is the output of the multiplier 38c. In the addition section 38n, "1" is added to the output of the multiplication section 38k, that is, the inverted output of the voltage command _Vm . The output of the adder 38m is input to the comparator 38g as a first voltage command V _m3 for driving the two switching

elements

51 and 53 of the upper arm among the plurality of switching elements 51 to 54. The output of the adder 38n is input to the comparator 38h as a second voltage command V _m4 for driving the two switching

elements

52 and 54 of the lower arm.

The comparison unit 38g compares the first voltage command V _m3 and the amplitude of the carrier. The output of the output inverter 38i, which inverts the output of the comparator 38g, becomes the PWM signal Q1 to the switching element 51, and the output of the comparator 38g becomes the PWM signal Q2 to the switching element 52. Similarly, the comparison unit 38h compares the second voltage command V _m4 and the amplitude of the carrier. The output of the output inverting section 38j, which inverts the output of the comparing section 38h, becomes the PWM signal Q3 to the switching element 53, and the output of the comparing section 38h becomes the PWM signal Q4 to the switching element 54. The output inversion section 38i prevents the switching element 51 and the switching element 52 from being turned on at the same time, and the output inversion section 38j prevents the switching element 53 and the switching element 54 from being turned on at the same time.

FIG. 8 is a diagram showing an example of waveforms of main parts when operating using the carrier comparator 38B shown in FIG. 7. FIG. 8 shows the waveform of the first voltage command V _m3 output from the adder 38m, the waveform of the second voltage command V _m4 output from the adder 38n, the waveforms of the PWM signals Q1 to Q4, and the inverter output. A voltage waveform is shown. In addition, in FIG. 8, for convenience, the waveform part of the first voltage command V _m3 whose amplitude value is larger than the peak value of the carrier, and the waveform part of the second voltage command V _m4 whose amplitude value is larger than the peak value of the carrier are shown. The waveform portion is represented by a flat straight line.

The PWM signal Q1 becomes "Low" when the first voltage command V _m3 is larger than the carrier, and becomes "High" when the first voltage command V _m3 is smaller than the carrier. PWM signal Q2 is an inverted signal of PWM signal Q1. The PWM signal Q3 becomes "Low" when the second voltage command V _m4 is larger than the carrier, and becomes "High" when the second voltage command V _m4 is smaller than the carrier. PWM signal Q4 is an inverted signal of PWM signal Q3. In this way, the circuit shown in FIG. 7 is configured with "Low Active", but even if it is configured with "High Active" where each signal has the opposite value. good.

As shown in FIG. 8, the waveform of the inverter output voltage shows a voltage pulse due to a voltage difference between PWM signal Q1 and PWM signal Q4, and a voltage pulse due to a voltage difference between PWM signal Q3 and PWM signal Q2. These voltage pulses are applied to the single-phase motor 12 as motor applied voltage.

In the waveform shown in FIG. 8, the switching operations of the switching

elements

51 and 52 are stopped in one half cycle T/2 of one cycle _{T of the voltage command V m} _, and During the other half cycle T/2, the switching operations of the switching

elements

53 and 54 are at rest.

Further, in the waveform shown in FIG. 8, the switching element 52 is controlled to be always on in one half cycle T/2 of one cycle _T of the voltage command V _m , and During the other half period T/2 of the period T, the switching element 54 is controlled to be always on. Note that FIG. 8 is an example, and in one half cycle T/2, the switching element 51 is controlled to be always on, and in the other half cycle T/2, the switching element 53 is always on. There may also be cases where it is controlled as follows. That is, the waveform shown in FIG. 8 has a feature that at least one of the switching elements 51 to 54 is controlled to be in the on state during half period T/2 of the voltage command V _m .

Further, in FIG. 8, the waveform of the inverter output voltage is unipolar modulated, changing by three potentials every cycle T of the voltage command _Vm . As mentioned above, bipolar modulation may be used instead of unipolar modulation, but in applications where the motor current waveform needs to be controlled more sinusoidally, it is preferable to employ unipolar modulation.

FIG. 9 is a block diagram showing a functional configuration for calculating the advance phase θ _v input to the carrier comparator 38 shown in FIG. 4. The function of calculating the advance phase θ _v can be realized by the rotational speed calculation section 42 and the advance phase calculation section 44, as shown in FIG. The rotation speed calculation unit 42 calculates the rotation speed ω of the single-phase motor 12 based on the detected value of the motor current I _m detected by the current detector 22. Further, the rotational speed calculation unit 42 calculates the reference phase θ _e based on the detected value of the motor current I _m . As described above, the reference phase θ _e is the phase obtained by converting the rotor mechanical angle, which is the angle from the reference position of the rotor 12a, into an electrical angle. The rotor mechanical angle is a calculated value calculated inside the rotational speed calculating section 42.

The advance phase calculation unit 44 calculates the advance phase θ _v based on the rotational speed ω, the reference phase θ _e , and the motor induced voltage. The motor induced voltage can be obtained from the detected value of the alternating current voltage V _ac . As described above, the detected value of the AC voltage V _ac includes the motor applied voltage that the inverter 11 applies to the single-phase motor 12 and the motor induced voltage induced by the single-phase motor 12 . Among these voltages, the motor induced voltage can be detected during the gate-off period when the inverter 11 is not outputting any voltage.

Next, the first point of drive control of the motor drive device 2 according to the first embodiment will be explained with reference to FIGS. 10 and 11. FIG. 10 is a diagram showing an example of an operation waveform used to explain the first control, which is the drive control at low speed in the first embodiment. FIG. 11 is a diagram showing an example of an operation waveform used to explain the second control, which is the drive control at high speed in the first embodiment. Note that "low speed" or "high speed" here refers to the relative relationship between the two, and the first control shown in FIG. 10 and the second control shown in FIG. Switch at the rotation speed. That is, when the preset rotational speed is set as the "first rotational speed" and the rotational speed of the single-phase motor 12 is less than the first rotational speed, the single-phase motor is 12. When the rotational speed of the single-phase motor 12 is equal to or higher than the first rotational speed, the single-phase motor 12 is driven under the second control shown in FIG.

The waveform of the motor induced voltage is shown in the upper part of FIG. In the middle part of FIG. 10, the waveform of the motor applied voltage and the waveform of the motor induced voltage are shown. The lower part of FIG. 10 shows changes in the electrical angle phase in which the phase of the motor induced voltage is expressed in electrical angle. In the middle part of FIG. 10, the gate-on period during which the inverter 11 is gate-on is shown by a coarse hatching pattern, and the gate-off period during which the inverter 11 is gate-off is shown by a fine hatching pattern. The gate-on period T1 is a period in which the polarity of the voltage applied to the motor is positive, and the gate-on period T2 is a period in which the polarity of the voltage applied to the motor is negative. A gate-off period T3 exists between the gate-on period T1 and the gate-on period T2.

Further, T4 represents a period of 1/2 of the rotation period of the single-phase motor 12, that is, a rotation half period. The electrical angle phase of 0 to 180 [deg] corresponds to the rotation half period T4. There is a relationship between T4=T1+T3 and T4=T2+T3 between the rotation half period T4, the gate-on periods T1 and T2, and the gate-off period T3.

Note that in this paper, the voltage applied to the motor during the gate-on periods T1 and T2 may be referred to as a "first voltage." Furthermore, in this paper, the gate-on periods T1 and T2 may be referred to as a "first period", and the gate-off period T3 may be referred to as a "second period". By this definition, the first and second time periods occur alternately, and the first and second time periods are repeated in this order.

Furthermore, as described above, the polarity of the first voltage is reversed between the gate-on period T1 and the gate-on period T2. The inverter 11 switches the polarity of the first voltage every time the first period arrives. By switching the polarity of the first voltage, the single-phase motor 12 can continue rotating in the intended rotation direction. Note that although FIG. 10 illustrates a case where the first voltage is a one-pulse voltage, the present invention is not limited to this. The first voltage may be a voltage of a plurality of PWM-controlled pulse trains.

As described above, a positive polarity voltage is applied during the gate-on period T1. The gate-on period T1 starts from a zero cross point where the polarity of the motor induced voltage switches from negative to positive. Further, during the gate-on period T2, a negative polarity voltage is applied. The gate-on period T2 starts from a zero cross point where the polarity of the motor induced voltage switches from positive to negative.

Additionally, in this paper, the phase in which the polarity of the voltage applied to the motor switches between positive and negative is referred to as the "polarity switching phase." The polarity switching phase is a point where the electrical angle phase changes from 180 [deg] to 0 [deg], and is indicated by symbol A in FIG. 10. In the first control at low speed, point A, which indicates the polarity switching phase, coincides with the zero cross point.

In the first control, switching of the polarity of the first voltage is performed based on the rotational speed of the single-phase motor 12 and the motor induced voltage. During the gate-off period T3, since the inverter 11 is gate-off, the voltage detector 21 can detect the motor induced voltage. Therefore, it is also possible to detect the zero cross point of the motor induced voltage. Note that the zero cross point is a phase obtained by converting the mechanical angle of the rotor into an electrical angle, and it is also possible to use the reference phase θ _e determined by calculation.

When the rotational speed of the single-phase motor 12 is less than the first rotational speed, the zero-crossing point of the motor induced voltage is set as the polarity switching point of the first voltage. That is, when the rotation speed is less than the first rotation speed, the threshold value for switching the polarity of the first voltage is set to a zero value. Therefore, the gate-on period T1 or the gate-on period T2 starts at the zero-crossing point of the motor induced voltage. Then, by repeating the gate-on periods T1 and T2, rotational torque is applied to the single-phase motor 12, and the single-phase motor 12 rotates with acceleration.

The length of the gate-on periods T1 and T2 and the amplitude of the voltage applied to the motor can be determined based on the duty ratio, modulation rate, and rotation speed. The duty ratio is the ratio of the gate-on periods T1 and T2 to the rotation half period T4.

Note that instead of directly detecting the motor induced voltage with the voltage detector 21, the motor induced voltage may be calculated based on the detected value of the voltage detector 20 or the detected value of the current detector 24. Note that when the detected value of the voltage detector 20 is used, a control means for zeroing the output voltage of the battery 10 or a mechanism for disconnecting the electrical connection between the battery 10 and the inverter 11 is required.

Next, the second control will be explained with reference to FIG. 11. Similar to FIG. 10, the upper part of FIG. 11 shows the waveform of the motor induced voltage, the middle part shows the waveform of the motor applied voltage and the waveform of the motor induced voltage, and the lower part shows the waveform of the motor induced voltage. The change in electrical angle phase, where the phase is expressed in electrical angle, is shown. The hatching patterns attached to the gate-on period and the gate-off period are the same as in FIG. 10.

In the second control, switching of the polarity of the first voltage is performed based on the rotational speed of the single-phase motor 12 and the motor induced voltage. During the gate-off period τ3, since the inverter 11 is gate-off, the motor induced voltage can be detected by the voltage detector 21.

During the gate-on period τ1, a first voltage of positive polarity is applied. The gate-on period τ1 starts when the absolute value of the amplitude of the motor induced voltage reaches ΔV. Further, during the gate-on period τ2, a first voltage of negative polarity is applied. The gate-on period τ2 starts when the absolute value of the amplitude of the motor induced voltage reaches ΔV. That is, in the second control, the value of ΔV to be compared with the absolute value of the amplitude of the motor induced voltage is set as the threshold value. As shown in FIG. 11, the threshold value ΔV is a positive value. The inverter 11 inverts the polarity of the voltage applied to the single-phase motor 12 every time the absolute value of the amplitude of the motor induced voltage reaches the threshold value ΔV. Therefore, in the second control at high speed, the polarity of the motor applied voltage is switched between positive and negative at point B where the absolute value of the amplitude of the motor induced voltage becomes the threshold value ΔV.

Note that in FIG. 11, as in FIG. 10, the gate-on periods τ1 and τ2 may be referred to as "first periods", and the gate-off period τ3 may be referred to as a "second period". Moreover, the relationship between the rotation half period τ4, the gate-on periods τ1 and τ2, and the gate-off period τ3 is τ4=τ1+τ3 and τ4=τ2+τ3. Furthermore, the lengths of the gate-on periods τ1 and τ2 and the amplitude of the voltage applied to the motor can be determined based on the duty ratio, modulation rate, and rotation speed.

Furthermore, similarly to FIG. 10, the polarity of the first voltage is reversed between the gate-on period τ1 and the gate-on period τ2. The inverter 11 switches the polarity of the first voltage every time the first period arrives. By switching the polarity of the first voltage, the single-phase motor 12 can continue rotating in the intended rotation direction. Note that although FIG. 11 illustrates a case where the first voltage is a one-pulse voltage, the present invention is not limited to this. The first voltage may be a voltage of a plurality of PWM-controlled pulse trains.

Next, the significance of the duty ratios T1/T4 and T2/T4 in the first control, the duty ratios τ1/τ4 and τ2/τ4 in the second control, and the threshold value ΔV in the second control of the first embodiment will be explained. do.

First, the duty ratios T1/T4, τ1/τ4, T2/T4, τ2/τ4 contribute to the motor applied voltage, and the threshold value ΔV contributes to the advance phase θ _v , which is the phase difference of the motor applied voltage with respect to the motor induced voltage. . During acceleration, the reactance component (ωL) is smaller at low rotational speeds than at high speeds. For this reason, the motor current flowing through the single-phase motor 12 has a smaller phase lag in the motor applied voltage with respect to the motor current at low speeds than at high speeds. A small phase lag means a large power factor. If the power factor is large, it becomes possible to apply effective motor torque to the single-phase motor 12.

Furthermore, at high speeds, the reactance component (ωL) becomes large. At this time, the phase delay of the motor applied voltage with respect to the motor current becomes large, but by increasing the advance phase θ _v , it is possible to suppress the power factor from becoming small. Thereby, the acceleration torque applied to the single-phase motor 12 can be efficiently obtained, and the electric power supplied to the single-phase motor 12 can be effectively utilized. Furthermore, the motor induced voltage generated in the single-phase motor 12 increases as the rotational speed increases. When the motor induced voltage is large, overcurrent can be suppressed even if the inverter output voltage is increased. Therefore, by increasing the inverter output voltage in accordance with the increase in rotational speed, it is possible to reduce the acceleration time while suppressing overcurrent.

Next, the second key point of drive control of the motor drive device 2 according to the first embodiment will be explained with reference to FIG. 12. FIG. 12 is a diagram used to explain the third control common to both low-speed and high-speed drive control in the first embodiment.

The horizontal axis in FIG. 12 represents the power supply voltage. In addition, in FIG. 12, the upper part shows a waveform representing the change in rotational speed with respect to the power supply voltage, the middle part shows a waveform representing the change in polarity switching phase with respect to the power supply voltage, and the lower part shows a waveform representing the change in the rotation speed with respect to the power supply voltage. A waveform representing a change in energization width is shown. As described above, the polarity switching phase is a phase in which the polarity of the voltage applied to the motor is switched between positive and negative. Further, the energization width represents the application period of the first voltage in terms of electrical angle phase.

As explained in the [Problems to be Solved by the Invention] section, in the prior art, the lower limit of the battery voltage that is permissible for operation is set large in consideration of the battery voltage, which is the power supply voltage. The problem was that the average driving time was short. In response to this problem, in the first embodiment, the following control is performed.

First, control is performed to increase the energization width in response to a decrease in the power supply voltage V _dc . This control is performed until the power supply voltage V _dc reaches a preset lower limit value V1. That is, when the power supply voltage V _dc decreases and reaches the lower limit value V1, the increase in the energization width is stopped. The lower limit value V1 is set based on the energization width limit value W1. Further, the limit value W1 of the energization width is set based on the upper limit value W2 of the energization width. Specifically, the limit value W1 is set so as not to exceed the upper limit value W2, that is, to be less than or equal to the upper limit value W2.

Furthermore, in the lower part of FIG. 12, the right side of the vertical axis shows a range in which the induced voltage detection period cannot be secured. As described above, in order to detect the motor induced voltage, it is necessary to gate off the inverter 11. Therefore, the energization width is limited in order to detect the induced voltage. As shown in FIG. 12, the upper limit W2 of the energization width is the lower limit of the energization width range in which the induced voltage detection period cannot be secured.

Note that if the inverter is a three-phase inverter that drives a three-phase motor, only one of the three phases needs to be gated off when detecting the motor induced voltage. Therefore, with the three-phase inverter, it is possible to detect the motor induced voltage while continuing to supply voltage to the three-phase motor. On the other hand, in the case of a single-phase inverter, it is necessary to gate off all elements when detecting the motor induced voltage, which is a restriction on control in the single-phase inverter. This point is a major difference between three-phase inverter control and single-phase inverter control, and is a major reason why three-phase inverter control techniques cannot be directly applied to single-phase inverter control.

Returning to the explanation of FIG. 12, after the power supply voltage V _dc reaches the lower limit value V1, that is, in the region where the power supply voltage V _dc is below the lower limit value V1, the energization width is maintained.

Further, in the first embodiment, the polarity switching phase is controlled based on the lower limit value V1. Specifically, as shown in the middle part of FIG. 12, after the power supply voltage V _dc reaches the lower limit value V1, that is, in the region where the power supply voltage V _dc is below the lower limit value V1, the power supply voltage V _dc reaches the lower limit value. Control is performed to make the polarity switching phase smaller than before reaching V1. Reducing the polarity switching phase means controlling the polarity switching phase in the advancing direction. Furthermore, controlling the polarity switching phase in the advancing direction means that point A shown in FIG. 10 and point B shown in FIG. 11 move to the left.

When the polarity switching phase is controlled in the forward direction, an effect that weakens the magnetic flux generated when driving the single-phase motor 12 occurs. That is, controlling the polarity switching phase in the advancing direction means strongly exerting magnetic flux weakening control on the single-phase motor 12. By strongly exerting the flux weakening control, it becomes possible to supply the single-phase motor 12 with electric power commensurate with the decrease in drive power caused by the decrease in the power supply voltage V _dc .

Note that when driving a three-phase motor without a position sensor, it is common to perform rotation vector control, and in this type of vector control, when the duty of the PWM signal increases, flux weakening control is performed. However, in the case of a single-phase motor, a rotation vector cannot be generated and an effect of weakening the magnetic flux always occurs, so it is difficult to apply the drive technology of a three-phase motor as is. Regarding this point, the inventors of the present invention have focused on the fact that in the case of a single-phase motor, advancing the phase of the voltage applied to the motor leads to lowering the average amount of weakening of the magnetic flux generated in the single-phase motor 12. This led to the above idea.

In the upper part of FIG. 12, the broken line is the operation waveform when the energization width and polarity switching phase are not controlled, and the solid line is the operation when at least one of the energization width and polarity switching phase is controlled. It is a waveform. As shown in FIG. 12, when at least one of the energization width and the polarity switching phase is controlled, it is possible to suppress a decrease in rotational speed that may occur due to a decrease in the power supply voltage V _dc . becomes.

As explained above, the motor drive device according to Embodiment 1 detects a physical quantity correlated with the motor induced voltage induced in the single-phase motor and the inverter to which the power supply voltage output from the DC power supply is applied. A detector. The inverter converts the power supply voltage into AC voltage and applies the converted AC voltage to the single-phase motor. Further, the inverter is gated on in a first period to apply a first voltage to the single-phase motor, and gated off in a second period after application of the first voltage. The motor drive device increases the energization width, which represents the application period of the first voltage in electrical angle phase, in response to a decrease in the power supply voltage, and when the power supply voltage further decreases and reaches the lower limit value of the power supply voltage. stops increasing the energization width. Such control can suppress a decrease in rotational speed that may occur due to a decrease in power supply voltage. Thereby, the motor drive device according to Embodiment 1 can obtain the effect that the operating time per power supply capacity can be made longer than before.

Further, the motor drive device according to the first embodiment increases the energization width, which represents the application period of the first voltage in terms of electrical angle phase, in response to a decrease in the power supply voltage, and further increases the energization width. When the limit value is reached, the increase in the energization width is stopped. Such control can suppress a decrease in rotational speed that may occur due to a decrease in power supply voltage. Thereby, the motor drive device according to Embodiment 1 can obtain the effect that the operating time per power supply capacity can be made longer than before.

In the above control, when the power supply voltage decreases and reaches the lower limit value of the power supply voltage, the motor drive device changes the polarity switching phase for switching between positive and negative polarity of the first voltage until the power supply voltage reaches the lower limit value of the power supply voltage. Control may be performed in the forward direction compared to before the value is reached. Alternatively, when the energization width increases and reaches the energization width limit value, the polarity switching phase for switching between positive and negative polarity of the first voltage may be changed in the advancing direction compared to before the energization width reaches the energization width limit value. May be controlled. By controlling in this way, it becomes possible to further enhance the effect of suppressing a decrease in rotational speed that may occur due to a decrease in power supply voltage. Note that the lower limit value of the power supply voltage and the limit value of the energization width can be set based on the upper limit value of the energization width.

Next, the effect of driving the single-phase motor 12 without a position sensor in the motor drive device 2 according to the first embodiment will be described.

First, in the case of a configuration without a position sensor, it is possible to start up without a position sensor, so costs such as material costs and processing costs for the position sensor can be reduced. Furthermore, since there is no position sensor, it is possible to eliminate the effect on performance due to misalignment of the position sensor. Thereby, stable performance can be ensured.

Furthermore, if the applied example is a vacuum cleaner and a magnetic pole position sensor is used as the position sensor, the distance between the permanent magnet provided in the rotor and the substrate provided with the magnetic pole position sensor becomes short. In this case, the substrate will be placed in a position that obstructs the flow of air generated by the blades, increasing pressure loss in the air path. The increase in pressure loss is a factor that deteriorates the suction power of the vacuum cleaner and reduces the suction power. On the other hand, in the position sensorless system, since no position sensor is provided, the degree of freedom in arranging the substrate increases, so that the board can be arranged parallel to the air path. Thereby, since the substrate does not block the air path, pressure loss in the air path can be suppressed and suction power can be improved. As a result, it becomes possible to improve the suction power of the vacuum cleaner.

Furthermore, when the applied example is an electric blower and the gas sucked by the electric blower contains a large amount of moisture, the amount of moisture that directly collides with the substrate increases. In this case, when a voltage is applied to the substrate, there is a concern that ion migration may occur, in which ionized metal moves between the electrodes, causing a short circuit. Furthermore, there is a concern about short circuits caused by dirt or dust accumulating on the substrate. As a countermeasure against this problem, methods are adopted such as applying a moisture proofing agent to the substrate or isolating the substrate from the air path, but both of these methods result in an increase in manufacturing costs. On the other hand, in a position sensorless system, since no position sensor is provided, the degree of freedom in arranging the substrate increases, so that the board can be placed avoiding the air path. This reduces the amount of moisture that directly collides with the substrate, thereby suppressing the occurrence of ion migration and reducing the amount of moisture proofing agent. Furthermore, since the degree of freedom in board placement is increased, the quality of the board can be improved by arranging the board outside the casing.

Additionally, since the position sensor is a sensitive sensor, high accuracy is required regarding the installation position of the position sensor. Further, after installation, adjustment is required depending on the installation position of the position sensor. On the other hand, in the case of a configuration without a position sensor, the position sensor itself becomes unnecessary, and the process of adjusting the position sensor can also be eliminated. This makes it possible to significantly reduce manufacturing costs. Furthermore, since the position sensor is not affected by aging, the quality of the product can be improved.

Furthermore, in the position sensorless system, since a position sensor is not required, the inverter and single-phase motor can be configured separately. This makes it possible to reduce restrictions when applying the product. For example, if the application is a product used in a water fountain or the like, the inverter can be placed isolated from the water fountain or the like. This makes it possible to reduce the possibility that the inverter will fail, thereby increasing the reliability of the device.

Additionally, in the case of a position sensorless system, the configuration includes a current detector. The current detector can detect motor abnormalities such as shaft lock or phase loss by detecting motor current. This allows the vehicle to be stopped safely even without a position sensor.

Note that in order to detect motor abnormality, for example, a threshold value for determining overcurrent is set. Then, when the shunt voltage reaches the threshold value, it is determined that the motor is abnormal. Further, if it is determined that the motor is abnormal, the output of the inverter is cut off. In this way, motor abnormality can be detected and the operation of the product can be safely stopped.

Embodiment 2.
In Embodiment 2, an application example of the motor drive device 2 described in Embodiment 1 will be described. The motor drive device 2 described above can be used, for example, in a vacuum cleaner. In the case of a product such as a vacuum cleaner that is used immediately after the power is turned on, the effect of high-speed rotation control of the motor drive device 2 according to the first embodiment is large.

FIG. 13 is a configuration diagram of a vacuum cleaner 61 according to the second embodiment. A vacuum cleaner 61 shown in FIG. 13 is a so-called stick-type vacuum cleaner. In FIG. 13, a vacuum cleaner 61 includes a battery 10 shown in FIG. 1, a motor drive device 2 shown in FIG. 1, an electric blower 64 driven by the single-phase motor 12 shown in FIG. It includes a chamber 65, a sensor 68, a suction port body 63, an extension tube 62, and an operating section 66.

A user using the vacuum cleaner 61 holds the operation unit 66 and operates the vacuum cleaner 61. The motor drive device 2 of the vacuum cleaner 61 drives the electric blower 64 using the battery 10 as a power source. By driving the electric blower 64, dust is sucked in through the suction port body 63. The sucked-in dust is collected into the dust collection chamber 65 via the extension pipe 62.

Although a stick-type vacuum cleaner is illustrated in FIG. 13, the present invention is not limited to stick-type vacuum cleaners. The technology of the present disclosure can be applied to any product as long as it is an electrical device equipped with an electric blower.

Furthermore, although FIG. 13 shows a configuration in which the battery 10 is used as a power source, the present invention is not limited to this. Instead of the battery 10, an AC power source supplied from an outlet may be used.

Furthermore, the above-mentioned motor drive device can be used, for example, in a hand dryer. In the case of a hand dryer, the shorter the time between inserting the hand and driving the electric blower, the better the user experience will be. Therefore, the effects of the high-speed rotation control of the motor drive device 2 according to the first embodiment are greatly exhibited.

FIG. 14 is a configuration diagram of a hand dryer 90 according to the second embodiment. In FIG. 14, the hand dryer 90 includes the motor drive device 2 shown in FIG. It includes a port 98 and an electric blower 95 driven by the single-phase motor 12 shown in FIG. Here, the sensor 97 is either a gyro sensor or a human sensor. In the hand dryer 90 , when a hand is inserted into the hand insertion part 99 at the top of the water receiver 93 , water is blown away by the air blown by the electric blower 95 , and the blown water is collected in the water receiver 93 . After that, it is collected in the drain container 94.

As described above, in Embodiment 2, a configuration example in which the motor drive device 2 according to Embodiment 1 is applied to a vacuum cleaner and a hand dryer has been described, but the present invention is not limited to these examples. The motor drive device 2 can be widely applied to electrical equipment equipped with a motor. Examples of electrical equipment equipped with motors are incinerators, crushers, dryers, dust collectors, printing machines, cleaning machines, confectionery machines, tea machines, woodworking machines, plastic extruders, cardboard machines, packaging machines, and hot air generators. , OA equipment, and electric blowers. An electric blower is a blowing means for transporting objects, collecting dust, or general ventilation.

Note that the configuration shown in the embodiment above is an example, and it is possible to combine it with another known technology, or omit or change a part of the configuration without departing from the gist. It is also possible.

1 Motor drive system, 2 Motor drive device, 5A, 5B leg, 6A, 6B connection end, 10 Battery, 11, 11A inverter, 12 Single phase motor, 12a Rotor, 12b Stator, 16a, 16b DC bus, 18a, 18b connection line, 20, 21 voltage detector, 22, 24 current detector, 25 control unit, 30 analog-digital converter, 30a digital output value, 31 processor, 32 drive signal generation unit, 33 carrier generation unit, 34 memory, 38, 38A, 38B carrier comparison section, 38a absolute value calculation section, 38b division section, 38c, 38d, 38f, 38k multiplication section, 38e, 38m, 38n addition section, 38g, 38h comparison section, 38i, 38j output inversion section, 42 rotation Speed calculation unit, 44 Advance angle phase calculation unit, 51, 52, 53, 54 Switching element, 51a, 52a, 53a, 54a Body diode, 55a, 55b Shunt resistor, 61 Vacuum cleaner, 62 Extension pipe, 63 Suction port body , 64, 95 electric blower, 65 dust collection chamber, 66 operation unit, 68, 97 sensor, 90 hand dryer, 91 casing, 92 hand detection sensor, 93 water receiver, 94 drain container, 96 cover, 98 air intake port, 99 Hand insertion part.

Claims

A motor drive device that drives a single-phase motor without a position sensor,
an inverter to which a power supply voltage output from a DC power supply is applied, converts the power supply voltage into an AC voltage, and applies the converted AC voltage to the single-phase motor;
a detector that detects a physical quantity correlated with a motor induced voltage induced in the single-phase motor;
Equipped with
The inverter gates on in a first period to apply a first voltage to the single-phase motor, and gates off in a second period after application of the first voltage,
In response to the decrease in the power supply voltage, the energization width in which the application period of the first voltage is expressed in electrical angle phase is increased, and when the power supply voltage further decreases and reaches the lower limit value of the power supply voltage, , a motor drive device that stops increasing the energization width.
the first and second periods arrive alternately;
The inverter switches the polarity of the first voltage between positive and negative each time the first period arrives,
When the power supply voltage decreases and reaches the lower limit value, a polarity switching phase for switching between positive and negative polarity of the first voltage is controlled in a forward direction compared to before the power supply voltage reaches the lower limit value. The motor drive device according to item 1.
The motor drive device according to claim 1 or 2, wherein the lower limit value is set based on the upper limit value of the energization width.
A motor drive device that drives a single-phase motor without a position sensor,
an inverter to which a power supply voltage output from a DC power supply is applied, converts the power supply voltage into an AC voltage, and applies the converted AC voltage to the single-phase motor;
a detector that detects a physical quantity correlated with a motor induced voltage induced in the single-phase motor;
Equipped with
The inverter gates on in a first period to apply a first voltage to the single-phase motor, and gates off in a second period after application of the first voltage,
In response to a decrease in the power supply voltage, the energization width, which represents the application period of the first voltage in electrical angle phase, is increased, and when the energization width further increases and reaches the limit value of the energization width, A motor drive device that stops increasing the energization width.
the first and second periods arrive alternately;
The inverter switches the polarity of the first voltage between positive and negative each time the first period arrives,
When the energization width increases and reaches the limit value, the polarity switching phase for switching the polarity of the first voltage between positive and negative is controlled in a forward direction compared to before the energization width reaches the limit value. The motor drive device according to item 4.
The motor drive device according to claim 4 , wherein the limit value is set based on an upper limit value of the energization width.
The inverter has a plurality of switching elements connected in a bridge,
The motor drive device according to any one of claims 1 to 6, wherein at least one of the plurality of switching elements is formed of a wide bandgap semiconductor.
The motor drive device according to claim 7, wherein the wide bandgap semiconductor is silicon carbide, gallium nitride, gallium oxide, or diamond.
An electric blower comprising the motor drive device according to any one of claims 1 to 8.
A vacuum cleaner comprising the electric blower according to claim 9.
A hand dryer comprising the electric blower according to claim 9.