JP6739691B1 - Motor drive device, electric blower, vacuum cleaner and hand dryer - Google Patents

Abstract

The motor drive device (2) includes an inverter (11), a current detector (22), and a control unit (25). The inverter (11) has first and second legs (5A, 5B), converts a DC voltage output from the battery (10) into an AC voltage, and applies the AC voltage to the single-phase motor (12). The control unit (25) is a motor that flows to the single-phase motor (12) connected between the connection point (6A) of the first leg (5A) and the connection point (6B) of the second leg (5B). The conduction of the switching elements (51, 52, 53, 54) is controlled based on the detected current value. The control unit (25) stops application of the AC voltage to the single-phase motor (12), and the motor current flows in the first direction from the connection point (6A) to the connection point (6B). In the first period, after the switching elements (51, 54) are controlled to be non-conducting, the first control is performed to control the switching elements (52, 53) to be conductive.

Description

The present invention relates to a motor drive device that drives a single-phase motor, an electric blower, an electric vacuum cleaner, and a hand dryer.

Conventionally, there are various types of motors such as a DC motor with a brush, an induction motor, and a permanent magnet (PM) motor, and the number of phases of the motor is also a single phase or a three-phase type. Among these various motors, the single-phase PM motor has a brushless structure that does not use a brush, which is a mechanical structure, as compared with a DC motor with a brush, and therefore wear of the brush does not occur. Due to this feature, the single-phase PM motor can ensure long life and high reliability. Further, the single-phase PM motor is a highly efficient motor because no secondary current flows through the rotor, as compared with the induction motor.

The single-phase PM motor has the following advantages even when compared with a three-phase PM motor having a different number of phases.
(1) A three-phase PM motor requires a three-phase inverter, whereas a single-phase PM motor requires a single-phase inverter.
(2) If a full-bridge inverter, which is generally used as a three-phase inverter, is used, six switching elements are required, whereas in the case of a single-phase PM motor, four switching elements are used even if the full-bridge inverter is used. Can be configured with.
(3) Due to the features of (1) and (2), the single-phase PM motor can be downsized as compared with the three-phase PM motor.
Patent Document 1 discloses a technique relating to a drive system of a single-phase PM motor.

JP, 2017-200433, A

In Patent Document 1, when the motor current flowing through the motor winding exceeds the threshold value, the voltage application to the motor winding is stopped. The operation of stopping the voltage application is defined as "free wheel" in Patent Document 1. During this freewheel period, one of the four switching elements connected to both ends of the motor winding is controlled to conduct, and a circuit in which a current flows is formed between the switching element and the diode of the corresponding switching element. It During this time, the motor current circulates between the motor winding and the two switching elements.

As described above, in Patent Document 1, the switching element is used to control the motor current. While circulating the motor current, the value of the motor current does not rise, but all the energy of the motor current is consumed as heat by the resistance component of the winding. Therefore, in the technique of Patent Document 1, the motor current cannot be used efficiently, so that the power consumption increases. In particular, in the case of a configuration in which a storage battery such as a battery is used as a power source, there is a problem that the usable time becomes short due to the large amount of power consumption.

The present invention has been made in view of the above, and an object of the present invention is to obtain a motor drive device that can efficiently use a motor current to reduce power consumption.

In order to solve the above-mentioned problems and achieve the object, a motor drive device according to the present invention includes an inverter, a current detector, and a control unit. The inverter has first and second legs, converts a DC voltage output from the power supply into an AC voltage, and applies the AC voltage to the single-phase motor. The first leg is a leg in which the first switching element of the upper arm and the second switching element of the lower arm are connected in series. The second leg is a leg in which the third switching element of the upper arm and the fourth switching element of the lower arm are connected in series. The second leg is connected in parallel with the first leg. The current detector detects the motor current flowing through the single-phase motor. The control unit controls conduction of the first, second, third and fourth switching elements based on the detection value of the current detector. The single-phase motor is connected between the first connection point and the second connection point. The first connection point is a connection point between the first switching element and the second switching element. The second connection point is a connection point between the third switching element and the fourth switching element. The control unit performs the first control of controlling the second and third switching elements to be conductive after controlling the first and fourth switching elements to be non-conductive in the first period. The first period is a period in which the application of the AC voltage to the single-phase motor is stopped and the motor current is flowing in the first direction from the first connection point toward the second connection point.

According to the motor drive device of the present invention, it is possible to effectively use the motor current and reduce the power consumption.

Configuration diagram of a motor drive system including the motor drive device according to the first embodiment Circuit configuration diagram of the inverter shown in FIG. A block diagram showing a functional part that generates a pulse width modulation (PWM) signal among the functional parts of the control unit shown in FIG. 1. Block diagram showing an example of the PWM signal generation unit shown in FIG. A time chart showing an example of waveforms of main parts in the PWM signal generation unit shown in FIG. Block diagram showing another example of the PWM signal generation unit shown in FIG. A time chart showing an example of waveforms of main parts in the PWM signal generation unit shown in FIG. The figure which shows the structure of the power converter circuit which is a circuit containing the inverter and single phase motor which are shown in FIG. The figure which shows the switching pattern defined by the combination of the drive signal which controls conduction|electrical_connection or non-conduction of each switching element of the power converter circuit shown by FIG. Diagram used for explaining the characteristic circuit operation in the power conversion circuit shown in FIG. The figure which shows the path|route of the electric current which flows into a power converter circuit when controlled by the pattern 1 defined in FIG. The figure which shows the path|route of the electric current which flows into a power converter circuit when controlled by the pattern 3 defined in FIG. The figure which shows the path|route of the electric current which flows into a power converter circuit when controlled by the pattern 5 defined in FIG. The figure which shows the path|route of the electric current which flows into a power converter circuit when controlled by the pattern 7 defined in FIG. The figure which shows the path|route of the electric current which flows into a power converter circuit when controlled by the pattern 8 defined in FIG. The figure which shows the equivalent circuit of the switching element of the power converter circuit shown in FIG. The figure which shows the equivalent circuit of the power converter circuit shown in FIG. 8 when a switching element is controlled by conduction. The figure which shows the equivalent circuit of the power converter circuit shown in FIG. 8 when a switching element is controlled by non-conduction. A time chart showing a waveform example of a PWM signal generated by the PWM signal generation unit shown in FIG. 6 is a flowchart showing an operation flow when a PWM signal is generated by the PWM signal generation unit shown in FIG. Time chart showing an example of the waveform of the PWM signal generated by the PWM signal generation unit shown in FIG. 4 is a flowchart showing an operation flow when a PWM signal is generated by the PWM signal generation unit shown in FIG. FIG. 3 is a block diagram showing an example of a hardware configuration that realizes the function of the control unit in the first embodiment. FIG. 3 is a block diagram showing another example of the hardware configuration that realizes the function of the control unit according to the first embodiment. Configuration diagram of an electric blower according to a second embodiment The block diagram of the electric vacuum cleaner provided with the electric blower which concerns on Embodiment 2. Configuration diagram of a hand dryer including an electric blower according to a second embodiment

Hereinafter, a motor drive device, an electric blower, an electric vacuum cleaner, and a hand dryer according to embodiments of the present invention will be described in detail with reference to the accompanying drawings. The present invention is not limited to the embodiments described below. Further, in the following description, the electrical connection and the physical connection will be simply referred to as “connection” without distinction.

Embodiment 1.
FIG. 1 is a configuration diagram of a motor drive system 1 including a motor drive device 2 according to the first embodiment. The motor drive system 1 shown in FIG. 1 includes a single-phase motor 12, a motor drive device 2, a battery 10, a voltage detector 20, and a switch 102.

The motor drive device 2 supplies AC power to the single-phase motor 12 to drive the single-phase motor 12. The battery 10 is a power source that outputs a DC voltage V _dc to the motor drive device 2. The voltage detector 20 detects the DC voltage V _dc output from the battery 10 to the motor drive device 2.

The single-phase motor 12 is used as a rotating electric machine that rotates an electric blower (not shown). The single-phase motor 12 and the electric blower are installed in devices such as an electric vacuum cleaner and a hand dryer.

Although the voltage detector 20 detects the DC voltage V _dc in the first embodiment, the detection target of the voltage detector 20 is not limited to the DC voltage V _dc output from the battery 10. The detection target of the voltage detector 20 may be the inverter output voltage which is the output voltage of the motor drive device 2. “Inverter output voltage” is synonymous with “motor applied voltage” described later.

The motor drive device 2 includes an inverter 11, a controller 25, and a drive signal generator 32. The inverter 11 is connected to the single-phase motor 12 and applies an AC voltage to the single-phase motor 12. A capacitor (not shown) (not shown in FIG. 1) may be inserted between the battery 10 and the inverter 11 for voltage stabilization. The control unit 25 controls the AC voltage output by the inverter 11.

Further, the motor driving apparatus 2 is provided with a current flowing to the single-phase motor 12, i.e., a current detector 22 for detecting the motor current I _m. The current detector 22, if detected current corresponding to the motor current I _m or the motor current I _m,, where may be disposed. Specifically, in the motor drive device 2, the current detector 22 may be arranged in series with the wiring of the single-phase motor 12, or may be arranged in series with the switching element of the inverter 11. It may be arranged on the power supply line or the ground line of the inverter 11.

Regarding the method of detecting the current in the current detector 22, the method of calculating the current value according to Ohm's law by inserting a resistor having a known resistance value and detecting the voltage value, the detection method using a transformer, and the Hall effect are used. However, any method may be used as long as the motor current I _m can be detected. The first embodiment describes a method of inserting a transformer current sensor in series with the wiring of the single-phase motor 12. Although the current detector 22 detects the current in the motor drive device 2, the current drive may be converted into a voltage to perform the control described below. Further, the inverter 11 is assumed to be a single-phase inverter, but any inverter that can drive the single-phase motor 12 is sufficient.

The control unit 25 includes a detection value of the DC voltage V _dc detected by the voltage detector 20, a detection value of the motor current I _m detected by the current detector 22, a command value output from the switch 102, and an unillustrated A protection signal etc. is input. Examples of the command value include active current command value Ip ^* due to torque, rotation speed command value ω ^{*, and the} like. Control unit 25, the detected value of the DC voltage _{V dc,} the detected value of the motor current _{I m,} based on the command values, PWM signals Q1, Q2, Q3, Q4 (hereinafter to as "Q1~Q4" Notation) is generated.

The switch 102 is, for example, a physical switch, and is a switch for switching between strong driving and weak driving at hand which is used in an electric vacuum cleaner or the like. There are various kinds of shapes of the switch 102 such as a push button type and a toggle type, but any shape can be used as long as it can transmit the user's request to the control unit 25. Further, the switch 102 is not limited to the physical switch, and may be a software process when the command value is automatically switched according to the usage time and the state.

The drive signal generation unit 32 drives the drive signals S1, S2, S3, S4 for driving the switching elements of the inverter 11 based on the PWM signals Q1 to Q4 output from the control unit 25 (hereinafter, “S1 to S4” as appropriate). Is generated). The drive signal generation unit 32 converts each of the PWM signals Q1 to Q4 output from the control unit 25 into drive signals S1 to S4 for driving the inverter 11, and outputs the drive signals S1 to S4 to the inverter 11. It should be noted that the drive signal generation unit 32 is merely an example in FIG. 1 and may have a structure built in the inverter 11 or a structure integrated with the control unit 25.

An example of the single-phase motor 12 is a brushless motor. When the single-phase motor 12 is a brushless motor, the rotor 12a of the single-phase motor 12 has a plurality of permanent magnets (not shown) arranged in the circumferential direction. That is, the single-phase motor 12 has a permanent magnet. The plurality of permanent magnets are arranged so that the magnetization direction is alternately inverted in the circumferential direction, and form a plurality of magnetic poles of the rotor 12a. A winding (not shown) is wound around the stator 12b of the single-phase motor 12. An alternating current flows through the winding. The current flowing through the winding of the single-phase motor 12 is appropriately referred to as "motor current". Although the number of magnetic poles of the rotor 12a is assumed to be four in the first embodiment, the number of magnetic poles of the rotor 12a may be other than four.

FIG. 2 is a circuit configuration diagram of the inverter 11 shown in FIG. The inverter 11 has a plurality of bridge-connected switching elements 51, 52, 53, 54 (hereinafter appropriately referred to as “51 to 54”). The switching elements 51 and 52 form the first leg 5A. In the first leg 5A, the switching element 51 and the switching element 52 are connected in series. The switching elements 53 and 54 form the second leg 5B. In the second leg 5B, the switching element 53 and the switching element 54 are connected in series. The first leg 5A and the second leg 5B are connected in parallel with each other.

The switching elements 51 and 53 are located on the high potential side, and the switching elements 52 and 54 are located on the low potential side. In an inverter circuit, the high potential side is generally called the "upper arm" and the low potential side is called the "lower arm". In the following description, the switching element 51 of the first leg 5A is referred to as "upper arm first element" or "first switching element", and the switching element 53 of the second leg 5B is referred to as "upper arm second element". Alternatively, it may be referred to as a “third switching element”. The switching element 52 of the first leg 5A is referred to as a "lower arm first element" or "second switching element", and the switching element 54 of the second leg 5B is referred to as a "lower arm second element" or "second arm". 4 switching element".

The single-phase motor 12 and the current detector 22 are connected between the connection point 6A which is the first connection point in the inverter 11 and the connection point 6B which is the second connection point in the inverter 11. The connection point 6A is a connection point between the switching element 51 and the switching element 52. The connection point 6B is a connection point between the switching element 53 and the switching element 54. The connection points 6A and 6B form an AC end in the bridge circuit. As described above, the current detector 22 may be inserted anywhere as long as the current flowing through the single-phase motor 12 can be detected, and the method for detecting the current in the current detector 22 is not limited.

A MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor), which is a metal oxide film semiconductor field effect transistor, is used for each of the plurality of switching elements 51 to 54. The MOSFET is an example of FET (Field-Effect Transistor). A MOSFET is a semiconductor element having a function of bidirectionally flowing a current, that is, a function of flowing a current from a drain to a source or from a source to a drain. This function is sometimes called the "reverse conduction function". That is, the MOSFET is a semiconductor element in which the transistor element itself has a reverse conduction function. Another example of the semiconductor device having the reverse conduction function is a reverse-conducting insulated gate bipolar transistor (RC-IGBT).

A body diode 51a connected in parallel is formed between the drain and the source of the switching element 51 in the switching element 51. A body diode 52a connected in parallel is formed between the drain and the source of the switching element 52 in the switching element 52. A body diode 53a connected in parallel is formed between the drain and the source of the switching element 53 in the switching element 53. A body diode 54a connected in parallel is formed between the drain and the source of the switching element 54 in the switching element 54. Each of the plurality of body diodes 51a, 52a, 53a, 54a is a parasitic diode formed inside the MOSFET and is used as a freewheeling diode. A free wheel diode may be separately connected. Further, although the MOSFET is illustrated in the first embodiment, the RC-IGBT described above may be used.

Further, the plurality of switching elements 51 to 54 are not limited to MOSFETs formed of a silicon-based material, and may be MOSFETs formed of a wide band gap semiconductor such as silicon carbide, gallium nitride, gallium oxide or diamond.

Generally, wide band gap semiconductors have higher withstand voltage and heat resistance than silicon semiconductors. Therefore, by using the wide band gap semiconductors for the plurality of switching elements 51 to 54, the withstand voltage property and the permissible current density of the switching elements are increased, and the semiconductor module incorporating the switching elements can be miniaturized. In addition, since wide band gap semiconductors have high heat resistance, it is possible to downsize the heat dissipation part for dissipating the heat generated in the semiconductor module, and also to simplify the heat dissipation structure for dissipating the heat generated in the semiconductor module. It is possible.

FIG. 3 is a block diagram showing a functional part that generates a PWM signal among the functional parts of the control unit 25 shown in FIG.

In FIG. 3, a phase θ _v used when generating a voltage command V _m described later is input to the PWM signal generation unit 38. Further, the phase θ _v , the carrier generated by the carrier generation unit 33, the DC voltage V _dc, and the voltage amplitude command V* that is the amplitude value of the voltage command V _m are input to the PWM signal generation unit 38. .. The PWM signal generation unit 38 generates PWM signals Q1 to Q4 based on the carrier, the phase θ _v , the DC voltage V _dc, and the voltage amplitude command V*.

FIG. 4 is a block diagram showing an example of the PWM signal generator 38 shown in FIG. FIG. 4 shows the detailed configurations of the PWM signal generator 38A and the carrier generator 33. In FIG. 4, the carrier generation unit 33 is set with a carrier frequency f _C [Hz] that is the frequency of the carrier. As an example of the carrier waveform, a triangular wave carrier that goes up and down between “0” and “1” is shown at the tip of the arrow of the carrier frequency f _C. The PWM control of the inverter 11 includes synchronous PWM control and asynchronous PWM control. In the case of synchronous PWM control, it is necessary to synchronize the carrier with the phase θ _v . On the other hand, in the case of asynchronous PWM control, it is not necessary to synchronize the carrier with the phase θ _v .

As shown in FIG. 4, the PWM signal generation unit 38A has an absolute value calculation unit 38a, a division unit 38b, a multiplication unit 38c, a multiplication unit 38d, a multiplication unit 38f, an addition unit 38e, a comparison unit 38g, a comparison unit 38h, and an output. It has an inversion unit 38i, an output inversion unit 38j, and an output selection unit 38p.

The absolute value calculator 38a calculates the absolute value |V*| of the voltage amplitude command V*. The division unit 38b divides the absolute value |V*| by the DC voltage _Vdc detected by the voltage detector 20. In the configuration shown in FIG. 4, the output of the division unit 38b becomes the modulation rate. The battery voltage, which is the output voltage of the battery 10, fluctuates as the current continues to flow. On the other hand, by dividing the absolute value |V*| by the DC voltage V _dc , the value of the modulation rate can be adjusted so that the voltage applied to the motor does not decrease due to the decrease in the battery voltage.

The multiplication unit 38c calculates the sine value of the phase θ _v . The multiplication unit 38c multiplies the sine value of the phase θ _v by the modulation rate that is the output of the division unit 38b. The multiplication unit 38d multiplies the voltage command V _m output from the multiplication unit 38c by “½”. The adder 38e adds "1/2" to the output of the multiplier 38d. The multiplication unit 38f multiplies the output of the addition unit 38e by "-1". The output of the addition unit 38e is input to the comparison unit 38g as a positive voltage command V _m1 for driving the two switching elements 51 and 53 of the upper arm among the plurality of switching elements 51 to 54. The output of the multiplication unit 38f is input to the comparison unit 38h as a negative voltage command V _m2 for driving the two switching elements 52 and 54 of the lower arm.

The comparison unit 38g compares the positive side voltage command V _m1 with the carrier amplitude. The output of the output inversion unit 38i, which is the output of the comparison unit 38g, becomes the PWM signal Q1′ to the output selection unit 38p, and the output of the comparison unit 38g becomes the PWM signal Q2′ to the output selection unit 38p. Similarly, the comparison unit 38h compares the negative-side voltage command V _m2 with the carrier amplitude. The output of the output inversion unit 38j, which is the output of the comparison unit 38h, becomes the PWM signal Q3′ to the output selection unit 38p, and the output of the comparison unit 38h becomes the PWM signal Q4′ to the output selection unit 38p.

As described above, the PWM signals Q1′, Q2′, Q3′, Q4′ (hereinafter appropriately referred to as “Q1′ to Q4′”) are input to the output selection unit 38p. Further, the phase θ _v is input to the output selection unit 38p as shown in FIG. Output selection unit 38p includes a phase theta _v, the PWM signal Q1'～Q4 ', the direction of the motor current _{I m} flowing through the single-phase motor 12, based on, generates a PWM signal Q1 to Q4. Details of the processing by the output selection unit 38p will be described later. When there is no need to change the PWM signals Q1′ to Q4′ input to the output selection unit 38p, the PWM signals Q1′ to Q4′ are output as they are as the PWM signals Q1 to Q4.

FIG. 5 is a time chart showing a waveform example of a main part in the PWM signal generation unit 38A shown in FIG. In FIG. 5, the waveform of the positive voltage command V _m1 output from the adder 38e, the waveform of the negative voltage command V _m2 output from the multiplier 38f, and the waveforms of the PWM signals Q1′ to Q4′ are shown. It is shown. Further, the waveform of the inverter output voltage when the switching elements 51 to 54 are controlled by each of the PWM signals Q1' to Q4' is shown in the lowermost part of FIG.

PWM signal Q1 'is "high (High)" when "low (Low)" next when the positive voltage command _{V m1} is greater than the carrier, the positive voltage command _{V m1} is smaller than the carrier. The PWM signal Q2' is an inverted signal of the PWM signal Q1'. PWM signal Q3 'is "high (High)" when "low (Low)" becomes when negative voltage instruction _{V m2} is larger than the carrier, the negative-side voltage instruction _{V m2} smaller than the carrier. The PWM signal Q4' is an inverted signal of the PWM signal Q3'. As described above, although the circuit shown in FIG. 4 is configured by “Low Active”, even if it is configured by “High Active” in which respective signals have opposite values, Good.

As shown in FIG. 5, the waveform of the inverter output voltage includes a voltage pulse due to the difference voltage between the PWM signal Q1′ and the PWM signal Q4′ and a voltage pulse due to the difference voltage between the PWM signal Q3′ and the PWM signal Q2′. Appears. These voltage pulses are applied to the single-phase motor 12 as a motor applied voltage.

Bipolar modulation and unipolar modulation are known as modulation methods used when generating the PWM signals Q1' to Q4'. Bipolar modulation is a modulation method that outputs a voltage pulse that changes with a positive or negative potential for each cycle of the voltage command V _m . Unipolar modulation is a modulation method that outputs a voltage pulse that changes with three potentials in each cycle of the voltage command V _m , that is, a voltage pulse that changes between a positive potential, a negative potential, and a zero potential. The waveform shown in FIG. 5 is based on unipolar modulation. In the motor drive device 2 of the first embodiment, any modulation method may be used. It should be noted that in applications where it is necessary to control the motor current waveform to a more sinusoidal wave, it is preferable to employ unipolar modulation having a smaller harmonic content rate than bipolar modulation.

In addition, the waveforms shown in FIG. 5 have switching elements 51 and 52 that configure the first leg 5A and a switching element 53 that configures the second leg 5B in the section of the half cycle T/2 of the voltage command V _m. , 54 of four switching elements. This method is called “two-sided PWM” because the switching operation is performed by both the positive side voltage command V _m1 and the negative side voltage command V _m2 . On the other hand, in one half cycle of one cycle T of the voltage command V _m , the switching operation of the switching elements 51 and 52 is stopped, and in the other half cycle of one cycle T of the voltage command V _m , There is also a method of suspending the switching operation of the switching elements 53 and 54. This method is called "one-sided PWM". Hereinafter, "one side PWM" will be described.

FIG. 6 is a block diagram showing another example of the PWM signal generation unit 38 shown in FIG. FIG. 6 shows an example of a PWM signal generation circuit by the above-mentioned “one-sided PWM”, and specifically shows detailed configurations of the PWM signal generation unit 38B and the carrier generation unit 33. The configuration of the carrier generation unit 33 shown in FIG. 6 is the same as or equivalent to that shown in FIG. Further, in the configuration of the PWM signal generation unit 38B shown in FIG. 6, the same or equivalent components as those of the PWM signal generation unit 38A shown in FIG. 4 are denoted by the same reference numerals.

As shown in FIG. 6, the PWM signal generator 38B has an absolute value calculator 38a, a divider 38b, a multiplier 38c, a multiplier 38k, an adder 38m, an adder 38n, a comparator 38g, a comparator 38h, and an output. It has an inversion unit 38i, an output inversion unit 38j, and an output selection unit 38q.

The absolute value calculator 38a calculates the absolute value |V*| of the voltage amplitude command V*. The division unit 38b divides the absolute value |V*| by the DC voltage _Vdc detected by the voltage detector 20. Also in the configuration of FIG. 6, the output of the division unit 38b becomes the modulation rate.

The multiplication unit 38c calculates the sine value of the phase θ _v . The multiplying unit 38c multiplies the sine value of the phase θ _v by the modulation rate that is the output of the dividing unit 38b. Multiplying unit 38k multiplies "-1" to the voltage command _{V m} is the output of the multiplication unit 38c. The adding unit 38m adds “1” to the voltage command V _m output from the multiplying unit 38c. The adder 38n adds "1" to the output of the multiplier 38k, that is, the inverted output of the voltage command V _m . The output of the addition unit 38m is input to the comparison unit 38g as a first voltage command V _m3 for driving the two switching elements 51 and 53 of the upper arm among the plurality of switching elements 51 to 54. The output of the adder 38n is input to the comparator 38h as the second voltage command V _m4 for driving the two switching elements 52 and 54 of the lower arm.

The comparison unit 38g compares the first voltage command V _m3 with the carrier amplitude. The output of the output inversion unit 38i, which is the output of the comparison unit 38g, becomes the PWM signal Q1′ to the output selection unit 38q, and the output of the comparison unit 38g becomes the PWM signal Q2′ to the output selection unit 38q. Similarly, the comparison unit 38h compares the second voltage command V _m4 with the amplitude of the carrier. The output of the output inversion unit 38j, which is the output of the comparison unit 38h, becomes the PWM signal Q3′ to the output selection unit 38q, and the output of the comparison unit 38h becomes the PWM signal Q4′ to the output selection unit 38q.

As described above, the PWM signals Q1' to Q4' are input to the output selection unit 38q. Further, the phase θ _v is input to the output selection unit 38q as shown in FIG. Output selection unit 38q includes a phase theta _v, the PWM signal Q1'～Q4 ', the direction of the motor current _{I m} flowing through the single-phase motor 12, based on, generates a PWM signal Q1 to Q4. Details of the processing by the output selection unit 38q will be described later. When there is no need to change the PWM signals Q1′ to Q4′ input to the output selection unit 38q, the PWM signals Q1′ to Q4′ are output as they are as the PWM signals Q1 to Q4.

FIG. 7 is a time chart showing a waveform example of a main part in the PWM signal generation unit 38B shown in FIG. FIG. 7 shows the waveform of the first voltage command V _m3 output from the adder 38m, the waveform of the second voltage command V _m4 output from the adder 38n, and the waveforms of the PWM signals Q1′ to Q4′. It is shown. Further, the waveform of the inverter output voltage when the switching elements 51 to 54 are controlled by each of the PWM signals Q1' to Q4' is shown in the lowermost part of FIG. In FIG. 7, for convenience, the waveform portion of the first voltage command V _m3 whose amplitude value is larger than the peak value of the carrier and the second voltage command V _m4 whose amplitude value is larger than the peak value of the carrier are shown. The corrugated portion is represented by a flat straight line.

PWM signal Q1 'is "low (Low)" next when the first voltage command _{V m3} is greater than the carrier, the first voltage command _{V m3} is "high (High)" when less than the carrier. The PWM signal Q2' is an inverted signal of the PWM signal Q1'. PWM signal Q3 'is "low (Low)" next when a second voltage command _{V m4} is greater than the carrier, the second voltage command _{V m4} is "high (High)" when less than the carrier. The PWM signal Q4' is an inverted signal of the PWM signal Q3'. As described above, although the circuit shown in FIG. 6 is configured by “Low Active”, even if it is configured by “High Active” in which respective signals have opposite values, Good.

As shown in FIG. 7, the waveform of the inverter output voltage includes a voltage pulse due to the difference voltage between the PWM signal Q1′ and the PWM signal Q4′ and a voltage pulse due to the difference voltage between the PWM signal Q3′ and the PWM signal Q2′. Appears. These voltage pulses are applied to the single-phase motor 12 as a motor applied voltage.

In the waveforms shown in FIG. 7, in one half cycle of one cycle T of the voltage command V _m, the switching operation of the switching elements 51 and 52 are at rest, the other of the one period T of the voltage command V _m In the half cycle, the switching operation of the switching elements 53 and 54 is stopped.

Further, as shown in FIG. 7, the waveform of the inverter output voltage is unipolar modulation that changes with three potentials for each cycle T of the voltage command V _m . As described above, the bipolar modulation may be used instead of the unipolar modulation, but it is preferable to use the unipolar modulation in the application where the motor current waveform needs to be controlled to be more sinusoidal.

The carrier used to generate these PWM signals is a triangular wave of peaks and valleys from the desired carrier frequency f _C. The carrier frequency f _C is variable according to the rotation speed of the motor, and the higher the rotation speed of the motor, the higher the frequency. It is desirable to generate carriers at a frequency twice or more the rotation speed. The carrier to be generated may be a sawtooth wave or the like instead of the triangular wave.

Next, the operation of the main part of the motor drive device 2 according to the first embodiment will be described with reference to FIGS. 8 to 18. FIG. 8 is a diagram showing a configuration of a power conversion circuit 50 which is a circuit including the inverter 11 and the single-phase motor 12 shown in FIG. FIG. 9 is a diagram showing a switching pattern defined by a combination of drive signals controlling conduction or non-conduction of each switching element of the power conversion circuit 50 shown in FIG. FIG. 10 is a diagram used for explaining a characteristic circuit operation in the power conversion circuit 50 shown in FIG. FIG. 11 is a diagram showing a path of a current flowing through the power conversion circuit 50 when controlled by the pattern 1 defined in FIG. FIG. 12 is a diagram showing a path of a current flowing through the power conversion circuit 50 when controlled by the pattern 3 defined in FIG. FIG. 13 is a diagram showing a path of a current flowing through the power conversion circuit 50 when controlled by the pattern 5 defined in FIG. FIG. 14 is a diagram showing a path of a current flowing through the power conversion circuit 50 when controlled by the pattern 7 defined in FIG. FIG. 15 is a diagram showing a path of a current flowing through the power conversion circuit 50 when controlled by the pattern 8 defined in FIG. FIG. 16 is a diagram showing an equivalent circuit of the switching elements of the power conversion circuit 50 shown in FIG. FIG. 17 is a diagram showing an equivalent circuit of the power conversion circuit 50 shown in FIG. 8 when the switching element is controlled to be conductive. FIG. 18 is a diagram showing an equivalent circuit of the power conversion circuit 50 shown in FIG. 8 when the switching element is controlled to be non-conductive.

First, in FIG. 8, the arrow pointing to the right from the connection point 6A to the connection point 6B is described as “+I _m ”. In the first embodiment, the direction of this current is defined as the “first direction”. Further, the opposite of the first direction is defined as "second direction". That is, the first direction is the direction in which the motor current I _m goes from the connection point 6A to the connection point 6B. The second direction is the direction in which the motor current I _m goes from the connection point 6B to the connection point 6A.

Further, in FIG. 8, in the circuit configuration diagram of the inverter 11 shown in FIG. 2, the single-phase motor 12 connected between the connection point 6A and the connection point 6B is equivalently represented by the motor winding 41 and the voltage source 42. It represents. L _m is the inductance component of the motor winding 41. R _m is a resistance component of the motor winding 41. e _m is the induced voltage generated in the motor windings 41. In Figure 8, it represents the single-phase motor 12 as a power source with a voltage source 42 of the induced voltage e _m. Further, in FIG. 8, a circuit including the inverter 11 and the single-phase motor 12 is defined as a power conversion circuit 50.

Further, FIG. 8 shows a power supply capacitor 40 connected in parallel to each of the first leg 5A and the second leg 5B. The power supply capacitor 40 is a power storage device that can store electric power. The power supply capacitor 40 holds the DC power supplied from the battery 10. Further, the power supply capacitor 40 can collect electric power supplied from the single-phase motor 12 and hold it as DC power when the single-phase motor 12 operates as a generator.

In the power conversion circuit 50, one switching element can be in two states of conducting and non-conducting. Therefore, in the power conversion circuit 50 including the four switching elements 51 to 54, one switching element has two states of conduction and non-conduction, and the number of elements is 4, so 2×4=16. Street conditions are assumed. Further, in each state, the motor current I _m is considering whether the flow to either the first or second direction, 16 × 2 = 32 kinds state is assumed.

In FIG. 9, the outputs of the drive signals S1 to S4 to the respective switching elements are shown. "1" is a drive signal for controlling each switching element to be conductive, and "0" is a drive signal for controlling each switching element to be non-conductive.

In FIG. 9A, patterns 1 to 9 are shown. In FIG. 9B, patterns 10 to 18 are shown. In FIG. 9C, patterns 19 to 25 are shown. In FIG. 9D, patterns 26 to 32 are shown. Each of pattern 1 to pattern 9 and pattern 19 to pattern 25 is a pattern when the motor current _Im is flowing in the first direction. Further, each of the pattern 10 to the pattern 18 and the pattern 26 to the pattern 32 is a pattern when the motor current I _m is flowing in the second direction.

In FIG. 9, pattern 1 and pattern 10 are patterns defined as power running. Power running is a state in which a voltage for flowing a current necessary to rotate the single-phase motor 12 is applied to the single-phase motor 12. Using the above parameters, the motor current I _m is expressed by the following equation (1) using the Laplace operator s.

_{I m = (v mi -e m} ) / (R m + sL m) ...... (1)

In the above formula (1), “v _mi ”is a motor applied voltage that the inverter 11 applies to the single-phase motor 12.

As described above, the motor winding 41 of the single-phase motor 12 has the inductance component L _m and the resistance component R _m . Therefore, when the inverter 11 continuously applied voltage, as shown in equation (1) increases the voltage applied to the motor v _mi, by the difference between the induced voltage e _m, the motor current I _m is the primary delay Keep doing If this state continues, the resistance component R _m of the motor winding 41 is small, so the current value of the motor current I _m becomes excessively large, which may cause a problem of heat generation of the single-phase motor 12. If the single-phase motor 12 is a PM motor, it may cause a problem such as demagnetization of the rotor magnet. The PWM control described above is a method of providing the single-phase motor 12 with a section where a voltage is applied and a section where a voltage is not applied. That is, by using the PWM control, it is possible to adjust the motor current I _m so as not to become excessive.

In FIG. 9, patterns 1 and 10 are patterns of a section where a voltage is applied, and are sections of power running. In addition, the patterns other than the patterns 1 and 10 are patterns in a section where no voltage is applied, and are patterns in a section other than the power running.

Of the patterns shown in FIG. 9, patterns 19 to 32 are basically not used. This is because these patterns are patterns in which both the switching elements of the same leg connected in series in the inverter 11 are conductive. Specifically, it is a pattern in which both the switching element 51 and the switching element 52 are conductive, and a pattern in which both the switching element 53 and the switching element 54 are conductive. In these cases, the power supply capacitor 40 connected to the inverter 11 and the ground are short-circuited through the switching elements of the upper and lower arms that are in a conductive state. This phenomenon is called "upper and lower arm short circuit". When the upper and lower arm short circuits occur, a large current flows through the inverter 11 regardless of the direction of the motor current I _m , and in the worst case, the switching element is damaged.

Further, controlling not only the switching elements of the same leg but also a plurality of switching elements to be conductive and non-conductive at the same time is a control mode to be avoided. This is obvious in consideration of variations in the timing of conduction of the switching elements, delays of signals transmitted from the processor controlling conduction to each switching element, and the like.

For example, in the pattern shown in FIG. 9, consider the switching control when the state of pattern 1 shifts to the state of pattern 4. The upper part of FIG. 10 shows the waveform of the drive signal in the case of shifting from the pattern 1 state to the pattern 4 state. In the lower part of FIG. 10, an enlarged waveform of a section surrounded by a broken line is shown.

As shown in FIG. 9, in order to shift from the state of pattern 1 to the state of pattern 4, control for making the switching element 53 conductive and control for making the switching element 54 non-conductive are carried out. Ideally, the conduction of the switching element 53 and the non-conduction of the switching element 54 are performed at the same time. At this time, the transition from the pattern 1 state to the pattern 4 state is performed without passing through other patterns. However, in reality, the timings at which the switching elements 53 and 54 change do not completely match when viewed in a minute time. The lower part of FIG. 10 shows a situation in which the timing at which the switching element 53 becomes conductive is delayed from the timing at which the switching element 54 becomes non-conductive.

In the case of the example shown in the lower part of FIG. 10, the state of pattern 1 is changed to the state of pattern 4 via the state of pattern 2. Also, unlike FIG. 10, when the timing at which the switching element 54 becomes conductive is delayed from the timing at which the switching element 53 becomes conductive, the state of pattern 1 goes through the state of pattern 23 from the state of pattern 1 to the state of pattern 4. It will be transferred after that.

As described above, the pattern 23 is a pattern in which both switching elements of the same leg connected in series are conductive. Therefore, it is not preferable to go through the pattern 23.

Therefore, in order to appropriately switch between conduction and non-conduction of each switching element, it is important to switch between conduction and non-conduction one by one in consideration of variations of each switching element, delay of signal transmission, and the like. Further, when switching the conduction and non-conduction of each of the switching elements one by one, it is important that the patterns to be passed do not include the patterns 19 to 32.

Looking at the pattern of FIG. 9 based on the above description, the transition from the power running state to the next state is narrowed down to two patterns. Specifically, the transfer destination from pattern 1, which is one of the power running patterns, is either pattern 2 or 3. Further, the transfer destination from the pattern 10 which is another powering pattern is any one of the patterns 11 and 12.

Here, when each switching element of the inverter 11 is a gate voltage application element such as a MOSFET, a configuration in which a bootstrap circuit is used as a gate power source is generally used for the switching elements 51 and 53 located on the high potential side. is there. In the bootstrap circuit, the gate voltage is raised by charging the boot capacitor with electric charges, but it is necessary to periodically charge the boot capacitor. Therefore, it is difficult to keep the switching elements 51 and 53 constantly turned on. The length of time that the switching elements 51 and 53 can be kept on depends on the capacity of the boot capacitor. Considering the cost, we would like to avoid increasing the boot capacitor. Therefore, basically, it is preferable to avoid the state where the switching elements 51 and 53 are kept on. From the above, pattern 3 is more practical as the transfer destination of pattern 1. Further, as the transfer destination of the pattern 10, the pattern 12 is more practical. Needless to say, the patterns 2 and 11 may be used, as this is not a technical problem.

Next, the circuit operation associated with the pattern transition will be described. First, the circuit operation when shifting from pattern 1 to pattern 3 will be described. By the control according to the pattern 1, the motor current I _m flowing in the first direction increases. As shown in FIG. 11, the motor current I _m flows from the power supply capacitor 40 through the switching element 51, the motor winding 41 and the switching element 54, and returns to the power supply capacitor 40. If the switching element 51 is controlled to be off in this state, the state shifts to the state of pattern 3 shown in FIG. As described above, since the inductance component L _m exists in the motor winding 41, the motor current I _m tries to keep flowing in the first direction. When the switching element 51 becomes non-conductive, the current loop from the motor winding 41 via the switching element 54 and the power supply capacitor 40 disappears. On the other hand, a body diode 52a is connected in parallel to the switching element 52. Therefore, as nowhere to go in the motor current I _m is not lost, a new current loop via the body diode 52a of the switching element 52 is formed.

As described above, in the pattern 3, the motor current I _m flows in the body diode 52a connected in parallel to the switching element 52, so that a current loop is formed. When the motor current I _m passes through the body diode, a voltage drop occurs due to the forward voltage. The rate of loss caused by this voltage drop is high. Therefore, it is desirable to form a current loop that avoids the body diode as much as possible. Therefore, in the first embodiment, the pattern 3 is changed to the pattern 5. The transition from the pattern 3 to the pattern 5 may be achieved by making the switching element 52 conductive. As a result, as shown in FIG. 13, the motor current I _m flows inside the main body of the switching element 52, that is, through the channel of the switching element 52. As a result, the current flowing in the current loop via the body diode 52a of the switching element 52 becomes small.

As described above, the voltage is applied to the motor winding 41 in the pattern 1, and the pattern 5 is transferred via the pattern 3. This makes it possible to stop the voltage application to the motor winding 41 and control the current flowing through the motor winding 41. However, in this method, the motor current I _m flowing while the voltage application is stopped is consumed as heat of the resistance component. Therefore, in the first embodiment, control is performed not to consume the motor current I _m , but to store electric power based on the motor current I _{m in} the power supply capacitor 40 as a battery. By performing such control, in addition to the effect of avoiding the motor current I _m is dissipated as heat, when re-applying a voltage to the motor windings 41, suppression of the power supplied from the power supply Is possible. As a result, in the motor drive device in which the power source is a storage battery such as a battery, the battery can be used for a longer period of time than before and can be used for a long time.

In order to perform the above control, in the first embodiment, the state of pattern 5 shown in FIG. 13 is further shifted to the state of pattern 7 shown in FIG. Specifically, the switching element 54 is controlled to be non-conductive from the state shown in FIG. When the switching element 54 is controlled to be non-conducting, the current loop through the switching element 52, the motor winding 41 and the switching element 54 disappears. On the other hand, a body diode 53a is connected in parallel to the switching element 53. Thereby, as shown in FIG. 14, a new current loop is formed via the body diode 53a of the switching element 53.

As described above, the current loop passing through the body diode has a large loss. Therefore, in order to form a current loop that avoids the body diode, the state is changed to the pattern 8. Specifically, the switching element 53 is controlled to be conductive from the state shown in FIG. As a result, as shown in FIG. 15, the motor current I _m flows inside the main body of the switching element 53, that is, through the channel of the switching element 53.

Here, the effect of forming the current loop avoiding the body diode will be described with reference to FIGS. 16 to 18. As shown on the right side of FIG. 16, the switching element in which the body diode is connected in anti-parallel has an on-resistance R _on which is a resistance component at the time of conduction and a forward voltage V _f which is a forward voltage drop component. It can be replaced with an equivalent circuit connected in parallel. FIG. 17 shows an equivalent circuit when the power conversion circuit 50 is controlled by the pattern 8. Further, FIG. 18 shows an equivalent circuit when the power conversion circuit 50 is controlled by the pattern 7. 17 and FIG. _{18, R on2} is on resistance of the switching element _{52, R on3} is the on resistance of the switching element 53. _{Also, R m, L m, I} m, the _{e m} are as described above. In the equivalent circuit of FIG. 17 and FIG. 18, the induced voltage generated by rotation of the single-phase motor 12 expressed as a voltage source as e _m, represents the motor current I _m flowing through the motor windings 41 by a current source.

In the switching element, the voltage drop caused by the on-resistance R _on is extremely smaller than the forward voltage V _f of the body diode. The relation when the power consumption is compared when the switching element is conducting and when it is non-conducting is represented by the following equation (2).

V _f ·I _m >R _on3 (I _m ) ² (2)

As can be understood from the above formula (2), the power loss of the pattern 8 is smaller than that of the pattern 7. Therefore, in the first embodiment, control is performed to shift from pattern 5 to pattern 8 to pattern 8.

The flow of a series of operations up to this point will be described with reference to FIGS. 19 and 20. FIG. 19 is a time chart showing an example of waveforms of the PWM signals Q1 to Q4 generated by the PWM signal generation unit 38B shown in FIG. FIG. 20 is a flowchart showing an operation flow when the PWM signal is generated by the PWM signal generator 38B shown in FIG.

In the upper part of FIG. 19, the waveform of the first voltage command V _m3 output from the addition unit 38m of the PWM signal generation unit 38B and the second voltage command V _m4 output from the addition unit 38n of the PWM signal generation unit 38B. And the waveforms of the PWM signals Q1 to Q4 generated by the output selection unit 38q of the PWM signal generation unit 38B. In addition, the lower part of FIG. 19 shows an enlarged waveform of the part surrounded by the broken-line square frame K1 in the upper part of FIG. Further, each of the two dashed vertical lines penetrating the inside of the frame K1 corresponds to each of the two dashed vertical lines shown in the lower part.

As shown in FIGS. 19 and 20, the output selection unit 38q switches the processing between the A section and the B section. The section A is a section in which the first voltage command V _m3 is a flat straight line that does not cross the carrier and the second voltage command V _m4 changes so as to cross the carrier. The section B is a section in which the first voltage command V _m3 changes so as to cross the carrier and the second voltage command V _m4 is a flat straight line that does not cross the carrier. The determination as to whether it is the A section or the B section is performed using the phase θ _v input to the output selection unit 38q. Specifically, when the phase θ _v is 0°≦θ _v <180°, it is determined as “A section”. When the phase θ _v is 180°≦θ _v <360°, it is determined as “B section”. The output selection unit 38q switches the processing according to the phase θ _v .

Next, the processing of the PWM signal generation unit 38B in the first embodiment will be described with reference to the waveform of FIG. 19 and the flowchart of FIG. The main body of the operation in the flowchart of FIG. 20 is the PWM signal generation unit 38B. In the following description, the motor current I _m is assumed to flow in the second direction. Further, in the flowchart of FIG. 20, it is assumed that the state immediately after the start is the state in which the inverter 11 is controlled by the pattern 10 which is the powering pattern.

20, in step S100, it is determined whether or not the phase θ _v is 0°≦θ _v <180°. If the phase θ _v is 0°≦θ _v <180°, the process proceeds to step S101. If the phase θ _v is 180°≦θ _v <360°, the process proceeds to the opposite phase flowchart (not shown). Note that the reverse-phase flowchart is the same process as the flowchart shown in FIG. 20, and the description in this document is omitted.

In step S101, it is determined whether or not the phase θ _v is 0°<θ _v <θ1 or θ2<θ _v <180°. θ1 and θ2 are set values that satisfy θ1<θ2. If the phase θ _v is 0°<θ _v <θ1 or θ2<θ _v <180°, the process proceeds to step S102. In step S102, the PWM signals Q1′ to Q4′ input to the output selection unit 38q are output as they are as the PWM signals Q1 to Q4. The processing of steps S101 and S102 means that the processing of steps S104 to S118, which will be described later, is not performed in the switching portion between the A section and the B section. When the process of step S102 ends, the process returns to step S100.

If the phase θ _v is θ1≦θ _v ≦θ2, the process proceeds to step S103. In step S103, the carrier and the second voltage command V _m4 are compared. When the carrier is not lower than the second voltage command V _m4 , the process proceeds to step S102. If the carrier is below the second voltage command V _m4 , the process proceeds to step S104. In step S104, the process of shifting to pattern 11 is performed. In step S105, it is determined whether the time Ts11 has elapsed. By steps S104 and S105, the state of the pattern 11 is continued for the time Ts11. When the process of step S105 ends, the process proceeds to step S106. In the lower waveform of FIG. 19, “T _sx ”described between the pattern 10 and the pattern 11 is the delay time required for switching the conduction of the switching element.

In step S106, the process of shifting to pattern 13 is performed. In step S107, it is determined whether the time Ts13 has elapsed. By steps S106 and S107, the state of the pattern 13 is continued for the time Ts13. When the process of step S107 ends, the process proceeds to step S108.

In step S108, the process of shifting to pattern 15 is performed. In step S109, it is determined whether the time Ts15 has elapsed. By steps S108 and S109, the state of the pattern 15 is continued for the time Ts15. When the process of step S109 ends, the process proceeds to step S110.

In step S110, the process of shifting to pattern 17 is performed. In step S111, it is determined whether time Ts17 has elapsed. With steps S110 and S111, the state of the pattern 17 is continued for the time Ts17. When the process of step S111 ends, the process proceeds to step S112.

In step S112, the process of shifting to pattern 15 is performed. In step S113, it is determined whether the time Ts15' has elapsed. By steps S112 and S113, the state of the pattern 15 is continued for the time Ts15'. When the process of step S113 ends, the process proceeds to step S114.

In step S114, the process of shifting to pattern 13 is performed. In step S115, the carrier is compared with the second voltage command V _m4 . When the carrier does not exceed the second voltage command V _m4 , the process returns to step S114 and the state of the pattern 13 is continued. When the carrier exceeds the second voltage command V _m4 , the process proceeds to step S116.

In step S116, the process of shifting to pattern 11 is performed. In step S117, it is determined whether the time Ts11' has elapsed. By steps S116 and S117, the state of the pattern 11 is continued for the time Ts11'. When the process of step S117 ends, the process proceeds to step S118.

In step S118, the process of shifting to pattern 10 is performed. When the process of step S118 ends, the process returns to step S100. After that, the processing from step S100 is repeated.

As described above, in the process according to the flowchart of FIG. 20, after the state of the pattern 10 is changed to the states of the pattern 11 and the pattern 13 in this order, the state is further changed to the states of the pattern 15 and the pattern 17 in this order. I am trying. That is, instead of recirculating the motor current I _m between the motor windings 41 and each of the switching elements, by shifting to the state of the pattern 15 and pattern 17, recovering power by the motor current I _m to the power supply capacitor 40 I am making it. This makes it possible to reduce the power consumption of the motor drive device 2.

In addition, in the processing of FIG. 20, the time Ts11, Ts13, Ts15, Ts17, Ts15', Ts13', and Ts11' when each pattern is continued vary depending on the motor parameter. Therefore, it is necessary to examine or investigate an appropriate time in advance by experiments or simulations.

The times Ts11 and Ts11′, the times Ts13 and Ts13′, and the times Ts15 and Ts15′ may be the same or different. Further, it is necessary to make the total time of these times and the delay time T _sx required for switching the switching element shorter than the half cycle of the carrier. Furthermore, it is desirable that the times Ts13 and Ts15 in which the motor current I _m is consumed as heat be made as short as possible. Since the pattern 15 has a larger loss due to the body diode of the switching element than the pattern 17, the time Ts15 is preferably shorter than the time Ts17.

Further, the delay time T _sx is the time required for turning on or off the switching element, and although it varies depending on the switching element, it cannot be omitted. Therefore, it is necessary to secure at least the delay time T _sx for each pattern.

Further, when the current value of the motor current I _m flowing through the motor windings 41 is small, the process proceeds from the pattern 15 and 17 to short the motor windings 41 to the pattern 11 or pattern 13, thereafter the pattern 15 or pattern 17 You may bring it back. By shorting the motor windings 41, it is possible to increase the motor current I _m which the energy of the induced voltage generated during the rotation of the single-phase motor 12 through the motor windings 41. This is an operation for temporarily boosting the current flowing through the motor winding 41 to generate a boosted voltage in the motor winding 41. This operation can be performed based on the detection value of the current detector 22. Specifically, when the detected value of the motor current I _m is below the specified current, it is sufficient to shift to a pattern of short-circuiting the motor winding 41.

Further, the voltage of the power supply capacitor 40 is excessively increased by the motor current I _m flows to the power supply capacitor 40 side, there is a risk that the equipment may be damaged beyond the breakdown voltage of the device connected to the power supply capacitor 40. Therefore, it is preferable to monitor the power supply capacitor 40 by the voltage detector 20 or the like. When the detected value of the voltage detector 20 rises to the specified voltage, the switching element is controlled so as to shift from the state of the pattern 15 or the pattern 17 to the state of another pattern. For example, if to reflux motor current I _m in the interior of the power conversion circuit 50, the excess energy can be consumed by the body diode or the internal resistance, it is possible to avoid damage to the equipment in advance ..

Note that FIG. 20 shows a processing flow when the motor current I _m is flowing in the second direction. However, when the motor current I _m is flowing in the first direction, each pattern is Is changed to a corresponding pattern, and the similar processing flow is obtained. When the motor current _{I m} flows in the first direction, instead of the respective patterns 10,11,13,15,17, pattern 1,3,5,7,8 is used.

The examples of FIGS. 19 and 20 were examples of operation by the PWM signal generator 38B shown in FIG. 6, that is, examples of application to the one-sided PWM system. The method of the first embodiment can also be applied to the double-sided PWM method. Hereinafter, an application example to the double-sided PWM system will be described with reference to FIGS. 21 and 22. FIG. 21 is a time chart showing an example of waveforms of the PWM signals Q1 to Q4 generated by the PWM signal generation unit 38A shown in FIG. FIG. 22 is a flowchart showing an operation flow when the PWM signal generation unit 38A shown in FIG. 4 generates a PWM signal.

In the upper part of FIG. 21, the waveform of the positive side voltage command V _m1 output from the adder 38e of the PWM signal generator 38A and the negative side voltage command V _m2 output from the multiplier 38f of the PWM signal generator 38A are shown. And the waveforms of the PWM signals Q1 to Q4 generated by the output selection unit 38p of the PWM signal generation unit 38A. Further, the lower part of FIG. 21 shows an enlarged waveform of the part surrounded by the broken-line square frame K2 in the upper part of FIG. Further, each of the two dashed vertical lines penetrating the inside of the frame K2 corresponds to each of the two dashed vertical lines shown in the lower part. As shown in FIGS. 21 and 22, the output selection unit 38p switches the processing between the A section and the B section. The section A is a section in which the amplitude of the positive voltage command V _m1 is a positive value. The section B is a section in which the amplitude of the negative voltage command V _m2 is a positive value. The determination as to whether it is the A section or the B section is performed by using the phase θ _v input to the output selection unit 38p. Specifically, when the phase θ _v is 0°≦θ _v <180°, it is determined as “A section”. When the phase θ _v is 180°≦θ _v <360°, it is determined as “B section”. The output selection unit 38p switches the processing according to the phase θ _v .

Next, the processing of the PWM signal generation unit 38A in the first embodiment will be described with reference to the waveform of FIG. 21 and the flowchart of FIG. The main body of the operation in the flowchart of FIG. 22 is the PWM signal generation unit 38A. In the following description, the motor current I _m is assumed to flow in the second direction. Further, in the flowchart of FIG. 22, it is assumed that the state immediately after the start is the state in which the inverter 11 is controlled by the pattern 10 which is the powering pattern.

First, in FIG. 22, among the processes of steps S100 to S134, the processes of steps S100 to S118 generate the PWM signals Q1 to Q4 of the part surrounded by the frame K3 in the upper part of FIG. .. Further, the PWM signals Q1 to Q4 of the portion surrounded by the frame K2 in the upper part of FIG. 21 are generated by the processing of steps S119 to S134. In addition, in FIG. 22, the processing from step S100 to step S118 is the destination when the second voltage command V _m4 is replaced with the negative side voltage command V _m2 and the determination in step S103 is “No”. 20 is that step S119 is different from FIG. Except for these differences, the processing is the same as that in FIG. 20, and the description here is omitted.

In step S118, the process of shifting to pattern 10 is performed. When the process of step S118 ends, the process proceeds to step S119.

In step S119, the carrier and the positive voltage command V _m1 are compared. When the carrier does not exceed the positive voltage command V _m1 , the process returns to step S102. When the carrier exceeds the positive voltage command V _m1 , the process proceeds to step S120. In step S120, the process of shifting to pattern 12 is performed. In step S121, it is determined whether the time Ts12 has elapsed. By steps S120 and S121, the state of the pattern 12 is continued for the time Ts12. When the process of step S121 ends, the process proceeds to step S122.

In step S122, the process of shifting to pattern 14 is performed. In step S123, it is determined whether the time Ts14 has elapsed. By steps S122 and S123, the state of the pattern 14 is continued for the time Ts14. When the process of step S123 ends, the process proceeds to step S124.

In step S124, the process of shifting to pattern 16 is performed. In step S125, it is determined whether the time Ts16 has elapsed. By steps S124 and S125, the state of the pattern 16 is continued for the time Ts16. When the process of step S125 ends, the process proceeds to step S126.

In step S126, the process of shifting to pattern 17 is performed. In step S127, it is determined whether the time Ts17 has elapsed. By steps S126 and S127, the state of the pattern 17 is continued for the time Ts17. When the process of step S127 ends, the process proceeds to step S128.

In step S128, the process of shifting to pattern 16 is performed. In step S129, it is determined whether the time Ts16' has elapsed. By steps S128 and S129, the state of the pattern 16 is continued for the time Ts16'. When the process of step S129 ends, the process proceeds to step S130.

In step S130, the process of shifting to pattern 14 is performed. In step S131, the carrier is compared with the positive voltage command V _m1 . When the carrier is not below the positive voltage command V _m1 , the process returns to step S130 and the state of the pattern 13 is continued. If the carrier is below the second voltage command V _m4 , the process proceeds to step S132.

In step S132, the process of shifting to pattern 12 is performed. In step S133, it is determined whether the time Ts12' has elapsed. By steps S132 and S133, the state of the pattern 12 is continued for the time Ts12'. When the process of step S133 ends, the process proceeds to step S134.

In step S134, the process of shifting to pattern 10 is performed. When the process of step S134 ends, the process returns to step S100. After that, the processing from step S100 is repeated.

As described above, in the process according to the flowchart of FIG. 22, after the state of the pattern 10 is changed to the states of the pattern 11 and the pattern 13 in this order, the state is further changed to the states of the pattern 15 and the pattern 17 in this order. I am trying. That is, instead of recirculating the motor current I _m between the motor windings 41 and each of the switching elements, by shifting to the state of the pattern 15 and pattern 17, recovering power by the motor current I _m to the power supply capacitor 40 I am making it. This makes it possible to reduce the power consumption of the motor drive device 2.

Further, in the process according to the flowchart of FIG. 22, after shifting from the state of the pattern 10 to the respective states of the pattern 12 and the pattern 14 in this order, the state further shifts to the respective states of the pattern 16 and the pattern 17 in this order. There is. That is, instead of recirculating the motor current I _m between the motor windings 41 and each of the switching elements, by shifting to the state of the pattern 16 and pattern 17, recovering power by the motor current I _m to the power supply capacitor 40 I am making it. This makes it possible to reduce the power consumption of the motor drive device 2.

In the process of FIG. 22, the time Ts12, Ts14, Ts16, Ts17, Ts16', Ts14', Ts12' when each pattern is continued changes appropriately depending on the motor parameter. Therefore, it is necessary to examine or investigate an appropriate time in advance by experiments or simulations.

The times Ts12 and Ts12', the times Ts14 and Ts14', and the times Ts16 and Ts16' may be the same or different. Further, it is necessary to make the total time of these times and the delay time T _sx required for switching the switching element shorter than the half cycle of the carrier. Furthermore, it is desirable that the times Ts14 and Ts16 in which the motor current I _m is consumed as heat be made as short as possible. Since the pattern 16 has a larger loss due to the body diode of the switching element than the pattern 17, the time Ts16 is preferably shorter than the time Ts17.

As described above, even in the double-sided PWM system, it is possible to carry out the same processing as in the single-sided PWM system.

When the parasitic inductance between the battery 10 as a power source and the power supply capacitor 40 connected in parallel is large, the supply of power from the battery 10 to the power supply capacitor 40 is delayed due to the parasitic inductance. In this case, when the output to the load is large and large, a large amount of power is required, and the delay of the parasitic inductance cannot be ignored. Even in such a case, by applying the method of the first embodiment, the electric power of the motor current I _m can be reused without being consumed as heat. As a result, it becomes possible to stably supply electric power to the load.

When an external storage battery such as a battery is used as the power supply and the power supply capacitor is mounted on the inverter board as in the example of the first embodiment, the physical wiring distance between the battery and the inverter board is reduced. Occurs. In the case of such a design, since the parasitic inductance is always generated, the effect of using the method of the first embodiment can be expected. Further, since the electric power of the motor current I _m can be efficiently used, it is possible to reduce the capacity of the power supply capacitor mounted on the inverter board. Therefore, it can also contribute to the size reduction and weight reduction of the inverter board.

In addition, the control according to the first embodiment is assumed to be performed regardless of whether the single-phase motor 12 is operating or is stopped. In particular, during operation, the operation during the dead time in switching the switching element is included. In the case of high carrier control such that the ratio of the carrier cycle to the motor rotation cycle is 5 times or more, the dead time becomes a large loss. On the other hand, the first embodiment is a method of intentionally using the drive pattern of the switching element with a small current loss, and the period corresponding to the dead time can be shortened. This makes it possible to reduce the loss even with high carrier control.

In the first embodiment, the PWM control for comparing the applied voltage of the inverter 11 with the carrier amplitude and the voltage command value has been described, but the present invention is not limited to this. There are many methods of controlling the motor other than the PWM control, and any method may be used. The method according to the first embodiment positively controls the recovery of the motor current _Im into the power supply capacitor 40 when the voltage application to the motor winding 41 is stopped. Therefore, any type of motor control can be implemented.

As described above, the motor drive device according to the first embodiment includes the inverter that converts the DC voltage output from the power supply into the AC voltage and applies the AC voltage to the single-phase motor. The inverter includes a first leg in which a first switching element of an upper arm and a second switching element of a lower arm are connected in series, a third switching element of an upper arm and a fourth switching element of a lower arm. And have a second leg connected in series. In the first period, the motor drive device performs the first control of controlling the second and third switching elements to be conductive after controlling the first and fourth switching elements to be non-conductive. The first period is a period in which the application of the AC voltage to the single-phase motor is stopped and the motor current is flowing in the first direction from the first connection point toward the second connection point. The first connection point is a connection point between the first switching element and the second switching element. The second connection point is a connection point between the third switching element and the fourth switching element. By the first control performed by the motor drive device, the energy of the motor current flowing through the single-phase motor can be efficiently recovered. This makes it possible to obtain a motor drive device that can efficiently use the motor current and reduce power consumption.

In the first period, the motor drive device controls the first, second, and third switching elements to be non-conductive and the fourth switching element to be conductive, and then controls the second switching element. The first control may be performed by controlling to be conductive, then controlling the fourth switching element to be non-conductive, and then controlling the third switching element to be conductive. By doing so, it is possible to reduce the loss in the body diode while avoiding the phenomenon of simultaneously conducting a plurality of switching elements. Thereby, the energy of the motor current can be efficiently collected while preventing the upper and lower arm short circuits.

Further, when the detected value of the motor current falls below the specified current during the execution of the first control, the motor drive device controls the fourth switching element to be conductive after controlling the third switching element to be non-conductive. After that, the fourth switching element may be controlled to be non-conductive, and then the third switching element may be controlled to be conductive. By doing so, a boosted voltage can be generated in the motor winding 41. Thereby, the energy of the motor current can be efficiently recovered.

Further, when the detection value of the voltage detector rises to the specified voltage during the execution of the first control, the motor drive device controls the first and third switching elements to be non-conducting, and the second and fourth switching elements are turned off. The switching element may be controlled to be conductive. By doing so, it is possible to prevent the voltage of the electric storage device that recovers the energy of the motor current from rising. As a result, it is possible to prevent the voltage of the device connected to the electric storage device from exceeding the withstand voltage, and prevent damage to the device in advance.

Further, the motor drive device according to the first embodiment controls the first and fourth switching elements to be conductive after controlling the second and third switching elements to be non-conductive in the second period. Control is performed. The second period is a period in which the application of the AC voltage to the single-phase motor is stopped and the motor current is flowing in the second direction from the second connection point toward the first connection point. By the second control performed by the motor drive device, the energy of the motor current flowing through the single-phase motor can be efficiently recovered. As a result, it is possible to obtain a motor drive device that can efficiently use the energy of the motor current and reduce power consumption.

In the second period, the motor drive device controls the first, third, and fourth switching elements to be non-conductive, and the second switching element to be conductive, and then the fourth switching element. The second control may be performed by controlling to be conductive, then controlling the second switching element to be non-conductive, and then controlling the first switching element to be conductive. By doing so, it is possible to reduce the loss in the body diode while avoiding the phenomenon of simultaneously conducting a plurality of switching elements. Thereby, the energy of the motor current can be efficiently collected while preventing the upper and lower arm short circuits.

Further, when the detected value of the motor current falls below the specified current during the execution of the second control, the motor drive device controls the second switching element to be conductive after controlling the first switching element to be non-conductive. After that, the second switching element may be controlled to be non-conductive, and then the first switching element may be controlled to be conductive. By doing so, a boosted voltage can be generated in the motor winding 41. Thereby, the energy of the motor current can be efficiently recovered.

In addition, when the detection value of the voltage detector rises to the specified voltage during the execution of the second control, the motor drive device controls the first and third switching elements to be non-conducting, and the second and fourth switching elements are turned off. The switching element may be controlled to be conductive. By doing so, it is possible to prevent the voltage of the electric storage device that recovers the energy of the motor current from rising. As a result, it is possible to prevent the voltage of the device connected to the electric storage device from exceeding the withstand voltage, and prevent damage to the device in advance.

Next, a hardware configuration for realizing the function of the control unit 25 in the first embodiment will be described with reference to the drawings of FIGS. 23 and 24. FIG. 23 is a block diagram showing an example of a hardware configuration that realizes the function of the control unit 25 according to the first embodiment. FIG. 24 is a block diagram showing another example of the hardware configuration that realizes the function of the control unit 25 in the first embodiment.

When a part or all of the functions of the control unit 25 in the first embodiment is realized, as shown in FIG. 23, a processor 200 that performs an operation, a memory 202 that stores a program read by the processor 200, And an interface 204 that inputs and outputs signals.

The processor 200 may be an arithmetic unit such as an arithmetic unit, a microprocessor, a microcomputer, a CPU (Central Processing Unit), or a DSP (Digital Signal Processor). Further, the memory 202 is a nonvolatile or volatile semiconductor memory such as a RAM (Random Access Memory), a ROM (Read Only Memory), a flash memory, an EPROM (Erasable Programmable ROM), and an EEPROM (registered trademark) (Electrically EPROM). Examples thereof include a magnetic disk, a flexible disk, an optical disk, a compact disk, a mini disk, and a DVD (Digital Versatile Disc).

The memory 202 stores a program that executes the function of the control unit 25 according to the first embodiment. The processor 200 sends and receives necessary information via the interface 204, the processor 200 executes the program stored in the memory 202, and the processor 200 refers to the table stored in the memory 202 to perform the above-described processing. It can be carried out. The calculation result by the processor 200 can be stored in the memory 202.

Further, when a part of the functions of the control unit 25 in the first embodiment is realized, the processing circuit 203 shown in FIG. 24 can be used. The processing circuit 203 corresponds to a single circuit, a composite circuit, an ASIC (Application Specific Integrated Circuit), an FPGA (Field-Programmable Gate Array), or a combination thereof. Information input to the processing circuit 203 and information output from the processing circuit 203 can be obtained via the interface 204.

Note that a part of the processing in the control unit 25 may be executed by the processing circuit 203, and processing not executed by the processing circuit 203 may be executed by the processor 200 and the memory 202.

Embodiment 2.
In the second embodiment, an application example of the motor drive device 2 described in the first embodiment will be described.

FIG. 25 is a diagram showing a configuration example of the electric blower 64 according to the second embodiment. The electric blower 64 includes the motor drive device 2 described in the first embodiment, and the propeller 69 is attached to the single-phase motor 12 driven by the motor drive device 2. The electric blower 64 has a structure in which the motor drive device 2 rotates the single-phase motor 12 to blow out or suck the wind.

26: is a figure which shows the structural example of the electric vacuum cleaner 61 provided with the electric blower 64 which concerns on Embodiment 2. As shown in FIG. The electric vacuum cleaner 61 includes a battery 67 corresponding to the battery 10 shown in FIG. 1, a motor drive device 2 shown in FIG. 1, and an electric blower 64 driven by the single-phase motor 12 shown in FIG. .. Further, the electric vacuum cleaner 61 includes a dust collection chamber 65, a sensor 68, a suction port body 63, an extension pipe 62, and an operation unit 66.

A user who uses the electric vacuum cleaner 61 holds the operation unit 66 and operates the electric vacuum cleaner 61. The motor drive device 2 of the electric vacuum cleaner 61 drives the electric blower 64 using the battery 67 as a power source. By driving the electric blower 64, dust is sucked from the suction port body 63. The sucked dust is collected in the dust collecting chamber 65 via the extension pipe 62.

The electric vacuum cleaner 61 is a product that uses a battery 67 as a power source. By using the method of the first embodiment described above, it is possible to efficiently use the motor current I _m and reduce the power consumption. As a result, the battery 67 can be used for a long time, and the vacuum cleaner 61 that can be used for a long time can be obtained.

Further, in the electric vacuum cleaner 61 according to the second embodiment, the switching elements 51 to 54 of the inverter 11 are formed of wide bandgap semiconductors, so that the heat dissipation parts can be simplified and the size and weight can be reduced.

FIG. 27 is a diagram showing a configuration example of a hand dryer 90 including the electric blower 64 according to the second embodiment. The hand dryer 90 includes a casing 91, a hand detection sensor 92, a water receiving portion 93, a drain container 94, a cover 96, a sensor 97, an intake port 98, and an electric blower 64. Here, the sensor 97 is either a gyro sensor or a motion sensor. In the hand dryer 90, the water is blown off by the air blow by the electric blower 64 by inserting the hand into the hand insertion part 99 located above the water receiving part 93, and the blown off water is collected by the water receiving part 93. After that, it is stored in the drain container 94.

The hand dryer 90 is a product that is used all year round regardless of the season. Therefore, by adopting the product to which the method of the first embodiment is applied, the effect of continuously reducing the power consumption can be obtained.

As described above, in the present embodiment, the configuration example in which the motor drive device 2 is applied to the electric vacuum cleaner 61 and the hand dryer 90 has been described. However, the motor drive device 2 is applied to an electric device equipped with a motor. can do. Electric devices equipped with motors include incinerators, crushers, dryers, dust collectors, printing machines, cleaning machines, confectionery machines, tea making machines, woodworking machines, plastic extruders, cardboard machines, packaging machines, hot air generators, and OA. Equipment, electric blowers, etc. The electric blower is a blowing means for object transportation, dust suction, or general ventilation.

The configurations described in the above embodiments are examples of the content of the present invention, and can be combined with other known techniques, and the configurations of the configurations are not departing from the scope of the present invention. It is also possible to omit or change parts.

DESCRIPTION OF SYMBOLS 1 motor drive system, 2 motor drive device, 5A 1st leg, 5B 2nd leg, 6A, 6B connection point, 10 battery, 11 inverter, 12 single-phase motor, 12a rotor, 12b stator, 20 voltage detector, 22 current detector, 25 control section, 32 drive signal generation section, 33 carrier generation section, 38, 38A, 38B PWM signal generation section, 38a absolute value calculation section, 38b division section, 38c, 38d, 38f, 38k multiplication section, 38e, 38m, 38n adding section, 38g, 38h comparing section, 38i, 38j output inverting section, 38p, 38q output selecting section, 40 power supply capacitor, 41 motor winding, 42 voltage source, 50 power conversion circuit, 51, 52, 53, 54 switching element, 51a, 52a, 53a, 54a body diode, 61 vacuum cleaner, 62 extension tube, 63 suction port body, 64 electric blower, 65 dust collection chamber, 66 operation part, 68, 97 sensor, 90 hand Dryer, 91 Casing, 92 Hand detection sensor, 93 Water receiver, 94 Drain container, 96 cover, 98 Air inlet, 99 Hand insertion part, 102 switch, 200 processor, 202 memory, 203 processing circuit, 204 interface.

Claims (19)

A first leg in which a first switching element of the upper arm and a second switching element of the lower arm are connected in series; and a third switching element of the upper arm that is connected in parallel to the first leg. An inverter having a second leg in which a fourth switching element of a lower arm is connected in series and converting a DC voltage output from a power supply into an AC voltage and applying the AC voltage to a single-phase motor. When,
A current detector for detecting a motor current flowing through the single-phase motor,
Before SL first, a second, control unit for controlling conduction of the third and fourth switching elements,
Equipped with
The single-phase motor has a first connection point that is a connection point between the first switching element and the second switching element, and a connection point between the third switching element and the fourth switching element. Connected to a second connection point,
In the first period, the control unit performs the first control of controlling the second and third switching elements to be conductive after controlling the first and fourth switching elements to be non-conductive.
In the first period, application of the AC voltage to the single-phase motor is stopped, and the motor current flows in a first direction from the first connection point to the second connection point. It is a period that is a motor drive. Controls the single-phase motor based on the detection value of the current detector
The motor drive device according to claim 1. A voltage detector connected in parallel to the first leg and the second leg;
Controls the single-phase motor based on the detection value of the voltage detector
The motor drive device according to claim 1. In the first period,
The control unit controls the first, second, and third switching elements to be non-conductive, controls the fourth switching element to be conductive, and then controls the second switching element to be conductive, and then ,
The fourth switching element is controlled to be non-conductive, and thereafter,
The motor drive device according to claim 1, wherein the first control is performed by controlling the third switching element to be conductive. When the detected value of the motor current is lower than the specified current during the execution of the first control,
The control unit controls the fourth switching element to be conductive after controlling the third switching element to be non-conductive, and thereafter,
The fourth switching element is controlled to be non-conductive, and thereafter,
The motor drive device according to claim 4 , wherein the third switching element is controlled to be conductive. A power storage device that stores electric power supplied from the power source and the single-phase motor,
A voltage detector for detecting the voltage of the capacitor,
When the detection value of the voltage detector rises to a specified voltage during the execution of the first control,
The motor drive according to any one of claims 1 to 5 , wherein the control unit controls the first and third switching elements to be non-conductive and the second and fourth switching elements to be conductive. apparatus. The motor drive device according to claim 6 , wherein the control unit controls the second switching element to be non-conductive after controlling the second and fourth switching elements to be conductive. A first leg in which a first switching element of the upper arm and a second switching element of the lower arm are connected in series; and a third switching element of the upper arm that is connected in parallel to the first leg. An inverter having a second leg in which a fourth switching element of a lower arm is connected in series and converting a DC voltage output from a power supply into an AC voltage and applying the AC voltage to a single-phase motor. When,
A current detector for detecting a motor current flowing through the single-phase motor,
Before SL first, a second, control unit for controlling conduction of the third and fourth switching elements,
Equipped with
The single-phase motor has a first connection point that is a connection point between the first switching element and the second switching element, and a connection point between the third switching element and the fourth switching element. Connected to a second connection point,
The control unit performs second control of controlling the first and fourth switching elements to be conductive after controlling the second and third switching elements to be non-conductive in a second period,
During the second period, the application of the AC voltage to the single-phase motor is stopped, and the motor current flows in the second direction from the second connection point toward the first connection point. It is a period that is a motor drive. Controls the single-phase motor based on the detection value of the current detector
The motor drive device according to claim 8. A voltage detector connected in parallel to the first leg and the second leg;
Controls the single-phase motor based on the detection value of the voltage detector
The motor drive device according to claim 8. In the second period,
The control unit controls the first, third, and fourth switching elements to be non-conductive, controls the second switching element to be conductive, and then controls the fourth switching element to be conductive, and then ,
The second switching element is controlled to be non-conductive, and thereafter,
The motor drive device according to claim 10 , wherein the second control is performed by controlling the first switching element to be conductive. When the detected value of the motor current is lower than the specified current during execution of the second control,
The control unit controls the second switching element to be conductive after controlling the first switching element to be non-conductive, and thereafter,
The second switching element is controlled to be non-conductive, and thereafter,
The motor drive device according to claim 11 , wherein the first switching element is controlled to be conductive. A power storage device that stores electric power supplied from the power source and the single-phase motor,
A voltage detector for detecting the voltage of the capacitor,
When the detection value of the voltage detector rises to a specified voltage during the execution of the second control,
Wherein, said first and third switching elements is controlled in non-conductive, the motor drive according to any one of claims 10 to 12 for controlling the second and fourth switching elements to a conducting apparatus. The motor drive device according to claim 13 , wherein the control unit controls the fourth switching element to be non-conductive after controlling the second and fourth switching elements to be conductive. It said first, second, third and at least one motor driving apparatus according to any one of claims 1 to 14, which is formed by the wide band gap semiconductor of the fourth switching element. The motor drive device according to claim 15 , wherein the wide band gap semiconductor is silicon carbide, gallium nitride, gallium oxide, or diamond. An electric blower provided with the motor drive device according to any one of claims 1 to 16 . An electric vacuum cleaner comprising the electric blower according to claim 17 . A hand dryer comprising the electric blower according to claim 17 .

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