Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04R—LOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
  • H04R1/00—Details of transducers, loudspeakers or microphones
  • H04R1/20—Arrangements for obtaining desired frequency or directional characteristics
  • H04R1/32—Arrangements for obtaining desired frequency or directional characteristics for obtaining desired directional characteristic only
  • H04R1/40—Arrangements for obtaining desired frequency or directional characteristics for obtaining desired directional characteristic only by combining a number of identical transducers
  • H04R1/403—Arrangements for obtaining desired frequency or directional characteristics for obtaining desired directional characteristic only by combining a number of identical transducers loud-speakers
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04R—LOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
  • H04R1/00—Details of transducers, loudspeakers or microphones
  • H04R1/20—Arrangements for obtaining desired frequency or directional characteristics
  • H04R1/32—Arrangements for obtaining desired frequency or directional characteristics for obtaining desired directional characteristic only
  • H04R1/40—Arrangements for obtaining desired frequency or directional characteristics for obtaining desired directional characteristic only by combining a number of identical transducers
  • H04R1/406—Arrangements for obtaining desired frequency or directional characteristics for obtaining desired directional characteristic only by combining a number of identical transducers microphones
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04R—LOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
  • H04R3/00—Circuits for transducers, loudspeakers or microphones
  • H04R3/12—Circuits for transducers, loudspeakers or microphones for distributing signals to two or more loudspeakers
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04R—LOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
  • H04R2201/00—Details of transducers, loudspeakers or microphones covered by H04R1/00 but not provided for in any of its subgroups
  • H04R2201/40—Details of arrangements for obtaining desired directional characteristic by combining a number of identical transducers covered by H04R1/40 but not provided for in any of its subgroups
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04R—LOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
  • H04R2430/00—Signal processing covered by H04R, not provided for in its groups
  • H04R2430/20—Processing of the output signals of the acoustic transducers of an array for obtaining a desired directivity characteristic
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04R—LOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
  • H04R3/00—Circuits for transducers, loudspeakers or microphones
  • H04R3/005—Circuits for transducers, loudspeakers or microphones for combining the signals of two or more microphones

Definitions

• the present invention is concerned with the calculation of FIR filter coefficients for beamforming filters of a transducer array, such as e.g. an array of microphones or speakers.
• Beamforming technologies such as those used in the audio field, specify - in the case of a microphone array for the evaluation of the individual signals of the microphones or in the case of a loudspeaker array for the reproduction of the signals of the individual speakers - as the signals of an individual Filtering with a respective discrete-time filter are subjected.
• For broadband applications such as For this purpose, from the specification of the optimal frequency response coefficients for these discrete-time filters are determined.
• the resulting FIR filters accurately map the given frequency response in the frequency raster given by the DFT, but between the raster points the frequency response can take on arbitrary values. This often leads to useless designs with strong oscillations of the resulting frequency response. Furthermore, in frequency sampling design, the length of the FIR filter automatically results from the resolution of the frequency response specification (and vice versa). Filters created using Frequency Sampling Design are susceptible to temporal aliasing (time-domain aliasing), ie a periodic convolution of impulse responses (eg [Smi11]). Additional techniques such as zero-padding the DFTs or windowing the generated FIR filters may need to be used.
• An alternative approach is to determine the FIR coefficients in a one-step process directly in the tiling area [MDK11].
• the radiation behavior of the array for a given pattern of frequencies is represented directly as a function of the FIR coefficients of all transducers (ie loudspeakers / microphones) and formulated as a single optimization problem, by the solution of the optimal filter coefficients for all beamforming filters are determined simultaneously.
• the problem here is the size of the optimization problem, both in terms of the number of variables to be optimized (filter length times the number of beamforming filters) and with respect to the dimension of the determination equations and, if necessary, secondary conditions.
• the latter dimension is usually proportional to both the number of frequency raster points and the spatial resolution used to determine the desired beamformer response.
• the object of the present invention is to provide a concept for calculating FIR filter coefficients for the beamforming filters of a transducer array that is more effective, such as in relation to the ratio between achieved beamforming quality and the amount of computation required. This object is solved by the subject matter of the appended independent claim.
• One idea on which the present application is based is to have recognized that the effectiveness of calculating FIR filter coefficients for beamforming filters for transducer arrays, such as arrays of microphones or loudspeakers, can be increased if the calculation is performed in two stages is, on the one hand by calculating frequency-domain filter weights of the beamforming filters in a predetermined frequency raster, ie coefficients describing the transmission function of the beamforming filters in the frequency domain or for a respective frequency or for a sinusoidal input signal with a respective frequency, to target frequency responses for the beamforming filters such that application of the beamforming filters to the array approximates a desired direction selectivity, and followed by a calculation of the FIR filter coefficients for the beamforming filters, ie coefficients representing the impulse response of the beamforming filters in the array Describe time domain such that frequency responses of the beamforming filters approximate the intermediate frequency responses.
• the two-step nature allows for independent choice of the frequency resolution resulting from the discrete Fourier transform of the Implus responses described by the Fl R filter coefficients. Furthermore, specific secondary conditions can be specified both in the calculation of the beamforming drive weights in the frequency domain and in the calculation of the time domain FIR filter coefficients in order to influence the respective calculation in a targeted manner.
• Fig. 1 shows a schematic block diagram of a loudspeaker array with beamforming filters for which the embodiments of the present application could be used
• Fig. 2 shows a schematic block diagram of an icon array with beamforming filters for which embodiments of the present application could be used;
• Fig. 3 is a block diagram of an apparatus for calculating Fl R filter coefficients for the beamforming filters according to an embodiment
• FIG. 4 schematically illustrates how, in accordance with an embodiment of FIG. 3, the optimization-based calculation of the target frequency responses of the beamforming filters is performed stepwise via the modeling of a DSB design;
• Fig. 5 schematically illustrates how, according to an embodiment, the modification means arranged between the two calculation means in Fig. 3 makes the optimization target for the time domain optimization of the second calculation means more appropriate;
• FIG. 6 schematically illustrates how, in one embodiment, the delays taken in the delay adjustment module of FIG. 3 by phase leveling may be re-integrated into the calculated FIR filter coefficients
• the target frequency response is composed of an optimized component into a low-frequency segment and a DSB transfer function in a high-frequency segment.
• FIG. 1 shows first an example of an array 10 of loudspeakers 12 which, by the use of beamforming filters (BFF) 14, is to be enabled to have a desired directional selectivity, ie to radiate in a certain direction 16, for example.
• BFF beamforming filters
• the number N of speakers 12 may be two or more.
• the loudspeaker 12 n is connected to a common audio input 18 via its corresponding beamforming filter 14 n .
• the audio signal s () at the input 18 is a discrete-time audio signal consisting of a sequence of audio samples and the beamforming filters 14 n are configured as FIR filters and thus fold the audio signal with the impulse response of the respective beamforming filter 14 n as defined by the FIR filter coefficients of the respective beamforming filter 14 ".
• Fig. 1 illustrates only by way of example that the loudspeakers 12 n are equidistant in a line are arranged and the array 10 is a linear array of speakers.
• a two-dimensional arrangement of loudspeakers would also be conceivable, as well as a non-uniform distribution of the loudspeakers 12 in the array 10 and also an arrangement deviating from an arrangement along a straight line or a plane.
• the emission direction 16 can be measured, for example, by an angular deviation of the direction 16 from a mid-perpendicular of the straight lines or the surface along which the loudspeakers 12 are arranged.
• the abstraction should preferably be audible at a particular location in front of the array 10.
• the filter coefficients h of the beamforming filters 14 n can also be chosen more precisely such that the directional characteristic or directional selectivity of the array 10 not only reaches a maximum in a particular direction 16 when it comes to abstraction, but also satisfies other desired criteria, such as an angular beamwidth, a particular frequency response in a direction 16 of maximum abstraction or even a certain frequency response as far as an area comprising the direction 16 and directions around it is concerned.
• Fig. 2 shows such a microphone array.
• the microphone array of Fig. 2 is exemplarily also provided with the reference numeral 10, but in each case consists of microphones 20 1 ... 20 N together.
• each microphone generates a received audio signal s BFFi and is connected via a respective beamforming filter 14 n to a common output node 22 for outputting the received output signal s ', so that the filtered audio signals S' BFFII of the beamforming filter 14 n additively to the audio signal s contribute.
• an adder 24 is connected between the outputs of the beamforming filters 14 n and the common output node 22.
• the beamforming filters are configured again as FIR filters and form from the respective audio signal of the respective microphone 20 n , ie S 'BFF " ⁇ the filtered audio signal s BFF according to, for example
• the microphone array 10 in FIG. 2 makes it possible for the microphone array 10 in FIG. 2 to have a desired directional selectivity or directional characteristic in order to primarily or exclusively record the sound scene from a particular direction 16, so that it is sensitive in the output Signal s' is reflected, wherein the direction 16 again as in the case of Fig. 1 by the angulare deviation ⁇ or in the two-dimensional case ⁇ and ⁇ can be defined by a perpendicular line of the array 10 and the desired direction selectivity may optionally be more accurate than merely an indication of a direction of maximum sensitivity, namely more precisely with regard to the spatial dimension or frequency dimension.
• 3 shows an embodiment of a device for calculating FIR filter coefficients for the beamforming filters of a transducer array, such as an array of microphones, as shown for example in FIG. 2, or loudspeakers, as shown for example in FIG was shown.
• the device is indicated generally at 30 and may be implemented, for example, in software executed by a computer, in which case all of the devices and modules described below may be, for example, different sections of a computer program.
• An implementation in the form of a dedicated hardware e.g. in the form of an ASIC or in the form of a programmable logic circuit, e.g. an FPGA, however, is also possible.
• the device 30 calculates the Fl R filter coefficients 32, such as those just mentioned above for the beamforming filters 14 n , specifically for the array 10, for which purpose the device 30 has interfaces for obtaining information about the array 10 or information about the array 10 To obtain desired directional selectivity.
• FIG. 3 shows by way of example that the device 30 receives transducer data 34 from outside, which will be described in more detail below, for example the positions and orientations of the transducer elements, ie for example the loudspeakers or microphones, and their individual direction-selective sensitivity or Specify emission characteristics and / or the frequency response. Other information relates, for example, to the desired directional selectivity.
• FIG. 3 shows by way of example that the device 30 receives transducer data 34 from outside, which will be described in more detail below, for example the positions and orientations of the transducer elements, ie for example the loudspeakers or microphones, and their individual direction-selective sensitivity or Specify emission characteristics and / or the frequency response.
• the device 30 receives data 36 which indicate the desired directional behavior of the array 10, such as a direction of maximum radiation and, if appropriate, more precise information, such as the radiation behavior or the sensitivity around it mentioned maximum radiation / sensitivity around.
• the data 36 are supplemented, for example, by further data 38 which can be specified from the outside of the device 30 and, for example, the desired transfer characteristic or the frequency response of the array 10 in the direction of the radiation or sensitivity of the array 10, ie the frequency-dependent desired description the sensitivity or emission intensity of the array adjusted with the final FIR filter coefficients in a certain direction or directions.
• Fl R filter coefficients 32 Other information may also be provided to the device 30 for calculating the Fl R filter coefficients 32, such as specifications regarding a robustness of the calculated FIR filter coefficients to be maintained against deviations of the transducer data 34 from actual physical conditions of an actually constructed array 10, which 3 are denoted by the reference numeral 40, as well as data on a frequency restriction, 42, whose exemplary meaning for the calculation will be described below and possibly related to the transducer data 34.
• the device 30 could also be designed specifically for a particular array setup, and it would also be possible for the device to be designed specifically for particular settings of the other data. In the case of an input facility, this may be implemented via an input interface, such as via user input interfaces of a computer or read interfaces of a computer, such that, for example, the data is read from one or more particular files.
• the device of FIG. 3 comprises a first calculation device 44 and a second calculation device 46.
• the first calculation device 44 calculates frequency range filter weights of the beamforming filters, ie complex-valued samples of the transfer function of the beamforming filters.
• the first calculating means 44 calculates the frequency-domain driving weights in a frequency raster defined by certain frequencies ⁇ ( ... ⁇ ⁇ , which are not necessarily equidistant from each other, so that they describe a transfer function H m , ⁇ of the beamforming filters which
• the first calculation means uses, for example, a suitable optimization algorithm, such as e.g. a method for solving linear, quadratic or convex optimization problems.
• the frequency grid may be in accordance with requirements of the beamforming application, such as e.g. different requirements for the accuracy of the radiation specification in certain frequency ranges, or according to other requirements, such as. of the subsequent and hereinafter mentioned FIR time domain design method, such as e.g. depending on a necessary sampling rate for the definition of the desired frequency response.
• the second calculation means 46 is to determine the FIR filter coefficients of the beamforming filter, which describe the impulse response of the beamforming filter.
• the second calculation device 46 carries out the calculation in such a way that the frequency responses of the beamforming filters, as they correspond to the FIR filter coefficients via the relationship between transfer function and impulse response, approximate the target frequency responses predetermined by the first calculation device 44.
• the second calculation device 46 also uses optimization according to the following embodiment description, which can again be implemented as a method for solving linear, quadratic or convex optimization problems.
• the first calculation device 44 performs the calculation by solving a first optimization problem, according to which a deviation between a directional efficiency of the array, as reflected by the frequency range control weights H B / . (co k ), and the desired direction selectivity that may be predetermined by the data 34 and / or 38 is minimized.
• the first calculation means 44 may use the robustness specifications 40 as a constraint of the optimization problem, and the conversion data 34 may be used to determine the relationship between the optimization variables, namely the frequency range drive weights H , (j k ).
• the resulting directional selectivity on the other hand, set or define.
• the subsequent description will then turn to possible implementations of the second calculation means 46, which will result in the second calculation means also being able to solve an optimization problem to make the calculation.
• a deviation from the target frequency responses H BFF (iu k ) in the frequency range is minimized.
• the FIR filter coefficients to be calculated by the second calculator 46 correspond to the impulse response, and the means 46 tries to calculate them by optimization according to subsequent embodiments such that the transfer functions corresponding to these impulse responses correspond to the transfer functions H HFr ( ⁇ ⁇ ) as close as possible, as calculated by the first calculating means 44.
• the device 30 can optionally also include an optional modification device 50 for modifying the calculated FIR filter coefficients, as calculated by the second calculation device 46, so that the respective modification is taken into account.
• an optional modification device 50 for modifying the calculated FIR filter coefficients, as calculated by the second calculation device 46, so that the respective modification is taken into account.
• 3 will also exemplarily describe a possible modular structure for the first calculation device 44 and the target frequency response modification device 48, but the respective modu lar structure is only an example.
• the device 30 of Fig. 3 for finding the time domain FIR filters and the FIR filter coefficients for the beamforming filters, respectively can use solutions of optimization problems. Time-domain FIR filter computation using optimization-based filter design techniques, as described below, eliminates both the disadvantages of frequency sampling design and the complexity and hence the demands on computational time and resources, such as memory, for direct time-domain design of the filters. as described in the introductory part of the present application.
• the beamforming filters of FIG. 3 are designed in a two-stage process:
• the frequency grid does not have to be chosen equidistantly, but can also be irregular.
• an FIR filter is generated by applying its FIR filter coefficients h 'B , FF "are calculated. Optimization methods can also be used here to achieve the best possible approximation of the desired target frequency response for the given FIR filter arrangement, a freely selectable filter standard and optionally a number of additional secondary conditions.
• the frequency resolution of the FIR filter design defined by, for example, the Nyquist frequency of the FIR filter divided by half the FIR filter length, or, to a lesser extent, the Nyquist frequency (half the sampling rate) of the time-discrete system in which the Beamformer and thus the FIR filters are implemented, can be selected differently to the frequency resolution of the frequency domain resolution.
• the filter design process as implemented by the device 30, provides for a variety of individual interrelated measures and precautions in accordance with the embodiments described below. All in all, they enable the generation of particularly stable, robust control filters or beamforming filters. The operation of the device 30 will now be described in detail. However, some of the measures can also be left out depending on the application.
• the transducer characteristics ie the properties of, for example, microphones or loudspeakers
• the transducer data 34 describes the transducer characteristics that are typically derived from measurements or modeling, such as simulation.
• the transducer data 34 may represent, for example, the directional and frequency dependent transfer function of the transducers of (in the case of loudspeakers) and to (in the case of sensors or microphones, respectively) different points in space.
• a module 52 of the calculation device 44 may perform a direction characteristic interpolation, ie an interpolation of the transducer data 34, about the transfer function of the transducers from / to points or Allow directions that are not included under the original data 34, ie, not in the original records.
• a direction characteristic interpolation ie an interpolation of the transducer data 34
• the converter data of the module 52 thus obtained are used in two function blocks or modules 54 and 56 of the first calculation device 44, namely a delay and sum beamformer module and an optimization module 56.
• the delay and sum (delay and summation Beamformer module 54 calculates a delay and an amplitude weight, ie, for each transducer n, based on a desired specification for the directional behavior of the transducer array, which specifies, for example, a desired magnitude in the emission direction or emission directions, using the individual transducers of the array in the respective direction frequency-independent quantities, such as a time delay and a gain factor per converter 12 and 14.
• the optimization 56 operates in the frequency domain.
• transducer data 34 obtained by measurements often have delays, such as acoustic propagation, and a delay extraction module 58 connected between the directional characteristic interpolation module 52 and the optimization module 56 could be provided to remove common delay times of all transducer data , This simplifies the optimization process in the optimization module 56, because the delays no longer have to be included in the desired optimization target function, or the beamforming filters achieved do not have to compensate for the array delay common to the transducers. It will become apparent from the following description that an advantage of the use of the delay-and-sum block is that a design specification for the frequency response of the converter array in the desired radiation pattern which can be implemented with the converters used with high robustness is provided by the same. can be obtained with high sensitivity.
• the data 36 is converted, for example, by a module 60 into a target pattern specification, ie a mathematical formulation of the desired directional behavior.
• the objective function output by the target pattern specification 60 describes, for example, the desired complex sound radiation in different spatial directions ⁇ or ⁇ and ⁇ .
• the objective function can be either frequency-independent or frequency-dependent, ie with different specifications for different frequencies or frequency ranges.
• the mathematical formulation of the directional behavior may include one or more of the following: one or more preferred radiating directions or points;
• Areas in which the sound radiation is to be minimized optionally with a weighting function, to adjust the priority of individual partial areas.
• the desired complex sound radiation which is described by the objective function, is not necessarily limited to directions. Other arguments are also possible, for example, such as the desired radiation along a line or over a surface / volume.
• robustness denotes the property of only a relatively small deterioration of the abstraction behavior in the case of deviations of the transducer array 10 or of the transmission system, eg due to deviations of the control filters from ideal behavior, positioning errors of the transducers in the array or deviations from the modeled transmission behavior to show.
• a measure of robustness frequently used for, for example, microphone arrays is the so-called "White Noise Gain” [BW01, MSK09], ([WNG]), which is the quotient of the signal magnitude in the one-way direction and the L 2 standard of the drive weights for the This measure can also be usefully used for loudspeaker arrays [MK07], where the magnitude of the signal in the desired direction of radiation assumes the role of the magnitude in the direction of incidence.
• the magnitude in the direction of radiation (respectively direction of incidence) has a direct effect on the WNG and thus on the robustness in relation to an allowed standard of the drive weights.
• the achievable level in the emission direction depends both on the maximum permissible amount of the drive weights and on the emission characteristics of the converters. It is therefore necessary to specify the magnitude (or amplitude of the desired radiation pattern) so that both robustness and radiation amplitude requirements are achieved. To get a good starting point for this specification, it is possible to use the following procedure:
• module 54 Based on the desired emission direction or the data 36 for the desired directional behavior, the transfer function of the drive filters for the delay and sum beamformer (DSB) in the module 54 is created. That is, module 54 is based on simple filters which depend only on the position of the transducers and the emission direction and only on a frequency-independent basis
• Amplification value and a frequency-independent delay, each per transducer element exist. Only such frequency-independent values per converter are calculated by the module 54, starting from the desired directional behavior 36 and including the converter data 34.
• a DSB thus corresponds to the structure of FIGS. 1 and 2, although simpler BFFs are used, namely only those that have a time delay and perform a frequency-independent amplification. While the directivity of such a DSB is low, especially for low frequencies, it has high WNG values and thus good robustness.
• this DSB control filter set consisting of only the frequency-independent gain value and the frequency-independent delay per transducer element
• the radiation of the array in the desired emission direction is calculated / simulated. In the calculation of these frequency-independent value pairs per transducer element by the module 54, as already mentioned, the modeled or measured transducer characteristics flow from the data 34.
• the frequency response of the transducer array in the emission direction resulting from the DSB drive filter set of the module 54 may be referred to as the reference frequency response (or magnitude response) and used in the subsequent steps of the calculation by the calculation device 44.
• the advantage of this approach is that it provides a magnitude constraint that can be realized by the transducer array within the given maximum drive levels for the individual transducers, and that has (because of a DSB design) good robustness characteristics can, that it has good robustness properties.
• the calculation device 44 has yet another module, namely a module 58, which from the obtained reference magnitude response of the module 54 in combination with the specifications 38 for the desired frequency response or desired frequency response, the final specification of the frequency response the transducer array determined in the emission direction. That is, the starting point for the determination of the module 58 is the frequency response of the transducer array, as determined by the DS B values of the module 54, ie the frequency response that results for the transducer array with the DSB values in the corresponding direction . Based on this Magnituden- gear, 58 modifications are made by the module, for example, modifications to the reference magnitude response are made, for example, to equalize the frequency response.
• a module 58 which from the obtained reference magnitude response of the module 54 in combination with the specifications 38 for the desired frequency response or desired frequency response, the final specification of the frequency response the transducer array determined in the emission direction. That is, the starting point for the determination of the module 58 is the frequency response of the transducer array, as determined by the DS B values of the
• the directivity of the array can be increased within certain limits (either globally or for certain frequencies) by reducing the magnitude response in the direction of emission from the reference magnitude response.
• the use of the DSB reference design and its WNG value allows a good estimation of the robustness properties of the final design specification.
• psychoacoustic findings flow into the frequency response determination 58.
• optimization is then performed in module 56.
• the beamforming filter is designed here in the frequency domain for a number of discrete frequencies w k .
• optimizing methods based on convex optimization are preferably used [M07, MSK09]. These allow a best possible approximation of the prescribed or selected by the module 58 Abstrahi plausibleizing as starting from the data 36 by the modules 60 , 54 and 58, with respect to a selectable error standard, such as l_ 2 - (least squares) or norm (Chebyshev, Minimax norm).
• the result of the optimization in the module 56 is a complex drive value for each discrete frequency, so that a vector H n (u) k ) of more complex other weights per converter n results.
• measured or modeled transducer data or data 34 may be included to obtain drive frequency responses H n optimized for frequency response and emission characteristics.
• the optimization-based approach allows numerous constraints that can relate to both the radiation produced and the drive weights. For example, a limitation on the minimum white noise gain can be set. In the same way it is possible to set maximum amounts for the control weights in order to limit the control of the individual transducers.
• FIG. 4 The previous description of the possible implementation of the mode of operation of the first calculation device 44 is summarized again in an illustrative manner, reference is made to FIG. 4.
• the starting point of the target frequency response calculation by the calculation device 44 forms the desired direction selectivity, which is indicated in FIG. 4 by ⁇ . is written and provided with the reference numeral 70.
• the desired direction selectivity ⁇ is illustrated here by way of example as a function ⁇ dependent on the emission angle ⁇ . However, as stated above, the directionality may be defined other than angular.
• FIG. 4 indicates by a dashed d that the desired direction selectivity 70 may be defined in the space sense, not just in one plane. At the top right in Fig. 4 is indicated how the angle ⁇ and ⁇ could be defined.
• a frequency dependence could already be included, ie ⁇ could depend on ⁇ .
• a "frequency response" ⁇ has often been used because directionally certain frequencies are attenuated less or more, but this frequency response ⁇ is to be distinguished from the frequency response H n ⁇ u) k ) for the individual beamforming filters Both act as a filter with a transfer function as determined by the dependence on ⁇ , but the frequency response ⁇ is affected by the finally calculated frequency responses H n of the individual beamforming filters.
• the desired direction selectivity 70 is now to be achieved with the particular transducer array.
• loudspeakers have been used as examples of the elements of the array in the upper right, but as already mentioned, an array of other transducers, such as microphones, is likewise possible.
• the array is thus determined from certain transducer positions, transducer orientations, a transducer frequency response, wherein the frequency response may in turn be direction-dependent, and / or a directional dependence of the radiation or the sensitivity, which in turn may be frequency dependent.
• module 54 a pair of values ⁇ ⁇ and a n , namely frequency-independent delay ⁇ and a gain value a, which is also frequency-independent, are now determined for each transducer n, so that assuming only these frequency-independent values in the BFFs the transducer n are applied, the directional selectivity ⁇ '72 results, which is direction-dependent, that is, for example, depending on ⁇ and optionally ⁇ and frequency-dependent, ie depending on ⁇ .
• the determination in module 54 is performed so that the desired direction selectivity 70 is achieved or approximated where possible. Of course, this is only possible to a limited extent since only frequency-independent delay or amplification is determined per converter.
• ⁇ '72 now serves as a starting point for the actual desired direction selectivity 74, which will later be the basis of optimization 56.
• Directional selectivity 72 uses the prior knowledge that it Its DSB property is robust.
• the module 58 now modifies the directionality ⁇ '72 to be closer to the desire for a particular frequency dependence of directional selectivity. For example, in the module 58 by the transfer characteristic 38, a frequency dependence of the directional selectivity ⁇ in a predefined direction cp 0 or ⁇ 0 , ⁇ 0 specified, such as in the direction of maximum radiation or maximum selectivity, ie the direction for the ⁇ at 70 is maximum.
• the optimization target 74 of the optimization 56 is thus a frequency- and direction-dependent directional selectivity D 2ie , and the optimization 56 is carried out such that it finds target frequency responses or transfer functions H n (oo k ) for the beamforming filters n their use in the beamforming filters of the transducer array 10, the optimization target 74 is achieved or approximated as well as possible, ie a deviation is minimized according to a certain criterion.
• the optimization 56 could thus be considered as a fine adjustment of beamforming filter transfer functions 76 which, when used for the beamforming filters, are equivalent to the frequency independent delays and gains.
• the DSB design is actually used only for the optimization target formulation and the frequency domain optimization 56 may start independently of the DSB design.
• the variables to be optimized are therefore because the transfer function n H n is a complex-valued function, 2 NK, where N is the number of transducers and K is the number of frequency samples for which optimization 56 is performed.
• the optimized target frequency responses 78, which result from the optimization 56, can be achieved by optionally additionally subjecting the optimization to additional constraints, such as, for example, ancillary conditions regarding compliance with certain robustness criteria, as specified by the data 40. Optimization 56 can therefore be a quadratic pro- program with a constraint that does not fall below a certain robustness measure.
• the frequency responses of the individual drive filters n result from the drive weights ⁇ ⁇ ( ⁇ ⁇ ⁇ ) obtained in the optimization 56, since the weights of the filter are taken in each case.
• These filters often contain a significant delay or delay, which manifests itself for example by the phase or group delay. This delay is a hindrance for the further processing stages, such as, in particular, the subsequent optimization in the second calculation device 46.
• the optional smoothing step described below is made more difficult or requires a significantly higher resolution of the frequency raster in the optimization 56 in the first calculation device.
• phase unwrapping since the smoothing involves a determination of the continuous phase by a "phase unwrapping." The greater the increase in the phase function contained in the frequency response, the more difficult is the correct detection and subsequent compensation of the phase jumps, which has a negative effect on the correctness of the phase phase. unwrapping algorithms.
• the optimization step in the second calculation means 46 if the local optimization target, i. Target frequency response 78, is present in a version that is as close as possible to a zero-phase frequency response, in which so the phase terms caused by delays are eliminated as possible. Further requirements of the optimization step in the calculation device 46 will be described in more detail below. In general, the following aspects should be considered:
• the causality of the resulting filters is not relevant at this stage of the design process. It is possible to work with non-causal desired frequency responses for the drive filters, which are close to zero-phase transfer functions.
• the causality can be made causal again after the FIR design (by reinserting the extracted delays, possibly supplemented by additional delays).
• the following procedure can be used to adjust the gain values. - The adaptation is done for each filter BFF n individually.
• Phase in phase unwrapping determines the continuous phase of the frequency response - the linear portion (ie the slope) of the phase function is determined by a least squares fit with a first order polynomial, from which the linear portion of the delay can be determined be determined.
• the linear delay component is rounded to an integer multiple of the sampling period. This can simplify the subsequent recombination which then requires only a shift of the impulse response (eg by prefixing a corresponding number of zeros or by implementing these delays in the form of a delay line.) On the basis of this linear term, a vector of complex Exponential calculated, which has a negated to this linear phase term phase profile.
• FIG. 5 once again illustrates the mode of operation of the delay adjustment module 80 of the modification device 48.
• the starting point is the set of target frequency responses 78, if necessary, to be modified, ie H n (oa k ).
• 5 shows illustratively the phase curve 82 of ⁇ ⁇ ( ⁇ ⁇ ). It has exemplary phase jumps 84.
• the 27-degree phase-shift-adjusted phase profile is shown at 86 and can be approximated by a linear curve 88, such as a least square fit, where the linear component 88 has a slope corresponding to a frequency-independent delay ⁇ ' ⁇ .
• the modification of the target frequency response 78 by the module 80 now provides that this linear component 88 is eliminated or reduced, ie the 2n phase-jump-corrected phase characteristic is leveled or straightened, wherein FIG. 5 shows the phase variation of the thus modified Target frequency responses H ' n (oj k )
• the delays ⁇ ' ⁇ are reserved or stored.
• Another module of the modification device 48 is the optional frequency range smoothing module! 92.
• the frequency domain smoothing by the module 92 has the following meaning.
• the frequency responses 78 and H ' n (u) k ) of the drive filters n generated by the optimization-based filter design typically exhibit strong fluctuations of the magnitude and also of the phase.
• Such design specifications are difficult to implement in an FIR filter design or require a very high FIR filter order or FIR length of the beamforming filters. In the latter case, although a good agreement with the given interfaces can be achieved, however, strong overshoot phenomena often occur between the interpolation points .omega., Which deteriorate the frequency response of the resulting beamformer.
• the desired frequency responses 78 of the drive filters are subjected to a smoothing algorithm. This is done, for example, for psychoacoustic considerations with a frequency-dependent window width of, for example, 1/3 octave or 1/6 octave [HN00].
• the smoothing is carried out separately for magnitude and phase, ie a separate smoothing of the magnitude transfer function (to be more precise of the "zero-phase" frequency response (cf., for example, [Sar93, SI07]) and the continuous ( It is possible that magnitude and phase are generated by a "phase unwrapping" algorithm in the module 92 from the complex frequency response H n (to k ) or H ' n (io k ) and independently by convolution with a frequency-dependent smoothing filter, also referred to as "window.”
• the "phase un-wrapping" in the module 92 can be given in the case of the presence of the module 80.
• FIG. 6 shows by way of example by means of a double arrow that the FIR filter coefficients obtained by the optimization in the calculation device 46 are recombined, such as the zero insertion described below in more detail h BFF beamforming filter mode describe the impulse response of the respective beamforming filter n and pass over or correspond to the transfer function ⁇ ⁇ ( ⁇ ) of the respective beamforming filter via FFT or Fourier transformations the impulse response and at 98 exemplarily the 2 jr phase shift-adjusted phase characteristic of the transfer function.
• each beamforming filter n could not only be defined in the beamforming filter n by the beamforming FIR filter coefficients h BFF would be, but also by the frequency independent
• Delay ⁇ ' ⁇ the latter being considered in beamforming filters of Figures 1 and 2 by a simple delay element connected in series with the FIR filter.
• FIG. 3 represents the optional possibility, according to which restrictions 42 are made for the optimization target with regard to the frequency response of these frequency sections.
• the frequency sections may be selected depending on the characteristics of the transducers.
• the frequency restriction 42 is included by specifying a maximum magnitude as a constraint on the convex optimization problem: under the constraint that ⁇ ( ⁇ ) ⁇ ⁇ ) ⁇ V ⁇ e X,
• X represents a discretized representation of the transition or do not care bands, ie those frequency sections for which there is no optimization goal in the optimization in the second calculator 46 and ⁇ ⁇ ⁇ the maximum allowable magnitude of the frequency response at the frequency ⁇ within the transition bands X denotes.
• the aim of the optimization in the second calculation device 46 is derived from the frequency responses obtained by the frequency domain design for the beamformer, which are described above as H n (u) k ) and H ' n (u> k ) and ⁇ " ⁇ ( ⁇ ⁇ ), respectively. and hereinafter referred to as the desired frequency response with the variable name H ⁇ to generate FIR filters, with which the filtering of the source signals, ie the loudspeaker signals in the case of a loudspeaker array, as shown in Fig. 1 2 and the microphone output signals may be in the case of a microphone array as shown in Fig. 2.
• a mathematical optimization method may be used, which may be a convex optimization method, for example designed FIR
• Filter h (i) is determined so that H ⁇ is approximated as best as possible, ie that the error with respect to a selectable norm p becomes minimal.
• the optimization problem can generally be represented in the following form:
• the suffix ⁇ constraint (s)> is optional. Additional conditions do not have to exist, but can be present as described above with regard to the high-frequency restrictions. A single constraint is also possible. In general, these constraints represent a variety of possible constraints, including, but not limited to, the frequency response or coefficients of the FIR filter.
• the target frequency responses for the time domain optimization in the second calculation device 46 which arise in the context of the frequency domain optimization 56 (or with modification 80 and / or 92) are generally complex-valued and have a non-trivial, in particular neither linear nor minimal-phase, frequency response.
• the optimization problem of the above equation (2) corresponds to a filter design problem for FIR filters with arbitrary phase characteristics.
• a large number of methods are described in the literature, for example in [PR95, KM95; KM99].
• the delays contained in the filters H 1 and H 2 ie the linear term of the phase response which has been adjusted by 2 n-phase jumps, are of particular importance.
• [KM99] the use of arbitrary phase responses results in very poorly conditioned optimization probes or degenerate solutions. This is especially true when the standard formulation of a causal FIR filter with the frequency response
• the desired function H ⁇ should be adjusted so that the linear component of the phase is as close to 0 as possible. This is effectively realized by the modifications 80 and 50.
• the modification device 50 optionally integrates the previously compensated delay components back into the drive filters.
• the integration of the delays ⁇ ' ⁇ into the filters n is bypassed by the pure delays ⁇ ' ⁇ at the transit time of the beamforming Application can be applied to the input or output signals of the control filter by means of suitable signal processing devices, such as digital delay lines (De / ay Lines). In this case it is only necessary to ensure that the impulse responses of the obtained FIR filters are causal, ie the indices of the impulse responses begin at 0.
• Such a modification does not require active arithmetic operations at run-time, but merely corresponds to the introduction of a constant implementation-related delay for all drive filters n. It should be ensured that this delay is constant for all drive filters of a beamformer.
• the delay lines can be used for integer delays, as described with respect to FIG. 6, which do not require filtering operations but only require indexed access to the signal and also do not cause distortion.
• any delay values can be mapped.
• this requires delay lines with access to arbitrary delays (Fractional Delay Lines), which can cause distortion, require processing power, and possibly require additional latency or delay.
• frequency domain optimization 56 could also use a hybrid design approach.
• an optimization-based approach for obtaining the frequency domain drive functions H m (to k ) as described so far is combined with a design corresponding to the DSB design as calculated in the module 54, wherein the DSB Design approach is used for the high frequencies.
• the aim is to reduce the required filter order while improving robustness. It is exploited that the emission characteristic of the transducer array can no longer be completely controlled due to the spatial aliasing for high frequencies.
• a DSB design approach is used for frequencies above a fixed fundamental frequency, such as a frequency that is relatively close to the spatial aliasing frequency of the transducer array.
• the frequency domain specification of the complete filter is composed of two parts: the frequency responses obtained by optimization up to the cutoff frequency and the frequency responses corresponding to those of the DSB for the frequencies above it.
• the combination of the two methods is achieved by the subsequent smoothing as described above. tion and the optimization-based FIR design.
• a critical step here is the matching of the signal delays of both design approaches. For example, it is possible to use a least-square fitting to determine a delay offset for the DSB in such a way that the delay jumps of the individual control filters are minimized in the root mean square.
• the hybrid design approach allows for more robust high frequency radiation, resulting in less erratic fluctuations in performance without significant loss of performance, with partially improved low frequency directivity, and consistent filter ordering.
• the reason for this can be assumed that the degrees of freedom provided by a certain filter order are better used with the hybrid design approach for the frequency ranges in which the characteristic can be influenced, while for high frequencies in which due to spatial aliasing hard restrictions for the Suppression of unwanted radiation, less resources are spent.
• the transfer function to be used for time domain optimization in the second calculation means 46 is composed of the transfer function obtained according to the frequency domain optimization 56 as far as a portion of lower audio frequencies 100 is concerned, the frequency independent value pairs corresponding transfer functions H n in the higher audio frequency section 102 are used.
• Section 100 and section 102 can adjoin one another at a cut-off frequency ⁇ 9 ⁇ which, for example, corresponds to the spatial limit aliasing frequency of the transducer array or deviates from the latter by less than 10%. It is also possible that, as indicated by dashed dots, low-frequency section and high-frequency sections 100 and 102 overlap each other.
• the time domain optimization transfer function finally to be used could be obtained, for example, by averaging between both transfer functions (DSB design and optimization result of 56).
• DSP design and optimization result of 56 the above embodiments thus described a possibility for the creation of a design of robust FIR filters for beamforming applications.
• Complex-valued frequency responses of the individual beamforming filters can be used to produce FIR filters with arbitrary phase responses.
• robustness properties of the beamformer can be obtained.
• the present invention can be used in a variety of beamforming applications, such as e.g.
• loudspeaker arrays for room-selective sounding for producing "quiet zones” or for reproducing surround material via loudspeaker lines (soundbars)
• the above exemplary embodiments can also be used by icon arrays to record sound directionally selectively
• the bandwidths required there are much lower than for audio applications, so that an implementation as an FIR filter or the need for a design approach for broadband filters can be estimated here only with difficulty.
• the inventive set of FIR filter coefficients 32 for the beamforming filters may be stored on a digital storage medium or may be transmitted on a transmission medium such as a wireless transmission medium or a wired transmission medium such as the Internet.
• embodiments of the invention may be implemented in hardware or in software.
• the implementation may be using a digital storage medium, such as a floppy disk, a DVD, a Blu-ray Disc, a CD, a ROM, a PROM, an EPROM, an EEPROM or FLASH memory, a hard disk, or other magnetic or optical memory are stored on the electronically readable control signals, which can cooperate with a programmable computer system or cooperate such that the respective method is performed. Therefore, the digital storage medium can be computerized.
• some embodiments according to the invention include a data carrier having electronically readable control signals capable of interacting with a programmable computer system such that one of the methods described herein is performed.
• embodiments of the present invention may be implemented as a computer program product having a program code, wherein the program code is operable to perform one of the methods when the computer program product runs on a computer.
• the program code can also be stored, for example, on a machine-readable carrier.
• Other embodiments include the computer program for performing any of the methods described herein, wherein the computer program is stored on a machine-readable medium.
• an embodiment of the method according to the invention is thus a computer program which has a program code for performing one of the methods described herein when the computer program runs on a computer.
• a further embodiment of the inventive method is thus a data carrier (or a digital storage medium or a computer-readable medium) on which the computer program is recorded for carrying out one of the methods described herein.
• a further embodiment of the method according to the invention is thus a data stream or a sequence of signals, which represent the computer program for performing one of the methods described herein.
• the data stream or the sequence of signals may be configured, for example, to be transferred via a data communication connection, for example via the Internet.
• Another embodiment includes a processing device, such as a computer or a programmable logic device, that is configured or adapted to perform one of the methods described herein.
• a processing device such as a computer or a programmable logic device, that is configured or adapted to perform one of the methods described herein.
• Another embodiment includes a computer on which the computer program is installed to perform one of the methods described herein.
• Another embodiment according to the invention comprises a device or system adapted to transmit a computer program for performing at least one of the methods described herein to a receiver.
• the transmission can be done for example electronically or optically.
• the receiver may be, for example, a computer, a mobile device, a storage device or a similar device.
• the device or system may include a file server for transmitting the computer program to the recipient.
• a programmable logic device eg, a field programmable gate array, an FPGA
• a field programmable gate array may cooperate with a microprocessor to perform one of the methods described herein.
• the methods are performed by any hardware device. This may be a universal hardware such as a computer processor (CPU) or hardware specific to the process, such as an ASIC.
• HM00 Panagiotis D. Hatziantoniou and John N. Mourjopoulos. Generalized fractional-octave smoothing of audio and acoustic responses. Journal of the Audio Engineering Society, 48 (4): 259-280, April 2000 [KM95] Lina J. Karam and James H. McCileil. Complex Chebyshev approximation for FIR filter design. IEEE Transactions on Circuits and Systems II: Analog and Digital Signal Processing, 42 (3): 207-216, March 1995.

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