WO2015158293A1 - 前导符号的生成、接收方法和频域符号的生成方法及装置 - Google Patents

前导符号的生成、接收方法和频域符号的生成方法及装置 Download PDF

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Publication number
WO2015158293A1
WO2015158293A1 PCT/CN2015/076812 CN2015076812W WO2015158293A1 WO 2015158293 A1 WO2015158293 A1 WO 2015158293A1 CN 2015076812 W CN2015076812 W CN 2015076812W WO 2015158293 A1 WO2015158293 A1 WO 2015158293A1
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WIPO (PCT)
Prior art keywords
time domain
generating
sequence
preamble symbol
prefix
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PCT/CN2015/076812
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English (en)
French (fr)
Inventor
张文军
邢观斌
黄戈
徐洪亮
Original Assignee
上海数字电视国家工程研究中心有限公司
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Priority claimed from CN201410153040.XA external-priority patent/CN105007145B/zh
Priority claimed from CN201410168180.4A external-priority patent/CN105007146B/zh
Priority claimed from CN201410177035.2A external-priority patent/CN105024952B/zh
Priority claimed from CN201410184919.0A external-priority patent/CN105024791B/zh
Priority claimed from CN201410185112.9A external-priority patent/CN105024963A/zh
Priority claimed from CN201410229558.7A external-priority patent/CN105323048B/zh
Priority claimed from CN201410259080.2A external-priority patent/CN105282076B/zh
Priority claimed from CN201410274626.1A external-priority patent/CN105282078B/zh
Priority claimed from CN201410326504.2A external-priority patent/CN105245479B/zh
Priority claimed from CN201410753506.XA external-priority patent/CN105743624B/zh
Priority claimed from CN201510039510.4A external-priority patent/CN105991495B/zh
Priority claimed from CN201510061935.5A external-priority patent/CN105991266B/zh
Priority claimed from CN201510076155.8A external-priority patent/CN105991501B/zh
Priority claimed from CN201510076216.0A external-priority patent/CN105991502B/zh
Priority claimed from CN201510076151.XA external-priority patent/CN105991500B/zh
Priority to KR1020197033488A priority Critical patent/KR102191859B1/ko
Priority to US15/304,856 priority patent/US10574494B2/en
Priority to CA2945854A priority patent/CA2945854A1/en
Application filed by 上海数字电视国家工程研究中心有限公司 filed Critical 上海数字电视国家工程研究中心有限公司
Priority to KR1020167032055A priority patent/KR102048221B1/ko
Priority to KR1020207035510A priority patent/KR102234307B1/ko
Publication of WO2015158293A1 publication Critical patent/WO2015158293A1/zh
Priority to US16/726,927 priority patent/US11012275B2/en
Priority to US16/726,928 priority patent/US10958494B2/en

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2602—Signal structure
    • H04L27/2605—Symbol extensions, e.g. Zero Tail, Unique Word [UW]
    • H04L27/2607—Cyclic extensions
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2602—Signal structure
    • H04L27/261—Details of reference signals
    • H04L27/2613—Structure of the reference signals
    • H04L27/26136—Pilot sequence conveying additional information
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2657—Carrier synchronisation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2662—Symbol synchronisation
    • H04L27/2663—Coarse synchronisation, e.g. by correlation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04W—WIRELESS COMMUNICATION NETWORKS
    • H04W56/00—Synchronisation arrangements
    • H04W56/0005—Synchronisation arrangements synchronizing of arrival of multiple uplinks
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
    • H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
    • H04L27/38—Demodulator circuits; Receiver circuits
    • H04L27/3818—Demodulator circuits; Receiver circuits using coherent demodulation, i.e. using one or more nominally phase synchronous carriers

Definitions

  • the present invention relates to the field of communications technologies, and in particular, to a method and a device for generating and receiving preamble symbols and a method for generating frequency domain symbols.
  • the OFDM system in order for the receiving end of the OFDM system to correctly demodulate the data transmitted by the transmitting end, the OFDM system must implement accurate and reliable time synchronization between the transmitting end and the receiving end. At the same time, because the OFDM system is very sensitive to the carrier frequency offset, the receiving end of the OFDM system needs to provide an accurate and efficient carrier frequency estimation method to accurately estimate and correct the carrier frequency offset.
  • an OFDM system is composed of physical frames, and each frame usually has a synchronization frame header, which is called a preamble symbol or a bootstrap, and implements time synchronization between the transmitting end and the receiving end.
  • the preamble symbol is a sequence of symbols known to both the transmitting end and the receiving end of the OFDM system.
  • the preamble symbol is used as the beginning of the physical frame and usually contains the P1 symbol.
  • the use of the P1 symbol (preamble) or the bootstrap symbol includes:
  • the problem solved by the present invention is that in the current DVB_T2 standard and other standards, the DVB_T2 time domain structure cannot be applied to coherent detection, and the preamble symbol fails in the DBPSK differential decoding under the complex frequency selective fading channel, and the receiving algorithm detects the probability of failure.
  • the embodiments of the present invention provide a method for generating a preamble symbol, a receiving method, and a method for generating a frequency domain symbol and related devices.
  • An embodiment of the present invention provides a method for generating a preamble symbol, which includes the following steps: generating a prefix according to a partial time domain body signal intercepted from a time domain body signal; generating according to all or part of the partial time domain body signal Super-prefix; based on a cyclic prefix, a time domain body signal, and a super-prefix, generating a time-domain symbol, the preamble symbol includes at least one of the time-domain symbols.
  • the method further includes: wherein the preamble symbol includes at least one of the time domain based on the cyclic prefix, the time domain body signal, and the super prefix generated time domain symbols. symbol.
  • the method further includes: the step of generating the prefix and the super prefix includes: the prefix is directly intercepted from the back of the time domain body signal, and the super prefix is All or part of the partial time domain body signal corresponding to the prefix is modulated.
  • the method further includes: the step of generating the prefix and the super prefix, including: the part that is intercepted from the back of the time domain body signal according to the first predetermined
  • the processing rule is processed to be formed as a prefix, and the portion intercepted from the rear of the time domain body signal is processed into a super prefix according to a second predetermined processing rule
  • the first predetermined processing rule includes: direct copying; or multiplication by an identical fixed coefficient Or predetermined different coefficients
  • the second predetermined processing rule includes: performing modulation processing when the first predetermined processing rule is a direct copy; or multiplying the corresponding when the first predetermined processing rule is multiplied by an identical fixed coefficient or a predetermined different coefficient Modulation processing is performed after the coefficient.
  • the length of the super prefix does not exceed the length of the prefix.
  • the method further includes: generating a frequency shift sequence; multiplying part or all of the time domain body signal by the frequency shift sequence To get the super prefix of the time domain body signal.
  • the method further includes: wherein the frequency offset sequence of the frequency offset sequence is based on a frequency domain subcarrier spacing corresponding to the time domain body signal or according to a length of the super prefix. It is determined that the frequency shift sequence can arbitrarily select the initial phase value.
  • the method further includes: wherein the preamble symbol transmits signaling information by determining a combination of a length of a prefix and a length of a super prefix, When generating a super prefix, different signaling information is transmitted by intercepting partial time domain symbols with different starting positions.
  • the method further includes: the length of the time domain body signal is 2048 sampling periods, the length of the prefix is 520 sampling periods, and the length of the super prefix is 504. For the sampling period, the starting position of the super prefix intercepted in the time domain body signal is the 1528th sample.
  • the preamble symbol contains the prefix, the time domain body signal and the super prefix signal.
  • the time domain expression for the time domain symbol is:
  • the feature further comprises: a length N A of the time domain body signal is 2048, a length of the cyclic prefix Len C is 520, and a length of the super prefix Len B is 504, the time domain expression of the preamble symbol including the cyclic prefix, the time domain body signal, and the super prefix time domain symbol is:
  • the method further includes: wherein the time domain body signal carries the emergency broadcast identifier by using at least one bit signaling, and the modulated signal is used to modulate the signal length in the time domain body. Different starting positions intercepted in the signal to carry the emergency broadcast identification.
  • the method further includes: wherein the time domain body signal is processed based on the frequency domain symbol, optionally, in the method for generating the provided preamble symbol Further, the method further includes the step of: generating the frequency domain symbols by: separately generating the fixed sequence and the signaling sequence in the frequency domain, and then filling the effective subcarriers.
  • an embodiment of the present invention further provides a method for generating a frequency domain symbol, which includes the following steps: separately generating a fixed sequence and a signaling sequence in a frequency domain, and then filling the effective subcarriers. Used to form frequency domain symbols.
  • the method further includes: determining an average power ratio of the fixed sequence and the signaling sequence, and generating a fixed sequence and a signaling sequence according to the average power ratio.
  • the average power ratio between the fixed sequence and the signaling sequence is a value of 2.
  • the fixed sequence and the signaling sequence are arranged in a predetermined staggered arrangement rule, and the predetermined arrangement rule includes any one of the following two rules: arrangement in parity interleave or even odd interleave; or placing a part of the signaling sequence on odd subcarriers Another part of the signaling sequence is placed on even subcarriers, and a part of the fixed sequence is placed on the odd subcarriers, and another part of the fixed sequence is placed on the even subcarriers.
  • the method further includes: generating a same sequence generation formula based on a length and a number of the preset signaling sequence. And selecting different phase base values based on the same sequence generation formula to generate different constant envelope zero autocorrelation sequences; and selecting a signaling sequence from each of the obtained constant envelope zero autocorrelation sequences according to the determined length of the signaling sequence.
  • the method further includes: the generating step of the signaling sequence includes: generating a plurality of sequences based on the length and the number of the preset signaling sequence. a sequence generation formula; generating a formula for each sequence, selecting a different phase base value to generate a constant envelope zero autocorrelation sequence; and obtaining a constant envelope zero autocorrelation sequence from the determined length of the signaling sequence Select the signaling sequence.
  • the method further includes: wherein, for the generated constant envelope zero autocorrelation sequence, the method further includes the following steps: zero autocorrelation of the generated constant envelope The sequence is further cyclically shifted.
  • an embodiment of the present invention further provides a method for receiving a preamble symbol, including the steps of: processing a received signal; determining whether there is a preamble symbol that is expected to be received in the processed signal; When yes, the location of the preamble symbol is determined and the signaling information carried by the preamble symbol is solved.
  • the receiving method of the provided preamble symbol further characterized by: determining whether there is a preamble symbol that is expected to be received in the processed signal, and determining a position of the preamble symbol when the determination is yes
  • the step of solving the signaling information carried by the preamble symbol includes the following Any at least one of the steps: initial timing synchronization mode, integer multiple frequency offset estimation mode, precise timing synchronization mode, channel estimation mode, decoding analysis mode, and fractional multiple frequency offset estimation mode.
  • the receiving method of the provided preamble symbol further characterized by: determining, by using a result of any at least one of the following, whether there is a preamble symbol that is expected to be received in the processed signal: initial timing synchronization, Integer multiple frequency offset estimation, precise timing synchronization, channel estimation, decoding analysis and fractional multiple frequency offset estimation.
  • the method further includes: the step of: determining whether there is a preamble symbol to be received in the baseband signal, including: initially determining the preamble symbol by using an initial timing synchronization manner a location; and based on a result of the initial timing synchronization mode, determining whether there is a preamble symbol that is expected to be received in the processed signal.
  • the method further includes: the initial timing synchronization manner includes the following: a first initial timing synchronization manner: using a cyclic prefix, a time domain body signal, and a super prefix
  • the processing value is used for initial timing synchronization, the position of the preamble symbol is initially determined, and/or the second initial timing synchronization mode: when any time domain body signal in the current pilot symbol contains a known signal, the time domain body signal is according to a predetermined N
  • the difference value is subjected to a difference operation, and the time domain signal corresponding to the known information is also subjected to a difference operation, and then the two are cross-correlated to obtain a differential correlation result of the N sets and the N difference values one by one, based on the N
  • the method further includes: performing the step of: determining the location of the preamble symbol and solving the signaling information carried by the preamble symbol, including: using a preamble All or part of the time domain waveform of the symbol and/or all or part of the time domain of the preamble symbol
  • the frequency domain signal obtained by transforming the waveform is used to solve the signaling information carried by the preamble symbol.
  • the method further includes: when the received frequency domain symbols used to generate the preamble symbols are respectively arranged by the fixed sequence and the signaling sequence, and then filled in
  • the step of valid subcarriers further includes: performing integer multiple frequency offset estimation or channel estimation by using a fixed sequence, and performing the step of performing integer multiple frequency offset estimation or channel estimation by using the fixed sequence, including: according to the initially determined initial preamble symbol Positioning, intercepting a signal including all or part of the fixed subcarriers; and computing the signal including all or part of the fixed subcarriers with a frequency domain fixed subcarrier sequence or a time domain signal corresponding to the frequency domain fixed subcarrier sequence to obtain an integer Frequency doubling estimation or channel estimation.
  • an embodiment of the present invention provides a preamble symbol generating apparatus, including: a prefix generating unit that generates a prefix according to a partial time domain body signal intercepted from a time domain body signal; and a super prefix generating unit according to the All or part of the partial time domain body signal generates a super prefix; the preamble symbol generating unit generates a time domain symbol based on at least one of a cyclic prefix, a time domain body signal and a super prefix, and the preamble symbol includes at least one of the time domain symbols.
  • an embodiment of the present invention provides a frequency domain symbol generating apparatus, including: a sequence generating unit, configured to separately generate a fixed sequence and a signaling sequence in a frequency domain; and a frequency domain symbol generating unit, The fixed sequence and the signaling sequence are arranged and padded onto the effective subcarriers for forming frequency domain symbols.
  • an embodiment of the present invention provides a receiving apparatus for a preamble symbol, including: a receiving processing unit that processes a received signal; and a determining unit that determines whether there is a preamble symbol that is expected to be received in the processed signal; And a positioning parsing unit, configured to determine a location of the preamble symbol and obtain signaling information carried by the preamble symbol when the determination result is yes.
  • the part is used as a prefix to implement coherent detection, which solves the problem of non-coherent detection performance degradation and DBPSK differential decoding failure under complex frequency selective fading channel, and all or part of the time domain subject signal based on the above-mentioned intercepted cyclic prefix length
  • the modulated signal is generated such that the generated preamble symbol has good fractional frequency offset estimation performance and timing synchronization performance.
  • a time domain symbol having a three-segment structure may be selected as a preamble symbol according to requirements of transmission efficiency and robustness; when the current pilot symbol includes a time domain symbol having a three-segment structure, based on the same OFDM symbol body, Signaling may be transmitted by utilizing different starting points of the second portion from the first portion, such as emergency broadcasts, hook information, transmitter flag information, or other transmission parameters.
  • FIG. 1 is a schematic diagram of time domain symbols of a first three-segment structure in an embodiment of the present invention
  • FIG. 2 is a schematic diagram of a time domain symbol of a second three-segment structure in an embodiment of the present invention
  • FIG. 3 is a schematic diagram of an acquisition process of a time domain symbol based on a first three-segment structure in an embodiment of the present invention
  • FIG. 4 is a schematic diagram of an acquisition process of a time domain symbol based on a second three-segment structure in an embodiment of the present invention
  • FIG. 5 is a schematic diagram showing an arrangement of a frequency domain structure according to a first predetermined staggered arrangement rule in an embodiment of the present invention
  • FIG. 6 is a schematic diagram showing an arrangement of a frequency domain structure according to a second predetermined staggered arrangement rule in an embodiment of the present invention
  • FIG. 7 is a logic diagram of a correlation result to be detected corresponding to a three-segment structure CAB in a method for receiving a preamble symbol in an embodiment of the present invention.
  • FIG. 8 is a diagram showing a method of receiving a preamble symbol in a method corresponding to a three-segment structure BCA in an embodiment of the present invention. A schematic diagram of the detection of related results.
  • a method for generating a preamble symbol includes the following steps:
  • the preamble symbol includes at least one of the time domain symbols based on at least one of the cyclic prefix, the time domain body signal, and the modulated signal.
  • 1 is a schematic diagram of time domain symbols of a first three-segment structure in an embodiment of the present invention.
  • 2 is a schematic diagram of time domain symbols of a second three-segment structure in an embodiment of the present invention.
  • a time domain symbol having a second three-segment structure having a second three-segment structure.
  • the time domain structure of the time domain symbols included in the above preamble symbols will be described below with reference to FIGS. 1 and 2.
  • the time domain structure comprises a three-segment structure, and the three-segment structure has two cases, a first three-segment structure and a second three-segment structure.
  • the first three-segment structure is: a time domain body signal (part A), a prefix (part C) generated based on a partial time domain body signal intercepted from a time domain body signal, based on the portion A modulated signal (Part B) generated by part or all of the domain body signal.
  • the second three-segment structure is: a time domain body signal (part A), a prefix (part C) generated based on a partial time domain body signal intercepted from the time domain body signal, according to the portion The super prefix generated by the domain body signal (Part B).
  • a time domain body signal (indicated by A in the figure) is taken as the first part, and the last part of the first part is taken out according to a predetermined acquisition rule, and processed according to the first predetermined processing rule. And copying to the front part of the first part to generate a third part (indicated by C in the figure) as a prefix, and at the same time, taking a part from the back part of the first part according to a predetermined acquisition rule, processing and copying according to the second predetermined processing rule Go to the back of the first part or process and copy to the front of the prefix to generate the second part (marked by B in the figure) to respectively correspond to the suffix or super prefix, thereby respectively generating B as the suffix as shown in Figure 1.
  • the first three-segment structure (CAB structure) and B are super-prefixed as the second three-segment structure (BCA structure) as shown in FIG.
  • the first predetermined processing rule includes: direct copying; or multiplying each sampling signal in the extracted part by an identical fixed coefficient or Different coefficients are predetermined.
  • the second predetermined processing rule includes: performing modulation processing when the first predetermined processing rule is a direct copy; or when the first predetermined processing rule is multiplying each of the sampled signals in the extracted portion by an identical fixed coefficient or a predetermined different coefficient The modulation process is performed after multiplying by the corresponding coefficient.
  • the third part when the third part is a direct copy as a prefix, the second part is modulated and then used as a suffix or super prefix, and when the third part is multiplied by the corresponding coefficient, the second part also needs to be multiplied by the coefficient and Modulation processing is performed as a suffix or super prefix.
  • FIG. 3 is a schematic diagram of an acquisition process of a time domain symbol based on a first three-segment structure in an embodiment of the present invention.
  • the C segment is a direct copy of the A segment
  • the B segment is the modulated signal segment of the A segment.
  • the length of A is 1024
  • the length of intercept C is 520
  • the length of B is 504.
  • each sample of the signal can be multiplied by a fixed coefficient, or each sample is multiplied by a different coefficient.
  • the data range of B does not exceed the data range of C, that is, the range of the portion A selected for the modulated signal segment B does not exceed the range of the portion A intercepted as the prefix C.
  • the sum of the length of B and the length of C is the length of A.
  • N A be the length of A
  • Len C be the length of C
  • Len B be the length of the modulated signal segment B.
  • the sampling point number of A be 0,1,...N A -1.
  • N1 be the first sampling point number of the first part A corresponding to the starting point of the second part B of the modulated signal segment
  • N2 is the selective copying modulation.
  • the second sample point number of the first portion A corresponding to the end point of the second portion B of the signal segment.
  • the first sample point sequence number and the second sample point sequence number satisfy the following predetermined constraint relationship:
  • the modulation performed on the second portion B segment is a modulation frequency offset, that is, a frequency shift sequence, a modulation M sequence or other sequences, etc.
  • the modulation frequency offset is taken as an example, and when P1_A(t) is A, The domain expression, then the time domain expression of the first CAB three-segment structure is
  • the modulation frequency offset value f SH may be selected as a frequency domain subcarrier spacing corresponding to the time domain OFDM main symbol, that is, 1/N A T, where T is a sampling period, and N A is a time domain.
  • the frequency shift sequence can be arbitrarily selected as the initial phase.
  • f SH can also be selected as 1/(Len B T).
  • the autocorrelation delay of the CA segment containing the same content is N A
  • the autocorrelation delay of the CB segment containing the same content is N A +Len B
  • the autocorrelation delay of the AB segment containing the same content is Len B .
  • the length of the C segment is exactly the same as the B segment, that is, the B segment can be regarded as a complete adjustment frequency offset segment of the C segment.
  • a cyclic prefix C is spliced at the front of the time domain OFDM symbol A as a guard interval
  • the modulated signal segment B is spliced at the rear of the OFDM symbol as a modulation frequency offset sequence to generate a first type
  • FIG. 4 is a schematic diagram of a time domain symbol acquisition process based on a second three-segment structure in an embodiment of the present invention.
  • the time domain expression of the second three-segment structure time domain symbol is, note that in order to make the receiving end processing method as consistent as possible, in the BCA structure, the modulation frequency offset value is exactly opposite to the CAB structure, and modulation The initial phase can be arbitrarily selected.
  • the autocorrelation delay of the CA segment containing the same content is N A
  • the autocorrelation delay of the BC segment containing the same content is Len B
  • the autocorrelation delay of the BA segment containing the same content is N A +Len B .
  • the current pilot symbol includes a symbol of a three-segment structure, whether it includes the first three-segment structure or the second three-segment structure, based on the same OFDM symbol body, the following method may also be adopted.
  • the domain structure transmits signaling.
  • Such as emergency broadcast, hook information, transmitter logo information or other transmission parameters Such as emergency broadcast, hook information, transmitter logo information or other transmission parameters.
  • the predetermined length is 1024
  • Len C is 512
  • Len B is 256.
  • N1 can take 512+i*160 ⁇ i ⁇ 16, which can represent 16 different methods and transmit 4bit signaling parameters.
  • Different transmitters can transmit the corresponding identifier of the transmitter by taking different N1, the same transmitter can also transmit the transmission parameter by changing N1 in a time-sharing manner.
  • the emergency broadcast identity EAS_flag is transmitted using 1-bit signaling.
  • N1 512-L, that is, the corresponding number of the OFDM symbol with N A is 1024 is 512-L ⁇ 1023-2L, and the frequency offset sequence is modulated to generate B, which is placed at the back of A. .
  • N1 512+L, that is, the corresponding number of the OFDM symbol with N A is 1024 is 512+L ⁇ 1023 and modulate the frequency offset sequence to generate B, and put it to the back of A.
  • N A 1024
  • Len C is 520
  • Len B is 504
  • a different three-segment structure may be used to identify the emergency broadcast.
  • the detection of the time domain symbols of a single three-segment structure utilizes the delay autocorrelation of the CB segment, the CA segment and the BA segment to obtain the peak value, in the case of using two three-segment structure time domain symbol splicing, in order to make the two three-segment structure time domain symbols
  • the autocorrelation values can be added to obtain more robust performance, and the parameter N1 of each of the two three-segment structure time domain symbols (ie, N1 is the sampling point number of A corresponding to the start of the modulation signal segment B) needs to be satisfied.
  • N1 of the first symbol be N1_1
  • N1 of the second symbol be N1_2
  • it is necessary to satisfy N1_1+N1_2 2N A -(Len B +Len c ).
  • the modulation used for the B segment is the modulation frequency offset, the frequency offset value is exactly the opposite.
  • the preamble or bootstrap includes: a time domain symbol having a first three-segment structure; or a time domain symbol having a second three-segment structure. It should be particularly noted that the preamble symbol or bootstrap of the present invention is not limited to a structure including only C-A-B or B-C-A, and may also include other time domain structures, such as a conventional CP structure.
  • the present invention also provides a method for generating a frequency domain symbol.
  • a method for generating a frequency domain symbol by a frequency domain OFDM symbol having the following frequency domain structure 1 will be described.
  • the time domain main signal (Part A) is inversely Fourier transformed by the frequency domain OFDM symbol. Obtained by forming a time domain OFDM symbol.
  • the method for generating the frequency domain symbols provided by the present invention is not limited to use only in the symbols in which the time domain is the three-segment structure as shown in FIG. 1 to FIG. 7 described above, and may be applied to Symbols of other arbitrary time domain structures.
  • P1_X be the corresponding frequency domain OFDM symbol
  • P1_X i be the inverse discrete Fourier transform to obtain the time domain OFDM symbol
  • M is the power sum of valid non-zero subcarriers.
  • the frequency domain structure of the first type of P1_X is described, which is defined as the frequency domain structure one.
  • the method for generating the frequency domain symbols includes the following steps:
  • the fixed sequence and the signaling sequence are arranged and padded to a valid subcarrier for forming a frequency domain symbol.
  • the P1_X frequency domain structure that is, the frequency domain OFDM symbols respectively include a virtual subcarrier and a signaling sequence (called It is a sub-carrier of SC) and a fixed sequence (called FC) subcarrier.
  • the predetermined staggered rule includes any of the following two rules:
  • a first predetermined interleaving rule arranged in a parity interleave or an even odd interleave;
  • the second predetermined interleaving rule placing a part of the signaling sequence on the odd subcarriers, another part of the signaling sequence on the even subcarriers, and placing a part of the fixed sequence on the odd subcarriers, and another part of the fixed sequence on the even subcarriers.
  • the first predetermined interleaving rule is SC and FC parity interleaving or even odd interleaving emissions, such that FC is used as pilot rule emissions;
  • the second predetermined interleaving rule requires partial SC sequences to be placed on odd subcarriers, and remaining SC sequences in even numbers. Subcarriers; at the same time, some FC sequences need to be placed on odd subcarriers, and the remaining FC sequences are placed on even subcarriers, so that FC or SC are all placed on odd or even subcarriers, and all will fall under certain special multipaths. And such emissions will increase the negligible complexity of channel estimation and are therefore a better choice.
  • FIG. 5 is a schematic diagram showing the arrangement of signaling sequence subcarriers, fixed sequence subcarriers, and virtual subcarriers according to a first predetermined staggered arrangement rule according to an embodiment of the present invention.
  • the step includes: filling a certain zero sequence subcarriers on both sides of the effective subcarrier to form a frequency domain OFDM symbol of a predetermined length.
  • the length N A of the time domain body signal A in the time domain structure described above is 1024, and the length formed by the Fourier transform FFT in the frequency domain is N FFT of 1024.
  • the N FFT has a predetermined length of 1024
  • the two sides are padded with (1024-LP)/2 zero sequence subcarriers.
  • the (11) fixed sequence generating step the fixed sequence is composed of 353 complex numbers whose modulus is constant, and the nth value of the fixed sequence subcarrier is expressed as:
  • R is the power ratio of FC and SC
  • SC i mode is constant to 1
  • the radians value ⁇ n of the fixed sequence subcarriers is determined by the first predetermined fixed subcarrier radians value table in Table 1 below;
  • the (12) signaling sequence generating step includes two types, that is, the following first signaling sequence generating manner and the second signaling sequence generating manner.
  • the signaling sequence generated in the frequency domain may adopt any one of the following two manners. The specific manner of generating the signaling sequence is described in detail below.
  • the first signaling sequence is generated:
  • the root value is chosen to be the length of the signaling sequence.
  • the sequence length L For example, determine the sequence length L and the number of signaling. For example, to transmit N bits, the signaling number num is 2N and the root value of exp(j ⁇ qn(n+1)/root) in the CAZAC sequence generation formula is selected.
  • the sequence length L is less than or equal to the root value, and the root value is greater than or equal to 2*num.
  • the root value is a prime number.
  • k is the number of bits of the cyclic shift.
  • the selected q i (0 ⁇ i ⁇ num-1) must satisfy the following conditions: any two q i , q j (0 ⁇ i , j ⁇ num-1) Meet q i +q j ⁇ root.
  • each of the num sequences is truncated as a contiguous partial sequence or a full sequence of length L as a signaling sequence.
  • the value of q ranges from 1 to 352, and the number of cyclic shift bits per sequence ranges from 1 to 353.
  • the following 128 groups are preferred, the q value and the cyclic shift bits are as shown in the q value value table of Table 2 and the cyclic shift bit table of Table 3, respectively:
  • the second signaling sequence is generated in the following way:
  • the root value is chosen to be the length of the signaling sequence.
  • the sequence length L determines the sequence length L and the number of signaling. For example, to transmit N bits, the number of signaling num is 2N, and select some K root k (0 ⁇ k ⁇ K-1) of exp(j ⁇ qn(n+1)/root) in the CAZAC sequence generation formula. .
  • the signaling sequence length L is less than or equal to the minimum value of all root k , and the sum of several root k is greater than or equal to 2*num, that is, Usually root k is a prime number.
  • the selected q i (0 ⁇ i ⁇ num k -1) must satisfy the following conditions: any two q i , q j (0 ⁇ i , j ⁇ num k - 1) Satisfy q i +q j ⁇ root k .
  • each of the num sequences is cyclically truncated into a contiguous partial sequence of length L or all sequences as a signaling sequence.
  • each sequence is truncated to a length of 353.
  • the step generated by the (12)th signaling sequence generates a total of 512 signaling sequences, that is, Seq 0 , Seq 1 , . . . Seq 511 , and each signaling sequence Seq 0 ⁇ Seq according to the second signaling sequence generation manner.
  • 511 further takes the opposite number, that is, -Seq 0 to -Seq 511 , and the receiving end uses the positive and negative of the correlation value to distinguish whether it is a positive sequence or an inverse sequence, that is, a total of 10 bits of signaling information is transmitted
  • 512 signaling sequences can be further divided into 4 groups, each group of 128 signaling sequences, each group of 128 signaling sequence generation sub-steps are as follows:
  • Sub-step 1 Generating a reference sequence zc i (n), which is a Zadoff-Chu sequence zc(n) of length N:
  • Sub-step 2 Producing a length of 2N by copying zc i (n) twice
  • Step 3 From A particular starting position k i intercepts a sequence of length 353, yielding SC i (n):
  • the N values, u i and shift values k i of each set of signaling sequences Seq 0 to Seq 127 are respectively determined by respective respective Tables 4 to 7 predetermined signaling sequence parameter tables.
  • the N values of the first set of sequences Seq 0 to Seq 127 , u i and the shift value k i are as shown in Table 4 below.
  • the steps of generating the second set of sequences Seq 128 to Seq 255 are the same as those of the first set of sequences, and the N values, u i and shift values k i are as shown in Table 5 below.
  • the steps of generating the third set of sequences Seq 256 to Seq 383 are the same as those of the first set of sequences, and the N values, u i and shift values k i are as shown in Table 6 below.
  • the generation steps of the fourth set of sequences Seq 384 to Seq 511 are the same as those of the first set of sequences, and the N values, u i and shift values k i are shown in Table 7 below.
  • FIG. 6 is a schematic diagram showing the arrangement of signaling sequence subcarriers, fixed sequence subcarriers, and virtual subcarriers according to a second predetermined interleaving rule in the embodiment of the present invention.
  • the signaling sequence of the first half of the figure on the left side of the dotted line is placed on the odd subcarriers, and the signaling sequence of the other half of the right side of the dotted line is placed on the even subcarriers, and is located on the left side of the dotted line.
  • the first half of the fixed sequence is placed on the even subcarriers, and the latter part of the fixed sequence on the right side of the dotted line is placed on the odd subcarriers. That is, P1_X 0 , P1_X 1 , . . . , P1_X 1023 are generated according to the second predetermined interleaving rule.
  • the odd carrier In the first half of the SC, the odd carrier is placed, the FC is put into the even carrier, and the second half of the SC is placed in the even carrier, and the FC is placed in the odd carrier.
  • the signaling sequence, the parity of the fixed sequence is exchanged.
  • Fixed sequence subcarrier Signaling sequence subcarrier The parity positions are interchangeable and have no effect on transmission performance.
  • the lengths of the zero-sequence sub-carriers filled in the left and right sides may also be different, but it is not suitable for too much difference.
  • the frequency domain OFDM symbol generated according to the second predetermined interleaving rule includes the following steps:
  • a (21) fixed sequence generating step wherein the fixed sequence generating step is the same as in the (11) fixed sequence generating step, and only the fixed sequence subcarrier radians value ⁇ n is obtained by using a second predetermined fixed subcarrier radians value table. Determining; wherein the second predetermined fixed subcarrier radians value table is as shown in Table 8 below:
  • Table 8 fixed subcarrier radians value table (according to the second predetermined staggered arrangement rule)
  • the signaling sequence and the fixed sequence obtained by the (21) step and the (22) step are alternately arranged in a parity and even odd-even manner, and after zero-carriers are filled on the left and right sides, the frequency is formed according to the following formula. Domain OFDM symbol,
  • a method for receiving a preamble symbol is also provided, and the method for receiving the preamble symbol is applicable to a preamble symbol generated by a sending end by using a predetermined generation rule.
  • the generated preamble symbol includes all the technical elements related to the first three-segment structure and/or the second three-segment structure as explained above from the time domain angle as in the present embodiment. And/or including all of the technical elements involved in the frequency domain description as described above in the embodiment from the frequency domain, and are not repeated here, so in short, the applicable predetermined generation rules are not lost.
  • the method for generating a preamble symbol described above from the perspective of the time domain and the method for generating a frequency domain symbol explained from the frequency domain are generally included.
  • the preamble symbols generated for the predetermined generation rule respectively satisfy the time domain symbols having the above-described three-segment structure and the reception method of the preamble symbols corresponding to the frequency domain symbols having the above-described frequency domain structure one.
  • This embodiment provides a method for receiving a preamble symbol, including the following steps:
  • Step S11 processing the received signal
  • Step S12 determining whether there is a preamble symbol including the three-segment structure that is expected to be received in the processed signal;
  • Step S13 If the result of the foregoing determination is yes, determine the location of the preamble symbol and solve the signaling information carried by the preamble symbol.
  • the received preamble symbol includes the first end of the first three segments according to a predetermined generation rule according to a predetermined generation rule.
  • the first three-segment structure as described above includes a time domain body signal, a prefix generated based on all or part of the time domain body signal, and a suffix generated based on all or part of the partial time domain body signal.
  • the second three-segment structure as described above includes a time domain body signal, a prefix generated based on all or part of the time domain body signal, and a super prefix generated based on all or part of the partial time domain body signal.
  • the received physical frame signal is processed to obtain a baseband signal as described in step S11.
  • the signal received by the receiving end is an analog signal, so it needs to be analog-to-digital converted to obtain a digital signal, and then subjected to filtering, downsampling, etc. to obtain a baseband signal.
  • the receiving end receives the intermediate frequency signal, it needs to perform spectrum shifting after the analog-to-digital conversion processing, and then performs filtering, downsampling and the like to obtain a baseband signal.
  • step S12 it is determined whether there is a preamble symbol including the three-segment structure that is desired to be received in the baseband signal.
  • the receiving end determines whether there is a preamble symbol that is expected to be received in the received baseband signal, that is, whether the received signal meets the receiving standard. For example, if the receiving end needs to receive data of the DVB_T2 standard, it is necessary to determine the received data. Whether the signal contains the leading symbol of the DVB_T2 standard, similarly, it is necessary to judge whether the received signal contains the time domain symbol of the CAB and/or BCA three-segment structure.
  • the above steps S12 and S13 include any at least one of the following steps: initial timing synchronization, integer multiple frequency offset estimation, precise timing synchronization, channel estimation, decoding analysis, and fractional multiple frequency offset estimation.
  • the reliability judgment may be freely combined by using any one of the following methods or any two at least two ways, that is, determining whether there is a preamble symbol that is expected to be received in the processed signal: an initial timing synchronization manner, an integer multiple frequency offset estimation manner, and an accurate timing Synchronization method, channel estimation method, decoding result analysis method and fractional multiple frequency offset estimation method.
  • the step S12 includes an initial timing synchronization mode of S12-1, for initially determining that the preamble symbol is in the physical frame.
  • the location in the frame further includes a result of determining, by the S12-2 based on the initial timing synchronization mode, whether the preamble symbol including the three-segment structure that is desired to be received is present in the baseband signal.
  • the initial timing synchronization mode may take the following initial synchronization by any one or both of the following (1) initial timing synchronization method and (2) initial timing synchronization mode.
  • the (1) initial timing synchronization method includes the following steps:
  • the obtained processing value is used for initial timing synchronization, and the position of the leading symbol is initially determined.
  • the necessary inverse processing and/or signal solution is performed on the baseband signal.
  • three accumulative correlations between the third portion C and the first portion A, the first portion A and the second portion B, and the third portion C and the second portion B in the three-segment structure are obtained.
  • the value is any one of U ca '(n), U cb '(n), U ab '(n) or any at least two.
  • a correlation value to be detected is obtained based on at least one of the above-described accumulated correlation values.
  • the three-segment structure is a C-A-B structure.
  • the received signal is subjected to a delayed sliding autocorrelation, and the delay correlation expression Uca (n) and the delay-related accumulated value Uca '(n) are as follows:
  • the received signal is subjected to delay sliding autocorrelation and demodulation frequency offset, paying attention to the delay correlation expression U cb (n) and the delay correlation accumulated value U cb '(n) is as follows:
  • the received signal is subjected to delay sliding correlation based on the processing relationship of the second portion B and the first portion A and the modulation frequency offset value, and the delay correlation expression U ab (n) and the delay correlation accumulated value U ab '(n) are as follows:
  • corr_len can take 1/f SH T to avoid continuous wave interference or take Len B to make the peak sharp.
  • the delay-related accumulated values U ca '(n), U cb '(n), U ab '(n) are used to perform the required delay matching and perform mathematical operations, the mathematical operations including multiplication or addition, such as U cb '(n ) ⁇ U ab '* (n), or
  • FIG. 7 is a logic diagram of a correlation result to be detected corresponding to a three-segment structure CAB in an embodiment of the present invention.
  • C, A, and B in the figure represent the lengths of the C segment, the A segment and the B segment signal, respectively, and the moving average filter may be a power normalization filter.
  • A is N A
  • B is Len B
  • C is Len C .
  • the three-segment structure is a B-C-A structure.
  • the received signal is subjected to a delayed sliding autocorrelation, and the delay correlation expression Uca (n) and the delay-related accumulated value Uca '(n) are as follows:
  • the received signal is subjected to delay sliding autocorrelation and demodulation frequency offset, paying attention to the delay correlation expression U cb (n) and delay correlation accumulation.
  • the value U cb '(n) is as follows:
  • the received signal is subjected to delay sliding correlation based on the processing relationship of the second portion B segment and the first portion A segment and the modulation frequency offset value, and the delay correlation expression U ab (n) and the delay correlation accumulated value U ab '(n) as follows:
  • corr_len can take 1/f SH T to avoid continuous wave interference or take Len B to make the peak sharp.
  • the mathematical operations including addition or multiplication, such as U cb '* (nN A ) ⁇ U ab '(n), or U ca '(n) ⁇ U cb '* (nN A ) ⁇ U ab '(n) to obtain the calculated value, that is, the correlation value 2 to be detected.
  • FIG. 8 is a logic diagram of a correlation result to be detected corresponding to a three-segment structure BCA in an embodiment of the present invention.
  • a correlation value of the preliminary timing synchronization is formed based on the correlation result 1 to be detected and/or the correlation result 2 to be detected.
  • the transmission preamble symbol includes the following two cases (a) and (b),
  • the initial timing synchronization is completed by any one or a combination of the above (1) initial timing synchronization method and the following (2) initial timing synchronization method. Wherein, based on the two completions, the first preliminary synchronization operation value obtained in the (1) initial timing synchronization manner and the second preliminary synchronization operation value obtained in the (2) initial timing synchronization manner are further weighted, The weighted operation value completes the initial timing synchronization.
  • the (2) initial timing synchronization method will be specifically described below.
  • the subject signal A of any CAB and/or BCA contains a known signal such as a fixed subcarrier, or a preamble symbol, for example, a time domain symbol of a three-segment structure of several CABs and/or BCAs, one of the time domains
  • the main signal A of the symbol is a known signal, that is, when any of the preamble symbols
  • the time domain body signal includes a known signal
  • the (2) initial timing synchronization mode may perform a differential operation on the time domain body signal A according to a predetermined N difference values, and a time domain signal corresponding to the known information.
  • the difference operation is also performed, and then the two are cross-correlated to obtain a differential correlation result of the N sets and the N difference values one by one, and the initial synchronization is performed based on the result of the N sets of differential correlations, and the processed value is obtained for preliminary determination.
  • the position of the leading symbol where N ⁇ 1.
  • the baseband data is subjected to differential operation by differential value.
  • the phase rotation caused by the carrier frequency offset becomes a fixed carrier phase e j2 ⁇ D ⁇ f , where ⁇ f represents the carrier frequency deviation.
  • the local time domain sequence (such as the fixed subcarrier is filled according to the corresponding position and the remaining positions are 0 and then IFFT is obtained to obtain the corresponding time domain sequence)
  • the correlation peak can be well given and the peak value is not affected by the carrier deviation.
  • the frame synchronization/timing synchronization position is obtained by the following equation
  • the differential correlation algorithm can resist the influence of any large carrier frequency offset, but the signal noise is enhanced due to the differential operation of the received sequence first, and the noise enhancement is very serious at low SNR. , causing the signal to noise ratio to deteriorate significantly.
  • N the value of N is 64, and 64 sets of differential correlation are implemented.
  • D(0), D(1), ..., D(N-1) are the selected N different difference values.
  • any one of the following two may be adopted based on performance requirements of the transmission system:
  • the difference value D(i) arbitrarily selects N different values and satisfies D(i) ⁇ L, where L is the length of the local time domain sequence corresponding to the known information.
  • a predetermined processing operation is performed on the N results (64) to obtain a final correlation result.
  • the predetermined processing operation herein, which are separately explained.
  • the first predetermined processing operation is a predetermined processing operation
  • the N different differential correlation results are subjected to predetermined processing operations by the following formula to obtain a final differential result.
  • the following equation is an example in which absolute values are added to obtain a final differential result.
  • the difference value selected according to this rule is obtained as After the difference correlation value, the adjacent two sets of differential correlation values are conjugate multiplied, and the conjugate multiplied by the N-1 group is obtained by the following formula.
  • the obtained N-1 group RM i,m can be weighted vector addition. Or the average difference result is averaged to get better performance than the first predetermined processing operation.
  • the following equation is an example of vector addition to obtain the final difference result.
  • the conjugate multiplication value can be obtained not only in the second predetermined processing operation but also the weight vector addition or average.
  • the final correlation result may also be obtained by adding or averaging the at least two differential correlation results directly by weighted absolute values in the first predetermined processing operation described above.
  • the position within a certain range of the maximum value position of the correlation value of the initial timing synchronization can be utilized. To initially determine the position of the leading symbol in the physical frame. The value corresponding to the position is used to further determine whether the received preamble symbol is included in the received signal, or the subsequent positional estimation and/or decoding operation is performed by using the position to further receive whether the signal includes the desired preamble symbol.
  • the preamble symbol including the three-segment structure desired to be received exists in the processed signal, that is, the baseband signal. Specifically, the detection is performed based on the result of the initial timing synchronization. If the result of the detection satisfies the preset condition, it is determined that there is a preamble symbol including the three-segment structure that is expected to be received in the baseband signal.
  • the satisfaction of the preset condition herein may mean that the result of the initial synchronization according to the initial timing itself satisfies the preset condition determination, and may also mean that the result of the synchronization according to the initial timing is not sufficient, and then according to other subsequent steps such as an integer multiple.
  • the frequency offset estimation and/or decoding result is determined.
  • the direct determination is based on the initial timing synchronization result
  • the preset condition includes the initial timing synchronization result to perform a specific operation, and then it is determined whether the maximum value of the operation result exceeds the threshold threshold.
  • the predetermined acquisition rule between the C-part, the A-part, and the B-part of the first three-segment structure and the second three-segment structure may be used. And/or predetermined processing rules, obtaining two sets of delay-related accumulated values, each group of three values, based on at least one of three delay-related accumulated values of each of the two groups, generating two sets of related results to be detected, thereby This detects and determines whether the leading symbol contains a three-segment structure and which three-segment structure is included.
  • the first group of to-be-detected correlation results meet the preset condition, determining that there is a preamble symbol of the first three-segment structure that is expected to be received in the baseband signal; and if the second group of to-be-detected correlation results meets the preset condition, Determining, in the baseband signal, a preamble symbol of a second three-segment structure that is desired to be received; In the case where both groups are satisfied, it indicates that the leading symbol contains two three-segment structures at the same time.
  • the initial timing synchronization resolves the emergency broadcast by any combination of any one or any of the following: the third part and the Different delay relationships of the same content between the two parts; and different delay relationships of the same content between the first part and the second part to distinguish between sending emergency broadcasts and ordinary broadcasts.
  • the receiving end will implement the multi-branch step S12-1 included in the above step S12: an initial timing synchronization manner, which is used for initially determining the position of the preamble symbol; and determining whether the correlation is based on the plurality of to-be-detected correlation results.
  • an initial timing synchronization manner which is used for initially determining the position of the preamble symbol; and determining whether the correlation is based on the plurality of to-be-detected correlation results.
  • the delayed sliding autocorrelation defining N1 of a certain value described above is one branch.
  • Each branch contains the above three delay-related accumulated values.
  • the receiving end simultaneously performs the above-mentioned delayed sliding autocorrelation branch of 2Q different N1 values, and then from 2Q U 2 '(n) ⁇ U 3 '* (n) or U ca '(nN A +N1) ⁇ U cb In the absolute value of '(n) ⁇ U ab '* (n), it is determined whether or not there is a desired leading symbol.
  • any of the absolute values does not exceed the threshold threshold, then there is no signal expected to be received in the baseband signal.
  • the received signal delays 1024 sample points and performs sliding autocorrelation with the received signal
  • the received signal delays 1528 sample points and the received signal of the demodulated frequency offset is subjected to sliding autocorrelation
  • the received signal delays 504 samples and performs sliding autocorrelation with the received signal after demodulation frequency offset
  • the received signal delays 1024 sample points and performs sliding autocorrelation with the received signal after demodulation frequency offset
  • the received signal delays 1544 sample points and the received signal of the demodulated frequency offset is subjected to sliding autocorrelation
  • the received signal delays 520 sample points and performs a sliding autocorrelation with the received signal after the demodulation frequency offset.
  • threshold threshold is used as a preset condition to determine whether or not there is a preamble symbol that is expected to be received
  • the current pilot symbol contains only one of the first three-segment structure and the second three-segment structure to identify the non-emergency broadcast, and the other is used to identify the emergency broadcast, which is parsed by the following.
  • Step S12-1 may obtain two branches according to a predetermined acquisition rule and/or a predetermined processing rule between the C segment, the A segment, and the B segment of the first three-segment structure and the second three-segment structure.
  • each branch has 3 values
  • step S12-2 includes detecting the correlation value to be detected of each of the two branches.
  • the fractional frequency offset estimation can also be performed by using the preliminary timing synchronization result of the (1) mode and/or the (2) mode.
  • the second small offset value can be calculated, and then U cb '(n) and U ab '(n) are conjugated.
  • the third small offset value can be calculated by taking the angle corresponding to the maximum value.
  • the angles in the logic operation block diagrams in FIGS. 7 and 8 above are used to obtain the small partial deviation, and the small partial estimation can be performed based on the second small offset value and any one or two of the third small offset values.
  • the transmission preamble symbol includes the features required by the (1)th preliminary timing synchronization mode and the (2) preliminary timing synchronization mode, based on any one of the first, second, and third small offset values or any at least The combination of two to get a small bias estimate.
  • the delay number may be adjusted within a certain range, for example, the delay number of some delay correlators should be increased or decreased by one.
  • the three delays are added to and subtracted from each other, and then multiple delay sliding autocorrelations are performed according to the obtained adjusted multiple delays and the number of delays.
  • the sliding delay autocorrelation is implemented according to the three delay numbers, and then the correlation is selected. The most obvious result is that the timing deviation can be estimated at the same time.
  • Step S12-2 includes an initial timing synchronization mode for initially determining the position of the preamble symbol in the physical frame. Further, after the initial synchronization, the integer multiple frequency offset estimation manner may also be performed based on the result obtained by the initial timing synchronization manner.
  • the receiving end may further perform integer multiple frequency offset estimation by using the fixed sequence, that is, the receiving method of the preamble symbol of the present invention may further include the following integer multiple frequency offset estimation step:
  • the first integer multiple frequency offset estimation method includes: intercepting a time domain signal including at least all or part of the time domain main signal according to the result of the initial timing synchronization, and using the frequency sweeping method to differentiate the intercepted time domain signal
  • a plurality of N frequency-swept time domain signals corresponding to the frequency offset values are obtained, and the known time domain signals obtained by inverse Fourier transform of the known frequency domain sequence and each of the frequency sweep time domain signals are obtained.
  • the maximum correlation peaks of the N cross-correlation results are compared, and the frequency offset value of the swept time domain signal corresponding to the largest cross-correlation result is an integer multiple frequency offset estimation value; and/or
  • the second integer multiple frequency offset estimation method includes:
  • the time domain signal of the length of the main body time domain signal is intercepted according to the result of the initial timing synchronization, and the obtained frequency domain subcarrier is cyclically shifted according to different shift values in the frequency sweeping range, and the effective subcarrier is intercepted.
  • An integer multiple frequency offset estimation value is obtained by using the correspondence between the bit value and the integer multiple frequency offset estimation value.
  • the time domain main signal A corresponds to the frequency domain structure one, that is, the frequency domain OFDM symbols respectively include a virtual subcarrier, a signaling sequence (referred to as SC) subcarrier, and a fixed sequence (referred to as The FC) subcarrier three parts, then the known frequency domain sequence mentioned below is a fixed subcarrier.
  • SC signaling sequence
  • FC fixed sequence
  • the first integer multiple frequency offset estimation method intercepts all or part of the time domain waveform of the received preamble symbol according to the position of the preamble symbol detected by the initial timing synchronization, and adopts a frequency sweep method, that is, a step of changing at a fixed frequency.
  • the path for example, corresponding to the integer multiple frequency offset interval, after the partial time domain waveform is modulated with different frequency offsets, several time domain signals are obtained.
  • T is the sampling period and f s is the sampling frequency.
  • A2 will be known as the sliding correlation signal with each of A1 y, select occur that A1 y maximum correlation peak, then
  • the frequency offset value y modulated by it is an integer multiple frequency offset estimation value.
  • the sweep range corresponds to the frequency offset range that the system needs to combat.
  • the frequency offset of positive and negative 500K is needed, and the system sampling rate is 9.14M, and the main body of the leading symbol is 2K length, then the sweep frequency range is That is [-114,114].
  • the second integer multiple frequency offset estimation method intercepts the main time domain signal A according to the position of the preamble symbol detected by the initial timing synchronization, and performs FFT to perform different shift values of the frequency domain subcarrier after the FFT. Cyclic shift, and then intercept the received sequence corresponding to the effective subcarrier, perform some operation (usually conjugate multiplication, or division) with the received sequence and the known frequency domain sequence, and then perform the result IFFT, IFFT
  • the result is a specific operation, such as taking the maximum diameter energy, or taking a number of large-path energy accumulations. Then a number of shift values, after several IFFTs, each time get an operation result, you will get several sets of operation results. Based on the results of the plurality of groups, it is determined which shift value corresponds to the integer multiple frequency offset estimation, thereby obtaining an integer multiple frequency offset estimation value.
  • the usual judging method is to select the corresponding shift value of the group with the largest energy as the integer octave bias estimation value based on the results of several groups.
  • the domain main signal A corresponds to the above-mentioned frequency domain structure
  • the following generalized offset estimation method can also be adopted.
  • the time domain main signal A of the corresponding symbol in the preamble symbol is subjected to Fourier transform to obtain a frequency domain OFDM symbol, and the transformed frequency domain OFDM symbol is cyclically shifted by the above-mentioned frequency sweep range, and the FC is on the subcarrier.
  • the position and the interval between the two fixed-sequence sub-carriers are differentially multiplied by the interval, and the correlation is performed with the differential multiplication value of the known fixed-sequence sub-carriers to obtain a series of correlation values, and the maximum correlation value is selected. Cyclic shift, you can get the integer octave bias estimate.
  • the frequency offset is compensated, and then the transmission signaling is analyzed.
  • the precise timing synchronization mode is performed by using the known information in the preamble symbol.
  • the fixed subcarrier sequence FC included in one or more frequency domain symbols is used to perform accurate timing synchronization
  • the step of determining the position of the preamble symbol in the physical frame and solving the signaling information carried by the preamble symbol is described in detail.
  • the step includes the following:
  • Determining the location of the preamble symbol includes determining a location of the preamble symbol in a physical frame based on a result of the detection that satisfies the preset condition;
  • the position of the preamble symbol is determined according to the portion of the value or the maximum value at which the peak value of the correlation value to be detected is large.
  • the channel estimation method is also included in the step of parsing the transmission signaling.
  • channel estimation can be performed by using the received signal containing a fixed sequence subcarrier and a known frequency domain fixed sequence subcarrier and/or its time domain signal corresponding to the inverse Fourier transform.
  • the time domain is performed and/or performed in the frequency domain, and details are not described herein again.
  • the position of the subsequent signaling symbol or the position of the data symbol may be obtained according to the parameter content and the determined position of the preamble symbol and based on this Parsing signaling symbols or data symbols.
  • the step of parsing the signaling signal includes: using all or part of the time domain waveform of the preamble symbol and/or all of the preamble symbol Or the frequency domain signal obtained by the Fourier transform of the partial time domain waveform to solve the signaling information carried by the preamble symbol.
  • the signaling sequence subcarrier set is generated based on a known set of signaling sequences.
  • the signal including the signaling sequence subcarrier includes: all or part of the time domain waveform of the received preamble symbol, or one of the one or more main body OFDM symbols after Fourier transform is intercepted from the preamble symbol or Multiple frequency domain OFDM symbols.
  • the signaling sequence subcarrier set is a set formed by filling each signaling sequence in the signaling sequence set onto the effective subcarrier.
  • one or more frequency domain OFDM symbols obtained by performing Fourier transform on the time domain signals of the N A length corresponding to the ODFM symbol body are intercepted; then, the zero carrier is removed, and the carrier position is removed according to the signaling subcarrier position.
  • the frequency domain decoding function is completed by performing specific mathematical operations on the channel estimation values and the known signaling sequence subcarrier sets.
  • the foregoing process may also be performed in the time domain, and the frequency domain symbols of the corresponding length generated by the known signalling sequence subcarrier set after zero padding at the appropriate position are inversely transformed by Fourier transform.
  • the time domain signaling waveform set is directly related to the time domain receiving signal for acquiring the accurate location of the multipath, and the one with the largest absolute value of the correlation value can also solve the signaling information of the frequency domain transmission, which will not be described here.
  • the present invention also provides the apparatus for generating a preamble symbol, the apparatus for generating a frequency domain symbol, and the apparatus for receiving a preamble symbol, the apparatus for generating the preamble symbol, the apparatus for generating a frequency domain symbol, and the reception of a preamble symbol.
  • Device and method for generating preamble symbols and frequency domain symbols in the above embodiments The generation method and the receiving method of the preamble symbol respectively correspond to each other, and the structural and technical elements of the device may be formed by the corresponding conversion method of the generating method and the receiving method, and the description thereof is omitted here.

Abstract

本发明提供了一种前导符号的生成方法、接收方法、及所涉及的频域符号的生成方法和相关的装置，其特征在于：根据从时域主体信号截取的部分时域主体信号生成前缀；根据该部分时域主体信号的全部或部分生成超前缀；基于所述循环前缀、所述时域主体信号和所述超前缀中至之少一生成时域符号，所述前导符号包含至少一个该时域符号，因而，当基于时域主体信号的全部或一定长度的部分作为前缀，可实现相干检测，解决了非相干检测性能下降以及在复杂频率选择性衰落信道下差分解码失效的问题，并且基于上述截取的时域主体信号的全部或部分生成超前缀，使得生成的前导符号具有良好的小数倍频偏估计性能和定时同步性能。

Description

前导符号的生成、接收方法和频域符号的生成方法及装置 技术领域
本发明涉及通信技术领域，特别涉及前导符号的生成、接收方法和频域符号的生成方法及装置。
背景技术
通常为了使OFDM系统的接收端能正确解调出发送端所发送的数据，OFDM系统必须实现发送端和接收端之间准确可靠的时间同步。同时，由于OFDM系统对载波的频偏非常敏感，OFDM系统的接收端还需要提供准确高效的载波频率估计方法，以对载波频偏进行精确的估计和纠正。
目前，OFDM系统中由物理帧组成，每一帧都通常有一个同步帧头，称为前导符号或bootstrap，实现发送端和接收端时间同步。前导符号是OFDM系统的发送端和接收端都已知的符号序列，前导符号做为物理帧的开始，通常包含P1符号。P1符号(preamble)或者bootstrap符号的用途包括有：
1)使接收端快速地检测以确定信道中传输的是否为期望接收的信号；
2)提供基本传输参数(例如FFT点数、帧类型信息等)，以使接收端可以进行后续接收处理；
3)检测出初始载波频偏和定时误差，进行补偿后达到频率和定时同步；
4)紧急警报或广播系统唤醒。
现有例如DVB_T2标准中提出了基于已有时域结构的P1符号设计，较好地实现了上述功能。但是，在低复杂度接收算法上仍然有一些局限。例如，在1024、542、或者482个符号的长多径信道时，定时粗同步会发生较大偏差，导致频域上估计载波整数倍频偏出现错误。另外，在复杂频率选择性衰落信道时，例如长多径时，DBPSK差分解码也可能会失效。而且，由于DVB_T2时域结构中没有循环前缀，若和需要进行信道估计的频域结构组合，将造成其频域信道估计性能严重下降的问题。
发明内容
本发明解决的问题是目前DVB_T2标准及其他标准中，DVB_T2时域结构不能适用于相干检测，而且前导符号在复杂频率选择性衰落信道下DBPSK差分解码失效，接收算法检测出现失败概率的问题。
为解决上述问题，本发明实施例提供了以下前导符号的生成方法、接收方法、及所涉及的频域符号的生成方法和相关的装置。
<方法一>
本发明实施例提供了一种前导符号的生成方法，其特征在于，包括如下步骤：根据从时域主体信号截取的部分时域主体信号生成前缀；根据该部分时域主体信号的全部或部分生成超前缀；基于循环前缀、时域主体信号和超前缀中至之少一生成时域符号，前导符号包含至少一个该时域符号。
可选地，在所提供的前导符号的生成方法中，进一步具有这样的特征，其中，基于依次排列的循环前缀、时域主体信号和超前缀生成时域符号，前导符号包含至少一个该时域符号。
可选地，在所提供的前导符号的生成方法中，进一步具有这样的特征，其中，前缀、超前缀的生成步骤中，包含：前缀从时域主体信号后部直接截取得到，超前缀是对与前缀对应的部分时域主体信号的全部或部分进行调制得到。
可选地，在所提供的前导符号的生成方法中，进一步具有这样的特征，其中，前缀、超前缀的生成步骤中，包含：对从时域主体信号后部截取出的部分按照第一预定处理规则进行处理形成为前缀，对从时域主体信号后部截取出的部分按照第二预定处理规则进行处理形成为超前缀，第一预定处理规则包括：直接拷贝；或乘以一个相同固定系数或预定不同系数，第二预定处理规则包括：当第一预定处理规则为直接拷贝时进行调制处理；或者当第一预定处理规则为乘以一个相同固定系数或预定不同系数时也乘以相应的系数后进行调制处理。
可选地，在所提供的前导符号的生成方法中，进一步具有这样的特征，其 中，超前缀的长度不超过前缀的长度。
可选地，在所提供的前导符号的生成方法中，进一步具有这样的特征，其中，超前缀的生成步骤包括：设置频移序列；将时域主体信号的部分或者全部乘以该频移序列以得到该时域主体信号的超前缀。
可选地，在所提供的前导符号的生成方法中，进一步具有这样的特征，其中，频移序列的调制频偏值根据时域主体信号对应的频域子载波间隔或者根据超前缀的长度来确定，频移序列可任意选择初始相位值。
可选地，在所提供的前导符号的生成方法中，进一步具有这样的特征，其中，前导符号通过以下方式传输信令信息：确定一种前缀的长度和超前缀的长度的组合前提下，在生成超前缀时，通过以不同的起始位置截取部分时域符号来实现传输不同的信令信息。
可选地，在所提供的前导符号的生成方法中，进一步具有这样的特征，其中，时域主体信号的长度为2048个采样周期，前缀的长度为520个采样周期，超前缀的长度为504个采样周期，超前缀在时域主体信号中截取的起始位置为第1528个采样。
可选地，在所提供的前导符号的生成方法中，进一步具有这样的特征，其中，设P1_A(t)是时域符号的时域表达式，NA为时域主体信号的长度，设LenC为前缀的长度，LenB为超前缀的长度，fSH为对时域主体信号进行调制的调制频偏值，T为采样周期，则前导符号包含有前缀、时域主体信号和超前缀信号的时域符号的时域表达式为：
Figure PCTCN2015076812-appb-000001
可选地，在所提供的前导符号的生成方法中，进一步具有这样的特征，其 中，时域主体信号的长度NA为2048，循环前缀LenC的长度为520，超前缀LenB的长度为504，则前导符号包含有循环前缀、时域主体信号和超前缀的时域符号的时域表达式为：
Figure PCTCN2015076812-appb-000002
可选地，在所提供的前导符号的生成方法中，进一步具有这样的特征，其中，时域主体信号利用至少一个比特信令来承载紧急广播标识，利用调制信号以调制信号长度在时域主体信号中截取的不同起始位置，以实现承载紧急广播标识。
可选地，在所提供的前导符号的生成方法中，进一步具有这样的特征，其中，时域主体信号基于对频域符号进行处理得到，可选地，在所提供的前导符号的生成方法中，进一步具有这样的特征，其中，频域符号的生成步骤包含：将在频域上所分别生成固定序列和信令序列进行排列后填充至有效子载波上。
<方法二>
另外，本发明的实施例还提供了一种频域符号的生成方法，其特征在于，包括如下步骤：将在频域上所分别生成固定序列和信令序列进行排列后填充至有效子载波上，用于形成频域符号。
可选地，在所提供的频域符号的生成方法中，进一步具有这样的特征，还包括：确定固定序列和信令序列的平均功率比，依据平均功率比分别生成固定序列和信令序列。
可选地，在所提供的频域符号的生成方法中，进一步具有这样的特征，其中：固定序列和信令序列之间的平均功率比取值2。
可选地，在所提供的频域符号的生成方法中，进一步具有这样的特征，其 中：固定序列和信令序列以预定交错排列规则进行排列，预定排列规则包含以下两种规则中的任意一种：呈奇偶交错或者偶奇交错进行排列；或把一部分信令序列放在奇数子载波，另一部分信令序列放在偶数子载波，且把一部分固定序列放在奇数子载波，另一部分固定序列放在偶数子载波。
可选地，在所提供的频域符号的生成方法中，进一步具有这样的特征，其中：信令序列的生成步骤包括：基于所预设的信令序列的长度和个数生成同一序列生成公式；基于同一序列生成公式选择不同的相位基值产生不同恒包络零自相关序列；以及根据所确定的信令序列的长度从得到的每一个恒包络零自相关序列中选取信令序列。
可选地，在所提供的频域符号的生成方法中，进一步具有这样的特征，其中：信令序列的生成步骤包括：基于所预设的信令序列的长度和个数确定序列生成若干个序列生成公式；针对每一个序列生成公式，选择不同的相位基值相应产生恒包络零自相关序列；以及根据所确定的信令序列的长度从得到的每一个恒包络零自相关序列中选取信令序列。
可选地，在所提供的频域符号的生成方法中，进一步具有这样的特征，其中：对所产生恒包络零自相关序列，还包括以下步骤：对所产生的恒包络零自相关序列进一步循环移位。
<方法三>
另外，本发明的实施例还提供了一种前导符号的接收方法，其特征在于，包括如下步骤：对接收信号进行处理；判断处理后的信号中是否存在期望接收的前导符号；在判断结果为是时，确定该前导符号的位置并解出该前导符号所携带的信令信息。
可选地，在所提供的前导符号的接收方法中，进一步具有这样的特征，其中：在判断处理后的信号中是否存在期望接收的前导符号，及在判断为是时确定该前导符号的位置并解出该前导符号所携带的信令信息的步骤中，包含以下 任意至少一种步骤：初始定时同步方式、整数倍频偏估计方式、精准定时同步方式、信道估计方式、解码分析方式以及小数倍频偏估计方式。
可选地，在所提供的前导符号的接收方法中，进一步具有这样的特征，其中：利用以下任意至少一种的结果来判断处理后的信号中是否存在期望接收的前导符号：初始定时同步、整数倍频偏估计、精准定时同步、信道估计、解码分析及小数倍频偏估计方式。
可选地，在所提供的前导符号的接收方法中，进一步具有这样的特征，其中：判断基带信号中是否存在期望接收的前导符号的步骤中，包含：通过初始定时同步方式初步确定前导符号的位置；以及基于初始定时同步方式的结果，判断处理后的信号中是否存在期望接收的前导符号。
可选地，在所提供的前导符号的接收方法中，进一步具有这样的特征，其中：初始定时同步方式包含以下：第一种初始定时同步方式：利用循环前缀、时域主体信号和超前缀三者中任意两个间的处理关系，对处理后的信号进行必要反处理进行延迟滑动自相关来获取累加相关值；以及基于累加相关值进行延迟关系匹配和/或特定的数学运算后,将所得处理值用于初始定时同步，初步确定前导符号的位置，和/或第二种初始定时同步方式：当前导符号中任意时域主体信号包含已知信号时，将时域主体信号依照预定N个差分值进行差分运算,并将已知信息对应的时域信号也进行差分运算,再将两者进行互相关得到N组与该N个差分值一一对应的差分相关的结果,基于该N组差分相关的结果进行初始同步，得到处理值，用于初步确定前导符号的位置，其中N≥1，其中，当基于第一初始定时同步方式和第二初始定时同步方式完成时，则将分别所得的处理值再进行加权运算，基于该加权运算值完成初始定时同步。
可选地，在所提供的前导符号的接收方法中，进一步具有这样的特征，其中：在确定该前导符号的位置并解出该前导符号所携带的信令信息的步骤中，包含：利用前导符号的全部或部分时域波形和/或该前导符号的全部或部分时域 波形经过变换后得到的频域信号，以解出该前导符号所携带的信令信息。
可选地，在所提供的前导符号的接收方法中，进一步具有这样的特征，还包括：当所接收的用于生成前导符号的频域符号包含分别由固定序列和信令序列进行排列后填至有效子载波的步骤时，还包括：利用固定序列做整数倍频偏估计或信道估计，该利用固定序列做整数倍频偏估计或信道估计的步骤，包括：根据所初步确定的该前导符号的位置，截取包含全部或部分固定子载波的信号；将该包含全部或部分固定子载波的信号与频域固定子载波序列或该频域固定子载波序列对应的时域信号进行运算，以得到整数倍频偏估计或信道估计。
<装置一>
另外，本发明的实施例提供了一种前导符号的生成装置，其特征在于，包括：前缀生成单元，根据从时域主体信号截取的部分时域主体信号生成前缀；超前缀生成单元，根据该部分时域主体信号的全部或部分生成超前缀；前导符号生成单元，基于循环前缀、时域主体信号和超前缀中至之少一生成时域符号，前导符号包含至少一个该时域符号。
<装置二>
另外，本发明的实施例提供了一种频域符号的生成装置，其特征在于，包括：序列生成单元，用于在频域上所分别生成固定序列和信令序列；频域符号生成单元，将固定序列和信令序列进行排列后填充至有效子载波上，用于形成频域符号。
<装置三>
另外，本发明的实施例提供了一种前导符号的接收装置，其特征在于，包括：接收处理单元，对接收信号进行处理；判断单元，判断处理后的信号中是否存在期望接收的前导符号；定位解析单元，用于在判断结果为是时确定该前导符号的位置并解出该前导符号所携带的信令信息。
现有技术相比，本发明技术方案具有以下有益效果：
根据本发明实施例提供的前导符号的生成方法、接收方法、及所涉及的频域符号的生成方法和相关的装置，当时域主体信号是OFDM符号时，基于时域主体信号的全部或一定长度的部分作为前缀，可实现相干检测，解决了非相干检测性能下降以及在复杂频率选择性衰落信道下DBPSK差分解码失效的问题，并且基于上述截取的循环前缀长度的时域主体信号的全部或部分生成调制信号，使得生成的前导符号具有良好的小数倍频偏估计性能和定时同步性能。
进一步地，可根据传输效率和鲁棒性的要求选择发送具有三段结构的时域符号做为前导符号；当前导符号包含具有三段结构的时域符号时，基于同一个OFDM符号主体，还可以通过利用从第一部分中选取第二部分的不同起点来传输信令，比如紧急广播，hook信息、发射机标志信息或者其他传输参数。
附图说明
图1是本发明的实施例中第一种三段结构的时域符号的示意图；
图2是本发明的实施例中第二种三段结构的时域符号的示意图；
图3是本发明的实施例中基于第一种三段结构的时域符号的获取处理的示意图；
图4是本发明的实施例中基于第二种三段结构的时域符号的获取处理的示意图；
图5是本发明的实施例中频域结构一按照第一预定交错排列规则的排列示意图；
图6是本发明的实施例中频域结构一按照第二预定交错排列规则的排列示意图；
图7是本发明的实施例中前导符号的接收方法中对应于三段结构CAB的待检测相关结果的逻辑示意图。
图8是本发明的实施例中前导符号的接收方法中对应于三段结构BCA的待 检测相关结果的逻辑示意图。
具体实施方式
{生成方法}
在本实施例中，提供了一种前导符号的生成方法，该前导符号的生成方法包括如下步骤：
根据从时域主体信号截取的部分时域主体信号生成循环前缀；
基于该部分时域主体信号的部分或全部生成调制信号；
基于循环前缀、时域主体信号和调制信号中至之少一生成时域符号，前导符号包含至少一个该时域符号。
图1是本发明的实施例中第一种三段结构的时域符号的示意图。图2是本发明的实施例中第二种三段结构的时域符号的示意图。
所生成的前导符号包含：
具有第一种三段结构的时域符号；或
具有第二种三段结构的时域符号。
通过图1和图2对上述前导符号中所包含的时域符号所具有的时域结构进行以下说明。该时域结构包含三段结构，该三段结构存在两种情况，第一种三段结构、第二种三段结构。
如图1中所述，第一种三段结构为：时域主体信号(A部分)、根据从时域主体信号截取的部分时域主体信号所生成的前缀(C部分)、基于该部分时域主体信号的部分或全部所生成的调制信号(B部分)。
如图2中所述，第二种三段结构为：时域主体信号(A部分)、根据从时域主体信号截取的部分时域主体信号所生成的前缀(C部分)、根据该部分时域主体信号所生成的超前缀(B部分)。
具体来说，将一段时域主体信号(图中以A标示)作为第一部分，齐第一部分的最末端按照预定获取规则取出一部分，按照第一预定处理规则进行处理 并复制到该第一部分的前部来生成第三部分(图中以C标示)从而作为前缀，同时，从第一部分的后部按照预定获取规则取出一部分，按照第二预定处理规则进行处理并复制到该第一部分的后部或者处理并复制到前缀的前部来生成第二部分(图中以B标示)从而分别相应作为后缀或超前缀，从而，分别生成如图1所示的B作为后缀的第一种三段结构(CAB结构)和B作为超前缀的如图2所示的第二种三段结构(BCA结构)。
关于从第一部分中获取出第三部分、第二部分后进行处理具体规则来说，第一预定处理规则包括：直接拷贝；或者对所取出部分中的每个采样信号乘以一个相同固定系数或预定不同系数。第二预定处理规则包括：当第一预定处理规则为直接拷贝时进行调制处理；或者当第一预定处理规则为所取出部分中的每个采样信号乘以一个相同固定系数或预定不同系数时也乘以相应的系数后进行调制处理。也就是，当第三部分是直接拷贝作为前缀时，第二部分进行调制处理后再作为后缀或者超前缀，而当第三部分是乘以相应系数时，第二部分也需要进行乘以系数并进行调制处理，再作为后缀或者超前缀。
图3是本发明的实施例中基于第一种三段结构的时域符号的获取处理的示意图。
本实施例中C段为A段的直接拷贝，而B段为A段的调制信号段，如图3所示，比如A的长度为1024，截取C的长度为520，而B的长度为504，其中在对C和B进行一定的处理时，可以对信号的每个采样乘以一个固定的系数，或每个采样乘以一个不同的系数。
B的数据范围不超过C的数据范围，即选择给调制信号段B的那部分A的范围不会超出截取作为前缀C的那部分A的范围。优选地，B的长度和C的长度之和为A的长度。
设NA为A的长度，设LenC为C的长度，LenB为调制信号段B的长度。设A的采样点序号为0,1,…NA-1.设N1为选择复制给调制信号段第二部分B的起 点对应的第一部分A的第一采样点序号，N2为选择复制给调制信号段第二部分B的终点对应的第一部分A的第二采样点序号。其中，第一采样点序号和第二采样点序号满足以下预定约束关系：
N2＝N1+LenB-1      (公式1)
通常，对第二部分B段实施的调制为调制频偏即乘以一个频移序列，调制M序列或其他序列等，本实施中以调制频偏为例，设P1_A(t)是A的时域表达式，则第一种C-A-B三段结构的时域表达式为
Figure PCTCN2015076812-appb-000003
                        (公式2)
其中，若时域主体信号为OFDM符号，调制频偏值fSH可选取为时域OFDM主体符号对应的频域子载波间隔即1/NAT，其中T为采样周期，NA为时域OFDM主体符号的长度，在本例中，NA为1024，取fSH＝1/1024T。且频移序列可任意选择初相，为了使相关峰值尖锐，fSH也可以选择为1/(LenBT)。
如图3所示，NA＝1024；LenC＝520，LenB＝504，N1＝520。此时CA段包含相同内容的自相关延迟为NA，CB段包含相同内容的自相关延迟为NA+LenB，而AB段包含相同内容的自相关延迟为LenB。
另外的一个实施例，C段的长度和B段完全相同，即B段可看作为C段的完整调整频偏段。
具体地，将循环前缀C拼接在所述时域OFDM符号A的前部作为保护间隔，并将所述调制信号段B拼接在所述OFDM符号的后部作为调制频偏序列以生成第一种三段结构的时域符号。举例来说，当NA＝1024时，具体表达式可如 下，
Figure PCTCN2015076812-appb-000004
                        (公式3)
图4是本发明的实施例中基于第二种三段结构的时域符号获取处理的示意图。
同理，第二种三段结构时域符号的时域表达式为，注意，为了使得接收端处理方法尽可能一致，因此在B-C-A的结构中，调制频偏值正好与C-A-B结构相反，且调制可任意选择初相。
Figure PCTCN2015076812-appb-000005
                        (公式4)
如图4所示，NA＝1024；LenC＝520，LenB＝504，N1＝504，此时CA段包含相同内容的自相关延迟为NA，BC段包含相同内容的自相关延迟为LenB，而BA段包含相同内容的自相关延迟为NA+LenB。
进一步地，当前导符号包含一个三段结构的符号时，无论是包含第一种三段结构，还是包含第二种三段结构，基于同一个OFDM符号主体，还可以通过下述方法来通过时域结构传输信令。
利用从所述第一部分中选取所述第二部分的不同起点来传输信令，即在生成所述调制信号时，通过以不同的起始位置截取所述部分时域符号来实现传输不同的信令信息。
比如紧急广播，hook信息、发射机标志信息或者其他传输参数。
举例来说，对于第一种三段结构，例如，所述预定长度为1024,LenC为512，LenB为256。
其中，N1可取512+i*160≤i<16,则可表示16种不同的取法,传输4bit信令参数。不同的发射机可以通过取不同的N1来传输该发射机的对应的标识、同一个发射机也可以通过分时地改变N1来发送传输参数。
又比如，用1比特信令来传输紧急广播标识EAS_flag。
若EAS_flag＝1，取N1＝512-L，即把NA为1024的OFDM符号的对应序号为512-L～1023-2L的采样点并调制频偏序列后生成B，放到A的后部。
若EAS_flag＝0，取N1＝512+L，即把NA为1024的OFDM符号的对应序号为512+L～1023的采样点并调制频偏序列后生成B，放到A的后部。
L的取值为8。
具体地，NA＝1024，LenC为520，LenB为504，N1＝520时表示EAS_flag＝0，而N1＝504时表示EAS_flag＝1；或者N1＝504时表示EAS_flag＝0，而N1＝520时表示EAS_flag＝1；
又比如，NA＝2048，LenC为520，LenB为504，N1＝1544时表示EAS_flag＝0，而N1＝1528时表示EAS_flag＝1；或者N1＝1528时表示EAS_flag＝0，而N1＝1544时表示EAS_flag＝1；
以具体的表达式来说，
当EAS_flag＝0时，C-A-B三段结构的时域表达式为
Figure PCTCN2015076812-appb-000006
和
Figure PCTCN2015076812-appb-000007
当EAS_flag＝1时，C-A-B三段结构的时域表达式为
Figure PCTCN2015076812-appb-000008
和
Figure PCTCN2015076812-appb-000009
除了用从所述第一部分中选取所述第二部分的不同起点来表示紧急广播之外，当前导符号仅包含一种三段结构时，还可以采用不同的三段结构来标识紧急广播。比如可以用发送第一种三段结构C-A-B表示EAS_flag＝0，而用第二种三段结构B-C-A表示EAS_flag＝1；或者，用发送第一种三段结构C-A-B表示EAS_flag＝1，而用第二种三段结构B-C-A表示EAS_flag＝0。
单个三段结构时域符号的检测利用CB段，CA段和BA段的延迟自相关来获取峰值，在使用两个三段结构时域符号拼接时，为了使得2个三段结构时域符号的自相关值能够相加，获得更鲁棒的性能，则2个三段结构时域符号各自的参数N1(即N1为选择复制给调制信号段B的起点对应的A的采样点序号)需要满足某种关系，设第一个符号的N1为N1_1，第二个符号的N1为N1_2，需要满足N1_1+N1_2＝2NA-(LenB+Lenc)。且如果对B段采用的调制是调制频偏的话，频偏值要正好相反。
用序号1表示C-A-B结构的符号，用序号2表示B-C-A结构的符号。则设P1_A(t)是A1的时域表达式，P2_A(t)是A2的时域表达式，则具有第一种三段结构的时域符号的时域表达式为：
Figure PCTCN2015076812-appb-000010
                        (公式5)
举例来说，NA＝2048；LenC＝520，LenB＝504，fSH＝1/2048T。
Figure PCTCN2015076812-appb-000011
则具有第二种三段结构的时域符号的时域表达式为：
Figure PCTCN2015076812-appb-000012
                        (公式6)
仍以上例来说，，NA＝2048；LenC＝520，LenB＝504，fSH＝1/2048T
Figure PCTCN2015076812-appb-000013
除了上文中介绍的前导符号(preamble)或bootstrap包含：具有第一种三段结构的时域符号；或具有第二种三段结构的时域符号。需要特别说明的是，本发明的前导符号或bootstrap不限于只包含C-A-B或者B-C-A的结构，也可还包含其他时域结构，比如传统CP结构等。
本发明还提供了一种频域符号的生成方法，以下分别由具有下述的频域结构一的频域OFDM符号对该频域符号的生成方法进行说明。
另外，结合上文的三段时域结构来看，存在着这样时、频域之间的对应关系，通常情况下，时域主体信号(A部分)是由频域OFDM符号经傅立叶反变换后形成时域OFDM符号而得到。然而，注意的是，本发明所提供的频域符号的的生成方法，不限于只使用在时域是采用上述如图1至图7中所示的三段结构的符号中，也可以适用于其他任意时域结构的符号。
设P1_X为对应的频域OFDM符号，P1_Xi作离散傅里叶反变换后得到时域OFDM符号：
Figure PCTCN2015076812-appb-000014
      (公式7)
其中，M为有效非零子载波的功率和。
在本发明中，阐述2种不同类型的P1_X的频域结构。
【频域结构一】
首先阐述第一种类型的P1_X的频域结构，定义为频域结构一。针对频域结构一来说，频域符号的生成方法，包括如下步骤：
在频域上分别生成固定序列和信令序列；以及
将所述固定序列和所述信令序列进行排列后填充至有效子载波，用于形成频域符号。
P1_X频域结构一即频域OFDM符号分别包括虚拟子载波、信令序列(称 为SC)子载波和固定序列(称为FC)子载波三部分。
对信令序列子载波和固定序列子载波按照预定交错排列规则进行排列后，将虚拟子载波分布在其两侧。该预定交错排列规则包含以下两种规则中的任意一种：
第一预定交错排列规则：呈奇偶交错或者偶奇交错进行排列；以及
第二预定交错排列规则：把一部分信令序列放在奇数子载波，另一部分信令序列放在偶数子载波，且把一部分固定序列放在奇数子载波，另一部分固定序列放在偶数子载波。
第一预定交错排列规则是SC和FC奇偶交错或者偶奇交错排放，这样FC做为导频规则排放；第二预定交错排列规则则需要把部分SC序列放在奇数子载波，剩余SC序列放在偶数子载波；同时需要把部分FC序列放在奇数子载波，剩余FC序列放在偶数子载波，这样避免FC或者SC全部放在奇或者偶子载波上，在某些特殊多径下会全部衰落掉，且这样的排放会给信道估计提高忽略不计的复杂度，因此是更优的选择。
设固定序列的长度为L(即承载固定序列的有效子载波的个数为L)、信令序列的长度为P(即承载信令序列的有效子载波的个数为P)，在本实施例中，L＝P。需要说明的是，当固定序列和信令序列的长度不一致时(例如P>L)，可以通过补零序列子载波的方式来实现固定序列和信令序列按上述规则交错排列。
图5是本发明的实施例中信令序列子载波、固定序列子载波及虚拟子载波按照第一预定交错排列规则的排列示意图。
如图5所示，本优选的实施方式中，本步骤包括：在有效子载波两侧分别填充一定的零序列子载波以形成预定长度的频域OFDM符号。
与上述时域结构中的时域主体信号A的长度NA为1024所对应，进行傅里叶变换FFT在频域中所形成的长度为NFFT为1024。
以下沿用以NFFT预定长度为1024的例子，零序列子载波的长度的G＝1024-L-P，两侧填充(1024-L-P)/2个零序列子载波。例如，L＝P＝353，则G＝318，两侧各填充159个零序列子载波。
按第一预定交错排列规则生成的频域OFDM符号包含以下步骤：
第(11)固定序列生成步骤：固定序列由353个复数组成，其模恒定，所述固定序列子载波的第n个值表示为：
Figure PCTCN2015076812-appb-000015
      (公式8)
其中，R为FC和SC的功率比值，SCi模恒定为1
Figure PCTCN2015076812-appb-000016
      (公式9)
固定序列子载波的弧度值ωn通过下述的表1中的第一预定固定子载波弧度值表确定；
表1第一预定固定子载波弧度值表(第一预定交错排列规则)
5.43 2.56 0.71 0.06 2.72 0.77 1.49 6.06 4.82 2.10
5.62 4.96 4.93 4.84 4.67 5.86 5.74 3.54 2.50 3.75
0.86 1.44 3.83 4.08 5.83 1.47 0.77 1.29 0.16 1.38
4.38 2.52 3.42 3.46 4.39 0.61 4.02 1.26 2.93 3.84
3.81 6.21 3.80 0.69 5.80 4.28 1.73 3.34 3.08 5.85
1.39 0.25 1.28 5.14 5.54 2.38 6.20 3.05 4.37 5.41
2.23 0.49 5.12 6.26 3.00 2.60 3.89 5.47 4.83 4.17
3.36 2.63 3.94 5.13 3.71 5.89 0.94 1.38 1.88 0.13
0.27 4.90 4.89 5.50 3.02 1.94 2.93 6.12 5.47 6.04
1.14 5.52 2.01 1.08 2.79 0.74 2.30 0.85 0.58 2.25
5.25 0.23 6.01 2.66 2.48 2.79 4.06 1.09 2.48 2.39
5.39 0.61 6.25 2.62 5.36 3.10 1.56 0.91 0.08 2.52
5.53 3.62 2.90 5.64 3.18 2.36 2.08 6.00 2.69 1.35
5.39 3.54 2.01 4.88 3.08 0.76 2.13 3.26 2.28 1.32
5.00 3.74 1.82 5.78 2.28 2.44 4.57 1.48 2.48 1.52
2.70 5.61 3.06 1.07 4.54 4.10 0.09 2.11 0.10 3.18
3.42 2.10 3.50 4.65 2.18 1.77 4.72 5.71 1.48 2.50
4.89 4.04 6.12 4.28 1.08 2.90 0.24 4.02 1.29 3.61
4.36 6.00 2.45 5.49 1.02 0.85 5.58 2.43 0.83 0.65
1.95 0.79 5.45 1.94 0.31 0.12 3.25 3.75 2.35 0.73
0.20 6.05 2.98 4.70 0.69 5.97 0.92 2.65 4.17 5.71
1.54 2.84 0.98 1.47 6.18 4.52 4.44 0.44 1.62 6.09
5.86 2.74 3.27 3.28 0.55 5.46 0.24 5.12 3.09 4.66
4.78 0.39 1.63 1.20 5.26 0.92 5.98 0.78 1.79 0.75
4.45 1.41 2.56 2.55 1.79 2.54 5.88 1.52 5.04 1.53
5.53 5.93 5.36 5.17 0.99 2.07 3.57 3.67 2.61 1.72
2.83 0.86 3.16 0.55 5.99 2.06 1.90 0.60 0.05 4.01
6.15 0.10 0.26 2.89 3.12 3.14 0.11 0.11 3.97 5.15
4.38 2.08 1.27 1.17 0.42 3.47 3.86 2.17 5.07 5.33
2.63 3.20 3.39 3.21 4.58 4.66 2.69 4.67 2.35 2.44
0.46 4.26 3.63 2.62 3.35 0.84 3.89 4.17 1.77 1.47
2.03 0.88 1.93 0.80 3.94 4.70 6.12 4.27 0.31 4.85
0.27 0.51 2.70 1.69 2.18 1.95 0.02 1.91 3.13 2.27
5.39 5.45 5.45 1.39 2.85 1.41 0.36 4.34 2.44 1.60
5.70 2.60 3.41 1.84 5.79 0.69 2.59 1.14 5.28 3.72
5.55 4.92 2.64              
第(12)信令序列生成步骤：信令序列生成步骤含两种，即下述的第一信令序列生成方式、第二信令序列生成方式。在本实施例中，在频域上生成信令序列可以采用如下两种方式中的任一种，下面详细描述这两种生成信令序列的具体方式。
第一信令序列生成方式：
1.1确定信令序列的长度和个数；
1.2基于所述信令序列的长度和个数确定CAZAC序列生成公式中的root值；其中，信令序列的长度小于或者等于root值，且root值大于或者等于信令序列的个数的两倍。优选地，root值选取为信令序列的长度。
例如，确定序列长度L以及信令个数。比如，要传N个bit，则信令个数num为2N并选择CAZAC序列生成公式中exp(jπqn(n+1)/root)的root值。其中，序列长度L小于或者等于root值，且root值要大于等于2*num。通常root值为质数。
1.3选择不同的q值产生CAZAC序列，其中q值的个数等于信令序列的个数，且任意两个q值之和不等于root值；且所产生的CAZAC序列需要经过循环移位，循环移位的位数由相应的root值和q值决定。
例如，选择num个不同的q0、q1、……、qnum-1产生CAZAC序列：
s(n)＝exp(jπqn(n+1)/root),n＝0,...root-1，      (公式10)
经过循环移位后的序列为：
sk(n)＝[s(k),s(k+1),...,s(L-1),s(0),...,s(k-1)]    (公式11)
其中，k是循环移位的位数。
需要说明的是，在本实施例中，选出的qi(0≤i≤num-1)必须满足下述条件：任何2个qi、qj(0≤i,j≤num-1)满足qi+qj≠root.
在上述条件下，优先选择使得整体频域OFDM符号PAPR低的序列。且如果L大于等于2*num，优先选择root＝L.这样序列的自相关值为零。
1.4根据所确定的信令序列的个数从所有的CAZAC序列中选取所述信令序列。需要说明的是，若L＝root，则不需要截取，所得到的CAZAC序列即可作为信令序列。
例如，将num个序列中每一个序列截取长度为L的连续部分序列或者全部序列作为信令序列。
举例来说，信令序列长度L＝353，数量num＝128，则可选择root为最接近的质数353。q的取值范围为1～352，每个序列循环移位位数的取值范围为1～353。在所有可选的信令序列中，优选出如下128组，其q值和循环移位位分别如下表2的q值取值表和表3的循环移位位数表所示：
表2：q值取值表格
1 9 10 16 18 21 28 29 32 35 49 51 53 54 55
57 59 60 61 65 68 70 74 75 76 77 78 82 84 85
86 88 90 95 96 103 113 120 123 125 126 133 134 135 137
138 140 141 142 145 147 148 150 151 155 156 157 161 163 165
167 170 176 178 179 181 182 184 185 187 194 200 201 204 209
210 217 222 223 224 225 229 232 234 235 237 239 241 244 246
247 248 249 251 252 253 254 255 262 270 272 273 280 282 290
291 306 307 308 309 311 313 314 315 317 320 326 327 330 331
333 336 338 340 342 345 347 349              
表3：循环移位位数表格
105 244 172 249 280 251 293 234 178 11 63 217 83 111 282
57 85 134 190 190 99 180 38 191 22 254 186 308 178 251
277 261 44 271 265 298 328 282 155 284 303 113 315 299 166
342 133 115 225 13 26 326 148 195 145 185 121 58 162 118
151 182 230 39 249 305 309 144 188 181 265 140 212 137 10
298 122 281 181 267 178 187 177 352 4 353 269 38 342 288
277 88 124 120 162 204 174 294 166 157 56 334 110 183 131
171 166 321 96 37 261 155 34 149 156 267 332 93 348 300
245 101 186 117 329 352 215 55              
第二信令序列生成方式：
2.1确定信令序列的长度和个数；
2.2基于所述信令序列的长度和个数确定CAZAC序列生成公式中若干个root值；其中，信令序列的长度小于或者等于所选择的若干个root值中的最小值，且所选择的若干个root值之和大于或者等于信令序列的个数的两倍。优选地，root值选取为信令序列的长度。
例如，确定序列长度L以及信令个数。比如，要传N个bit，则信令个数num为2N，并选择CAZAC序列生成公式中exp(jπqn(n+1)/root)的若干K个rootk(0≤k≤K-1)。其中，信令序列长度L小于或者等于所有rootk中的最小值，并且若干个rootk的和大于等于2*num，即
Figure PCTCN2015076812-appb-000017
通常rootk值为质数。
2.3针对每一个root值，选择不同的q值产生CAZAC序列，其中q值的个数小于或者等于相应的root值的1/2，且任意两个q值之和不等于相应的root值；且所产生的CAZAC序列需要经过循环移位，循环移位的位数由相应的root值和q值决定。
例如，针对每个rootk(0≤k≤K-1)，选择numk个不同的q0、q1、
Figure PCTCN2015076812-appb-000018
产生CAZAC序列exp(jπqn(n+1)/rootk),n＝0,...rootk-1。其中，
Figure PCTCN2015076812-appb-000019
且
Figure PCTCN2015076812-appb-000020
在本第二信令序列生成方式中，针对每一个root值，选择不同的q值产生CAZAC序列，以及所产生的CAZAC序列需要经过循环移位的方式可以参照上述方式1的描述，在此不再赘述。
需要说明的是，在本实施例中，选出的qi(0≤i≤numk-1)必须满足下述条件：任意2个qi、qj(0≤i,j≤numk-1)满足qi+qj≠rootk。
在上述条件下，优先选择使得整体频域OFDM符号PAPR低的序列。且可优先选择其中一个root＝L。这样该root产生的序列的自相关值为零。
2.4根据所确定的信令序列的个数从得到的每一个CAZAC序列中选取所述信令序列。需要强调的是，若其中某个root＝L，则根据选取为信令序列的长度的root值所产生的CAZAC序列确定所述信令序列。
例如，将num个序列中每一个序列循环截取长度为L的连续部分序列或者全部序列作为信令序列。
举例来说，例如，L＝353，num＝128。按第一信令序列生成方式优先选择root为353。然后，选择q＝1,2,…128。满足qi+qj≠353,(0≤i,j≤128-1)。最后，将每个序列截取到长度为353。
又例如，L＝350，num＝256。按第二信令序列生成方式选择root1为353、root2＝359,然后针对root1＝353，选出q＝1,2,3,…128共128个序列,qi+qj≠353。然后针对root2＝359，选出q＝100,101,102,…227共128个序列，总共256个序列.。最后将每个序列循环截取到长度为353。
以下，由第(12)信令序列生成的步骤具体根据第二信令序列生成方式共生成512个信令序列，即Seq0，Seq1，…Seq511，每个信令序列Seq0～Seq511再分别取相反数，即-Seq0～-Seq511，接收端利用相关值的正负来区分是正序列还是反序列，即共传送10bit信令信息，512个信令序列又可以进一步分为4组，每组128个信令序列，每组128个信令序列生成子步骤如下：
第1子步骤：生成基准序列zci(n)，其为长度为N的Zadoff-Chu序列zc(n)：
Figure PCTCN2015076812-appb-000021
    (公式12)
第2子步骤:通过拷贝两次zci(n)产生长度为2N的
Figure PCTCN2015076812-appb-000022
Figure PCTCN2015076812-appb-000023
    (公式13)
第3子步骤:从
Figure PCTCN2015076812-appb-000024
中某特定的起始位置ki截取长度为353的序列，产生SCi(n)：
SCi(n)＝zci *(ki-1+n),n＝0～352    (公式14)
每组信令序列Seq0～Seq127的N值，ui和移位值ki分别由各个相应的下述表4至表7预定信令序列参数表确定。
第一组序列Seq0～Seq127的N值，ui和移位值ki如下表4所示。
表4：第一组信令序列参数
Figure PCTCN2015076812-appb-000025
第二组序列Seq128～Seq255的生成步骤和第一组序列相同，其N值，ui和移位值ki如下表5所示。
表5：第二组信令序列参数
Figure PCTCN2015076812-appb-000026
第三组序列Seq256～Seq383的生成步骤和第一组序列相同，其N值，ui和移位值ki如下表6所示。
表6：第三组信令序列参数
Figure PCTCN2015076812-appb-000027
第四组序列Seq384～Seq511的生成步骤和第一组序列相同，其N值，ui和移位值ki下表7所示。
表7：第四组信令序列参数
Figure PCTCN2015076812-appb-000028
Figure PCTCN2015076812-appb-000029
第(13)排列填充步骤，将由所述(11)步骤和所述(12)步骤所得到的固定序列和信令序列按奇偶交错排放，填充虚拟子载波后，按如下公式形成所述频域OFDM符号，
Figure PCTCN2015076812-appb-000030
    (公式16)
图6是本发明的实施例中信令序列子载波、固定序列子载波及虚拟子载波按照第二预定交错排列规则的排列示意图。
如图6所示，把图中位于虚线左侧的前半部分的信令序列放在奇数子载波，图中虚线右侧的另一半部分的信令序列放在偶数子载波，且位于虚线左侧的前半部分固定序列放在偶数子载波，把位于虚线右侧的后部分固定序列放在奇数子载波。即P1_X0,P1_X1,…,P1_X1023按第二预定交错排列规则产生，在前半段SC放奇载波，FC放偶载波，而后半段SC放偶载波，FC放奇载波，将前后半部分的信令序列、固定序列的奇偶位置相交换。这样的固定序列子载波
Figure PCTCN2015076812-appb-000031
信令序列子载波
Figure PCTCN2015076812-appb-000032
所处的奇偶位置可以互换，对传输性能无任何影响。
填充虚拟载波即零序列子载波时，左右两侧填充的零序列子载波长度也可以不同，但不宜相差过多。
下面继续给出按第二预定交错排列规则优化生成的频域符号的具体实施例。按第二预定交错排列规则生成的频域OFDM符号包含以下步骤：
第(21)固定序列生成步骤，该固定序列生成步骤与上述第(11)固定序列生成步骤中相同，仅固定序列子载波弧度值ωn的取值通过第二预定固定子载波弧度值表来确定；其中，第二预定固定子载波弧度值表通过如下表8所示：
表8固定子载波弧度值表(按第二预定交错排列规则)
0.63 2.34 5.57 6.06 0.55 5.68 2.20 1.58 2.23 4.29
1.80 3.89 4.08 2.41 5.06 0.10 4.49 4.15 4.99 6.18
0.86 4.31 3.08 0.73 1.67 5.03 4.26 1.73 5.58 2.74
5.06 1.23 1.67 1.31 2.19 5.90 2.13 3.63 3.90 0.73
4.13 5.90 5.00 1.78 6.10 2.45 2.00 3.61 1.72 5.90
4.07 0.39 4.72 2.73 4.67 3.56 4.13 3.07 3.74 4.87
1.54 4.28 1.88 2.96 3.07 4.13 1.97 5.69 4.45 2.07
6.05 4.88 3.39 2.55 5.83 1.86 1.65 4.23 0.46 3.24
1.39 0.19 0.66 4.13 4.83 2.26 2.19 3.06 5.66 0.66
5.19 5.04 4.62 3.64 0.66 3.52 1.18 4.18 5.93 5.51
1.05 2.18 5.87 1.27 0.92 0.66 5.75 0.16 5.04 0.54
5.68 0.13 4.76 0.56 1.57 1.59 4.50 3.18 0.82 3.84
4.39 5.53 2.25 3.20 4.04 6.03 4.41 0.32 1.39 5.06
4.67 3.20 4.63 0.88 6.00 3.99 0.31 3.72 4.17 3.37
4.77 0.30 4.85 2.65 0.88 3.13 1.77 6.05 0.46 1.93
4.25 1.47 6.12 1.18 3.19 3.00 2.88 5.43 1.01 2.96
2.16 1.17 4.77 6.07 5.32 3.55 1.64 4.35 5.10 3.87
2.79 4.57 0.51 3.27 2.42 1.52 1.40 0.19 0.35 4.96
6.04 4.90 5.47 5.55 1.40 1.91 4.62 4.22 2.11 4.14
2.33 2.75 2.68 2.06 4.86 0.34 0.47 3.13 2.97 0.05
5.75 1.51 6.22 2.48 5.10 5.20 2.18 2.31 4.29 3.09
3.93 5.47 3.22 1.84 4.67 1.35 3.04 0.60 0.62 5.09
6.04 5.39 2.71 2.47 1.86 2.69 1.75 4.94 5.98 1.08
5.99 3.84 3.67 5.53 1.59 5.60 1.22 5.35 4.44 2.72
5.97 5.08 2.32 0.13 4.52 2.18 1.36 5.72 4.76 2.98
5.30 1.71 4.31 2.05 1.68 4.61 3.86 2.52 5.36 2.39
3.29 1.47 6.05 0.48 5.57 1.29 4.88 5.97 0.53 0.88
5.43 2.12 3.97 2.61 2.51 0.50 6.00 5.86 5.35 1.15
5.38 4.42 5.05 0.96 2.41 4.84 0.79 4.99 0.51 1.32
5.09 1.33 2.83 2.27 4.36 0.53 5.89 4.98 5.33 2.12
2.35 0.59 1.94 1.65 4.44 2.99 4.37 0.01 1.64 0.08
5.34 4.09 2.14 3.31 3.69 1.38 5.95 3.31 2.44 4.81
4.03 4.80 0.39 3.28 4.57 0.30 4.66 2.21 4.22 2.20
3.98 4.78 3.97 6.17 5.59 2.78 5.92 3.61 1.41 0.88
5.24 5.47 2.38 2.42 3.22 5.38 5.02 5.10 3.06 2.43
1.51 4.52 4.85              
第(22)信令序列生成步骤，该信令序列生成步骤与上述第(12)信令序列生成步骤相同，
第(23)排列填充步骤，由第(21)步骤和第(22)步骤所得到的信令序列和固定序列按奇偶再偶奇交错排列后，左右两侧填充零载波后，按如下公式形成频域OFDM符号，
Figure PCTCN2015076812-appb-000033
  (公式17)
{接收方法}
本实施例中还提供了前导符号的接收方法，该前导符号的接收方法，适用于发送端以预定生成规则所生成的前导符号。
在预定生成规则中，所生成的前导符号包含如本实施例中上述从时域角度所说明的例如与该第一种三段结构和/或该第二种三段结构涉及的所有的技术要素，和/或包含如本实施例中上述从频域角度所说明的例如与频域结构一涉及所有的技术要素，在此不进行重复赘述，因而简言之，所适用的预定生成规则不失一般性地包含上述从时域角度说明的前导符号的生成方法和从频域角度说明的频域符号的生成方法。
以下针对预定生成规则所生成的前导符号分别满足具有上述三段结构的时域符号、对应于具有上述频域结构一的频域符号来进行前导符号的接收方法的说明。
本实施例提供了一种前导符号的接收方法，包含以下步骤：
步骤S11：对接收信号进行处理；
步骤S12：判断处理后的信号中是否存在期望接收的上述包含三段结构的前导符号；
步骤S13：在上述判断结果为是的情况下，确定该前导符号位置并解出该前导符号所携带的信令信息，
其中，接收的前导符号包含发送端根据预定生成规则以任意数量第一种三段 结构和/或第二种三段结构自由组合生成的至少一个时域符号，
如上述的第一种三段结构包含：时域主体信号、基于该时域主体信号全部或部分生成的前缀、以及基于该部分时域主体信号的全部或部分生成的后缀。
如上述的第二种三段结构包含：时域主体信号、基于该时域主体信号的全部或部分生成的前缀、以及基于该部分时域主体信号的全部或部分生成的超前缀。
如步骤S11中所述，对接收到的所述物理帧信号进行处理以得到基带信号。通常接收端接收到的信号为模拟信号，因此需要先对其进行模数转换以得到数字信号，再进行滤波、下采样等处理后得到基带信号。需要说明的是，若接收端接收到的是中频信号，在对其经过模数转换处理后还需要进行频谱搬移，然后再进行滤波、下采样等处理后得到基带信号。
如步骤S12所述，判断所述基带信号中是否存在期望接收的上述包含三段结构的前导符号。
具体来说，首先，接收端将判断接收到的基带信号中是否存在期望接收的前导符号，即接收到的信号是否符合接收标准，例如接收端需要接收DVB_T2标准的数据，则需要判断接收到的信号是否包含DVB_T2标准的前导符号，同理，这里需要判断接收到的信号是否包含C-A-B和/或B-C-A三段结构的时域符号。
在判断得到的所述处理后的信号中是否存在期望接收的所述前导符号及在判断为是时，确定该前导符号的位置并解出该前导符号所携带的信令信息的步骤中即、上述步骤S12和步骤S13中，包含以下任意至少一种步骤：初始定时同步、整数倍频偏估计、精准定时同步、信道估计、解码分析以及小数倍频偏估计。
可利用以下任意一种方式或者任意至少两种方式自由组合进行可靠度判断，即来判断处理后的信号中是否存在期望接收的前导符号：初始定时同步方式、整数倍频偏估计方式、精准定时同步方式、信道估计方式、解码结果分析方式以及小数倍频偏估计方式。
该步骤S12包含S12-1初始定时同步方式，用于初步确定前导符号在物理帧 中的位置，还包含S12-2基于初始定时同步方式的结果，判断所述基带信号中是否存在期望接收的上述包含三段结构的前导符号。初始定时同步方式可采取以下通过下述第(①)初始定时同步方式和第(②)初始定时同步方式中的任意一种或两种组合来完成初始定时同步。
[第(①)初始定时同步方式]
下面具体介绍第(①)初始定时同步方式。第(①)初始定时同步方式包含以下步骤：
利用所述循环前缀、所述时域主体信号和所述调制信号三者中任意两个间的处理关系，对处理后的信号进行必要反处理和/或信号解调后进行延迟滑动自相关来获取累加相关值；以及
基于累加相关值进行延迟关系匹配和/或特定的数学运算后,将所得处理值用于初始定时同步，初步确定前导符号的位置。
其中，依照期望接收的三段结构中的第三部分C、第一部分A以及第二部分B两两之间的处理关系和/或调制关系，对基带信号进行必要的反处理和/或信号解调后进行延迟滑动自相关，得到三段结构中第三部分C和第一部分A之间、第一部分A和第二部分B、和第三部分C和第二部分B之间的三个累加相关值即Uca'(n)，Ucb'(n)，Uab'(n)中任意一个或任意至少两个。基于至少一个上述的累加相关值得到待检测相关值。
举例来说，假设三段结构为C-A-B结构，
基于第三部分C与第一部分A的延迟关系，将接收信号进行延迟滑动自相关，其延迟相关表达式Uca(n)及延迟相关累加值Uca'(n)如下：
Uca(n)＝r(n)r*(n-NA)
Figure PCTCN2015076812-appb-000034
    (公式18-1；18-2)
可选择对Uca'(n)进行能量归一化。
即
Figure PCTCN2015076812-appb-000035
    (公式19)
基于第二部分B与第三部分C的处理关系及调制频偏值，将接收信号进行延迟滑动自相关并解调频偏，注意其延迟相关表达式Ucb(n)及延迟相关累加值Ucb'(n)如下：
Figure PCTCN2015076812-appb-000036
Figure PCTCN2015076812-appb-000037
    (公式20-1；20-2)
同样可选择对Ucb'(n)进行能量归一化。
基于第二部分B与第一部分A的处理关系及及调制频偏值，将接收信号进行延迟滑动相关，其延迟相关表达式Uab(n)及延迟相关累加值Uab'(n)如下：
Figure PCTCN2015076812-appb-000038
Figure PCTCN2015076812-appb-000039
    (公式21-1；21-2)
同样可选择对Uab'(n)进行能量归一化。
其中，corr_len可取1/fSHT,以避免连续波干扰或者取LenB以使得峰值尖锐。
利用延迟相关累加值Uca'(n)、Ucb'(n)、Uab'(n)进行需要的延迟匹配并进行数学运算，数学运算包含相乘或相加，比如Ucb'(n)·Uab '*(n)，或者
Uca'(n-NA+N1)·Ucb'(n)·Uab '*(n)来得到运算值即待检测相关值1。
图7是本发明的实施例中对应于三段结构CAB的待检测相关结果的逻辑示意图。其中图中的C,A,B分别表示C段，A段和B段信号的长度，且滑动平均滤波器可以是功率归一化滤波器。其中A为NA，B为LenB，C为LenC。
再举例来说，假设三段结构为B-C-A结构，
基于第三部分C与第一部分A的延迟关系，将接收信号进行延迟滑动自相关，其延迟相关表达式Uca(n)及延迟相关累加值Uca'(n)如下：
Uca(n)＝r(n)r*(n-NA)
Figure PCTCN2015076812-appb-000040
    (公式22-1；22-2)
可选择对Uca'(n)进行能量归一化。
即
Figure PCTCN2015076812-appb-000041
    (公式23)
基于第二部分B段与第三部分的C段的处理关系及调制频偏值，将接收信号进行延迟滑动自相关并解调频偏，注意其延迟相关表达式Ucb(n)及延迟相关累加值Ucb'(n)如下：
Figure PCTCN2015076812-appb-000042
Figure PCTCN2015076812-appb-000043
    (公式24-1；24-2)
同样可选择对Ucb'(n)进行能量归一化。
基于第二部分B段与第一部分A段的处理关系及及调制频偏值，将接收信号进行延迟滑动相关，其延迟相关表达式Uab(n)及延迟相关累加值Uab'(n)如下：
Figure PCTCN2015076812-appb-000044
Figure PCTCN2015076812-appb-000045
    (公式25-1；25-2)
同样可选择对Uab'(n)进行能量归一化。
其中，corr_len可取1/fSHT,以避免连续波干扰或者取LenB以使得峰值尖锐。
利用延迟相关累加值Uca'(n)、Ucb'(n)、Uab'(n)进行需要的延迟匹配并进行数学运算，该数学运算包含相加或相乘，比如Ucb '*(n-NA)·Uab'(n)，或者Uca'(n)·Ucb '*(n-NA)·Uab'(n)来得到运算值即待检测相关值2。
图8是本发明的实施例中对应于三段结构BCA的待检测相关结果的逻辑示意图。
图7和图8中的相同的部分只需要一套接收资源，图示为了清晰表述故分开。其中图中的C,A,B分别表示C段，A段和B段信号的长度，且滑动平均滤波器可以是功率归一化滤波器。其中A为NA，B为LenB，C为LenC。
基于待检测相关结果1和/或待检测相关结果2形成初步定时同步的相关值。
进一步地，当发送前导符号同时包含以下两种情况(a)和(b)时，
(a)所述时域主体信号中包含已知信息；
(b)以及检测到所述时域符号具有所述C-A-B三段结构，
通过上述第(①)初始定时同步方式和下述第(②)初始定时同步方式中的任意一种或两种组合来完成初始定时同步。其中当基于两种完成时，则将第(①)初始定时同步方式所得的第一初步同步运算值和第(②)初始定时同步方式所得的第二初步同步运算值再进行加权运算，基于该加权运算值完成初始定时同步。
[第(②)初始定时同步方式]
下面具体介绍第(②)初始定时同步方式。
其中,当任一C-A-B和/或B-C-A的主体信号A包含已知信号比如固定子载波时,或者比如前导符号包含若干个C-A-B和/或B-C-A的三段结构的时域符号，其中某个时域符号的主体信号A为已知信号时，也就是当所述前导符号中任意 三段结构中时域主体信号包含已知信号时，第(②)初始定时同步方式可通过将时域主体信号A依照预定N个差分值进行差分运算,并将已知信息对应的时域信号也进行差分运算,再将两者进行互相关得到N组与该N个差分值一一对应的差分相关的结果,基于该N组差分相关的结果进行初始同步，得到处理值，用于初步确定前导符号的位置，其中N≥1。
下面描述第(②)初始定时同步方式中差分相关的具体过程，首先介绍单组差分相关的过程。
确定差分值，将接收基带数据进行按差分值的差分运算，将已知信息所对应的本地时域序列也进行按差分值的差分运算，然后再将这两个差分运算的结果进行互相关，得到对应于该差分值的差分相关结果。这单组的差分相关结果的运算过程为现有技术。设差分值为D，接收基带数据为rn，每一步具体公式描述如下；
首先，将接收基带数据进行按差分值的差分运算
Figure PCTCN2015076812-appb-000046
      (公式26)
经过差分运算后，载波频偏带来的相位旋转变成了固定的载波相位ej2πD△f，这里△f表示载波频率偏差。
同时将本地时域序列(比如固定子载波按相应位置填充而其余位置0后做IFFT后得到对应的时域序列)也进行差分运算
Figure PCTCN2015076812-appb-000047
    (公式27)
然后将差分之后的接收数据和本地差分序列进行互相关，得到
Figure PCTCN2015076812-appb-000048
    (公式28)
在系统没有多径，也没有噪声的情况下，
Figure PCTCN2015076812-appb-000049
    (公式29)
Figure PCTCN2015076812-appb-000050
可以很好地给出相关峰，且峰值不受载波偏差影响。帧同步/定时同步位置利用如下式得到
Figure PCTCN2015076812-appb-000051
    (公式30)
从上述单组差分相关运算过程可知，差分相关算法可以对抗任意大载波频偏的影响，但是由于先将接收序列进行差分运算，使得信号噪声增强，而且在低信噪比下，噪声增强非常严重，造成信噪比显著恶化。
为了避免上述问题，因此不止单组差分相关运算，可实施多组差分相关，比如N的取值为64，实施64组差分相关，得到
Figure PCTCN2015076812-appb-000052
其中D(0)，D(1)，…,D(N-1)为选择的N个不同的差分值。
对N个结果进行特定数学运算，得到最终相关结果。
本实施例中，针对多组差分相关(64组)按照预定差分选定规则被选定出的过程，可基于传输系统的性能需求采用以下两种中任意一种：
(1)第一预定差分选定规则：差分值D(i)任意选择N个不同值且满足D(i)<L，其中L为已知信息相对应的本地时域序列的长度。
(2)第二预定差分选定规则：差分值D(i)选择N个为等差数列的不同值且满足D(i)<L，即D(i+1)-D(i)＝K，K为满足
Figure PCTCN2015076812-appb-000053
的常整数，其中L为已知信息相对应的本地时域序列的长度。
对这N个结果(64个)进行预定处理运算，得到最终相关结果，这里的预定处理运算的优选实施例有两种，分别进行阐述。
第一种预定处理运算：
差分值D(i)可任意选择N个不同的值，满足D(i)<L。因为，所任意选择的差分值D(i)，每组差分相关后的相位ej2πD(i)△f i＝0,...,N-1各不相同，不能直接矢量相加，所以仅能够加权绝对值相加或平均。通过以下公式对N个不同的差分相关结果进行预定处理运算，得到最终差分结果。下式为绝对值相加得到最终差分结果的例子。
Figure PCTCN2015076812-appb-000054
    (公式31)
第二种预定处理运算：
差分值D(i)可任意选择N个不同的值，满足D(i)<L，且满足D(i)为等差数列，即D(i+1)-D(i)＝K，K为满足
Figure PCTCN2015076812-appb-000055
的常整数。
按此规则选择的差分值，得到如
Figure PCTCN2015076812-appb-000056
的差分相关值后，再将相邻2组差分相关值进行共轭相乘，通过以下公式得到N-1组共轭相乘后的值。
Figure PCTCN2015076812-appb-000057
    (公式32)
因为，通过此共轭相乘将原本每组不同的相位ej2πD(i)△f变成了相同的ej2πK△f，所以，得到的N-1组RMi,m可进行加权矢量相加或平均得到最终差分结果，以得到较之第一种预定处理运算更好的性能。下式为矢量相加得到最终差分结果的例子。
Figure PCTCN2015076812-appb-000058
    (公式33)
需要说明的是，当差分值D(i)是采用上述第二预定差分选定规则情况下，不仅可匹配述第二种预定处理运算中获得共轭相乘值再进行加权矢量相加或平均以得到最终相关结果，还可匹配按照上述第一预定处理运算中直接对至少两个差分相关结果通过加权绝对值相加或平均以得到最终相关结果。
基于运算Rdc,m得到初始定时同步的相关值。
无论是采用第(①)初始定时同步方式还是第(②)初始定时同步方式，假定接收信号中包含期望的前导符号，都可以利用初始定时同步的相关值的最大值位置的一定范围内的位置来初步确定前导符号在物理帧中的位置。利用此位置对应的值来进一步判定接收信号中是否包含期望的前导符号，或利用该位置再进行后续的整偏估计和/或解码等操作，来进一步接收信号中是否包含期望的前导符号。
在基于上述初始定时同步方式的结果，判断所述处理后信号即基带信号中是否存在期望接收的上述包含三段结构的前导符号。具体包括基于初始定时同步的结果进行检测，若检测的结果满足预设条件，则确定所述基带信号中存在期望接收的包含三段结构的前导符号。进一步地，这里的满足预设条件即可以指单一根据初始定时同步的结果本身满足预设条件确定，也可指仅根据初始定时同步的结果本身不足以确定，再根据后续的其他步骤比如整数倍频偏估计和/或解码结果确定。
假设根据初始定时同步结果直接判定时，可基于是否满足预设条件来判定，该预设条件包含初始定时同步结果进行特定运算，然后判断运算结果的最大值是否超过阈值门限来判定。
特别地，上述第(①)初始定时同步方式的具体实施中，可根据第一种三段结构和第二种三段结构的C部分、A部分以及B部分的两两之间的预定获取规则和/或预定处理规则，得到两组延迟相关累加值，每组3个值，基于这2组中每组的三个延迟相关累加值中的至少一个生成两组待检测相关结果，从而，对此进行检测并判断前导符号中是否包含三段结构以及包含哪一种三段结构。
比如，若第一组待检测相关结果满足预设条件，则确定所述基带信号中存在期望接收的第一种三段结构的前导符号；若第二组待检测相关结果满足预设条件，则确定所述基带信号中存在期望接收的第二种三段结构的前导符号；若 发生两组都满足的情况，则表明前导符号中同时包含两种三段结构。
当发送端用从所述第一部分中选取所述第二部分的不同起点来传输信令时，初始定时同步通过以下任意一种或任意两种相自由组合来解析紧急广播：第三部分与第二部分之间相同内容的不同延迟关系；以及第一部分与第二部分之间相同内容的不同延迟关系，以区别发送紧急广播和普通广播。
举例来说，接收端将实施多支路上述步骤S12中所包含的S12-1步骤：初始定时同步方式，用于初步确定前导符号的位置的步骤；再基于多个待检测相关结果判断出是否存在期望接收的前导符号以及所传输的时域信令。
例如，当前导符号利用不同的B从A截取的起点位置N1来传输Q比特信令时，定义上述的某个取值的N1的延迟滑动自相关为一个支路。每个支路包含上述3个延迟相关累加值。接收端同时进行2Q种不同N1取值的上述延迟滑动自相关支路，然后从2Q个U2'(n)·U3 '*(n)或Uca'(n-NA+N1)·Ucb'(n)·Uab '*(n)的绝对值中，判断是否存在期望的前导符号。
如果任意一个绝对值都没有超过阈值门限，则表明基带信号中不存在期望接收的信号。比如用N1为504或520来传输1比特紧急警报或广播系统标识时，其中N1＝520表示为正常前导符号，N1＝504表示为紧急警报或广播系统，则进行2个支路的上述S21-1步骤。
比如，紧急警报广播标志为0的支路，即N1＝520，采取：
接收信号延迟1024个采样点与接收信号进行滑动自相关；
接收信号延迟1528个采样点与解调频偏后的接收信号进行滑动自相关；
接收信号延迟504个采样点与解调频偏后的接收信号进行滑动自相关，及
比如，紧急警报广播标志为1的支路，即N1＝504，采取：
接收信号延迟1024个采样点与解调频偏后的接收信号进行滑动自相关；
接收信号延迟1544个采样点与解调频偏后的接收信号进行滑动自相关；
接收信号延迟520个采样点与解调频偏后的接收信号进行滑动自相关。
若采用阈值门限的方法作为预设条件来判定、进行检测是否存在期望接收的前导符号的情况下，
若N1＝520的支路待检测相关值最大值超过阈值门限，则表明基带信号为期望信号，且前导符号出现，EAS_flag＝0；相反，若N1＝504的待检测相关值最大值超过阈值门限，则表明EAS_flag＝1；若2组都没有超过阈值门限,则表明该基带信号不是期望信号。
当前导符号仅包含第一种三段结构和第二种三段结构中的其中一种来标识非紧急广播，则利用另一种来标识紧急广播，通过以下来进行解析。
步骤S12-1可根据第一种三段结构和第二种三段结构的C段、A段以及B段的两两之间的预定获取规则和/或预定处理规则，得到两个支路的上述S12-1步骤，每支路3个值，且步骤S12-2中包含对这两个支路中每支路的待检测相关值进行检测。其中若第一支路检测结果满足预设条件，则确定所述基带信号中存在期望接收的第一种三段结构,且表明EAS_flag＝0；若第二支路检测结果满足预设条件，则确定所述基带信号中存在期望接收的第二种三段结构,且表明EAS_flag＝1；若发生两个支路都满足的情况，需要另行判断，比如可以以两组的峰噪比的明显性来进行判断紧急广播。
进一步地，初步完成初始定时同步后,利用第(①)方式和/或第(②)方式的初步定时同步结果还可以进行小数倍频偏估计。
当第(①)初步定时同步方式时，取Uca'(n)中最大值的角度，可算出第2小偏值，再将Ucb'(n)和Uab'(n)共轭相乘(对应C-A-B结构)或者Uab'(n)和Ucb'(n-NA)共轭相乘(对应B-C-A结构)后，也取最大值对应的角度，可算出第3小偏值。如上图7和图8中逻辑运算框图中的角度用于求小偏的示意部分，可基于第2小偏值，第3小偏值的任意之一和之二来进行小偏估计。
针对小数倍频偏估计的算法，具体举例来说，当采用第(②)初步定时同步方式时，
Figure PCTCN2015076812-appb-000059
i＝0,...N-1，取其最大值，对应的相位为ej2πK△f，可算出△f并转换成相应第1小偏值。
当发送前导符号包含第(①))初步定时同步方式和第(②)初步定时同步方式实施所需的特征时，基于第1、第2、第3小偏值的任意之一或者任意至少之二的组合来得到小偏估计值。
需要说明的是，考虑到系统采样钟偏差的影响，在上述实施例中，可以将应有延迟数进行一定范围内的调整，例如将其中一些延迟相关器应有的延迟数加减一，形成本身和加减一后的三个延迟数，再依据所得调整后多个延迟数及应有延迟数进行多个延迟滑动自相关，例如依据这三个延迟数实施滑动延迟自相关，再选择相关结果最为明显的那个，同时可以估计出定时偏差。
不失一般性地，若当前导符号中除具有C-A-B或者B-C-A的结构，还包含其他时域特性时，除利用上述C-A-B或者B-C-A的结构特点的定时同步方法，再叠加上针对其他时域结构特点实施的其他的定时同步方法，并不脱离本发明所描述的精神。
步骤S12-2包含初始定时同步方式，用于初步确定前导符号在物理帧中的位置。进一步地，初始同步后，还可以基于所述初始定时同步方式所得的结果进行所述整数倍频偏估计方式。
进一步地，当时域主体信号A对应上述频域结构时，接收端还可以利用固定序列做整数倍频偏估计，即本发明的前导符号的接收方法还可以包括以下整数倍频偏估计步骤：
1)根据所确定该前导符号的位置，截取包含固定子载波的信号；
2)将该包含固定子载波的信号与频域固定子载波序列或该频域固定子载波序列对应的时域信号进行运算，以得到整数倍频偏估计。
接下来对基于初始定时同步结果的整数倍频偏估计方式进行说明，在进行整数倍频偏估计的步骤中，包括以下两种具体方式中任意一种或两种组合：.
第一整数倍频偏估计方式包含：根据初始定时同步的结果，截取至少包含全部或部分时域主体信号的一段时域信号，采用扫频方式对所截取出的该段时域信号以不同频偏进行调制后，得到若干N个与频偏值一一对应的扫频时域信号，将由已知频域序列进行傅里叶反变换所得的已知时域信号与每个扫频时域信号进行滑动互相关后，比较N个互相关结果的最大相关峰值，其最大的那个互相关结果所对应的扫频时域信号被调制的频偏值即为整数倍频偏估计值；和/或
第二整数倍频偏估计方式包含：
将根据初始定时同步的结果截取主体时域信号长度的时域信号进行傅里叶变换，将所得的频域子载波在扫频范围内按不同移位值进行循环移位，截取有效子载波所对应的接收序列，对该接收序列和已知频域序列进行预定运算再进行反变换，基于若干组移位值的一一对应的若干组反变换结果进行选择，得到最优的移位值，利用位值和整数倍频偏估计值之间的对应关系，获得整数倍频偏估计值。
下面举例具体描述整偏估计方式，比如时域主体信号A对应具有上述频域结构一，即频域OFDM符号分别包括虚拟子载波、信令序列(称为SC)子载波和固定序列(称为FC)子载波三部分，则下文所提到的已知频域序列即为固定子载波。
第一整数倍频偏估计方式，根据初始定时同步检测出的前导符号出现的位置，截取接收到的前导符号的时域波形的全部或者一部分，采用扫频的方式，即以固定的频率变化步径，比如对应整数倍频偏间隔，将该部分时域波形调制上不同的频偏后，得到若干个时域信号
Figure PCTCN2015076812-appb-000060
    (公式34)
其中，T为采样周期，fs为采样频率。而已知频域序列按预定子载波填充方 式后进行傅立叶反变换对应的时域信号为A2，将A2作为已知信号与每个A1y进行滑动相关，选取出现最大相关峰值的那个A1y，则对其所调制的频偏值y即为整数倍频偏估计值。
其中，扫频范围对应系统所需要对抗的频偏范围，比如需要对抗正负500K的频偏，而系统采样率为9.14M,前导符号主体为2K长度，则扫频范围为
Figure PCTCN2015076812-appb-000061
即[-114,114]。
第二整数倍频偏估计方式：根据初始定时同步检测出的前导符号出现的位置,截取主体时域信号A，并进行FFT，将FFT后的频域子载波进行扫频范围的不同移位值的循环移位，而后截取有效子载波所对应得接收序列，用接收序列和已知频域序列进行某种运算(通常为共轭相乘，或者相除)，将其结果进行IFFT，对IFFT的结果进行特定运算，比如取最大径能量，或者取若干大径能量累加。那么若干个移位值，经过若干次IFFT后，每次都得到一个运算结果，则会得到若干组的运算结果。基于这若干组结果判断出哪个移位值对应了整数倍频偏估计，由此得到整数倍频偏估计值。
通常的判断方法是基于若干组的结果，选择能量最大的那组对应的移位值，作为整数倍频偏估计值。
当时域主体信号A对应上述频域结构一时，还可以采用下述的整偏估计方法。
截取前导符号中对应的某个符号的时域主体信号A进行傅立叶变换后得到频域OFDM符号，将变换得到的频域OFDM符号进行上述扫频范围的循环移位，且按FC在子载波上的位置及前后2个固定序列子载波的间隔进行隔点差分相乘，且与已知固定序列子载波的隔点差分相乘值进行相关运算,得到一系列相关值，选取最大相关值对应的循环移位，即可相应得到可以得到整数倍频偏估计值。
整数倍频偏估计的具体算法有很多种,不再赘述。
进一步地，完成上述整数倍频偏估计后，对频偏进行补偿后进而对传输信令进行解析。
进一步可选择地，完成整数倍频偏估计后，利用前导符号中的已知信息进行精准定时同步方式。
比如在对应具有频域结构一时，用1个或多个频域符号所包含的固定子载波序列FC来进行精准定时同步方式；
以下对步骤S12-3中在上述判断结果为是的情况下，确定该前导符号在物理帧中的位置并解出该前导符号所携带的信令信息的步骤进行详细说明，该步骤包含以下：
确定该前导符号的位置包括：基于满足预设条件的检测的结果来确定该前导符号在物理帧中的位置；
若存在期望接收的前导符号，根据待检测相关值峰值大的那部分值或者最大值确定前导符号的位置。
在对传输信令进行解析的步骤中还包含信道估计方式，
比如具有频域结构一情况下，利用接收到的包含固定序列子载波的信号和已知频域固定序列子载波和/或其进行傅立叶反变换对应的时域信号完成信道估计，同样可以选择在时域进行和/或在频域进行，在此不再赘述。
进一步地，当解出前导符号中的帧格式参数和/或紧急广播内容后，可根据参数内容和已确定前导符号的位置来得到后续信令符号的位置或者数据符号的位置并基于此进行后续解析信令符号或数据符号。
继续对步骤S12-3中解出该前导符号所携带的信令信息的步骤进行说明，该解析信令信号的步骤包括：利用前导符号的全部或部分时域波形和/或该前导符号的全部或部分时域波形经过傅里叶变换后得到的频域信号，以解出该前导符号所携带的信令信息。
下面再针对频域结构一来解释信令解析过程。
通过包含信令序列子载波的信号与信令序列子载波集合或该信令序列子载波集合对应的时域信号进行运算，以解出该前导符号中由信令序列子载波所携带的信令信息。其中信令序列子载波集合基于已知的信令序列集合产生。
其中，包含信令序列子载波的信号包括：接收到的前导符号的全部或者部分时域波形，或者从前导符号中截取1个或多个主体OFDM符号经傅里叶变换后得到的1个或多个频域OFDM符号。信令序列子载波集合是由信令序列集合中各个信令序列填充至有效子载波上而形成的集合。
具体地，截取1个或多个对应ODFM符号主体的NA长度的时域信号进行傅立叶变换后得到的1个或多个频域OFDM符号；然后，去除零载波，根据信令子载波位置取出接收到的1个或多个频域信令子载波。将其与上述信道估计值以及已知的信令序列子载波集进行特定的数学运算，完成频域解码功能。
例如，设i＝0:M-1，M为信令子载波个数，j＝0:2P-1，P为频域所传信令比特数，即对应信令子载波集共有2P个元素，且每个元素对应长度为M的序列，Hi为每个信令子载波对应的信道估计值，SC_reci为接收到的频域信令子载波值，
Figure PCTCN2015076812-appb-000062
为信令序列子载波集中第j个元素中的第i个取值。则
Figure PCTCN2015076812-appb-000063
j＝0:2P-1，取max(corrj)所对应的j，即得到频域传输的信令信息。
在其他实施例中，上述过程也可以在时域上进行，利用已知信令序列子载波集经在适当位置补零后生成的相应长度的频域符号经傅里叶反变换后所对应的时域信令波形集直接与获取多径准确位置的时域接收信号进行同步相关，取相关值绝对值最大的那个，也可以解出频域传输的信令信息，这里不再赘述。
本实施中还提供了上述发明内容中所述的前导符号的生成装置、频域符号的生成装置和前导符号的接收装置，该前导符号的生成装置、频域符号的生成装置和前导符号的接收装置与上述实施例中前导符号的生成方法、频域符号的 生成方法和前导符号的接收方法所分别相对应，那么装置中所具有的结构和技术要素可由生成方法、接收方法相应转换形成，在此省略说明不再赘述。
本发明虽然已以较佳实施例公开如上，但其并不是用来限定本发明，任何本领域技术人员在不脱离本发明的精神和范围内，都可以利用上述揭示的方法和技术内容对本发明技术方案做出可能的变动和修改，因此，凡是未脱离本发明技术方案的内容，依据本发明的技术实质对以上实施例所作的任何简单修改、等同变化及修饰，均属于本发明技术方案的保护范围。

Claims (31)

  1. 一种前导符号的生成方法，其特征在于，包括如下步骤：
    根据从时域主体信号截取的部分时域主体信号生成前缀；
    根据该部分时域主体信号的全部或部分生成超前缀；
    基于所述循环前缀、所述时域主体信号和所述超前缀中至之少一生成时域符号，所述前导符号包含至少一个该时域符号。
  2. 如权利要求1所述的前导符号的生成方法，其特征在于：
    其中，基于依次排列的所述循环前缀、所述时域主体信号和所述超前缀生成所述时域符号，前导符号包含至少一个该时域符号。
  3. 如权利要求1所述的前导符号的生成方法，其特征在于：
    其中，所述前缀、所述超前缀的生成步骤中，包含：
    所述前缀从时域主体信号后部直接截取得到，
    所述超前缀是对与所述前缀对应的部分时域主体信号的全部或部分进行调制得到。
  4. 如权利要求1所述的前导符号的生成方法，其特征在于：
    其中，所述前缀、所述超前缀的生成步骤中，包含：
    对从所述时域主体信号后部截取出的部分按照第一预定处理规则进行处理形成为所述前缀，
    对从所述时域主体信号后部截取出的部分按照第二预定处理规则进行处理形成为所述超前缀，
    所述第一预定处理规则包括：直接拷贝；或
    乘以一个相同固定系数或预定不同系数，
    所述第二预定处理规则包括：当所述第一预定处理规则为直接拷贝时进行调制处理；或者
    当所述第一预定处理规则为乘以一个相同固定系数或预定不同系数时也乘以相应的系数后进行调制处理。
  5. 如权利要求1所述的前导符号的生成方法，其特征在于：
    其中，所述超前缀的长度不超过所述前缀的长度。
  6. 如权利要求1所述的前导符号的生成方法，其特征在于：
    其中，所述超前缀的生成步骤包括：
    设置频移序列；
    将所述时域主体信号的部分或者全部乘以该频移序列以得到该时域主体信号的所述超前缀。
  7. 如权利要求6所述的前导符号的生成方法，其特征在于：
    其中，所述频移序列的调制频偏值根据所述时域主体信号对应的频域子载波间隔或者根据超前缀的长度来确定，所述频移序列可任意选择初始相位值。
  8. 如权利要求1所述的前导符号的生成方法，其特征在于：
    其中，所述前导符号通过以下方式传输信令信息：
    确定一种所述前缀的长度和超前缀的长度的组合前提下，在生成所述超前缀时，通过以不同的起始位置截取所述部分时域符号来实现传输不同的信令信息。
  9. 如权利要求1所述的前导符号的生成方法，其特征在于：
    其中，所述时域主体信号的长度为2048个采样周期，所述前缀的长度为520个采样周期，所述超前缀的长度为504个采样周期，所述超前缀在所述时域主体信号中截取的起始位置为第1528个采样。
  10. 如权利要求1所述的前导符号的生成方法，其特征在于：
    其中，设P1_A(t)是时域符号的时域表达式，NA为所述时域主体信号的长度，设LenC为所述前缀的长度，LenB为所述超前缀的长度，fSH为对所述时域主体信号进行调制的调制频偏值，T为采样周期，则所述前导符号包含有所述前缀、所述时域主体信号和所述超前缀信号的时域符号的时域表达式为：
    Figure PCTCN2015076812-appb-100001
  11. 如权利要求10所述的前导符号的生成方法，其特征在于：
    其中，所述时域主体信号的长度NA为2048，所述循环前缀LenC的长度为520，所述超前缀LenB的长度为504，则所述前导符号包含有所述循环前缀、所述时域主体信号和所述超前缀的时域符号的时域表达式为：
    Figure PCTCN2015076812-appb-100002
  12. 如权利要求1所述的前导符号的生成方法，其特征在于：
    其中，所述时域主体信号利用至少一个比特信令来承载紧急广播标识，
    利用所述调制信号以所述调制信号长度在所述时域主体信号中截取的不同起始位置，以实现承载紧急广播标识。
  13. 如权利要求1所述的前导符号的生成方法，其特征在于：
    其中，所述时域主体信号基于对频域符号进行处理得到，
  14. 如权利要求13所述的前导符号的生成方法，其特征在于：
    其中，所述频域符号的生成步骤包含：将在频域上所分别生成固定序列和信令序列进行排列后填充至有效子载波上。
  15. 一种频域符号的生成方法，其特征在于，包括如下步骤：
    将在频域上所分别生成固定序列和信令序列进行排列后填充至有效子载波上，用于形成所述频域符号。
  16. 如权利要求15所述的频域符号的生成方法，其特征在于，还包括：
    确定所述固定序列和所述信令序列的平均功率比，依据平均功率比分别生成所述固定序列和所述信令序列。
  17. 如权利要求16所述的频域符号的生成方法，其特征在于，还包括：
    其中，所述固定序列和所述信令序列之间的所述平均功率比取值2。
  18. 如权利要求15所述的频域符号的生成方法，其特征在于：
    其中，所述固定序列和所述信令序列以预定交错排列规则进行排列，
    所述预定排列规则包含以下两种规则中的任意一种：
    呈奇偶交错或者偶奇交错进行排列；或
    把一部分信令序列放在奇数子载波，另一部分信令序列放在偶数子载波，且把一部分固定序列放在奇数子载波，另一部分固定序列放在偶数子载波。
  19. 如权利要求15所述的频域符号的生成方法，其特征在于：
    其中，所述信令序列的生成步骤包括：
    基于所预设的所述信令序列的长度和个数生成同一序列生成公式；
    基于同一序列生成公式选择不同的相位基值产生不同恒包络零自相关序列；以及
    根据所确定的信令序列的长度从得到的每一个恒包络零自相关序列中选取所述信令序列。
  20. 如权利要求15所述的频域符号的生成方法，其特征在于：
    其中，所述信令序列的生成步骤包括：
    基于所预设的信令序列的长度和个数确定序列生成若干个序列生成公式；
    针对每一个序列生成公式，选择不同的相位基值相应产生恒包络零自相关序列；以及
    根据所确定的信令序列的长度从得到的每一个恒包络零自相关序列中选取所述信令序列。
  21. 如权利要求19或20所述的频域符号的生成方法，其特征在于：
    其中，对所产生恒包络零自相关序列，还包括以下步骤：
    对所产生的恒包络零自相关序列进一步循环移位。
  22. 一种前导符号的接收方法，其特征在于，包括如下步骤：
    对接收信号进行处理；
    判断处理后的信号中是否存在期望接收的前导符号；
    在判断结果为是时，确定该前导符号的位置并解出该前导符号所携带的信令信息。
  23. 如权利要求22所述的前导符号的接收方法，其特征在于：
    其中，在判断处理后的信号中是否存在期望接收的所述前导符号，及在判断为是时确定该前导符号的位置并解出该前导符号所携带的信令信息的步骤中，包含以下任意至少一种步骤：
    初始定时同步方式、整数倍频偏估计方式、精准定时同步方式、信道估计方式、解码分析方式以及小数倍频偏估计方式。
  24. 如权利要求22所述的前导符号的接收方法，其特征在于：
    其中，利用以下任意至少一种的结果来判断所述处理后的信号中是否存在期望接收的所述前导符号：
    初始定时同步、整数倍频偏估计、精准定时同步、信道估计、解码分析及小数倍频偏估计方式。
  25. 如权利要求22所述的前导符号的接收方法，其特征在于：
    其中，所述判断所述基带信号中是否存在期望接收的前导符号的步骤中，包含：
    通过初始定时同步方式初步确定前导符号的位置；以及
    基于初始定时同步方式的结果，判断所述处理后的信号中是否存在期望接收的前导符号。
  26. 如权利要求25所述的前导符号的接收方法，其特征在于：
    其中，所述初始定时同步方式包含以下：
    第一种初始定时同步方式：
    利用所述循环前缀、所述时域主体信号和所述超前缀三者中任意两个间的 处理关系，对处理后的信号进行必要反处理进行延迟滑动自相关来获取累加相关值；以及
    基于累加相关值进行延迟关系匹配和/或特定的数学运算后,将所得处理值用于初始定时同步，初步确定前导符号的位置，和/或
    第二种初始定时同步方式：
    当所述前导符号中任意所述时域主体信号包含已知信号时，将时域主体信号依照预定N个差分值进行差分运算,并将已知信息对应的时域信号也进行差分运算,再将两者进行互相关得到N组与该N个差分值一一对应的差分相关的结果,基于该N组差分相关的结果进行初始同步，得到处理值，用于初步确定前导符号的位置，其中N≥1，
    其中，当基于第一初始定时同步方式和第二初始定时同步方式完成时，则将分别所得的所述处理值再进行加权运算，基于该加权运算值完成初始定时同步。
  27. 如权利要求22所述的前导符号的接收方法，其特征在于：
    其中，在确定该前导符号的位置并解出该前导符号所携带的信令信息的步骤
    中，包含：
    利用前导符号的全部或部分时域波形和/或该前导符号的全部或部分时域波形经过变换后得到的频域信号，以解出该前导符号所携带的信令信息。
  28. 如权利要求22所述的前导符号的接收方法，其特征在于，还包括：
    当所接收的用于生成所述前导符号的频域符号包含分别由固定序列和信令序列进行排列后填至有效子载波的步骤时，还包括：利用所述固定序列做整数倍频偏估计或信道估计，
    该利用所述固定序列做整数倍频偏估计或信道估计的步骤，包括：-
    根据所初步确定的该前导符号的位置，截取包含全部或部分固定子载波的信号；
    将该包含全部或部分固定子载波的信号与频域固定子载波序列或该频域固定子载波序列对应的时域信号进行运算，以得到整数倍频偏估计或信道估计。
  29. 一种前导符号的生成装置，其特征在于，包括：
    前缀生成单元，根据从时域主体信号截取的部分时域主体信号生成前缀；
    超前缀生成单元，根据该部分时域主体信号的全部或部分生成超前缀；
    前导符号生成单元，基于所述循环前缀、所述时域主体信号和所述超前缀中至之少一生成时域符号，所述前导符号包含至少一个该时域符号。
  30. 一种频域符号的生成装置，其特征在于，包括：
    序列生成单元，用于在频域上所分别生成固定序列和信令序列；
    频域符号生成单元，将所述固定序列和所述信令序列进行排列后填充至有效子载波上，用于形成所述频域符号。
  31. 一种前导符号的接收装置，其特征在于，包括：
    接收处理单元，对接收信号进行处理；
    判断单元，判断处理后的信号中是否存在期望接收的前导符号；
    定位解析单元，用于在判断结果为是时确定该前导符号的位置并解出该前导符号所携带的信令信息。
PCT/CN2015/076812 2014-04-16 2015-04-16 前导符号的生成、接收方法和频域符号的生成方法及装置 WO2015158293A1 (zh)

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