WO2013091351A1 - 一种跨导增强无源混频器 - Google Patents

一种跨导增强无源混频器 Download PDF

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Publication number
WO2013091351A1
WO2013091351A1 PCT/CN2012/076193 CN2012076193W WO2013091351A1 WO 2013091351 A1 WO2013091351 A1 WO 2013091351A1 CN 2012076193 W CN2012076193 W CN 2012076193W WO 2013091351 A1 WO2013091351 A1 WO 2013091351A1
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Prior art keywords
transconductance
stage
drain
output
gate
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PCT/CN2012/076193
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English (en)
French (fr)
Inventor
吴建辉
时霄
陈超
刘智林
赵强
温俊峰
王旭东
白春风
田茜
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东南大学
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Priority to US13/980,355 priority Critical patent/US8933745B2/en
Publication of WO2013091351A1 publication Critical patent/WO2013091351A1/zh

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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H11/00—Networks using active elements
    • H03H11/02—Multiple-port networks
    • H03H11/04—Frequency selective two-port networks
    • H03H11/0422—Frequency selective two-port networks using transconductance amplifiers, e.g. gmC filters
    • H03H11/0466—Filters combining transconductance amplifiers with other active elements, e.g. operational amplifiers, transistors, voltage conveyors
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/14—Balanced arrangements
    • H03D7/1425—Balanced arrangements with transistors
    • H03D7/1441—Balanced arrangements with transistors using field-effect transistors
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/14—Balanced arrangements
    • H03D7/1425—Balanced arrangements with transistors
    • H03D7/1466—Passive mixer arrangements

Definitions

  • the present invention relates to a transconductance enhanced mixerless mixer comprising a transconductance enhanced transconductance stage, a passive mixing switch pair and an output transimpedance amplifier.
  • the transconductance amplifier stage uses a pre-amplified transconductance enhancement structure that greatly enhances the transconductance so that the same transconductance value can be achieved at a lower bias current, and the RF current is modulated by the mixing stage to generate an output intermediate frequency current signal.
  • the voltage output is formed by the transimpedance amplifier, and finally the intermediate frequency voltage signal is obtained.
  • the transimpedance amplifier uses a transconductance enhancement structure, and the input impedance is further reduced, improving current utilization efficiency and port isolation.
  • the mixer structure features low power consumption, high conversion gain, and good port isolation.
  • the traditional Gilbert mixers have relatively balanced indicators, reliable operation, and good port isolation.
  • the performance of traditional Gilbert mixers is sometimes difficult to meet the current needs in many practical applications.
  • the mixing stage adopts an active mixing structure
  • the flicker noise will have an influence in the zero-IF receiver structure, and if the passive mixing structure can be adopted, since the passive mixer has no quiescent current, The flicker noise is also greatly reduced, and the linearity of the passive mixer is usually higher than that of the active mixer.
  • the signal received by the receiver is generally small, and the conventional transconductance is generally only the transconductance value of the input transistor at the low bias current.
  • the transconductance value is limited and the conversion gain is low. If the transconductance level can achieve a large transconductance value under the same bias current in one way, it is of great significance to improve the performance of the whole mixer.
  • the design structure is based on this and successfully designed.
  • the new transconductance circuit structure greatly enhances the transconductance value of the transconductance circuit.
  • An object of the present invention is to provide a passive mixer having an enhanced transconductance transconductance stage, which can achieve the same transconductance value at a lower bias current, and the radio frequency current is modulated by the mixing stage. Generate an output IF current signal. The voltage output is formed by the transimpedance amplifier, and finally the intermediate frequency voltage signal is obtained.
  • the transimpedance amplifier uses a transconductance enhancement structure, and the input impedance is further reduced, improving current utilization efficiency and port isolation.
  • the mixer structure features low power consumption, high conversion gain, and good port isolation.
  • the object of the present invention is achieved by the following method:
  • the mixer generally consists of a transconductance stage, a mixing stage, and a load output stage.
  • the traditional classic mixer has a single transconductance circuit structure with a small transconductance gain, which is generally the transconductance value of the input transistor g m ;) , as shown in Figure 1 (a).
  • the RF signal is converted into an RF current through the transconductance stage, and the RF current is then subjected to modulation of the mixing stage to generate a down-converted signal at the intermediate frequency at the output, and then converted into a voltage signal through the load output stage.
  • a transconductance stage circuit having a transconductance enhancement function is designed, and FIG. 1(b) shows the voltage-current conversion relationship after the transconductance enhancement of the circuit structure. The detailed working principle is explained below.
  • the transconductance enhanced passive mixer of the present invention as shown in FIG. 2, the RF current signal outputted at two points of the transconductance stages A and B is coupled to the mixing stage, and enters the transimpedance amplifier after the frequency conversion process of the mixing switch.
  • the load-level circuit uses a transconductance-enhanced common-gate input structure, making the resistance seen from the source side of PM6 low in the output band.
  • the two points of the transconductance stage output A and B are equivalent to the AC ground for the output low frequency signal, so that the IF signal is pendulum at this port.
  • the width is as low as possible to improve the IF-RF port isolation.
  • the C7 connected to the mixing output makes the transconductance output points A and B equally equivalent to the AC ground for the RF signal, which reduces the RF voltage swing at the node, improving current utilization efficiency and linearity. degree.
  • a resonant network composed of L0 and C4 is connected between the gates of Li 0 and Li 1 of the transconductance stage. The resonance of the parallel resonant network is near the input RF frequency, and the signals of the intermediate frequency and the low frequency are suppressed to ensure the positive effect. The stability of the feedback transconductance enhancement circuit.
  • the transconductance enhancement circuit In the input frequency band, due to the AC short circuit effect caused by the capacitor C7 and the low on-resistance of the switch pair, the transconductance enhancement circuit generates a low impedance node in the positive feedback loop, ensuring positive feedback at the frequency.
  • the loop gain is less than 1 for stability.
  • the implementation of the transconductance enhancement function is as follows: For the RF signal near 2.4 GHz, assume that the positive and negative inputs of the RF signal are respectively + ⁇ , - v RF .
  • the absolute value of the gain at (D, D) is ⁇ , then the current flowing through the resistor R2 is ⁇ (assuming that the transconductance values of all the manifolds are ⁇ ⁇ ), and the current direction is pulled out by the defect;
  • the negative end of the RF signal passes through PM1 and is amplified at point D.
  • C3 is coupled to the gate of ⁇ 0, the drain terminal of ⁇ 0 turns to current ⁇ , and the current direction is also pulled out from point A, then the RF current drawn from point A is 3 ⁇ 4 + ⁇ .
  • the resistors R4 and R5 are used to balance the phase delay of the current generated by the leakage current of PM0 and PM1 flowing through R2 and R3.
  • the two points A and B form a differential RF current output, which is coupled to the switching mixing stage via C5, C6, and the output stage.
  • the transconductance of the present invention increases its passive mixer, and C5, C6 are connected in series between the transconductance stage and the mixing stage to block the influence of the DC signal.
  • the mixing stage output is connected to C7, so that the mixed high frequency signal can be filtered out.
  • the output IF signal is amplified by the load stage.
  • Figure 3 shows the output waveform of the unused transconductance gain structure (smaller amplitude low frequency sine wave) and the output waveform of the transconductance gain structure (large amplitude low frequency sine wave), where the local oscillator frequency is 2. 45 GHz; It can be clearly seen from the figure that the conversion gain is greatly improved due to the improvement of the transconductance structure.
  • the transconductance enhances the transconductance stage of the passive mixer, which increases the RF current converted by the RF input voltage.
  • the main working principle is: The current is multiplied by the local oscillator signal through the passive double balance switch to realize the mixing function.
  • the switching stage uses passive mixing, there is no static power consumption and the flicker noise from the switching stage is eliminated.
  • the output stage uses transconductance enhancement technology to generate a lower input impedance in the output band, draw the mixed intermediate frequency current into the load stage, and finally form an intermediate frequency output voltage on the output load through the current mirror.
  • the mixing output terminates the capacitor so that the RF signal for the transconductance stage is approximately AC grounded, making it easier to draw the RF current generated by the transconductance stage into the mixer switch as much as possible.
  • the low input impedance across the resistance level makes the intermediate frequency voltage fluctuation at the input of the trans-blocking stage small, which reduces the voltage feedthrough of the intermediate frequency signal to the output of the transconductance stage, stabilizes the output voltage of the transconductance stage, and increases the current utilization efficiency. Increased linearity.
  • the transconductance enhanced passive mixer described above has the characteristics of transconductance transconductance high, low power consumption, and high conversion gain.
  • FIG. 1 is a schematic diagram of the principle of a transconductance enhancement portion of the present invention
  • FIG. 2 is a schematic diagram of a transconductance enhanced passive mixer circuit of the present invention
  • Figure 3 shows the waveform of the input RF signal (light curve), the output waveform without the transconductance gain structure (smaller amplitude low frequency sine wave), and the output waveform with the transconductance gain structure (large amplitude low frequency sine wave)
  • the transconductance enhances the transconductance stage of the passive mixer to increase the RF current converted by the RF input voltage.
  • the main working principle is: The current is multiplied by the local oscillator signal through the passive double balance switch to realize the mixing function.
  • the switching stage uses passive mixing, there is no static power consumption and the flicker noise from the switching stage is eliminated.
  • the output stage uses transconductance enhancement technology, which can generate lower input impedance in the output frequency band, draw the mixed intermediate frequency current into the load stage, and finally form the intermediate frequency output voltage on the output load through the current mirror.
  • the mixing output terminates the capacitor so that the RF signal for the transconductance stage is approximately AC grounded, facilitating as much of the RF current generated by the transconductance stage to be drawn into the mixer switch as much as possible.
  • the low input impedance across the resistance level makes the intermediate frequency voltage fluctuation at the input of the trans-blocking stage small, which reduces the voltage feedthrough of the intermediate frequency signal to the output of the transconductance stage, stabilizes the output voltage of the transconductance stage, and increases the current utilization efficiency. Increased linearity.
  • the transconductance enhanced passive mixer described above has the characteristics of transconductance transconductance high, low power consumption, and high conversion gain.
  • the main structure of the transconductance enhanced passive mixer of the present invention is mainly composed of a transconductance enhanced transconductance/amplifier stage, a passive mixing stage, a load output stage, a bias circuit and the like.
  • the transconductance/amplifier stage includes a P-type metal oxide field effect transistor (hereinafter referred to as a PMOS transistor) PM0, PM1, an N-type metal oxide field effect transistor (hereinafter referred to as an NMOS transistor) NM0, NMl, and a cross-coupling capacitor and an LC resonant circuit. .
  • the bias voltages of PM0 and PM1 are given by bias voltage 1 through R0, Rl, respectively.
  • L0, C2, C3, and C4 are used to enhance the transconductance.
  • the output RF current of the transconductance stage is derived from between R2 and R4 and between R3 and R5.
  • the mixing stage is composed of PM2-PM5 as the core circuit, and the mixed result is output from the drain of PM2 and PM5, wherein the PM2 and PM4 drains are shorted, and PM3 and PM5 are leaked. Extremely short.
  • the load output stage is mainly composed of PM6-PM15, NM2, and -NM3.
  • PM6, PM7, PM10, PM11, NM2, and NM3 form the first stage differential amplifier circuit of the load output stage, and NM2 and NM3 are biased by the bias voltage 2.
  • This stage is outputted by the leakage of PM6 and PM7, connected to the second stage differential source follower consisting of PM8, PM9, PM12 and PM13.
  • the signal is output from the drain of PM12 and PM13, and connected to PM14, PM15 and R6.
  • R7 constitutes the third-stage differential common-source amplifying circuit, and the intermediate frequency signal is finally outputted by the leakage of PM14 and PM15.
  • the upper plates of the capacitors C0 and C1 are respectively connected to the positive input signal terminal and the negative input signal terminal.
  • the lower plate of CO is connected to the gate of PMO; the lower plate of C1 is connected to the gate of PM1; the upper plate of capacitor C2 is connected to the drain end of PM0, the upper plate of capacitor C3 is connected to the drain end of PM1, and the lower plate of C2 is connected C4's lower plate, C3's lower plate is connected to C4's upper plate, C4's upper plate is connected to L0 positive terminal, C4 lower plate is connected to L0 negative terminal, L0 positive terminal and C4 upper plate are connected to NM1 gate at the same time.
  • L0 negative terminal and C4 lower plate are connected to NM0 gate at the same time; PM0, PM1 source is connected to power supply voltage, resistor R0 is positively connected to PM0 gate, negative terminal is connected to positive terminal of resistor R1, and negative terminal of R1 is connected to PM1 Gate.
  • the positive terminal of the resistor R2 is connected to the leakage of PM0, the negative terminal is connected to the positive terminal of R4, the negative terminal of R4 is connected to the drain of NM0; the positive terminal of the resistor R3 is connected to the leakage of PM1, and the negative terminal is connected to the positive terminal of R5, R5 Negatively terminate the drain of NM1.
  • the sources of NM0 and NM1 are grounded.
  • the upper pole of the RF coupling capacitor C5 is connected to the negative terminal of R2 and the positive terminal of R4.
  • the upper pole of the RF coupling capacitor C6 is connected to the negative terminal of R3 and the positive terminal of R5, and the lower plate of C5 is connected to the source of PM2 and PM3, C6.
  • the lower plate is connected to the source of PM4 and PM5.
  • the positive terminal of the true signal is connected to the gates of PM3 and PM4, and the negative terminal of the local oscillator signal is connected to the gates of PM2 and PM5.
  • the drains of PM2 and PM4 are connected to the upper plate of C7 at the same time, and the drains of PM3 and PM5 are connected to the lower plate of C7 at the same time.
  • the positive output of the switching stage that is, the upper plate of C7 is connected to the source of PM6 of the load stage, and is connected to the drain of PM10.
  • the negative output of the switching stage that is, the lower plate of C7 is connected to the source of PM7 of the load stage, and the drain of PM11 is connected. pole.
  • PM6 and PM7 are biased with a bias voltage of 4; the drain of PM6 Connect the drain of NM2 and connect the gate of PM8 at the same time; the drain of PM7 is connected to the drain of NM3 and the gate of PM9; the drain of PM8 and PM9 is grounded to form the source follower; the source of NM2 and NM3 The pole is grounded, and the gates of NM2 and NM3 are biased by a bias voltage of 2.
  • the source of PM10-PM15 is connected to the power supply voltage; the source of PM8 is connected to the drain of PM12 and is connected to the gate of PM 14; the source of PM9 is connected to the drain of PM13 and the gate of PM15; the drain of PM14 As the positive terminal of the output voltage, the positive terminal of the resistor R7 is connected, and the negative terminal of the resistor R7 is grounded; the drain of the PM15 serves as the negative terminal of the output voltage, and is connected to the positive terminal of the resistor R8, and the negative terminal of the resistor R8 is grounded.

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Abstract

一种跨导增强无源混频器，该混频器包含跨导放大级、混频级以及输出跨阻放大器；跨导放大级采用预放大跨导增强结构，使得跨导大大增强，从而在更低的偏置电流下可实现相同的跨导值，射频电流经过混频级的调制作用生成输出中频电流信号。经跨阻放大器形成电压输出，最终得到中频电压信号。该跨阻放大器使用了跨导增强结构，输入阻抗被进一步降低，提高了电流利用效率和端口隔离度。该混频器结构具有功耗低，转换增益高、端口隔离度好等特点。

Description

一种跨导增强无源混频器
技术领域
本发明涉及一种跨导增强的无混频器， 该混频器包含增强跨导的跨导级、 无源混频 开关对及输出跨阻放大器。 跨导放大级采用预放大跨导增强结构， 使得跨导大大增强， 从而在更低的偏置电流下可实现相同的跨导值， 射频电流经过混频级的调制作用生成输 出中频电流信号。 经跨阻放大器形成电压输出， 最终得到中频电压信号。 该跨阻放大器 使用了跨导增强结构， 输入阻抗被进一步降低， 提高了电流利用效率和端口隔离度。 该 混频器结构具有功耗低， 转换增益高、 端口隔离度好等特点。
背景技术
传统吉尔伯特混频器各项指标比较均衡， 而且工作可靠， 端口隔离度好。 但是随着 射频接收机的单片化的要求的不断提高和射频技术的不断进步， 很多实际应用情况下， 传统吉尔伯特混频器的性能有时难以满足当下需求。 比如混频级采用有源混频结构时， 在零中频接收机结构中， 其闪烁噪声会带来一定影响， 而若能够采用无源混频结构， 由 于无源混频器没有静态电流， 其闪烁噪声也大大减小了， 且无源混频器的线性度通常会 高于有源混频器。
经典混频结构中， 对于跨导级中将射频电压转换为射频电流时， 由于接收机接受的 信号一般很小， 加之传统的跨导一般只为输入晶体管的跨导值， 在低偏置电流下其跨导 值有限， 转换增益偏低。 若能够在跨导级通过一种途径在相同偏置电流下实现较大的跨 导值， 则对整个混频器性能的提高， 有着重要的意义， 本设计结构正是基于此出发， 成 功设计新的跨导电路结构， 使得跨导级电路的跨导值得到大大增强。
对于无源混频器的输出级， 因端口隔离度、 线性度、转换增益等一系列问题的考虑， 都需要尽可能降低输出级跨阻放大器的输入电阻。 在本发明中采用了跨导增强结构的跨 阻放大器， 实现了这一目的。
发明内容
技术问题： 本发明的目的在于提供一种具有增强跨导跨导级的无源混频器， 在更低的偏 置电流下可实现相同的跨导值，射频电流经过混频级的调制作用生成输出中频电流信号。 经跨阻放大器形成电压输出， 最终得到中频电压信号。 该跨阻放大器使用了跨导增强结 构， 输入阻抗被进一步降低， 提高了电流利用效率和端口隔离度。 该混频器结构具有功 耗低， 转换增益高、 端口隔离度好等特点。 技术方案： 本发明目的通过以下方法实现： 混频器一般由跨导级、 混频级、 负载输 出级组成。 传统经典混频器的跨导级电路结构单一， 跨导增益较小， 一般即为输入晶体 管的跨导值 gm;)， 如图 1 ( a)所示。 射频信号通过跨导级转换成射频电流， 射频电流再 通过混频级的调制作用在输出端产生位于中频的下变频信号， 然后通过负载输出级转换 为电压信号。 上述过程中， 如果在跨导级的输出端能够得到较大的电流信号， 则对混频 器的转换增益和噪声性能的提升是很有帮助的。 在本发明的无源混频器中， 设计了一种 具有跨导增强功能的跨导级电路， 图 1 (b ) 即显示了本电路结构跨导增强后的电压电流 转换关系。 详细工作原理解释如下。
本发明的跨导增强无源混频器， 如图 2所示， 跨导级 A、 B两点输出的射频电流信号 耦合至混频级， 经混频开关的变频过程后进入跨阻放大器。 负载级电路由于釆用了跨导 增强的共栅输入结构， 使得在输出频段从 PM6的源端看进去的电阻很低。 当信号频率相 对较低时， 而 PM6与 PM7的极低等效输入电阻使得跨导级输出 A、 B两点对输出低频信号 而言等效为交流接地，使得中频信号在这个端口处的摆幅尽量低以提高中频-射频端口隔 离度。对于输入射频信号， 接在混频输出端的 C7使得对于射频信号， 跨导级输出点 A、 B 同样等效为交流地， 降低了该节点处的射频电压摆幅， 提高了电流利用效率和线性度。 另外跨导级的丽 0与丽 1的栅极之间接由 L0、 C4组成的谐振网络， 该并联谐振网络谐振 在输入射频频率附近， 对中频及低频等的信号起抑制增益作用， 保证了正反馈跨导增强 电路的稳定性。而在输入频段上， 由于电容 C7的带来的交流短路效应以及开关对的低导 通电阻， 使得跨导增强电路在正反馈环路中产生一个低阻抗节点， 保证了在该频率上正 反馈环路增益低于 1 , 保证了稳定性。 跨导增强功能的实现过程如下： 对于 2. 4GHz附近 的射频信号， 假设射频信号正负端输入分别为 + ^、 - vRF。 PM0、 PMl放大管在(；、 D点的 增益绝对值为 Λ， 则流过电阻 R2的电流为^ （假设所有 ΡΜ管的跨导值均为^ ^ )， 电流方向由 Α点拉出； 射频信号负端经过 PM1， 在 D点放大为
Figure imgf000004_0001
, 经 C3耦合至丽 0 的栅极， 在丽 0的漏端转为电流 ^^ , 电流方向也从 A点拉出， 则从 A点拉出的射 频电流和为 ¾ +Λ^ 。 与此同时， 在 Β点有大小相同的射频电流， 方向为向 Β点 灌入。 其中， 电阻 R4、 R5用来平衡 PM0、 PM1的漏端电流流经 R2、 R3所产生的电流相位 延迟。 至此， A、 B两点形成差分射频电流输出， 经过 C5， C6耦合至开关混频级， 输出级 采用跨导增强技术， 可以在输出频段产生较低的输入阻抗， 将混频得到的中频电流全部 吸入负载级， 最终通过电流镜在输出负载上形成中频输出电压。
本发明的跨导增其无源混频器， 在跨导级与混频级之间串联 C5， C6， 阻断直流信号 的影响。 混频级输出接 C7， 使混频后高频信号得以滤掉。 对输出中频信号通过负载级进 行了放大。 图 3所示为未釆用跨导增益结构的输出端波形 （较小幅度低频正弦波） 和采 用跨导增益结构的输出端波形 （较大幅度低频正弦波）， 其中本振频率为 2. 45GHz ; 从图 中可以清楚看出由于跨导结构的改进， 转换增益得到了大大的提高。
有益效果： 该跨导增强无源混频器的跨导级， 能够提高由射频输入电压转化的射频 电流。其主要工作原理是： 电流经过无源双平衡开关对与本振信号相乘， 实现混频功能。 开关级采用无源混频的方式， 不存在静态功耗且消除了来自开关级的闪烁噪声。 输出级 采用跨导增强技术， 可以在输出频段产生较低的输入阻抗， 将混频得到的中频电流全部 吸入负载级， 最终通过电流镜在输出负载上形成中频输出电压。 混频输出端接电容以使 对于跨导级的射频信号近似交流接地， 便于将跨导级产生的射频电流尽可能多地吸入混 频开关。 而跨阻级的低输入阻抗， 使得跨阻级输入端中频电压波动很小， 这样降低了中 频信号往跨导级输出端的电压馈通， 稳定了跨导级输出电压， 增加了电流利用效率， 提 高了线性度。 以上所述的跨导增强无源混频器具有跨导级跨导高、 功耗低、 转换增益高 的特点。
附图说明
图 1为本发明的跨导增强部分的原理示意图；
图 2为本发明的跨导增强无源混频器电路原理图；
图 3为输入射频信号波形 (浅色曲线）、 未采用跨导增益结构的输出端波形 （较小幅 度低频正弦波）、 采用跨导增益结构的输出端波形 （较大幅度低频正弦波）
具体实施方式
该跨导增强无源混频器的跨导级， 能够提高由射频输入电压转化的射频电流。 其主 要工作原理是： 电流经过无源双平衡开关对与本振信号相乘， 实现混频功能。 开关级采 用无源混频的方式， 不存在静态功耗且消除了来自开关级的闪烁噪声。 输出级采用跨导 增强技术， 可以在输出频段产生较低的输入阻抗， 将混频得到的中频电流全部吸入负载 级， 最终通过电流镜在输出负载上形成中频输出电压。 混频输出端接电容以使对于跨导 级的射频信号近似交流接地， 便于将跨导级产生的射频电流尽可能多地吸入混频开关。 而跨阻级的低输入阻抗， 使得跨阻级输入端中频电压波动很小， 这样降低了中频信号往 跨导级输出端的电压馈通， 稳定了跨导级输出电压， 增加了电流利用效率， 提高了线性 度。 以上所述的跨导增强无源混频器具有跨导级跨导高、 功耗低、 转换增益高的特点。
本发明的跨导增强型无源混频器主体结构主要由跨导增强的跨导 /放大级、 无源混频 级、 负载输出级、 偏置电路等模块构成。 跨导 /放大级包括 P型金属氧化物场效应管 （以 下简称 PMOS管） PM0、 PM1 , N型金属氧化物场效应管 （以下简称 NMOS管） NM0、 NMl及交叉耦合电容和 LC谐振回路构成。 PM0与 PM1的偏置电压由偏置电压 1通过 R0， Rl分别给出。 L0， C2， C3， C4用于增强跨导， 跨导级的输出射频电流从 R2与 R4 之间、 R3与 R5之间分别引出。经过 C5、 C6分别耦合至混频开关级，混频级由 PM2-PM5 作为核心电路， 混频后的结果从 PM2， PM5的漏极输出， 其中 PM2、 PM4漏极短接， PM3、 PM5漏极短接。 负载输出级主要由 PM6-PM15, NM2,-NM3构成。 PM6、 PM7、 PM10、 PM11、 NM2、 NM3构成负载输出级的第一级差分放大电路， NM2和 NM3 由 偏置电压 2提供偏置。该级由 PM6、 PM7的漏断输出，接至由 PM8、 PM9、 PM12、 PM13 组成的第二级差分源极跟随器，信号由 PM12、 PM13的漏极输出，接至由 PM14、 PM15、 R6、 R7组成第三级的差分共源放大电路， 中频信号最终由 PM14、 PM15的漏断输出。
电容 C0、 C1的上极板分别接正输入信号端和负输入信号端。 CO的下极板接 PMO的 栅极； C1的下极板接 PM1的栅极； 电容 C2的上极板接 PM0漏端， 电容 C3的上极板接 PM1漏端， C2的下极板接 C4的下极板， C3的下极板接 C4的上极板， C4的上极板接 L0正端， C4下极板接 L0负端， L0正端与 C4上极板同时接 NM1栅极， L0负端与 C4 下极板同时接 NM0栅极； PM0、 PM1的源极接电源电压， 电阻 R0正端接 PM0的栅， 负端接电阻 R1的正端， R1的负端接 PM1的栅极。 电阻 R2的正端接 PM0的漏断， 负端 接 R4的正端， R4的负端接 NM0的漏极； 电阻 R3的正端接 PM1的漏断， 负端接 R5的 正端， R5的负端接 NM1的漏极。 NM0和 NM1的源极接地。射频耦合电容 C5上极板接 R2的负端与 R4的正端， 射频耦合电容 C6上极板接 R3的负端与 R5的正端， C5的下极 板接 PM2、 PM3的源极， C6的下极板接 PM4、 PM5的源极。 本真信号的正端接 PM3 与 PM4的栅极， 本振信号的负端接 PM2与 PM5的栅极。 PM2与 PM4的漏极同时接 C7 的上极板， PM3与 PM5的漏极同时接 C7的下极板。 开关级的正输出即 C7的上极板接 负载级 PM6的源极，同时接 PM10的漏极，开关级的负输出即 C7的下极板接负载级 PM7 的源极， 同时接 PM11的漏极。 PM6与 PM7的栅极接偏置电压 4进行偏置； PM6的漏极 接 NM2的漏极， 同时接接 PM8的栅极； PM7的漏极接 NM3的漏极， 同时接 PM9的栅 极； PM8与 PM9的漏极接地，构成源极跟随器； NM2与 NM3的源极接地， NM2与 NM3 的栅极接偏置电压 2进行偏置。 PM10-PM15的源极均接电源电压； PM8的源极接 PM12 的漏极，同时接 PM 14的栅极； PM9的源极接 PM13的漏极，同时接 PM15的栅极； PM14 的漏极作为输出电压的正端， 接电阻 R7的正端， 电阻 R7的负端接地； PM15的漏极作 为输出电压的负端， 接电阻 R8的正端， 电阻 R8的负端接地。
以上所述仅为本发明的较佳实施方式， 本发明的保护范围并不以上述实施方式为限， 但凡本领域普通技术人员根据本发明所揭示内容所作的等效修饰或变化， 皆应纳入权利 要求书中记载的保护范围内。

Claims

权利要求书
1、 一种跨导增强无源混频器， 其特征在于： 该混频器包含具有跨导增强功能的跨导 级， 无源混频开关对， 以及输出跨阻放大器。 具有跨导增强功能的跨导级将输入射频电 压转化为射频电流， 射频电流经过双平衡混频开关对实现混频， 混频后的电流通过跨导 增强的负载输出级， 转换为中频电压输出。
2、 根据权利要求 1 所述的跨导增强型无源混频器， 其特征在于： 其主体结构主要由 跨导 /放大级、 混频级、 偏置电路、 负载输出级等模块构成。 跨导 /放大级包括 P型金属氧 化物场效应管 （以下简称 PMOS 管） PM0、 PM1， N型金属氧化物场效应管 （以下简称 NMOS管） NM0、 NM1及交叉耦合电容和 LC谐振回路构成。 PM0与 PM1的偏置电压 由偏置电压 1通过 R0， R1分别给出。 L0， C2， C3， C4为增强跨导所采用的电路元件， 跨导级的输出射频电流从 R2与 R4之间、 R3与 R5之间分别引出。 经过 C5、 C6分别耦 合至混频开关级， 混频级由 PM2-PM5作为核心电路， 混频后的结果从 PM2, PM5 的漏 极输出， 其中 PM2、 PM4漏极短接， PM3、 PM5 漏极短接。 负载输出级主要由 PM6- PM15, NM2,-NM3构成。 PM6、 PM7、 PM10、 PM11、 NM2、 NM3构成负载输出级的第 一级差分放大电路， NM2和 NM3由偏置电压 2提供偏置。 该级由 PM6、 PM7的漏断输 出， 接至由 PM8、 PM9、 PM12、 PM13 组成的第二级差分源极跟随器， 信号由 PM12、 PM13 的漏极输出， 接至由 PM14、 PM15、 R6、 R7组成第三级的差分共源放大电路， 中 频信号最终由 PM14、 PM15的漏断输出。
电容 C0、 C1的上极板分别接正输入信号端和负输入信号端。 CO的下极板接 PM0的 栅极； C1的下极板接 PM1的栅极； 电容 C2的上极板接 PM0漏端， 电容 C3的上极板接 PM1漏端， C2的下极板接 C4的下极板， C3的下极板接 C4的上极板， C4的上极板接 L0正端， C4下极板接 L0负端， L0正端与 C4上极板同时接 NM1栅极， L0负端与 C4 下极板同时接 NM0栅极； PM0、 PM1的源极接电源电压， 电阻 R0正端接 PM0的栅， 负端接电阻 R1的正端， R1的负端接 PM1的栅极。 电阻 R2的正端接 PM0的漏断， 负端 接 R4的正端， R4的负端接 NM0的漏极； 电阻 R3的正端接 PM1的漏断， 负端接 R5的 正端， R5的负端接 NM1的漏极。 NM0和 NM1的源极接地。 射频耦合电容 C5上极板接 R2的负端与 R4的正端， 射频耦合电容 C6上极板接 R3的负端与 R5的正端， C5的下极 板接 PM2、 PM3的源极， C6的下极板接 PM4、 PM5的源极。 本真信号的正端接 PM3与 PM4的栅极， 本振信号的负端接 PM2与 PM5的栅极。 PM2与 PM4的漏极同时接 C7的 上极板， PM3与 PM5的漏极同时接 C7的下极板。 开关级的正输出即 C7的上极板接负载 级 PM6的源极， 同时接 PM10的漏极， 开关级的负输出即 C7的下极板接负载级 PM7的 源极， 同时接 PM11的漏极。 PM6与 PM7的栅极接偏置电压 4进行偏置； PM6的漏极接 NM2 的漏极， 同时接接 PM8 的栅极； PM7 的漏极接 NM3 的漏极， 同时接 PM9 的栅 极； PM8 与 PM9 的漏极接地， 构成源极跟随器； NM2与 NM3 的源极接地， NM2与 NM3 的栅极接偏置电压 2进行偏置。 PM10-PM15 的源极均接电源电压； PM8 的源极接 PM12的漏极， 同时接 PM 14的栅极； PM9的源极接 PM13的漏极， 同时接 PM15的栅 极； PM14的漏极作为输出电压的正端， 接电阻 R7的正端， 电阻 R7的负端接地； PM15 的漏极作为输出电压的负端， 接电阻 R8的正端， 电阻 R8的负端接地。
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