WO2011054233A1 - 信道估计方法和装置 - Google Patents

信道估计方法和装置 Download PDF

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Publication number
WO2011054233A1
WO2011054233A1 PCT/CN2010/076102 CN2010076102W WO2011054233A1 WO 2011054233 A1 WO2011054233 A1 WO 2011054233A1 CN 2010076102 W CN2010076102 W CN 2010076102W WO 2011054233 A1 WO2011054233 A1 WO 2011054233A1
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Prior art keywords
channel estimation
raised cosine
channel
estimation value
deconvolution
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PCT/CN2010/076102
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English (en)
French (fr)
Inventor
吴更石
花梦
张春玲
焦淑蓉
彭秀琴
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华为终端有限公司
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Publication of WO2011054233A1 publication Critical patent/WO2011054233A1/zh
Priority to US13/463,452 priority Critical patent/US8711914B2/en

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Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
    • H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
    • H04L27/38—Demodulator circuits; Receiver circuits
    • H04L27/3845—Demodulator circuits; Receiver circuits using non - coherent demodulation, i.e. not using a phase synchronous carrier
    • H04L27/3854—Demodulator circuits; Receiver circuits using non - coherent demodulation, i.e. not using a phase synchronous carrier using a non - coherent carrier, including systems with baseband correction for phase or frequency offset
    • H04L27/3863—Compensation for quadrature error in the received signal
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/69—Spread spectrum techniques
    • H04B1/707—Spread spectrum techniques using direct sequence modulation
    • H04B1/7097—Interference-related aspects
    • H04B1/711—Interference-related aspects the interference being multi-path interference
    • H04B1/7113—Determination of path profile
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H04L25/0212—Channel estimation of impulse response
    • H04L25/0216—Channel estimation of impulse response with estimation of channel length
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H04L25/0224—Channel estimation using sounding signals
    • H04L25/0228—Channel estimation using sounding signals with direct estimation from sounding signals
    • H04L25/023—Channel estimation using sounding signals with direct estimation from sounding signals with extension to other symbols
    • H04L25/0232—Channel estimation using sounding signals with direct estimation from sounding signals with extension to other symbols by interpolation between sounding signals
    • H04L25/0234—Channel estimation using sounding signals with direct estimation from sounding signals with extension to other symbols by interpolation between sounding signals by non-linear interpolation

Definitions

  • the equalization coefficient is directly obtained on the sampled data with sampling deviation, which has a great influence on the demodulation performance, especially in the high-order modulation mode, such as 64-bit quadrature amplitude modulation (Quarature Amplitude Modulation; the following tube is called: 64QAM) ).
  • 64QAM Quadrature amplitude modulation
  • the following describes the processing of sampling deviation by taking Wideband Code Division Multiple Access (WCDMA) system as an example.
  • WCDMA Wideband Code Division Multiple Access
  • the WCDMA system's wireless transmission channel environment is a frequency selective multipath channel. Multipath channels with different delays will cause interference.
  • the WCDMA receiver is usually a multi-point receiver (Rake) receiver or equalizer.
  • High Speed Downlink Packet Access (HSDPA) is an enhanced evolution technology for WCDMA that provides the high-speed data rate required to implement multimedia services in third-generation mobile communication systems. Improve the spectrum and code resource utilization of the system, effectively improving the performance and capacity of the wireless network.
  • the HSDPA signal is usually received by an equalizer, and it is necessary to perform channel estimation for all the samples in a certain period of time when obtaining the equalization coefficient.
  • the WCDMA downlink receiver first periodically samples the received signal at an integer multiple of the chip transmission frequency (typically 2 or 4 times).
  • the WCDMA system performs multipath search to complete different The timing estimation of the path component realizes the chip phase alignment of the local chip and the received signal, and then uses the Common Pilot Channel (CPICH) for channel estimation.
  • CPICH Common Pilot Channel
  • the timing error estimator is used to control the interpolation filter to interpolate the signals obtained by the independent samples to obtain The approximate value of the signal at the time of the best decision sampling. Since each sampling point is interpolated and the sampling rate is generally 2 or 4 times the chip rate, the amount of operation is large.
  • Embodiments of the present invention provide a channel estimation method and apparatus, so as to eliminate a sample deviation of a channel estimation value and reduce an operation amount.
  • An embodiment of the present invention provides a channel estimation method, including:
  • the channel estimation value at each radial position in the pilot channel is obtained according to the channel estimation value after the raised cosine deconvolution and the sampling deviation obtained by the multipath search.
  • the embodiment of the invention further provides a channel estimation apparatus, including:
  • a search module configured to perform multipath search on the pilot channel;
  • An interpolation module configured to perform interpolation on a channel estimation value at a radial position of the multipath search;
  • a raised cosine deconvolution module configured to perform a raised cosine deconvolution on the channel estimation value after the interpolation module is interpolated;
  • a channel estimation value obtaining module configured to perform a raised cosine deconvolution channel estimation value and a sampling deviation obtained by the multipath search according to the raised cosine deconvolution module, to obtain a path position in the pilot channel Channel estimate.
  • the channel estimation value at the path position of the multipath search is interpolated, and the interpolated channel estimation value is subjected to raised cosine deconvolution; then the channel estimation value and multipath search according to the raised cosine deconvolution are performed.
  • the channel estimation method and apparatus provided by the embodiments of the present invention eliminate the sampling deviation of the channel estimation value, improve the demodulation performance, and significantly reduce the calculation amount. .
  • FIG. 1 is a flow chart of an embodiment of a channel estimation method according to the present invention.
  • FIG. 2 is a flow chart of another embodiment of a channel estimation method according to the present invention.
  • FIG. 3 is a schematic diagram of an embodiment of generating an H array according to the present invention.
  • FIG. 4 is a schematic structural diagram of an embodiment of a channel estimation apparatus according to the present invention.
  • FIG. 5 is a schematic structural diagram of another embodiment of a channel estimation apparatus according to the present invention.
  • FIG. 6 is a schematic structural diagram of still another embodiment of a channel estimation apparatus according to the present invention. detailed description
  • FIG. 1 is a flow chart of an embodiment of a channel estimation method according to the present invention. As shown in FIG. 1, the embodiment includes:
  • Step 101 Perform multipath search on the pilot channel, and estimate the channel on the path position of the multipath search. The values are interpolated.
  • any interpolation method may be adopted for interpolating the channel estimation value at the radial position of the multipath search, for example, quadratic interpolation or Sine interpolation.
  • a Sine interpolation of a finite length truncation of a channel estimation value at a radial position of a multipath search is described.
  • Step 102 Perform a raised cosine deconvolution on the interpolated channel estimation value.
  • Step 103 Obtain a channel estimation value at each path position in the pilot channel according to the channel estimation value after the raised cosine deconvolution and the sampling deviation obtained by the multipath search.
  • the above embodiment interpolates the channel estimation values at the path positions of the multipath search, and performs the raised cosine deconvolution on the interpolated channel estimation values; and then obtains the channel estimation values and multipath search according to the raised cosine deconvolution.
  • the sampling deviation is obtained, and the channel estimation value at each path position in the pilot channel is obtained.
  • the channel estimation method provided in this embodiment eliminates the sampling deviation of the channel estimation value, improves the demodulation performance, and significantly reduces the calculation amount.
  • FIG. 2 is a flowchart of another embodiment of a channel estimation method according to the present invention. As shown in FIG. 2, the embodiment includes:
  • Step 201 Perform multipath search on the pilot channel, and interpolate the channel estimation value at the path position of the multipath search.
  • the window is opened by the path position of the multipath search, and the length of the window is multipath delay extension.
  • channel estimation is performed on the sample points in the window. Assuming that the sample rate is P, L x P channel estimation values can be obtained, and the channel estimation value has the influence of the sampling deviation obtained by the multipath search. Then, for the above L ⁇ Among the channel estimation values, the channel estimation values at the path positions of the multipath search are interpolated.
  • any interpolation method such as quadratic interpolation or Sine interpolation may be employed. This embodiment will be described by taking a Sine interpolation of a finite length truncation of channel estimation values at the radial position of the multipath search.
  • the channel estimates are interpolated in the time domain. According to Nyquist's law, as long as the sampling frequency is higher than twice the signal frequency, the ideal Sine interpolation can recover the original signal without distortion. Since the ideal Sine interpolation is not possible, in the specific implementation, the finite-length truncated Sine interpolation can be used to eliminate the sample deviation of the channel estimation value.
  • the Sine interpolation of the finite length truncation of the channel estimation value at the path position of the multipath search may be: adding a path position of the multipath search and a total of LXP channel estimation values of the adjacent path of the path position, The Sine coefficient after windowing is truncated.
  • the Sine coefficient after the window is truncated can be obtained by multiplying the Sine function by a window function; for example:
  • the ideal Sine coefficient can be:
  • a Kaiser window function is used to implement a low-pass filter, and a finite length truncation is performed on the ideal Sine coefficient.
  • Tap [i] is the Sine coefficient after windowing.
  • Step 202 Perform a Raised Cosine (hereinafter referred to as RC) deconvolution operation on the interpolated channel estimation value.
  • RC Raised Cosine
  • the interpolated The channel estimate is subjected to an RC deconvolution operation.
  • the interference of the adjacent path to the path in the interpolated channel estimation value is mainly caused by the impact response of the RC.
  • the multipath search is performed to obtain six effective paths, and the radial positions are Pi, P 2 , P 3 , P 4 , P 5 , and P 6 respectively ; and the channel estimation value at the radial position of the multipath search is finitely long.
  • the channel estimates obtained after truncated Sine interpolation are h;, h' 2 , h' 3 , h' 4 , h; and h' 6 ; set h;, h' 2 , h' 3 , h' 4 , h; and h' 6 are obtained after the RC deconvolution operation, the channel estimates are h 2 , h 3 , h 4 , h 5 and h 6 ; RC is the root Raised Cosine (hereinafter referred to as: RRC) convolution receives the filter impulse response of RRC.
  • RRC Raised Cosine
  • Step 203 Obtain a channel estimation value at each path position in the pilot channel according to the channel estimation value after the RC deconvolution and the sampling deviation obtained by the multipath search.
  • the channel estimation value after RC deconvolution may be performed first.
  • the sampling rate of RC convolution is the same as the sampling rate of RC deconvolution; then the channel estimation value after RC convolution is convolved with the Sine coefficient after windowing truncation to obtain the pilot channel.
  • Channel estimate at each path location Specifically, the channel estimation value after RC deconvolution may be placed according to the path position of the multipath search, and the remaining positions are set to zero, and an H array of LxP length is generated, which is denoted as H(t), and FIG. 3 is an H array generated by the present invention.
  • H(t) an H array generated by the present invention.
  • the length of Hout(i) is LxP+N-1, where N is the coefficient length of the RC filter; RC (i-t) is the coefficient of the RC filter, which can be obtained by equation (4).
  • Hout ( i ) is convoluted with the Sine coefficient after the window is truncated to obtain the channel estimation value at each path position in the pilot channel.
  • the purpose of convolving Hout (i) with the Sine coefficient after windowing is to add back the sampling deviation obtained in the multipath search process, and improve the channel estimation at each path position in the obtained pilot channel. The accuracy of the value.
  • the RC deconvolved channel estimation value may be first subjected to a higher sampling rate RC convolution, that is, the sampling rate of the RC convolution is higher than the sampling rate of the RC deconvolution. Then in the channel estimation value after the RC convolution, the channel estimation value at each path position in the pilot channel is selected according to the sampling deviation obtained by the multipath search.
  • the above embodiment performs Sine interpolation on the channel estimation value at the path position of the multipath search, and performs RC deconvolution on the channel estimation value after the Sine interpolation; and then obtains the channel estimation value and the multipath search according to the RC deconvolution.
  • the sampling bias is obtained, and the channel estimation value at each path position in the pilot channel is obtained.
  • the channel estimation method provided in this embodiment eliminates the sample deviation of the channel estimation value, improves the demodulation performance, and significantly reduces the calculation amount.
  • the foregoing program may be stored in a computer readable storage medium, and the program is executed when executed.
  • the foregoing steps include the steps of the foregoing method embodiments; and the foregoing storage medium includes: a medium that can store program codes, such as a ROM, a RAM, a magnetic disk, or an optical disk.
  • the channel estimation apparatus of this embodiment can be used as a receiver, or a part of a receiver, to implement the process of the embodiment shown in FIG. 1 of the present invention.
  • the channel estimation apparatus may include: a search module 41, an interpolation module 42, a raised cosine deconvolution module 43, and a channel estimation value obtaining module 44.
  • the search module 41 is configured to perform multipath search on the pilot channel.
  • the interpolation module 42 is configured to perform interpolation on the channel estimation value at the radial position of the multipath search.
  • any interpolation method may be adopted for interpolating the channel estimation value at the path position of the multipath search, for example, : Quadratic interpolation or Sine interpolation.
  • a Sine interpolation of a finite length truncation of a channel estimation value at a radial position of a multipath search is described.
  • the raised cosine deconvolution module 43 is configured to perform a raised cosine deconvolution on the channel estimation value interpolated by the interpolation module 42.
  • the channel estimation value obtaining module 44 is configured to obtain a channel estimation value obtained by raising the cosine deconvolution and a sampling deviation obtained by multipath searching according to the raised cosine deconvolution module 43 to obtain a channel estimation value at each radial position in the pilot channel. .
  • the interpolation module 42 interpolates the channel estimation value at the radial position of the multipath search, and the raised cosine deconvolution module 43 performs a raised cosine deconvolution on the interpolated channel estimation value; then the channel estimation value obtaining module
  • the channel estimation value at each path position in the pilot channel is obtained according to the channel estimation value after the raised cosine deconvolution and the sample deviation obtained by the multipath search.
  • the channel estimation apparatus provided in this embodiment eliminates the channel estimation value. The sample deviation improves the demodulation performance and significantly reduces the amount of calculation.
  • FIG. 5 is a schematic structural diagram of another embodiment of the channel estimation apparatus according to the present invention.
  • the channel estimation apparatus of this embodiment can be used as a receiver, or a part of a receiver, to implement the flow of the embodiment shown in FIG. 2 of the present invention.
  • the channel estimation apparatus may include: a search module 51, an interpolation module 52, a raised cosine deconvolution module 53 and a channel estimation value obtaining module 54.
  • the search module 51 is configured to perform multipath search on the pilot channel.
  • the interpolation module 52 is configured to perform interpolation on the channel estimation value at the radial position of the multipath search.
  • any interpolation method may be adopted for interpolating the channel estimation value at the path position of the multipath search, for example,
  • the interpolation module 52 may include: a Sine interpolation sub-module 521 for performing finite-length truncated Sine interpolation on the channel estimation value at the path position of the multipath search; specifically, Sine
  • the interpolation sub-module 521 may be a first Sine interpolation sub-module, configured to roll the channel estimation value of the path position of the multipath search and the channel estimation value of the adjacent path of the path position, and the Sine coefficient after the windowing is truncated.
  • Product where the Sine coefficient after the window is truncated can be obtained by multiplying the Sine function by a window function.
  • a raised cosine deconvolution module 53 for channel estimation values after interpolation of the Sine interpolation sub-module 521
  • the line raised cosine deconvolution; specifically, the raised cosine deconvolution module 53 can perform raised cosine deconvolution according to equation (1).
  • the channel estimation value obtaining module 54 is configured to obtain a channel estimation value obtained by raising the cosine deconvolution and a sampling deviation obtained by multipath searching according to the raised cosine deconvolution module 53, and obtaining channel estimation values at respective path positions in the pilot channel. .
  • the channel estimation value obtaining module 54 may include: a first raised cosine convolution sub-module 541 and a obtained sub-module 542.
  • the first raised cosine convolution sub-module 541 is configured to perform a raised cosine convolution on the channel estimated value after the raised cosine deconvolution, a sampling rate of the raised cosine convolution and a sampling rate of the raised cosine deconvolution
  • the first raised cosine convolution sub-module 541 can perform a raised cosine convolution according to the equation (2).
  • the purpose of convolving the channel estimation value after the raised cosine convolution with the Sine coefficient after windowing is to add back the sampling deviation obtained in the multipath search process, and to improve the position of each path in the obtained pilot channel. The accuracy of the channel estimate.
  • FIG. 6 is a schematic structural diagram of still another embodiment of the channel estimation apparatus according to the present invention.
  • the channel estimation apparatus of this embodiment can be used as a receiver or a part of a receiver to implement the flow of the embodiment shown in FIG. 2 of the present invention.
  • the channel estimation device of the present embodiment is different from the channel estimation device of the embodiment shown in FIG. 5 in this embodiment.
  • the channel estimation value obtaining module 54 may include: a second raised cosine convolution sub-module 543 and Sub-module 544 is selected.
  • the second raised cosine convolution sub-module 543 is configured to perform a raised cosine convolution on the channel estimated value after the raised cosine deconvolution, and the sampling rate of the raised cosine convolution is higher than that of the raised cosine deconvolution.
  • Rate; selection sub-module 544 used for the sampling bias obtained according to the multipath search in the channel estimation value after the raised cosine convolution The difference is the channel estimate at each path location in the pilot channel.
  • the interpolation module 52 interpolates the channel estimation values at the radial position of the multipath search, and the raised cosine deconvolution module 53 performs a raised cosine deconvolution on the interpolated channel estimation values; then the channel estimation value obtaining module
  • the channel estimation value at each path position in the pilot channel is obtained according to the channel estimation value after the raised cosine deconvolution and the sample deviation obtained by the multipath search.
  • the channel estimation apparatus provided in this embodiment eliminates the channel estimation value. The sample deviation improves the demodulation performance and significantly reduces the amount of calculation.
  • modules in the apparatus in the embodiments may be distributed in the apparatus of the embodiment according to the description of the embodiments, or the corresponding changes may be located in one or more apparatuses different from the embodiment.
  • the modules of the above embodiments may be combined into one module, or may be further split into a plurality of sub-modules.

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Description

信道估计方法和装置 本申请要求于 2009 年 11 月 03 日提交中国专利局、 申请号为 200910221113.3的中国专利申请的优先权，其全部内容通过引用结合在本申请 中。 技术领域 本发明实施例涉及通信技术领域， 尤其涉及一种信道估计方法和装置。 背景技术 在很多情况下，接收机对接收信号的最佳采样无法通过直接采样获得， 即 采样取得的信号与发送码片之间有小于采样间隔的偏差。在有采样偏差的采样 数据上直接进行均衡系数的求取，对解调性能有很大影响，尤其是高阶调制方 式下， 如 64位正交幅度调制 （ Quadrature Amplitude Modulation; 以下筒称： 64QAM )。 下面以宽带码分多址（ Wideband Code Division Multiple Access; 以 下简称： WCDMA ) 系统为例对采样偏差的处理进行说明。
WCDMA 系统的无线传输信道环境为频率选择性多径信道， 具有不同时 延的多径信道将引起千扰， WCDMA接收机通常为多点接收（ Rake )接收机 或均衡器。 高速下行分组接入（ High Speed Downlink Packet Access; 以下筒称： HSDPA )作为 WCDMA的一种增强型演进技术， 提供了在第三代移动通信系 统中实现多媒体服务所需的高速数据速率，并且大大提高系统的频谱和码资源 利用率， 有效地提升了无线网络性能和容量。 HSDPA信号通常采用均衡器接 收， 在求取均衡系数时需要对某一段时间内的所有釆样点进行信道估计。 WCDMA 下行接收机首先会对接收信号进行周期性的采样， 该采样频率 为码片发送频率的整数倍 (一般为 2倍或 4倍）。 因为接收机对信号的传播延 迟一般是未知的，为了对接收机的输出信号进行同步釆样，必须从接收信号中 获得符号定时， 现有技术中， WCDMA 系统进行多径搜索完成对各个不同多 径分量的定时估计， 实现本地码片和接收信号的码片相位对齐，然后采用公共 导频信道（ Common Pilot Channel; CPICH )进行信道估计。
在实现本发明过程中，发明人发现现有技术中至少存在如下问题： 现有技 术在进行信道估计时利用定时误差估计量控制内插滤波器对独立釆样得到的 信号进行插值运算， 以得到信号在最佳判决取样时刻的近似值。 由于对每个采 样点都要进行插值处理， 并且采样速率一般为码片速率的 2倍或 4倍， 因此运 算量很大。 发明内容 本发明实施例提供一种信道估计方法和装置，以实现消除信道估计值的釆 样偏差， 降低运算量。
本发明实施例提供一种信道估计方法， 包括：
对导频信道进行多径搜索，并对所述多径搜索的径位置上的信道估计值进 行插值；
对插值后的信道估计值进行升余弦反卷积；
根据升余弦反卷积后的信道估计值和所述多径搜索获得的采样偏差，获得 所述导频信道中各径位置上的信道估计值。
本发明实施例还提供一种信道估计装置 , 包括：
搜索模块 , 用于对导频信道进行多径搜索； 插值模块， 用于对所述多径搜索的径位置上的信道估计值进行插值； 升余弦反卷积模块，用于对所述插值模块插值后的信道估计值进行升余弦 反卷积；
信道估计值获得模块，用于根据所述升余弦反卷积模块进行升余弦反卷积 后的信道估计值和所述多径搜索获得的采样偏差，获得所述导频信道中各径位 置上的信道估计值。
本发明实施例对多径搜索的径位置上的信道估计值进行插值，并对插值后 的信道估计值进行升余弦反卷积；然后根据升余弦反卷积后的信道估计值和多 径搜索获得的采样偏差，获得导频信道中各径位置上的信道估计值； 本发明实 施例提供的信道估计方法和装置消除了信道估计值的采样偏差，提高了解调性 能， 并且明显降低了运算量。
附图说明
图 1为本发明信道估计方法一个实施例的流程图；
图 2为本发明信道估计方法另一个实施例的流程图；
图 3为本发明生成 H数组一个实施例的示意图；
图 4为本发明信道估计装置一个实施例的结构示意图；
图 5为本发明信道估计装置另一个实施例的结构示意图；
图 6为本发明信道估计装置再一个实施例的结构示意图。 具体实施方式
图 1为本发明信道估计方法一个实施例的流程图，如图 1所示，该实施例包 括：
步驟 101 , 对导频信道进行多径搜索， 并对多径搜索的径位置上的信道估 计值进行插值。
本实施例中 ,在对多径搜索的径位置上的信道估计值进行插值可以采用任 意一种插值方法， 例如： 二次插值或 Sine插值等。 本发明实施例以对多径搜索 的径位置上的信道估计值进行有限长截短的 Sine插值为例进行说明。
步骤 102, 对插值后的信道估计值进行升余弦反卷积。
本实施例中，在对多径搜索的径位置上的信道估计值进行插值之后， 为消 除插值后的信道估计值中相邻径对本径的干扰，需要对插值后的信道估计值进 行升余弦反卷积操作。
步骤 103 ,根据升余弦反卷积后的信道估计值和多径搜索获得的采样偏差， 获得导频信道中各径位置上的信道估计值。
上述实施例对多径搜索的径位置上的信道估计值进行插值，并对插值后的 信道估计值进行升余弦反卷积；然后根据升余弦反卷积后的信道估计值和多径 搜索获得的采样偏差， 获得导频信道中各径位置上的信道估计值； 本实施例提 供的信道估计方法消除了信道估计值的采样偏差，提高了解调性能， 并且明显 降低了运算量。
图 2为本发明信道估计方法另一个实施例的流程图，如图 2所示，该实施例 包括：
步骤 201 , 对导频信道进行多径搜索， 并对多径搜索的径位置上的信道估 计值进行插值。
本实施例中， 以多径搜索的径位置左右开窗， 窗口的长度为多径时延扩展
L, 对窗内的釆样点进行信道估计， 假设釆样率为 P, 则可以得到 L x P个信道 估计值， 该信道估计值带有多径搜索获得的采样偏差的影响。 然后， 对上述 L χ Ρ个信道估计值中， 多径搜索的径位置上的信道估计值进行插值， 在进行插 值时， 可以采用任意一种插值方法， 例如： 二次插值或 Sine插值等。 本实施例 以对多径搜索的径位置上的信道估计值进行有限长截短的 Sine插值为例进行 说明。
在时域上对信道估计值进行插值， 由奈奎斯特定律可知， 只要采样频率高 于信号频率的两倍以上， 理想的 Sine插值可以无失真的恢复出原始信号。 由于 理想的 Sine插值是无法实现的，因此在具体实现时，可以采用有限长截短的 Sine 插值消除信道估计值的釆样偏差。
具体地， 对多径搜索的径位置上的信道估计值进行有限长截短的 Sine插 值可以为： 将多径搜索的径位置和该径位置的相邻径的共计 L X P个信道估计 值， 与加窗截短之后的 Sine系数进行卷积。
其中， 加窗截短之后的 Sine系数可以通过将 Sine函数与一窗函数相乘获 得； 例如： 理想的 Sine系数可以为：
1， u (k) = 0
SincTaps「u(k)
nx u (k) ) 式（ 1 ) 中， u ( k )为采样点位置， SincTaps[u (k)]该采样点的理想的 Sine系 数。
本实施例采用 Kaiser窗函数来实现低通滤波器， 对理想 Sine系数进行有 限长截短。 假设 Kaiser窗为 X-1阶， 抽头系数为 WinKaiser[i] , i=0,l, ... ,X, 则 加窗截短之后的 Sine系数可以为： Tap[i] = SincTaps [i] x WinKaiser[i] ( 2 ) 式（2 ) 中， Tap [i]为加窗截短之后的 Sine系数。 步驟 202, 对插值后的信道估计值进行升余弦（Raised Cosine; 以下简称： RC)反卷积操作。 本实施例中 , 在对多径搜索的径位置上的信道估计值进行有限长截短的 Sine插值之后， 为消除插值后的信道估计值中相邻径对本径的干扰， 需要对插 值后的信道估计值进行 RC反卷积操作。 其中， 插值后的信道估计值中相邻径 对本径的干扰主要是由 RC的冲击响应造成的。 具体地， 设多径搜索获得 6条有效径， 径位置分别为 Pi、 P2、 P3、 P4、 P5 和 P6; 设对多径搜索的径位置上的信道估计值进行有限长截短的 Sine插值之后 获得的信道估计值为 h;、 h'2、 h'3、 h'4、 h;和 h'6; 设对 h;、 h'2、 h'3、 h'4、 h;和 h'6进行 RC反卷积操作后获得的信道估计值为 、 h2、 h3、 h4、 h5和 h6; RC是发 射根升余弦（ Root Raised Cosine; 以下简称： RRC )卷积接收 RRC的滤波器冲 击响应， 在忽略噪声的影响时， 可以通过式（3)所示的方程来求取 、 h2、 h3、 h4、 h5和 h6。
'K' 一 1 RC(T1>2) RC(T1>3) RC(T ) RC(T1>5) RC(T1>6)" — h〖
RC(i2a) 1 RC(i2>3) RC(i2>4) RC(i2>5) RC(i2>6) h2
RC( ) RCCc3'2) 1 RC( ) RC( 5) RC( 6) h3
A
RC(i4a) RC(i4>2) RC(i4>3) 1 RC(i4>5) RC(i4>6) h4 h; RC(i54) RC(i52) RC( 3) RC(i54) 1 RC(i56) h5
K RC( ) RC(T6>2) RC(T6 ) RC(T6>4) RC(i6i5) 1 (3) 式（3) 中， 为径间隔， 」= — Pj, 本实施例中， i, je {1, 2, 3, 4， 5, 6}, 且1≠」； C ( lij)可以通过式（4)获得：
Figure imgf000009_0001
式（4) 中， 可以取 = 0.22, T = 0.26042 s。
步骤 203, 根据 RC反卷积后的信道估计值和多径搜索获得的采样偏差， 获 得导频信道中各径位置上的信道估计值。
在本实施例的一种实现方式中， 可以先对 RC反卷积后的信道估计值进行
RC卷积， RC卷积的釆样率与 RC反卷积的釆样率相同； 然后将 RC卷积后的信 道估计值与加窗截短之后的 Sine系数进行卷积， 获得导频信道中各径位置上的 信道估计值。 具体地， 可以将 RC反卷积后的信道估计值按多径搜索的径位置 放置， 其余位置置零， 生成 LxP长度的 H数组， 记为 H (t) , 图 3为本发明生 成 H数组一个实施例的示意图。
再对该 H数组进行 RC卷积， 具体卷积公式如式（5)所示：
Hout(i) = JH(t)RC(i - 1) , i=0, 1 ,· · -,L χ Ρ+Ν-2 (5)
t=o
式（5)中， Hout(i)的长度为 LxP+N-1, 其中 N为 RC滤波器的系数长度； RC (i-t)为 RC滤波器的系数， 可以通过式（4)获得。
获得 Hout ( i )之后 , 再将 Hout ( i )与加窗截短之后的 Sine系数进行卷积 , 即可获得导频信道中各径位置上的信道估计值。 其中， 将 Hout (i)与加窗截 短之后的 Sine系数进行卷积的目的是为了将多径搜索过程中获得的采样偏差 加回去， 提高获得的导频信道中各径位置上的信道估计值的准确度。
在本实施例的另一种实现方式中， 可以先对 RC反卷积后的信道估计值进 行更高采样率的 RC卷积， 即 RC卷积的采样率高于 RC反卷积的采样率； 然后在 RC卷积后的信道估计值中 , 根据多径搜索获得的采样偏差选择导频信道中各 径位置上的信道估计值。
上述实施例对多径搜索的径位置上的信道估计值进行 Sine插值， 并对 Sine 插值后的信道估计值进行 RC反卷积； 然后根据 RC反卷积后的信道估计值和多 径搜索获得的采样偏差，获得导频信道中各径位置上的信道估计值； 本实施例 提供的信道估计方法消除了信道估计值的釆样偏差，提高了解调性能， 并且明 显降低了运算量。
本领域普通技术人员可以理解：实现上述方法实施例的全部或部分步驟可 以通过程序指令相关的硬件来完成，前述的程序可以存储于一计算机可读取存 储介质中， 该程序在执行时， 执行包括上述方法实施例的步驟； 而前述的存储 介质包括： ROM、 RAM, 磁碟或者光盘等各种可以存储程序代码的介质。
图 4为本发明信道估计装置一个实施例的结构示意图， 本实施例的信道估 计装置可以作为接收机， 或接收机的一部分， 实现本发明图 1所示实施例的流 程。 如图 4所示， 该信道估计装置可以包括： 搜索模块 41、 插值模块 42、 升余 弦反卷积模块 43和信道估计值获得模块 44。
其中， 搜索模块 41 , 用于对导频信道进行多径搜索。
插值模块 42, 用于对多径搜索的径位置上的信道估计值进行插值； 本实施 例中，在对多径搜索的径位置上的信道估计值进行插值可以采用任意一种插值 方法， 例如： 二次插值或 Sine插值等。 本发明实施例以对多径搜索的径位置上 的信道估计值进行有限长截短的 Sine插值为例进行说明。
升余弦反卷积模块 43，用于对插值模块 42插值后的信道估计值进行升余弦 反卷积。 信道估计值获得模块 44,用于根据升余弦反卷积模块 43进行升余弦反卷积 后的信道估计值和多径搜索获得的采样偏差，获得导频信道中各径位置上的信 道估计值。
上述实施例中， 插值模块 42对多径搜索的径位置上的信道估计值进行插 值，升余弦反卷积模块 43对插值后的信道估计值进行升余弦反卷积； 然后信道 估计值获得模块 44根据升余弦反卷积后的信道估计值和多径搜索获得的釆样 偏差， 获得导频信道中各径位置上的信道估计值； 本实施例提供的信道估计装 置消除了信道估计值的釆样偏差， 提高了解调性能， 并且明显降低了运算量。
图 5为本发明信道估计装置另一个实施例的结构示意图， 本实施例的信道 估计装置可以作为接收机， 或接收机的一部分， 实现本发明图 2所示实施例的 流程。 如图 5所示， 该信道估计装置可以包括： 搜索模块 51、 插值模块 52、 升 余弦反卷积模块 53和信道估计值获得模块 54。
其中， 搜索模块 51 , 用于对导频信道进行多径搜索。
插值模块 52, 用于对多径搜索的径位置上的信道估计值进行插值； 本实施 例中 ,在对多径搜索的径位置上的信道估计值进行插值可以采用任意一种插值 方法， 例如： 二次插值或 Sine插值等； 其中， 插值模块 52可以包括： Sine插值 子模块 521 , 用于对多径搜索的径位置上的信道估计值进行有限长截短的 Sine 插值； 具体地， Sine插值子模块 521可以为第一 Sine插值子模块， 用于将多径搜 索的径位置的信道估计值和该径位置的相邻径的信道估计值，与加窗截短之后 的 Sine系数进行卷积； 其中， 加窗截短之后的 Sine系数可以通过将 Sine函数与 一窗函数相乘获得。
升余弦反卷积模块 53 , 用于对 Sine插值子模块 521插值后的信道估计值进 行升余弦反卷积； 具体地， 升余弦反卷积模块 53可以根据式（ 1 )进行升余弦 反卷积。
信道估计值获得模块 54,用于根据升余弦反卷积模块 53进行升余弦反卷积 后的信道估计值和多径搜索获得的采样偏差，获得导频信道中各径位置上的信 道估计值。
其中， 信道估计值获得模块 54可以包括： 第一升余弦卷积子模块 541和获 得子模块 542。 具体地， 第一升余弦卷积子模块 541 , 用于对升余弦反卷积后的 信道估计值进行升余弦卷积，升余弦卷积的釆样率与升余弦反卷积的釆样率相 同； 具体地， 第一升余弦卷积子模块 541可以根据式（2 )进行升余弦卷积。 获 得子模块 542, 用于将升余弦卷积后的信道估计值与加窗截短之后的 Sine系数 进行卷积， 获得导频信道中各径位置上的信道估计值； 其中， 获得子模块 542 将升余弦卷积后的信道估计值与加窗截短之后的 Sine系数进行卷积的目的是 为了将多径搜索过程中获得的采样偏差加回去，提高获得的导频信道中各径位 置上的信道估计值的准确度。
图 6为本发明信道估计装置再一个实施例的结构示意图， 本实施例的信道 估计装置可以作为接收机， 或接收机的一部分， 实现本发明图 2所示实施例的 流程。 本实施例的信道估计装置与本发明图 5所示实施例的信道估计装置的不 同之处在于， 本实施例中， 信道估计值获得模块 54可以包括： 第二升余弦卷积 子模块 543和选择子模块 544。
具体地， 第二升余弦卷积子模块 543 , 用于对升余弦反卷积后的信道估计 值进行升余弦卷积，升余弦卷积的釆样率高于升余弦反卷积的釆样率； 选择子 模块 544, 用于在升余弦卷积后的信道估计值中， 根据多径搜索获得的采样偏 差选择导频信道中各径位置上的信道估计值。
其余模块的功能与本发明图 5所示实施例的信道估计装置中各模块的功能 相同， 在此不再赘述。
上述实施例中， 插值模块 52对多径搜索的径位置上的信道估计值进行插 值，升余弦反卷积模块 53对插值后的信道估计值进行升余弦反卷积； 然后信道 估计值获得模块 54根据升余弦反卷积后的信道估计值和多径搜索获得的釆样 偏差， 获得导频信道中各径位置上的信道估计值； 本实施例提供的信道估计装 置消除了信道估计值的釆样偏差， 提高了解调性能， 并且明显降低了运算量。
本领域技术人员可以理解附图只是一个优选实施例的示意图，附图中的模 块或流程并不一定是实施本发明所必须的。
本领域技术人员可以理解实施例中的装置中的模块可以按照实施例描述 进行分布于实施例的装置中，也可以进行相应变化位于不同于本实施例的一个 或多个装置中。上述实施例的模块可以合并为一个模块， 也可以进一步拆分成 多个子模块。
最后应说明的是： 以上实施例仅用以说明本发明的技术方案， 而非对其限 制； 尽管参照前述实施例对本发明进行了详细的说明， 本领域的普通技术人员 应当理解： 其依然可以对前述各实施例所记载的技术方案进行修改， 或者对其 中部分技术特征进行等同替换； 而这些修改或者替换， 并不使相应技术方案的 本质脱离本发明各实施例技术方案的精神和范围。

Claims

权 利 要 求 书
1、 一种信道估计方法， 其特征在于， 包括：
对导频信道进行多径搜索，并对所述多径搜索的径位置上的信道估计值进 行插值；
对插值后的信道估计值进行升余弦反卷积；
根据升余弦反卷积后的信道估计值和所述多径搜索获得的釆样偏差，获得 所述导频信道中各径位置上的信道估计值。
2、根据权利要求 1所述的方法， 其特征在于， 所述对所述多径搜索的径位 置上的信道估计值进行插值包括：
对所述多径搜索的径位置上的信道估计值进行有限长截短的 Sine插值。
3、根据权利要求 2所述的方法， 其特征在于， 所述对所述多径搜索的径位 置上的信道估计值进行有限长截短的 Sine插值包括：
将所述多径搜索的径位置上的信道估计值和所述径位置相邻径的信道估 计值， 与加窗截短之后的 Sine系数进行卷积。
4、根据权利要求 1所述的方法， 其特征在于， 所述根据升余弦反卷积后的 信道估计值和所述多径搜索获得的采样偏差，获得所述导频信道中各径位置上 的信道估计值包括：
对升余弦反卷积后的信道估计值进行升余弦卷积 ,所述升余弦卷积的采样 率与所述升余弦反卷积的釆样率相同；
将升余弦卷积后的信道估计值与加窗截短之后的 Sine系数进行卷积 , 获得 所述导频信道中各径位置上的信道估计值。
5、根据权利要求 1所述的方法， 其特征在于， 所述根据升余弦反卷积后的 信道估计值和所述多径搜索获得的采样偏差，获得所述导频信道中各径位置上 的信道估计值包括：
对升余弦反卷积后的信道估计值进行升余弦卷积，所述升余弦卷积的釆样 率高于所述升余弦反卷积的采样率；
在所述升余弦卷积后的信道估计值中，根据所述多径搜索获得的采样偏差 选择所述导频信道中各径位置上的信道估计值。
6、 一种信道估计装置， 其特征在于， 包括：
搜索模块， 用于对导频信道进行多径搜索；
插值模块， 用于对所述多径搜索的径位置上的信道估计值进行插值； 升余弦反卷积模块，用于对所述插值模块插值后的信道估计值进行升余弦 反卷积；
信道估计值获得模块，用于根据所述升余弦反卷积模块进行升余弦反卷积 后的信道估计值和所述多径搜索获得的采样偏差，获得所述导频信道中各径位 置上的信道估计值。
7、 根据权利要求 6所述的装置， 其特征在于， 所述插值模块包括：
Sine插值子模块， 用于对所述多径搜索的径位置上的信道估计值进行有限 长截短的 Sine插值。
8、 根据权利要求 7所述的装置， 其特征在于， 所述 Sine插值子模块包括第 一 Sine插值子模块， 用于将多径搜索的径位置的信道估计值和该径位置的相邻 径的信道估计值， 与加窗截短之后的 Sine系数进行卷积。
9、根据权利要求 6所述的装置， 其特征在于， 所述信道估计值获得模块包 括： 第一升余弦卷积子模块，用于对升余弦反卷积后的信道估计值进行升余弦 卷积 , 所述升余弦卷积的采样率与所述升余弦反卷积的采样率相同； 获得子模块， 用于将升余弦卷积后的信道估计值与加窗截短之后的 Sine系 数进行卷积， 获得所述导频信道中各径位置上的信道估计值。
10、 根据权利要求 6所述的装置， 其特征在于， 所述信道估计值获得模块 包括：
第二升余弦卷积子模块，用于对升余弦反卷积后的信道估计值进行升余弦 卷积， 所述升余弦卷积的釆样率高于所述升余弦反卷积的釆样率；
选择子模块， 用于在所述升余弦卷积后的信道估计值中，根据所述多径搜 索获得的采样偏差选择所述导频信道中各径位置上的信道估计值。
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