WO2008023683A1 - Dispositif de séparation de signaux et procédé de séparation de signaux - Google Patents

Dispositif de séparation de signaux et procédé de séparation de signaux Download PDF

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WO2008023683A1
WO2008023683A1 PCT/JP2007/066155 JP2007066155W WO2008023683A1 WO 2008023683 A1 WO2008023683 A1 WO 2008023683A1 JP 2007066155 W JP2007066155 W JP 2007066155W WO 2008023683 A1 WO2008023683 A1 WO 2008023683A1
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matrix
signal
stream
calculating
equation
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PCT/JP2007/066155
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English (en)
Japanese (ja)
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Kazunari Hashimoto
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Panasonic Corporation
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H04L25/0204—Channel estimation of multiple channels
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/02—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas
    • H04B7/04—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L2025/0335—Arrangements for removing intersymbol interference characterised by the type of transmission
    • H04L2025/03426—Arrangements for removing intersymbol interference characterised by the type of transmission transmission using multiple-input and multiple-output channels
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L2025/03592—Adaptation methods
    • H04L2025/03598—Algorithms
    • H04L2025/03605—Block algorithms

Definitions

  • the present invention relates to a signal separation device and a signal separation method, and more particularly to a signal separation device and a signal separation method for performing reception processing in a communication system in which multistreams and single streams are mixed.
  • MIMO Multi_I beating
  • transmission signals multi-streams
  • signals received by multiple reception antennas are separated to extract information.
  • FIG. 1 shows an example of the main configuration of a receiver that separates MIMO signals using QR-MLD.
  • Receiving apparatus 10 shown in FIG. 1 includes a receiving antenna 11 1, 11 2, the receiving unit 12 1, 12 2, the channel estimator 13, QR decomposition unit 14, and orthogonal (Q H multiplication) unit 15, A maximum likelihood determination (MLD) unit 16, an LLR (Log Likelihood Ratio) calculation unit 17, and a decoding unit 18 are provided.
  • MLD maximum likelihood determination
  • LLR Log Likelihood Ratio
  • a receiving unit may be provided depending on the situation.
  • H the channel estimation value
  • n the receiver noise component
  • the transmission signal ⁇ , reception signal y, channel estimation value H, and receiver noise component n are expressed by equations (1) to (4).
  • the QR decomposition unit 14 performs QR decomposition on the channel estimation value H as shown in Equation (6).
  • QR decomposition as shown in Eq. (6), channel estimation value H is decomposed so that matrix Q after QR decomposition is a unitary matrix ⁇ lj and row ⁇ IJR is an upper triangular matrix.
  • the unitary matrix is a complex matrix ( ⁇ ni. — 1 ) where the inverse matrix is obtained by complex conjugate transpose.
  • Equation (9) is obtained.
  • the maximum likelihood determination unit 16 performs demodulation by maximum likelihood determination by the QR-MLD method, using the reception signal z orthogonalized as shown in Equation (10).
  • Channel estimation value H is QR-decomposed by QR decomposition unit 14 as shown in equation (14) c
  • the received signal z after being orthogonalized by the orthogonalizing unit 15 is expressed as in Expression (15).
  • the maximum likelihood determination unit 16 performs maximum likelihood determination by the QR-MLD method from the received signal z that has been orthogonalized using Equation (15).
  • QR In the maximum likelihood determination by the MLD method, the received signal y is multiplied by the complex conjugate transpose matrix Q H and the orthogonal received signal z is multiplied by the transmit replica and the upper triangular matrix R is multiplied by the symbol replica candidate.
  • the distance between signal points is compared with a point, and a transmission signal is determined from a combination of transmission symbols having the smallest distance between the signal points.
  • the distance between signal points is usually the square Euclidean distance, but the Manhattan distance may be used to reduce the amount of computation.
  • Equation (16) is used to calculate the square Euclidean distance, and the transmission replica d having the smallest square Euclidean distance expressed by Equation (16) is provisionally determined as the transmission signal X.
  • the modulation scheme is QPSK
  • there are four symbol points so there are as many values as the transmission replica d can take, and a transmission replica d can be taken for each transmission replica d.
  • the transmission replication power d used in the second-stage squared Euclidean distance is the power used for all symbol candidates. The number of computations of this squared Euclidean distance is reduced.
  • QR-MLD method (QRM—MLD (Maximum Likelihood Detection with QRdecomposition and the M_algorithm method) / Q Transmit replica used at Euclidean distance d
  • the number of candidates is set to S in order of increasing squared Euclidean distance obtained in the first stage (S is d
  • the maximum likelihood determination unit 16 holds the transmission replica candidate (d 1, d 2) and the corresponding square Euclidean distance in ascending order of the square Euclidean distance calculated from the equation (17).
  • the LLR calculation unit 17 Based on the square Euclidean distance calculated by the maximum likelihood determination unit 16, the LLR calculation unit 17 obtains a log likelihood ratio (LLR: Log Likelihood Ratio) for each bit for channel decoding.
  • LLR Log Likelihood Ratio
  • e 2 represents the minimum value of e 2 when the b-th bit of the transmission antenna p is “1”. same
  • e 2 represents the minimum value of e 2 when the b-th bit is “0”.
  • QR Maximum likelihood by MLD method min, p, b, 0 2
  • the log likelihood ratio required for channel decoding is obtained in this way.
  • the decoding unit 18 decodes the transmission signal x by Turbo decoding or the like using the log likelihood ratio.
  • the channel estimation value is decomposed into the product of the unitary matrix Q and the upper triangular row IJR, and then the square Euclidean distance is calculated.
  • the equation for calculating the square Euclidean distance E directly from Equation (5) without using QR decomposition is as shown in Equation (19).
  • the transmit replicas d and d in equation (19) are
  • MIMO is capable of increasing the transmission speed S, and information is transmitted reliably.
  • single stream transmission by SIMO Single Input Multiple Output
  • SIMO Single Input Multiple Output
  • signals transmitted from a single antenna are received by multiple antennas, and the received signals are received or processed by selection or combining diversity, so information is reliably transmitted when the SN (Single to Noise) ratio is small. It is effective when you want to.
  • the SIMD reception method with the best reception performance in a low SNR environment is the ML D method.
  • the signal point distance between the actually received received signal y and the symbol replica candidate point obtained by multiplying the transmission replica by the channel estimation value is compared, and this signal is compared.
  • the transmission signal is determined from the combination of transmission symbols with the smallest point-to-point distance. For example, when the square Euclidean distance is used as the distance between signal points and a single stream is received by multiple receiving antennas and the maximum likelihood is determined, the square Euclidean distance expressed by equations (22) and (23) is used.
  • the transmission replica d to be minimized is determined as the transmission signal X.
  • Equation (24) the addition value e 2 of the square Euclidean distance for two antennas (Equation (24)) is set as the final square Euclidean distance, and the transmission replica candidates are selected in order of decreasing square Euclidean distance.
  • the MLD method differs from the QR-MLD method in that the final squared Euclidean distance is obtained by adding the squared Euclidean distance e 2 (equation (24)) of the two antennas.
  • the output is in two stages corresponding to the same number of transmission signals, and the result of the squared Euclidean distance in the first stage is used to calculate the squared Euclidean distance in the second stage.
  • the smaller one is selected from the squared Euclidean distances of the antennas rather than simply calculating the squared Euclidean distances by adding the squared Euclidean distances of the antennas as shown in Equation (24).
  • Equation (24) min (e 2 ) / 2)
  • the square Euclidean distance calculated in this way is held in correspondence with the transmission replica candidates in ascending order of their power in the same manner as in the case of maximum likelihood determination by the QR-MLD method.
  • the distance is used to calculate the log-likelihood ratio for each bit for channel decoding.
  • Equation 25 The ratio LLR is calculated using Equation (25) similar to Equation (18). [Equation 25]
  • LLR e ⁇ l 2 ⁇ e ⁇ 0 2 (2 5) where e 2 represents the minimum value of e 2 when the b-th bit is '1'. Similarly, e 2 is a bit
  • Equation (16) z is a received signal after orthogonalization, and d is a transmission replica. Also,
  • R are diagonal elements of the R matrix obtained by QR decomposition of the channel estimate H, and are real numbers.
  • Equation (26) the square Euclidean distance in the case of the 1 ⁇ 2 antenna is calculated using Equation (26) to Equation (28) by the MLD method.
  • Equation (26) and Equation (27) the transmission replica d is multiplied by complex numbers h and h.
  • Eqs. (26) and (27) appear to be equal to the Euclidean distance calculation formula (16) of the first stage of the QR-MLD method.
  • Non-Patent Literature 1 Complexity-reduced Maximum Likelihood Detection Based on Replica candidate Selection with Decomp osition Using Pilot-Assisted Channel Estimation and Ranking for MIMOMultiplexing Using OFCDM), IEICE Technical Report, RCS2003_312 (2004-3)
  • the MLD method requires two real multiplications and two real additions more than the QR-MLD method. Therefore, in this state, a MIMO mode receiver that separates a spatially multiplexed signal from a plurality of transmit antennas using the same frequency and time using a plurality of receive antennas, and a single antenna. Force to receive transmitted signals with multiple antennas and select the received signals or receive them by combining diversity. There is a problem that the receiver cannot be downsized.
  • An object of the present invention is to share reception processing in the MIMO mode and SIMO mode in a communication system in which multistreams and single streams are mixed, and perform reception processing corresponding to the multimode.
  • a signal separation device and a signal separation method are provided.
  • the present invention is based on a force in which a received signal is either a multistream or a single stream, a mode identifying means for identifying, and a channel estimation value.
  • the channel compensation coefficient for calculating the channel compensation matrix and the triangular matrix is switched by switching the method for calculating the channel compensation matrix and the triangular matrix between the case where the received signal is a multi-stream and the case of a single stream.
  • a calculating means for multiplying the received signal by a complex conjugate transpose matrix of the channel compensation matrix; a multiplication result of the received signal and the complex conjugate transposed matrix; the triangular matrix and a transmission replica;
  • Maximum likelihood determining means for calculating a distance between signal points of a multiplication result with a signal, and likelihood calculating means for calculating the likelihood of the transmission replica signal using the distance between the signal points.
  • the complex conjugate transposed matrix multiplied by the received signal and the triangular matrix multiplied by the transmission candidate signal can be switched between the multi-stream case and the single stream case.
  • the SIMO mode reception processing can be shared to reduce the size of the signal separation device that can handle both modes.
  • signals that are spatially multiplexed using the same frequency and time by a plurality of transmitting antennas are received by a plurality of receiving antennas. It is possible to share the reception processing between the MIMO mode that is received by using the SIMO mode and the SIMO mode that receives signals transmitted from a single antenna using multiple antennas, and can perform reception processing corresponding to the multimode.
  • FIG. 1 A block diagram showing a main configuration of a conventional receiving apparatus when receiving a MIMO signal using the QR—MLD method.
  • FIG. 2 is a block diagram showing a main configuration of the receiving apparatus according to the embodiment of the present invention.
  • FIG.3 Block diagram showing the main configuration of the QR decomposition unit when receiving and processing MIMO signals using the QR-MLD method
  • FIG. 4 The main configuration of the QR decomposition unit when receiving and processing SIMO signals using the MLD method.
  • FIG. 5 A block diagram showing the main configuration of the channel compensation coefficient calculation unit according to the above embodiment. Lock figure
  • FIG. 6 Block diagram showing the main configuration of the orthogonalization unit when receiving and processing MIMO signals using the QR-MLD method.
  • FIG.7 Block diagram showing the main configuration of the orthogonalization unit when receiving and processing SIMO signals using the MLD method
  • FIG.8 Block diagram showing the main configuration of the maximum likelihood decision unit when receiving and processing MIMO signals using the QR-MLD method
  • FIG. 10 is a block diagram showing a main configuration of a maximum likelihood determination unit according to the embodiment.
  • FIG. 11 is a diagram for explaining the heel mode, SIMO mode, and SISO mode.
  • the channel estimation values are respectively applied to the y and h d terms in Equation (26).
  • the square Euclidean distance is calculated, and the coefficient multiplied to the transmission replica is made real.
  • the square Euclidean distance can be calculated by performing real multiplication and complex multiplication (two real multiplications) and complex subtraction once.
  • the square Euclidean distance can be calculated with the same number of operations as the QR—MLD method.
  • the number of calculations is not counted. This is because, in the QR-MLD method, the received signal y is multiplied by the complex conjugate transpose matrix Q H to the number of operations required to calculate the square Euclidean distance by the QR-M LD method shown above, and the received signal z The number of operations required to find
  • Equation (34) is the same as equation (15). I understand.
  • Equation (32) by using Equation (33) as the complex conjugate transpose matrix Q H , when the square Euclidean distance by MLD method is performed using orthogonalization processing similar to QR-MLD method, Necessary h and h can be calculated.
  • Equation (16) for calculating the square Euclidean distance in the first stage by the QR-MLD method is the same as Equation (26) by the MLD method.
  • equation (17) is a force S that is a different calculation formula from equation (26), and rd and e output from the first stage to the second stage in equation (17). If you replace 2 with zero,
  • Expression (35) which is a calculation expression similar to Expression (26), can be obtained.
  • each stage is independent.
  • Equation (37) channel estimation value H (Equation (37)) in which each component of channel estimation value H is zero as shown in Equation (36), received signal y is obtained from Equation (38). It is something to be calculated 0 ⁇
  • FIG. 2 shows a main configuration of the receiving apparatus according to the embodiment of the present invention.
  • a receiving apparatus 100 shown in FIG. 2 includes receiving antennas 110-1 and 110-2, receiving sections 120-1 and 120-2, a decoding section 180, and a signal separating apparatus 190.
  • the number of receiving antennas N is not limited to 2, and reception is performed according to the number of receiving antennas N.
  • a communication unit may be provided.
  • the receiving antennas 110-1 and 110-2 are multi-streams transmitted from a communication partner (not shown). Stream or single stream is received, and the receivers 120-1 and 120-2 are output.
  • Receiving sections 120-1 and 120-2 perform reception and demodulation processing on the multistream or single stream received via receiving antennas 110-1 and 110-2, and receive signals obtained Output y to channel estimation unit 130 and orthogonalization unit 150
  • Channel estimation section 130 performs channel estimation from received signal y, and outputs a channel estimation result to channel compensation coefficient calculation section 140.
  • Mode identification section 135 identifies whether the transmitted signal is multistream or single stream from a control signal notified from a communication partner (not shown), and the identification result is used as a channel compensation coefficient.
  • the calculation unit 140 and the maximum likelihood determination unit 160 output.
  • QR decomposition processing calculation is performed as channel compensation coefficient calculation processing when receiving a MIMO signal using the QR-MLD method.
  • the QR decomposition unit 240 is explained.
  • Figure 3 shows the main configuration of the QR decomposition unit 240.
  • the QR decomposition unit 240 shown in Fig. 3 includes input terminals 240-1, 240-2 squared norems 241-1, 241-2, inverse square roots 242-1, 242-2, and real multiplication 243-1, 243-2, 244-1, 244-2, inner product 245, complex multiplication 246, complex subtraction 247, and output unit 248.
  • the QR decomposition unit 240 decomposes the channel estimation value H into a unitary matrix Q and an upper triangular matrix R using the QR-MLD method (Formula (14)).
  • each component of channel estimation value H and unitary matrix Q is placed as shown in equation (39), each component of upper triangular matrix R is calculated from equations (40) to (45).
  • Figure 3 also shows the input / output relationship.
  • the subscripts i and q used in equation (40) represent the real part and the imaginary part, respectively.
  • FIG. 4 shows a configuration example of a main part of the QR decomposition unit 340 for calculating a channel compensation coefficient when receiving and processing a SIMO signal using the MLD method.
  • the QR decomposition section 340 shown in FIG. 4 includes an input terminal 240-2, an inner product 245, and an output section 249.
  • Figure 4 also shows the input / output relationship.
  • the output unit 249 outputs a complex conjugate transpose of a matrix having channel estimation values h and h as diagonal elements.
  • Is output to the orthogonalization unit 150 and I h and I h I 2 are diagonal elements.
  • the sequence is output to maximum likelihood determination section 160.
  • FIG. 5 shows a configuration example of a main part of channel compensation coefficient calculation section 140 according to the embodiment of the present invention.
  • the channel compensation coefficient calculation unit 140 employs a configuration in which a switch 1 41 -1-141 -5 and an output unit 142 are added to the QR decomposition unit 240 shown in FIG.
  • Channel compensation coefficient calculation section 140 switches a signal to be output to the arithmetic unit in the subsequent stage of each switch according to mode selection information S output from mode identification section 135, and calculates a channel compensation matrix from the channel estimation value. And calculate the triangular matrix.
  • the mode selection information is information for identifying whether it is multistream or single stream, and is notified from a communication partner (not shown).
  • H (h, h) is input to the input terminal 240-2, while the mode selection information is a single stream.
  • H (h, h) is input to the input terminal 240-2.
  • switch 141-1 When the mode selection information indicates multi-stream, switch 141-1 outputs Q to inner product 245 in the subsequent stage.
  • the mode selection information is a single stream.
  • inner product 245 calculates r, and when mode selection information is single stream, inner product 245 is I h
  • switch 141-2 indicates that the mode selection information is multi-stream, Q is output to the output unit 142. If the mode selection information is single stream, h is output.
  • the switch 141-3 outputs r to the output unit 142 when the mode selection information indicates multi-stream, and when the mode selection information is single-stream,
  • the switch 141-4 When the mode selection information indicates multi-stream, the switch 141-4 outputs r to the output unit 142, and when the mode selection information is single-stream,
  • the switch 141-5 outputs Q to the output unit 142 when the mode selection information indicates multi-stream, and outputs h when the mode selection information is single-stream.
  • the switch 141— ;! to 141-5 can be switched according to the mode selection information S indicating whether the stream is multistream or single stream.
  • Channel compensation matrix and triangular matrix required for stream and single stream reception processing can be acquired.
  • channel compensation coefficient calculation section 140 calculates a unitary matrix as a channel compensation matrix by QR decomposition of the channel estimation value, and calculates an upper triangular matrix as a triangular matrix.
  • channel compensation coefficient calculation section 140 calculates a diagonal matrix having channel estimation values h and h as diagonal elements as a channel compensation matrix, and I h I 2 and I h
  • a diagonal matrix having I 2 as a diagonal element is calculated as a triangular matrix.
  • the output unit 142 in the case of multi-stream, and outputs the complex conjugate transposed matrix Q H of Yunitari matrix Q to the orthogonalization section 150, while outputting the upper triangular ascending ⁇ IJR to maximum likelihood determination unit 160
  • the diagonal with channel estimation values h and h as diagonal elements
  • the complex conjugate transpose matrix of the matrix is output to the orthogonalization unit 150 and I h is
  • FIG. 6 shows a main configuration of orthogonalizing section 250 that realizes equation (48).
  • the orthogonalization unit 250 includes a multiplier 251— ;! to 251—4, a delay unit 252— ;! to 252—4, 254— ;! to 254-4, and a calorie calculator 253— ;! to 253—. 4 and.
  • Equation (49) the complex conjugate transpose ⁇ * of the channel estimation value ⁇ is multiplied as a weight on the received signal y as shown in Equation (49). It is done.
  • FIG. 7 shows the main configuration of the orthogonalizing unit 350 that realizes the equation (49).
  • the orthogonalization unit 350 shown in FIG. 7 employs the same configuration as the orthogonalization unit 250 using the QR-MLD method shown in FIG. 251— ;! to 251-4, delay units 252— ;! to 252—4, 254—;! To 254-4, and Calorie calculator 2 53— ;! to 253-4. Therefore, as shown in Equation (49), the orthogonalization unit can be shared by multiplying the received signal y by the complex conjugate transpose H * of the channel estimation value H as a weight.
  • Eqs. (48) and (49) orthogonalization is performed with half the amount of computation compared to the QR-MLD method.
  • the maximum likelihood determination is performed in the two stages of the first stage and the second stage.
  • the square Euclidean distance calculation for each stage is calculated from Equation (16) and Equation (17), and the square Euclidean distance e 2 at each transmit replica d of the first stage.
  • FIG. 8 shows a main configuration of maximum likelihood determination section 260 when receiving and processing a MIMO signal using the QR-MLD method.
  • the maximum likelihood determination unit 260 shown in FIG. 8 includes transmission replica generation units 261-1 and 261-2, £ separation calculation 262 262-1 and 262-2, and surviving replica selection ⁇ 263-1 and 263 -2. It has.
  • the square Euclidean distance in the case of a 1 ⁇ 2 antenna is calculated from the equations (29) and (30).
  • the concept of the first stage and the second stage is not calculated, and the square Euclidean distance is calculated individually for each receiving antenna, and finally, the square Euclidean distance for each receiving antenna is added to obtain the final.
  • the square Euclidean distance is calculated.
  • FIG. 9 shows a main configuration of maximum likelihood determination section 360 using the MLD method.
  • the maximum likelihood determination unit 360 shown in FIG. 9 replaces the maximum likelihood determination unit 260 shown in FIG. 8 by replacing the transmission replica generation unit 261-2 with the transmission replica generation unit 261-1, and replacing the distance calculation unit 262-2 with the distance. Instead of the calculation unit 262-1, the surviving replica selection unit 263-2 is replaced with the surviving replica selection unit 263-1, and a symbol selection unit 361 is added.
  • the maximum likelihood determination unit 260 shown in FIG. 8 is configured to have two first stages.
  • Symbol selection section 361 calculates the squared Euclidean distance for each receiving antenna. Specifically, the square Euclidean distance shown in Expression (31) is calculated. However, since reciprocal calculation is required in Equation (31), even if the square Euclidean distance shown in Equation (50) is calculated, good.
  • FIG. 10 shows a main configuration of maximum likelihood determination section 160 according to the present embodiment.
  • the maximum likelihood determination unit shown in FIG. 10 adopts a configuration in which a symbol selection unit 361 and a switch 161 are added to FIG.
  • the switch 161 switches a signal to be output to the LLR calculation unit 170 in accordance with the mode selection information S indicating multistream or single stream. Specifically, when the mode selection information indicates that the mode selection information is multi-stream, the switch 161 outputs the square Euclidean distance calculated using Equation (17) to the LLR calculation unit 170, and the mode selection information is In the case of a single stream, the squared Uterid distance calculated by the symbol selection unit 361 is output to the LLR calculation unit 170.
  • surviving replica selection section 263-1 also outputs replica signal candidates to symbol selection section 3601.
  • the square Euclidean distance is calculated in two stages of the first stage and the second stage, and when the mode selection information is single stream, reception is performed. The squared Euclidean distance is calculated separately for each antenna.
  • the QR—MLD method performs multi-stream reception and the MLD method performs single-stream reception, so the number of bits for calculating the log-likelihood ratio differs for each mode, but the equations used in the LLR calculation logic (18), (25 ) Can be realized with the same configuration, so it is necessary to share the LLR calculation unit 170 with the force S.
  • the channel compensation coefficient calculation unit 140 multiplies the received signal by the channel compensation coefficient calculation unit 140 and the transmission replica signal in both the multi-stream case and the single-stream case.
  • the channel estimation value is QR-decomposed to obtain the unitary matrix and upper triangular matrix, and the unitary matrix is converted to the channel compensation matrix.
  • upper triangular matrix is triangular row ⁇ IJ, and in the case of a single stream, a diagonal matrix with the channel estimation value as the diagonal element is the channel compensation matrix, and the diagonal matrix is the product of the channel estimate and the complex conjugate of the channel estimation value. Is a triangular matrix.
  • the signal point distance between the multiplication result obtained by orthogonalizing the received signal and the complex conjugate transpose matrix of the channel compensation matrix and the multiplication result of the triangular matrix and the transmitted replica signal is The maximum likelihood is determined by calculating for each stage corresponding to the number of transmission signals, and the likelihood of the transmission replica signal is calculated using the distance between signal points. Since the received signal z after orthogonalization, which is obtained by multiplying the received signal y by the complex conjugate, is in a state in which the phase rotation has been restored, the resource for calculating the distance between signal points using the QR-MLD method for the multi-stream is reduced. The distance between signal points can be calculated using the MDL method.
  • a plurality of transmission antennas allow signals that are spatially multiplexed using the same frequency and time to be received by the plurality of reception antennas. It is possible to reduce the size of a receiver that can handle both modes by sharing the reception processing of the MIMO mode for reception and the SIMO mode for receiving signals transmitted from a single antenna by multiple antennas. it can.
  • maximum likelihood determination section 160 calculates the square Euclidean distance using only equation (16), and outputs the calculation result to LLR calculation section 170.
  • the square Euclidean distance in the first stage according to the QR-MLD method is equal to the square Euclidean distance between the received signal received via a single receiving antenna and the transmitted replica, so the QR for the multistream in MIMO mode —
  • the square Euclidean distance by the MLD method can be calculated for the SISO.
  • the SISO receives and processes a signal transmitted from a single antenna with a single antenna, power consumption can be reduced compared to MIMO and SIMO.
  • the circuit resources of the QR—MLD method for multistreams in the MIMO mode can be used for SIMO and SISO.
  • the square Euclidean distance can be calculated by the MLD method, and it becomes possible to support multimode (Fig. 11).
  • One aspect of the signal separation device of the present invention includes mode identifying means for identifying whether a received signal is a multistream or a single stream, a channel compensation matrix and a triangular matrix based on a channel estimation value.
  • Channel compensation coefficient calculating means for calculating the channel compensation matrix and the triangular matrix by switching the method of calculating the channel between the case where the received signal is a multi-stream and the case of a single stream, and the received signal Orthogonalizing means for multiplying the channel compensation matrix by the complex conjugate transpose matrix, a multiplication result of the received signal and the complex conjugate transpose matrix, and a multiplication result of the triangular matrix and the transmission replica signal.
  • a configuration comprising: a maximum likelihood determination unit that calculates a signal point distance; and a likelihood calculation unit that calculates the likelihood of the transmission replica signal using the signal point distance.
  • the complex conjugate transposed matrix multiplied by the received signal and the triangular matrix multiplied by the transmission candidate signal can be switched between the multi-stream case and the single stream case.
  • the SIMO mode reception processing can be shared to reduce the size of the signal separation device that can handle both modes.
  • the channel compensation coefficient calculating means performs QR decomposition on the channel estimation value to obtain a unitary matrix and an upper triangular matrix when the received signal is a multi-stream.
  • the diagonal matrix having the channel estimation value as a diagonal element is used as the channel compensation.
  • a matrix is used, and a diagonal matrix having a diagonal element as a product of the channel estimation value and the complex conjugate of the channel estimation value is used as the triangular matrix.
  • the maximum likelihood determination means includes a plurality of processing means for calculating the distance between the signal points corresponding to the same number of stages as the number of multi-stream transmission signals. Selection means for calculating the distance between the signal points used by the likelihood calculation means based on the distance between the signal points calculated in each stage; and when the received signal is a multi-stream, The distance between the signal points calculated in the final stage is selected as the distance between the signal points used by the likelihood calculating means, and when the received signal is a single stream, the signal point calculated by the selecting means. And a switching means for selecting the distance.
  • the same resource is used to calculate the distance between the signal points in the same number of stages as the number of multi-stream transmission signals.
  • the maximum likelihood determination by the MLD method can be performed by selecting the distance between the signal points used for the likelihood.
  • One aspect of the signal separation device of the present invention employs a configuration further comprising receiving means for receiving the received signal with any one of a plurality of receiving antennas.
  • each functional block used in the description of each of the above embodiments is typically realized as an LSI that is an integrated circuit. These may be individually integrated into one chip, or part or all of them. It may be integrated into a single chip to include Here, it may be called IC, system LSI, super LSI, or ultra LSI, depending on the difference in power integration as LSI. Also, the method of circuit integration is not limited to LSI, and may be realized by a dedicated circuit or a general-purpose processor. Field programmable gate arrays (FPGAs) that can be programmed after LSI manufacturing and reconfigurable processors that can reconfigure the connection and settings of circuit cells inside the LSI may be used. Furthermore, if integrated circuit technology that replaces LSI emerges as a result of advances in semiconductor technology or other derived technologies, it is naturally also possible to integrate functional blocks using this technology. For example, the possibility of applying technology is possible.
  • FPGAs Field programmable gate arrays
  • the signal separation device and signal separation method of the present invention share reception processing in the MIMO mode and the SIMO mode in a communication system in which multistreams and single streams are mixed, and receive signals corresponding to the multimodes.
  • reception processing in the MIMO mode and the SIMO mode in a communication system in which multistreams and single streams are mixed, and receive signals corresponding to the multimodes.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Power Engineering (AREA)
  • Radio Transmission System (AREA)

Abstract

L'invention concerne un dispositif de séparation de signaux dans lequel un processus de réception est partagé par le mode MIMO et le mode SIMO et le processus de réception est réalisé en accord avec un mode multiple dans un système de communication dans lequel un flux multiple et un flux unique sont mélangés. Le dispositif comprend une unité de calcul de coefficient de compensation de canal (140). Dans le cas du flux multiple, l'unité de calcul de coefficient de compensation de canal (140) calcule une matrice unitaire en guise de matrice de compensation de canal en décomposant par QR une valeur d'estimation de canal et calcule une matrice triangulaire supérieure en guise de matrice triangulaire. D'autre part, dans le cas du flux unique, l'unité de calcul de coefficient de compensation de canal (140) calcule une matrice diagonale ayant des valeurs d'estimation de canal h1, h2 comme éléments diagonaux en guise de matrice de compensation de canal et calcule une matrice diagonale ayant |h1|2, |h2|2 comme éléments diagonaux en guise de matrice triangulaire.
PCT/JP2007/066155 2006-08-22 2007-08-21 Dispositif de séparation de signaux et procédé de séparation de signaux WO2008023683A1 (fr)

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JP2010057071A (ja) * 2008-08-29 2010-03-11 Nec Corp 復調器および復調方法
TWI405416B (zh) * 2009-05-19 2013-08-11 Mediatek Inc 從複數個天線偵測資料的方法及系統
JP5645848B2 (ja) * 2009-12-25 2014-12-24 パナソニック株式会社 無線受信装置
CN111903081A (zh) * 2018-03-30 2020-11-06 日本电信电话株式会社 Oam多路复用通信系统以及模式间干扰补偿方法

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JP2004135328A (ja) * 2002-08-28 2004-04-30 Texas Instruments Inc 送信ダイバーシティ方式用の効率的な受信機構造
JP2005521358A (ja) * 2002-04-01 2005-07-14 インテル・コーポレーション 無線で送信された情報の送信モードをダイナミックに最適化するシステムおよび方法
JP2005354347A (ja) * 2004-06-10 2005-12-22 Sony Corp 通信システム、送信装置および受信装置
JP2006121348A (ja) * 2004-10-20 2006-05-11 Ntt Docomo Inc 信号分離装置及び信号分離方法

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JP2005521358A (ja) * 2002-04-01 2005-07-14 インテル・コーポレーション 無線で送信された情報の送信モードをダイナミックに最適化するシステムおよび方法
JP2004135328A (ja) * 2002-08-28 2004-04-30 Texas Instruments Inc 送信ダイバーシティ方式用の効率的な受信機構造
JP2005354347A (ja) * 2004-06-10 2005-12-22 Sony Corp 通信システム、送信装置および受信装置
JP2006121348A (ja) * 2004-10-20 2006-05-11 Ntt Docomo Inc 信号分離装置及び信号分離方法

Cited By (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2010057071A (ja) * 2008-08-29 2010-03-11 Nec Corp 復調器および復調方法
TWI405416B (zh) * 2009-05-19 2013-08-11 Mediatek Inc 從複數個天線偵測資料的方法及系統
JP5645848B2 (ja) * 2009-12-25 2014-12-24 パナソニック株式会社 無線受信装置
CN111903081A (zh) * 2018-03-30 2020-11-06 日本电信电话株式会社 Oam多路复用通信系统以及模式间干扰补偿方法
CN111903081B (zh) * 2018-03-30 2022-07-22 日本电信电话株式会社 Oam多路复用通信系统以及模式间干扰补偿方法

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