WO2002043256A1 - Procede de filtrage spatial de mise en correspondance et recepteur en reseau dans un systeme de radiocommunication - Google Patents

Procede de filtrage spatial de mise en correspondance et recepteur en reseau dans un systeme de radiocommunication Download PDF

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Publication number
WO2002043256A1
WO2002043256A1 PCT/CN2001/000693 CN0100693W WO0243256A1 WO 2002043256 A1 WO2002043256 A1 WO 2002043256A1 CN 0100693 W CN0100693 W CN 0100693W WO 0243256 A1 WO0243256 A1 WO 0243256A1
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Prior art keywords
digital
signal
array
beam output
output signal
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PCT/CN2001/000693
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English (en)
French (fr)
Inventor
Jiang Li
Jinlin Zhang
Qi Ding
Hebing Wu
Junfeng Guo
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Huawei Technologies Co., Ltd.
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Application filed by Huawei Technologies Co., Ltd. filed Critical Huawei Technologies Co., Ltd.
Priority to EP01937958A priority Critical patent/EP1341315B1/en
Priority to AU2001263744A priority patent/AU2001263744A1/en
Priority to DE60137963T priority patent/DE60137963D1/de
Publication of WO2002043256A1 publication Critical patent/WO2002043256A1/zh
Priority to US10/439,219 priority patent/US7006849B2/en

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Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/02—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas
    • H04B7/04—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas
    • H04B7/0408—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas using two or more beams, i.e. beam diversity
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/02—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas
    • H04B7/04—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas
    • H04B7/08—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the receiving station
    • H04B7/0837—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the receiving station using pre-detection combining
    • H04B7/0842—Weighted combining
    • H04B7/086—Weighted combining using weights depending on external parameters, e.g. direction of arrival [DOA], predetermined weights or beamforming
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/02—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas
    • H04B7/04—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas
    • H04B7/08—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the receiving station
    • H04B7/0868—Hybrid systems, i.e. switching and combining
    • H04B7/088—Hybrid systems, i.e. switching and combining using beam selection

Definitions

  • the present invention relates to an array processing method for an array receiver in a wireless communication system and an array receiver using the method, and more particularly, to a spatial-domain matched filtering method and an array receiver in a wireless communication system. Background technique
  • FDMA frequency division multiple access
  • TDMA time division multiple access
  • CDMA code division multiple access
  • Diversity technology mainly uses the irrelevant nature of the signals received by different antennas with a spacing greater than 10 carrier wavelengths, and uses the maximum ratio to combine the signals received by the antennas to improve the system's resistance to multipath fading.
  • the sectorization method is to divide a cell into 3, 6, 9, or 12 sectors, and each sector has a matching antenna and a predetermined spectrum range. Sectorization reduces communication channel interference to a certain extent, thereby improving the communication quality of the system.
  • Switched multi-beams form fixed beams in different directions in the cell.
  • the base station detects the signal quality of the desired signal in each beam and selects the best beam to receive.
  • One of the main reasons why it is called a switched multi-beam is that during the system's beam selection process, there are controllable switches on the path between each beam and each channel receiver, the so-called "switch matrix.” When a beam is selected, the path switches between the wave and the corresponding receiver are closed, while the switches of the other paths are opened.
  • the adaptive antenna array adaptively weights and combines the signals received by each antenna according to the maximum signal-to-noise ratio criterion, maximum likelihood criterion, and minimum mean square error criterion to effectively suppress interference and noise signals, thereby improving the wireless system. Overall performance.
  • the diversity method requires a larger spacing between antennas (generally greater than 10 wavelengths), the more antennas take up more space, the more space the base station can actually use is very limited. In addition, although the diversity method using maximum ratio combining has anti-multipath fading effects, it cannot effectively suppress interference signals.
  • the current common sectorization method is to use 3 sectors or 6 sectors.
  • the reason why more sectors are not used is because the more the sector is split, the less spectrum resources are available for each sector, which reduces the Following efficiency. And the more the sector is split, the more the beams overlap between different sectors, but the co-channel interference increases and the system performance decreases.
  • Switched multi-beam can also be regarded as a sectorization method in a sense, but the division of sectors is made by the dynamic combination of different beams. Because the "best" beam is always selected for reception, different from the sectorization method, the more overlap between the switched multi-beam beams will reduce the gain loss at the beam-to-beam interface.
  • the beams of existing switched multi-beam systems are coherently superimposed by a directional antenna or a radio frequency phase-shifted network (such as a Butler matrix) to form multiple narrow-beam coverage cells pointing in different directions in space. Theoretically, the narrower the beam, the better the spatial filtering performance of the beam-switched multi-beam antenna, and the stronger the interference suppression capability.
  • the existing switched multi-beam systems are limited in their ability to improve system communication capacity.
  • the switching matrix of the existing switched multi-beam system is implemented by a radio frequency switching device, which increases the hardware cost of the system.
  • the switching beam is usually selected based on the expected signal power in the beam. When there is a strong interference with the user and the expected user power is estimated for a short time, the beam selection error is often caused.
  • the adaptive antenna array uses adaptive algorithms to obtain the array weighting coefficients according to different criteria. Although the system performance can be optimized to a certain extent, the adaptive algorithms with good performance often require a large amount of calculation. For digital signal processing devices, The requirements are high, and many algorithms cannot be implemented with existing high-speed processing chips.
  • the object of the present invention is to provide a method for matching filtering in the digital baseband space domain in an array receiver for a wireless communication system.
  • Another object of the present invention is to provide an array receiver for a wireless communication system.
  • the digital baseband spatial domain matching filtering module used in the present invention uses the above-mentioned digital baseband spatial domain matching filtering method of the present invention. Characteristics, therefore, low hardware cost and good system performance. Summary of the invention
  • the digital baseband spatial domain matched filtering method includes the following steps:
  • the superscript "*" represents a complex conjugate operation, ⁇ is a constant, 0 ⁇ ⁇ 1;
  • Coeff) (k) Coeffi (k) + aCoeff i (k-1)
  • the superscript "*" indicates a complex conjugate operation
  • s (n) is the reference signal
  • N is the total number of sampling points
  • the product of the sampling time should be less than the coherence time
  • C 03 ⁇ 4 ⁇ (it) represents the kth time
  • the correlation coefficient obtained by coherent accumulation reflecting the desired signal energy in beam i is a constant, 0.5 ⁇ ⁇ 1;
  • the present invention further provides an array receiver, including an antenna array composed of multiple antennas;
  • An array digital signal generating module connected to the antenna array and configured to convert an antenna array analog signal received by the antenna array into an array digital signal
  • a digital baseband spatial domain matching filtering module connected to the array digital signal generating module, and configured to form one or more signal beams on each channel of the array digital signals of the array digital signal generating module;
  • a digital receiver module connected to the digital baseband space-domain matching filtering module, and configured to receive one or more signal beams formed on each channel by the digital baseband space-domain matching filtering module, in a time domain; Combining the signal beams;
  • the digital baseband spatial domain matching filtering module includes:
  • a digital baseband spatial domain matched filter bank receiving an array digital signal, and combining the array digital signal with
  • NB group weighting vector performs weighting operation to output NB group digital beam output signal
  • a splitter that forms the digital beam output signals S Bi (price) of the NB group digital beam output signals output by the digital baseband spatial domain matched filter bank to form digital beams corresponding to multiple channels;
  • a multi-beam selection module receives a digital beam output from the splitter, performs power normalization on the digital beam output signal as follows, and estimates the power value of the digital beam output signal (") within a relevant time.
  • the superscript "*" indicates a complex conjugate operation
  • ⁇ is a constant, 0 ⁇ ⁇ ⁇ 1; and then, the power normalized digital beam output signal ( «) and a reference signal are correlated, and it is estimated to reflect each The correlation coefficient C 03 ⁇ 4 T; of the desired signal energy in each beam, and outputs a selection instruction to select a digital beam output signal corresponding to the maximum correlation coefficient.
  • the estimated formula is:
  • Coeffi (k) Coeffi (k) + aCoeff i (k-1)
  • the superscript "*" represents a complex conjugate operation
  • s (n) is the reference signal
  • N is the total number of sampling points, and the product of it and the sampling time should be less than the coherence time
  • C 03 ⁇ 4 ⁇ (A :) represents the kth
  • the correlation coefficient obtained from the secondary coherent accumulation reflecting the desired signal energy in beam i is a constant, 0.5 ⁇ « ⁇ 1;
  • a digital switch matrix is respectively connected to the splitter and the multi-wave speed selection module, receives a selection instruction output by the multi-beam selection module, and outputs a digital beam output signal corresponding to the largest correlation coefficient.
  • FIG. 1 is a functional block diagram of an array receiver of the present invention
  • FIG. 2 shows a beam diagram of a coverage cell
  • Figure 3 shows the internal functional block diagram of the digital baseband airspace matched filtering module 104 in Figure 1;
  • Figure 4 shows a schematic diagram of a digital switch matrix
  • Figure 5 shows the basic principle diagram of a digital baseband spatial domain matched filter bank
  • FIG. 6 shows a flowchart of a digital baseband spatial domain matched filtering method according to the present invention
  • Fig. 7 shows simulation results when the present invention is applied to a CDMA system. The best embodiment of the present invention
  • FIG. 1 is a schematic diagram of an array receiver of the present invention.
  • the receiver mainly includes an antenna array 102, an array digital signal generating module 103, a digital baseband spatial domain matching filtering module 104, and a digital receiver module 105.
  • the main function of the array digital signal generating module 103 is to convert the antenna array analog signal received from the antenna array 102 into an array digital signal for digital processing, which includes a receiving unit (RX) 106 and an analog-to-digital conversion unit (A / D) 107.
  • the digital baseband spatial domain matching filtering module 104 can form one or more digital beams for a channel, for example, one code channel of the C band A system.
  • the digital baseband spatial domain matching filtering module 104 can form beams for different multipath signals of the same code channel.
  • the beamformed signal is sent to the corresponding finger of the digital receiver module 105 (a Rake receiver can be used), and the digital receiver module 105 combines the signals in the time domain. Since the analog signal is converted to a digital signal, the digital baseband space-domain matched filtering module 104 can be flexibly changed according to different requirements of the system.
  • the digital baseband spatial domain matching filter module 104 shown in FIG. 1 an array processing algorithm is used to process the received signal to form a so-called beam, so that the quality of the signal output in the beam is improved, thereby improving the performance of the entire receiver. Therefore, the array processing method adopted by the digital baseband spatial-domain matched filtering module 104 and the implementation complexity and stability are directly related to the system performance.
  • the invention proposes a simple, reliable and easy-to-implement array receiving method.
  • the digital signal output by the array digital signal generating module 103 in the array receiver shown in FIG. 1 can be expressed as:
  • X (t) ⁇ () A, (ts (t-7 ",) + n (t) (1)
  • 1,2,..., L is a multipath number
  • a () is M * l-dimensional vector, which represents the array response of the multipath signal related to direction 0 on the M antenna array elements, is the direction of arrival of the multiplier of the Jth multipath
  • (t) is the signal experienced by the multipath signal of the _ / th Fading
  • s (t) is the desired signal transmitted, r, is the delay of the Jth multipath signal
  • n (t) is the array interference and noise signal
  • the antenna array 102 such as a uniform linear array, a circular array, etc.
  • the value of N B should be such that the gain of the intersection point 201 between the beam formed by a ff () and the maximum gain of all beams is _3 ⁇ 0 dB That is, the marked line 202 in FIG. 2 is between -3 and 0 dB.
  • FIG. 3 illustrates an internal functional block diagram of the air-domain matched filtering module 104 in FIG. 1.
  • the air-domain matched filtering module 104 includes an air-domain matched filter bank 401, a splitter 402, a multi-beam selection module 403, and a digital switch matrix 404.
  • There are N sets of weighting vectors W in the spatially matched filter bank 401 ; a ff (), which receives the digital array signal X (t) output by the array digital signal generating module 103, and uses a set of weighting vectors to the digital array signal X ( t) weighting to obtain ⁇ digital beam output signals (0, the weighting result can be expressed by the following formula:
  • the multi-beam selection module 3 first calculate the power value of the digital beam output signal within the relevant time
  • This processing can greatly reduce the influence of strong interference signals on the correlation of beam selection.
  • the transmitted signals contain known information such as training sequences and pilot symbols that are known to the receiving end in advance.
  • training sequences 26 bits per Norma 1 burst
  • pilot symbols in each slot (Slot) of the DPCCH.
  • the receiving end can directly or indirectly obtain the so-called reference signal s (t) by using the known information.
  • the reference signal s (t) Correlation with the digital beam output signal (0), the correlation coefficient Coe ⁇ reflecting the desired signal energy in each beam output signal is obtained, and the specific calculation method is as follows:
  • Coeff t (k) Coeff, (k) + aCoeff, (k- ⁇ ) (6)
  • the multi-beam selection module 403 performs operations on the beam input from the splitter 402 as shown in formulas (3) to (6), and obtains a correlation coefficient ⁇ 3 ⁇ 4 ⁇ ⁇ ( ⁇ :). Then, as described above, the obtained correlation coefficients of the desired number of energies are compared, the largest correlation coefficient ⁇ 3 ⁇ 43 ⁇ 4 ⁇ is selected, and a selection instruction is output to the digital switch matrix 404.
  • the schematic of the digital switch matrix 404 is shown in Figure 4. Since a person of ordinary skill in the art can implement the digital switch matrix 404 according to the above requirements, only the digital switch matrix 404 shown in FIG. 4 is briefly described here.
  • the digital switch matrix 404 is mainly composed of a beam data line 601, an output data line 109, a digital switch 602, and a switch control signal line 603. It can be seen from FIG. 4 that each beam data line 601 can be connected to an output data line 109 corresponding to a different channel through a switch 602, and a control signal (previously called a selection instruction) output by the multi-beam selection module 403 is controlled by a switch control signal line 603 Controls the closing and opening of digital switches. Control signals can be represented by a control matrix.
  • the multi-beam selection module 403 can conveniently control the digital switch matrix 404 to select an appropriate beam to a designated output end.
  • the basic principle of the spatial domain matched filter bank 401 is shown in FIG. 5. It is mainly composed of a multiplier 502 and an adder 503.
  • the signal entering the spatial matched filter bank 401 is first subjected to complex multiplication with a preset weighting vector (501) by a multiplier 502, and then the complex signal is combined by an adder 503.
  • the combined signal is the output of the digital beam. Since the spatially matched filter bank 401 is mainly composed of a multiplier 502 and an adder 503, it can be conveniently implemented by an FPGA. Of course, the spatially matched filter bank 401 may also be configured by using other digital devices capable of performing multiplication and addition operations.
  • the implementation of the spatial matching filter block 104 of the present invention utilizes a new spatial matching filtering method.
  • the above formula (3), ( 4), (5), and (6) obtain correlation coefficients that reflect the desired signal energy in each beam, and then select and determine a digital beam based on the correlation coefficients.
  • FIG. 6 shows a flowchart of the spatial domain matching filtering method of the present invention.
  • step S1 the array digital signal X (t) is received in step S1, which corresponds to FIG. 1, that is, the airspace matching filtering module 104 receives the array digital signal from the array digital signal generating module 103.
  • step S2 a weighting operation is performed on the array digital signals and the NB group weighting vectors to obtain the N a group digital beam output signals.
  • This step S2 corresponds to the work performed by the spatial domain matched filter group 401 in FIG. 3.
  • step S3 the average power value of each beam and the power normalized signal digital wave output are estimated according to the digital beam output signal ( ⁇ , according to formulas (3) and (4)).
  • ⁇ the digital beam output signal
  • step S4 the correlation coefficient (3 ⁇ 4) that reflects the expected signal energy in each beam is estimated; the estimation is performed according to formulas (5) and (6), and the specific method can refer to the above description.
  • the next step is to compare the obtained correlation coefficients C 3 ⁇ 4 reflecting the desired signal energy in each beam; ⁇ :) to obtain the maximum correlation coefficient ⁇ , and output a number corresponding to the maximum correlation coefficient Beam output signal (see step S5).
  • steps S3 and S4 may be implemented by the multi-beam selection module 403 in the air-domain matching filtering module 104 shown in FIG. 3, and step S5 may be performed by the digital switch matrix 404 according to the multi-beam selection module 403. Selection instructions to achieve.
  • the inventor uses the spatial domain matching filtering method provided by the present invention to A simulation experiment was performed. The results are shown in Figure 7. The multi-path environmental parameters can be seen in the table below.
  • bit error rate digital receiver module output signal may be in the vicinity of 10-3, while the digital single-antenna receiver modules SNR required antenna terminal of the input signal is about 8.2dB can be made when the output signal reaches the error rate of 10-3.
  • the system performance is significantly improved.

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Description

无线通信系统中空域匹配滤波方法及其阵列接收机 技术领域
本发明涉及无线通信系统的阵列接收机的阵列处理方法及应用这种方法 的阵列接收机， 尤其涉及无线通信系统中空域匹配滤波方法及其阵列接收机。 背景技术
长期以来，无线通信系统始终面临着有限的频谱资源与不断快速增长的用 户之间的矛盾。 尽管频分多址（FDMA)、 时分多址（TDMA)以及码分多址（CDMA)技 术在一定程度上提高了的系统容量，但这还远不能满足日益增长的无线业务量 的需求。 因此人们开始利用信道的空域特性， 如采用分集、 扇区化以及最近采 用的开关多波束和自适应天线阵等方法来提高接收系统的容量。这些方法在不 同程度上改善了无线通信系统的通信质量， 提高了容量。
分集技术主要利用间距大于 10个载波波长的不同天线所接收的信号不相 关这一性质， 采用最大比合并各天线接收的信号， 使系统抗多径衰落性能得到 改善。
扇区化方法是将小区分成 3、 6、 9或 12个扇区， 每个扇区有配套的天线和 预定的频谱范围。 扇区化在一定程度上减小了通信道干扰， 因而提高了系统的 通信质量。
开关多波束是在小区不同方向上形成固定波束，基站检测每个波束中期望 信号的信号质量， 选择最好的波束进行接收。 之所以称为开关多波束， 其中一 主要原因是， 系统在选择波束的过程中， 每个波束与各信道接收机之间的通路 上有可控的开关， 即所谓"开关矩阵"。 当某一波束被选中后， 该波柬与相应接 收机之间的通路开关闭合， 而其它通路的开关断开。
自适应天线阵根据最大信噪比准则、最大似然准则以及最小均方误差准则 等， 自适应地对各天线接收的信号进行加权合并， 对干扰和噪声信号进行有效 地抑制， 从而提高无线系统的整体性能。
由于采用分集方法需要天线之间的间距比较大（一般大于 10个波长） ， 因 此天线越多占用的空间越大， 而基站实际能利用的空间是十分有限的。 另外， 采用最大比合并的分集方法虽然具有抗多径衰落效果，但不能有效抑制干扰信 号。
现在常见的扇区化方法是采用 3扇区或 6扇区， 之所以没有采用更多的扇区 是因为扇区分裂得越多， 每个扇区可用的频谱资源就越少， 降低了中继效率。 且扇区分裂得越多， 不同扇区之间重叠的波束就越多， 反而使同信道干扰增 加， 系统性能下降。
开关多波束在某种意义上也可以认为是扇区化方法，只是扇区的划分是由 不同波束动态组合而成的。 由于接收时总是选择 "最佳 "的波束进行接收， 所以 与扇区化方法不同， 开关多波束的波束之间重叠得越多反而会降低波束与波束 交界处的增益损失。现有开关多波束系统的波束由定向天线或采用射频相移网 络（如 Butler矩阵)进行场的相干叠加，形成多个指向空间不同方向的窄波束覆 盖小区。理论上讲，如果波束越窄，波束开关多波束天线的空域滤波性能越好， 抑制干扰能力越强。但由于定向天线的口径限制以及射频相移网络的相移精度 有限， 波束宽度有限， 波束之间的重叠也有限， 使现有开关多波束系统在系统 通信容量的改善能力上受到限制。 另外， 现有开关多波束系统的开关矩阵是由 射频开关器件实现的， 增加了系统的硬件成本。 开关波束在选择波束时， 通常 根据波束中期望信号功率大小进行， 当有强干扰用户而估计期望用户功率的时 间比较短时， 就往往会造成波束选择错误。
自适应天线阵根据不同准则采用自适应算法得到阵列加权系数， 虽然在某 种程度上可以使系统性能达到最优，但性能优良的自适应算法往往要求的计算 量比较大， 对数字信号处理器件的要求高， 许多算法采用现有的高速处理芯片 也无法实现。
基于上述原因， 并结合开关多波束和自适应天线阵技术， 本发明的目的在 于提供一种用于无线通信系统的阵列接收机中在数字基带空域匹配滤波方 法， 这种方法具有算法简便、 计算量较小的特点， 因此， 实现的硬件成本低、 系统性能佳。
本发明的另一个目的在于提供一种无线通信系统的阵列接收机，其中所使 用的数字基带空域匹配滤波模块釆用本发明上述的数字基带空域匹配滤波方 法， 具有算法简便、 计算量较小的特点， 因此， 实现的硬件成本低、 系统性能 佳。 发明内容
根据本发明的上述目的，本发明提供的数字基带空域匹配滤波方法包括下 列步骤：
(a)接收阵列数字信号；
(b)把所述阵列数字信号与蘭组加权向量进行加权运算， 得到 NB组数字波 束输出信号
(c)在相关时间内， 估算该数字波束输出信号的功率值
PSa (") = "A (") + )SBl (n - 1)¾ (" - 1)
并对所述波束输出信号 (w)进行功率归一化得到， 功率归一化数字波束 输出信号 » ^;
其中， 上标" *"表示复共轭运算， ^是常数， 0 <^< 1；
(d)将所述功率归一化数字波束输出信号 (； 和参考信号进行相关运 算， 估计反映每个波束中期望信号能量的相关系数 Co^; ; 其估计公式为：
CoeMk)
Figure imgf000005_0001
Coeff) (k) = Coeffi (k) + aCoeffi (k - 1)
其中， 上标" *"表示复共轭运算， s (n)为所述参考信号； N为采样点总数， 它与采样时间的乘积应小于相干时间， C0¾^ (it)表示第 k次相参积累得到的反映 波束 i中期望信号能量的相关系数， "是常数， 0.5 < α < 1 ;
(e)比较获得的反映各个波束中期望信号能量的相关系数 C¾e (yt)，得到最 大的相关系数 C0e 皿，输出与所述最大相关系数 <¾¾^皿相对应的数字波束输出 信号
根据本发明的另一目的， 本发明还提供一种阵列接收机， 包括- 由多个天线组成的天线阵列；
阵列数字信号生成模块， 与所述天线阵列相连， 用于将所述天线阵列接收 到的天线阵列模拟信号转换成阵列数字信号；
数字基带空域匹配滤波模块， 与所述阵列数字信号生成模块相连， 用于对 所述阵列数字信号生成模块的阵列数字信号在每个信道上形成一个或多个信 号波束；
数字接收机模块， 与所述数字基带空域匹配滤波模块相连， 用于接收所述 数字基带空域匹配滤波模块在每个信道上形成的一个或多个信号波束，在时域 上对所述信号波束进行合并；
所述数字基带空域匹配滤波模块， 包括：
数字基带空域匹配滤波器组， 接收阵列数字信号， 将所述阵列数字信号与
NB组加权向量进行加权运算， 输出 NB组数字波束输出信号 ；
分路器，把所述数字基带空域匹配滤波器组输出的所述 NB组数字波束输出 信号 SBi (ή)形成对应多个信道的数字波束；
多波束选择模块， 接收所述分路器输出的数字波束， 对所述数字波束输出 信号 进行如下的功率归一化，在相关时间内， 估算该数字波束输出信号的功率 值 (") ··
PSm (") = " A (") (") + (! - «P )¾, (" - 1)¾ (" - 1)
并对所述波束输出信号 («)进行功率归一化得到， 功率归一化数字波束 输出信号 ) ；
Figure imgf000006_0001
其中， 上标" *"表示复共轭运算， ^是常数， 0 <αρ< 1 ; 然后， 将所述功率 归一化数字波束输出信号 («)和参考信号进行相关运算， 估计反映每个波束 中期望信号能量的相关系数 C0¾T;， 并输出选择指令， 以选择与所述最大相关. 系数相对应的数字波束输出信号 所述估计公式为：
Figure imgf000006_0002
Coeffi (k) = Coeffi (k) + aCoeffi (k - 1)
其中， 上标" *"表示复共轭运算， s (n)为所述参考信号； N为采样点总数， 它与采样时间的乘积应小于相干时间， C0¾^ (A:)表示第 k次相参积累得到的反映 波束 i中期望信号能量的相关系数， "是常数， 0.5 < « < 1；
数字开关矩阵， 分别与所述分路器和所述多波速选择模块相连， 接收所述 多波束选择模块输出的选择指令，输出与最大的所述相关系数相对应的数字波 束输出信号。
如上所述， 由于本发明的数字基带空域匹配滤波方法采用了一种新的简便 的算法， 从而可算法硬件结构， 并同时提供较佳的系统性能。 附图概述
下面结合附图详细描述本发明的实施例。
图 1是本发明的阵列接收机的功能框图； 图 2示出了覆盖小区的波束示意图；
图 3示出了图 1中的数字基带空域匹配滤波模块 104的内部功能框图;
图 4示出了数字开关矩阵的原理图；
图 5示出了数字基带空域匹配滤波器组的基本原理图；
图 6示出了本发明的数字基带空域匹配滤波方法的流程图；
图 7示出了本发明应用于 CDMA系统时的仿真结果。 本发明的最佳实施方案
图 1是本发明的阵列接收机原理图。 该接收机主要包括天线阵列 102、 阵列 数字信号生成模块 103、 数字基带空域匹配滤波模块 104以及数字接收机模块 105。 阵列数字信号生成模块 103的主要作用是将从天线阵列 102接收到的天线 阵列模拟信号转换成可供数字处理的阵列数字信号， 其中包括有接收单元 (RX) 106及模拟到数字转换单元 (A/D) 107。 数字基带空域匹配滤波模块 104可 以对一个信道形成一个或多个数字波束， 例如对 C匪 A系统的一个码道， 数字基 带空域匹配滤波模块 104可以对相同码道的不同多径信号分别形成波束， 波束 形成后的信号送到数字接收机模块 105 (可以采用 Rake接收机）相应的手指， 由 数字接收机模块 105在时域对信号进行合并。 由于将模拟信号转换到数字信 号， 数字基带空域匹配滤波模块 104可以根据系统不同的要求灵活变化。
对于图 1所示的数字基带空域匹配滤波模块 104运用阵列处理算法对接收 的信号进行处理， 形成所谓波束， 使波束中输出的信号质量得到改善， 从而提 高整个接收机的性能。 因此， 数字基带空域匹配滤波模块 104所采用的阵列处 理方法以及实现的复杂度、 稳定性等直接关系到系统性能的好坏。 本发明提出 一种简单、 可靠、 易于实现的阵列接收方法。
图 1所示的阵列接收机中的阵列数字信号生成模块 103输出的数字信号可 以表示为：
X(t) = ^ ( )A, (t s(t - 7", ) + n(t) (1) 式中， 1，2， . . . ， L为多径数， a( ）是 M * l维向量， 表示与方向 0有 关的第 条多径信号在 M个天线阵元上的阵列响应， 是第 J条多径的波达方 向； (t)是第 _/条多径信号经历的衰落； s (t)是发射的期望信号， r,是第 J条多 径信号的时延； n (t)是阵列干扰和噪声信号；. 根据天线阵列 102的形式， 如均匀直线阵、 圆形阵等， 可以预先确定阵列对不 同方向上的信号具有的阵列响应 a( )(i = 1,2,...,NS)。 用 W; =aff( ）（上标 H表示共轭 转子运算)作为阵列各天线单元接收信号的加权， 相当于阵列在 方向形成波束， 对接收的阵列数字信号 X(t)进行空域滤波。为了兼顾空域匹配滤波的效果和系统数 字信号处理能力， NB的取值应使 aff ( )形成的波束与波束之间交会点 201的增益与 所有波束中的最大增益相差 _3~0 dB, 即图 2中的标线 202在- 3~0dB之间。
图 3示出了图 1中的空域匹配滤波模块 104的内部功能框图。如图 3所示，空域匹 配滤波模块 104包括有空域匹配滤波器组 401、分路器 402、多波束选择模块 403以及 数字开关矩阵 404。 空域匹配滤波器组 401内有 N组加权向量 W; =aff( )， 它接收 阵列数字信号生成模块 103输出的数字阵列信号 X(t)， 用 \^组加权向量 对数字 阵列信号 X(t)进行加权， 得到^个数字波束输出信号 (0， 其加权结果可以用下 式表示：
SBi (n) = W,. * X( = nH {Θ, )a(¾ )hd (t)S(t) + aH (θ{ )n(t) i=l, 2， 3...NBi (2) 空域匹配滤波器组 401 输出的^个数字波束输出信号 (t)通过分路器 402形成对应多个信道的数字波束， 即对于 Nc个信道， Ns个数字波束通过一 组分路器 402后共生成 Nc*N£个波束。这些波束分别传送给多波束选择模块 403 和数字开关矩阵 404。
在多波束选择模块 3中， 首先在相关时间内， 计算数字波束输出信号的功 率值
PSu (n) = apSBi (") (n) + (l-ap )SB/ (n - 1)¾ (n - 1) (3)
并对波束输出信号 («)进行功率归一化得到功率归一化数字波束输出信 号： ,("):
这样处理就可以在很大程度上降低强干扰信号对波束选择相关性的影 响。
在大多数通信系统中，发射的信号中都包含预先为接收端已知的训练序列、导 频符号等已知信息。 例如在 GSM系统中， 每个正常 (Norma) 1突发脉冲中有 26比特的 已知训练序列； 在 WCDMA系统中， DPCCH的每个时隙 (Slot)中有已知导频符号。接收 端可以利用这些已知信息直接或间接得到所谓参考信号 s(t)。 根据参考信号 s(t) 与数字波束输出信号 (0的相关性， 得到反映每个波束输出信号中期望信号能量 的相关系数 Coe^， 其具体运算方式如下：
W "=ι ( 5)
Coefft (k) = Coeff, (k) + aCoeff, (k - ΐ) ( 6 )
0^ ;(yt)表示第 k次相参积累得到的反映波束 i中期望信号能量的相关系 数， "是进行非相参积累时的遗忘因子， 它与小区内移动台的移动速度等因素 有关， 一般取 0 < « < 1。 从公式（4)中不难看出， NB个相关系数中最大的系数 €^χ所对应的方向^^即是与期望信号最接近的方向， 因此， 可以选择阵列 加权为 W = aff (^J的数字波束输出信号输出， 以获得最大增益。
对于图 3所示的空域匹配滤波模块来说， 多波束选择模块 403对由分路器 402输入的波束进行如公式（3)至（6)所示进行运算， 获得相关系数 <¾ς^(Λ：)。 然后， 如上所述， 比较获得的各个波束中期望号能量的相关系数， 选择其中最 大的相关系数 <¾¾^ ， 并向数字开关矩阵 404输出选择指令， 由数字开关矩阵
404输出与最大相关系数相对应的数字波束输出信号。
数字开关矩阵 404 的原理图可以参见图 4。 由于本技术人员的一般人员根据上 述要求都可以实现数字开关矩阵 404， 因此， 在此仅对图 4所示的数字开关矩阵 404 作一简单的介绍。
数字开关矩阵 404主要由波束数据线 601、 输出数据线 109、 数字开关 602 以及 开关控制信号线 603组成。 由图 4可以看出， 每条波束数据线 601可以通过开关 602 与对应不同信道的输出数据线 109相连， 多波束选择模块 403输出的控制信号 (前面 称为选择指令)通过开关控制信号线 603控制数字开关的闭合和断开。控制信号可以 用控制矩阵 表示。
Figure imgf000009_0001
可以设定矩阵 中的元素 为 1时表明数字开关闭合， 第 i个波束数据线 与第 j条输出数据线导通； ^为0时数字开关断开， 第 i个波束数据线与第 j 条数据线断开。 这样多波束选择模块 403可以方便地控制数字开关矩阵 404， 选择合适的波束到指定的输出端。 空域匹配滤波器组 401的基本原理如图 5所示。 它主要乘法器 502和加法器 503 组成。 进入空域匹配滤波器组 401的信号首先与预先设定的^个加权向量 (501) 通过乘法器 502进行复乘法运算，然后通过加法器 503完成复信号合并。合并后的信 号即为数字波束的输出。 由于空域匹配滤波器组 401主要由乘法器 502和加法器 503 组成， 所以它可以通过 FPGA方便地实现。 当然空域匹配滤波器组 401也可以通过采 用其他能够完成乘加运算的数字器件构成。
上面详细描述了本发明的阵列接收机的组成结构以及工作原理， 尤其着重 描述了本发明有独创之处的空域匹配滤波模块 104的构成以及工作原理。从上 面的揭示中可以看出，本发明的空域匹配滤波 块 104实现上利用了一种新的 空域匹配滤波方法， 在这种方法， 为获得数字波束， 利用了上述的公式（3)、 (4)、 ( 5 ) 和 （6 ) 获得反映每个波束中期望信号能量的相关系数， 然后根据 相关系数来选择确定数字波束。 为使该方法更为清楚， 图 6示出了本发明的空 域匹配滤波方法的流程图。
如图 6所示， 首先， 在步骤 S1接收阵列数字信号 X (t)， 对应于图 1来说， 即空域匹配滤波模块 104 从阵列数字信号生成模块 103 接收阵列数字信号
Χ (ΐ) ο
然后， 在步骤 S2， 把阵列数字信号与 NB组加权向量进行加权运算， 得到 Na组数字波束输出信号 该步骤 S2对应于图 3中空域匹配滤波器组 401 所完成的工作。
接着在步骤 S3，根据数字波束输出信号 (^ ，按照公式（3)和（4)估算每个 波束的平均功率值和功率归一化信号数字波柬输出 )， 具体的方法可以参 考上面的描述。
在步骤 S4， 估计反映每个波束中期望信号能量的相关系数 (¾ς ； 其估计 是依据公式（5)和（6)进行的， 具体的方法可以参考上面的描述。
如上面结合图 3所作的描述，然后的步骤是比较获得的反映各个波束中期 望信号能量的相关系数 C ¾ ;^:)， 得到最大的相关系数 ^^， 输出与最大相 关系数 相对应的数字波束输出信号 (参见步骤 S5)。
对应于这里描述的方法实施例中， 步骤 S3和 S4可以由图 3所示的空域匹 配滤波模块 104中的多波束选择模块 403来实现， 步骤 S5可以由数字开关矩 阵 404根据多波束选择模块 403的选择指令来实现。
本发明人针对 CDMA系统的上行链路用本发明提供的空域匹配滤波方法进 行了仿真实验， 其结果如图 7所示， 其多路径的环境参数可以参见下表。
Figure imgf000011_0001
可以看出， 对于 4阵元阵列接收机采用本发明， 在输入信噪比为 2.2dB左右时， 数字接收机模块输出信号的误码率可在 10— 3 附近，而单天线数字接收机模块需要天 线端输入信号的信噪比为 8.2dB左右时才可以使输出信号的误码率达到 10—3 。 显然 釆用本发明提供的方法， 系统性能得到显著提高。

Claims

权 利 要 求 书
1、 一种空域匹配滤波方法， 包括下列步骤：
(a)接收阵列数字信号；
(b)把所述阵列数字信号与 NB组加权向量进行加权运算， 得到 NB组数字波 束输出信号
(c)在相关时间内， 估算该数字波束输出信号的功率值 PSm (n)：
Ps3! (") = « A (" (") + )SB, (" - 1)¾ in -ΐ)
并对所述波束输出信号 («)进行功率归一化得到， 功率归一化数字波束 输出信号 »^ ; 其中， 上标" *"表示复共轭运算， 是常数， 0<^<1；
(d)将所述功率归一化数字波束输出信号 («)和参考信号进行相关运 算， 估计反映每个波束中期望信号能量的相关系数 ¾e .; 其估计公式为- )= |¾;(") (")||2
丄 n=\
Coefft (k) = Coeffi (k) + aCoejf, (k― 1)
其中， 上标" *"表示复共轭运算， s(n)为所述参考信号； N为釆样点总数， 它与采样时间的乘积应小于相干时间， Co¾^(:)表示第 k次相参积累得到的反映 波束 i中期望信号能量的相关系数， "是常数；
(e)比较获得的反映各个波束中期望信号能量的相关系数 CO^ ，得到最 大的相关系数 Co^T^，输出与所述最大相关系数 Coe g对应的数字波束输出 信号 0)。
2、 如权利要求 1所述的空域匹配滤波方法， 其特征在于， 所述遗忘因子 的值为 0.5<α<1。
3、 一种阵列接收机， 包括- 由多个天线组成的天线阵列；
阵列数字信号生成模块， 与所述天线阵列相连， 用于将所述天线阵列接收 到的天线阵列模拟信号转换成阵列数字信号；
数字基带空域匹配滤波模块， 与所述阵列数字信号生成模块相连， 用于对 所述阵列数字信号生成模块的阵列数字信号在每个信道上形成一个或多个信 号波束； 数字接收机模块， 与所述数字基带空域匹配滤波模块相连， 用于接收所述 数字基带空域匹配滤波模块在每个信道上形成的一个或多个信号波束， 在时域 上对所述信号波束进行合并；
所述数字基带空域匹配滤波模块， 包括：
数字基带空域匹配滤波器组， 接收阵列数字信号， 将所述阵列数字信号与
NB组加权向量进行加权运算， 输出 NB组数字波束输出信号
分路器，把所述数字基带空域匹配滤波器组输出的所述 NB组数字波束输出 信号 SBt (n)形成对应多个信道的数字波束；
多波束选择模块， 接收所述分路器输出的数字波束， 对所述数字波束输出 信号 S (w)进行如下的功率归一化，在相关时间内， 估算该数字波束输出信号的功率 值 ¾ )： . 并对所述波束输出信号 («)进行功率归一化得到， 功率归一化数字波束 输出信号 » 其中， 上标" *"表示复共轭运算， 是常数， 0< <1； 然后， 将所述功率 归一化数字波束输出信号 («)和参考信号进行相关运算， 估计反映每个波束 中期望信号能量的相关系数 C ^；， 并输出选择指令， 以选择与所述最大相关 系数相对应的数字波束输出信号 («); 所述估计公式为：
Figure imgf000013_0001
Coejfi (k) = Coeff. (k) + aCoejf, (k - 1)
其中， 上标" *"表示复共轭运算， s(n)为所述参考信号； N为采样点总数， 它与采样时间的乘积应小于相干时间， Coe^ t)表示第 k次相参积累得到的反映 波束 i中期望信号能量的相关系数， "是常数， 0.5<α<1;
数字开关矩阵， 分别与所述分路器和所述多波速选择模块相连， 接收所述 多波束选择模块输出的选择指令，输出与最大的所述相关系数相对应的数字波 束输出信号。
4、 如权利要求 2所述的阵列接收机， 其特征在于， 所述遗忘因子的值为 0·5<α<1。
PCT/CN2001/000693 2000-11-15 2001-05-08 Procede de filtrage spatial de mise en correspondance et recepteur en reseau dans un systeme de radiocommunication WO2002043256A1 (fr)

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AU2001263744A1 (en) 2002-06-03
EP1341315A1 (en) 2003-09-03
EP1341315A4 (en) 2007-10-03
CN1352490A (zh) 2002-06-05
CN1129237C (zh) 2003-11-26
ATE425597T1 (de) 2009-03-15
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