WO1999055124A1 - Procede permettant de faire fonctionner des lampes a decharge en atmosphere gazeuse, et configuration destinee a l'application d'un tel procede - Google Patents

Procede permettant de faire fonctionner des lampes a decharge en atmosphere gazeuse, et configuration destinee a l'application d'un tel procede Download PDF

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Publication number
WO1999055124A1
WO1999055124A1 PCT/EP1999/002556 EP9902556W WO9955124A1 WO 1999055124 A1 WO1999055124 A1 WO 1999055124A1 EP 9902556 W EP9902556 W EP 9902556W WO 9955124 A1 WO9955124 A1 WO 9955124A1
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WIPO (PCT)
Prior art keywords
voltage
transformer
lamp
arrangement according
arrangement
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PCT/EP1999/002556
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German (de)
English (en)
Inventor
Manfred Diez
Alfons Günther
Hubert Gauseweg
Johannes Rarbach
Original Assignee
Manfred Diez
Guenther Alfons
Hubert Gauseweg
Johannes Rarbach
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Application filed by Manfred Diez, Guenther Alfons, Hubert Gauseweg, Johannes Rarbach filed Critical Manfred Diez
Priority to EP99920680A priority Critical patent/EP1078558A1/fr
Publication of WO1999055124A1 publication Critical patent/WO1999055124A1/fr

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Classifications

    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
    • H05B41/282—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices
    • H05B41/2825—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices by means of a bridge converter in the final stage
    • H05B41/2828—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices by means of a bridge converter in the final stage using control circuits for the switching elements
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/36—Controlling
    • H05B41/38—Controlling the intensity of light
    • H05B41/39—Controlling the intensity of light continuously
    • H05B41/392—Controlling the intensity of light continuously using semiconductor devices, e.g. thyristor
    • H05B41/3921—Controlling the intensity of light continuously using semiconductor devices, e.g. thyristor with possibility of light intensity variations
    • H05B41/3925—Controlling the intensity of light continuously using semiconductor devices, e.g. thyristor with possibility of light intensity variations by frequency variation

Definitions

  • the invention relates to a method for operating a gas discharge lamp, and an arrangement for performing such a method.
  • gas discharge lamps e.g. those for the generation of UV radiation or for the generation of IR radiation.
  • UV lamps but always includes all gas discharge lamps.
  • this object is achieved by a method according to claim 1.
  • curve shapes of current and voltage at the radiator result, which result in a significantly lower distortion power and thus an improved lamp factor.
  • lamp factors of the order of 0.98 can be achieved with the invention.
  • the radiator also emits radiation during a percentage of each half wool of the supplied alternating voltage, since the current in the radiator - after a rapid increase - remains at a value, for example until the subsequent commutation, which enables radiation power to be emitted . Therefore, more radiation can be generated with the same radiator than with the known arrangements, i.e. the utilization of the radiator is improved and the radiation power of a given gas discharge radiator can be increased with a method according to the invention, e.g. a given size UV lamp.
  • the commutation follows the current maximum directly.
  • the gas in the radiator has become very hot due to the previous temperature rise, so that the temperature drop due to the commutation represents a cooling of the radiator, which prevents an uncontrolled rise in temperature and current in it, but a renewed rapid rise after commutation this 2
  • a preferred arrangement according to the invention is the subject of claim 15.
  • Fig. 1 is an overview circuit diagram of an arrangement according to the invention, in 3 which various voltage and current forms are entered for clarification,
  • FIG. 2 shows a schematic representation of a stray field transformer, as can be used above all at mains frequency
  • FIG. 3 shows a section, seen along the line III-III of FIG. 2,
  • FIG. 6 shows a variant of FIG. 1,
  • FIG. 7 shows a variant of FIG. 1,
  • FIG. 8 shows a variant of FIG. 1, which shows how a plurality of lamps 26, 26 'can be connected there,
  • Fig. 9 shows a variant of Fig. 1, which the use of a
  • FIG. 10 is a block diagram to explain a preferred control arrangement for operating a radiator 26 with different powers
  • 11A and 11B is a schematic representation of the primary
  • Fig. 13 is a schematic representation of a preferred design of a 4
  • Stray field transformer which is particularly suitable for arrangements according to the invention which are to be operated at higher operating frequencies
  • 15 is a block diagram to explain a preferred arrangement for power control using digital components
  • FIG. 16 is a flow chart for explaining FIG. 15;
  • FIG. 17 shows a supplementary flow chart with a subroutine for calculating the lamp factor
  • 19 is a block diagram for explaining an influencing of the inverter 30 via a PWM actuator
  • 21 is a block diagram for explaining an influence of the inverter 30 via a block controller
  • 25 is a circuit diagram of a preferred embodiment of a pulse generator for driving an IGBT transistor, 26A-C curves for explaining Fig. 25,
  • 29 shows the output voltage of the converter shown in FIG. 28, at a frequency of 400 Hz and measured on the secondary side of the transformer 446,
  • FIG. 30 shows a schematic illustration of a calculation process in the design of the transformer 446
  • Fig. 31 representations of secondary voltage (top) and secondary current
  • Fig. 32 is a representation analogous to Fig. 31, but with the lamp
  • 35 shows a measurement curve 471 which shows the power supplied to a UV lamp in the frequency range 400 to 1400 Hz
  • 36 shows a schematic illustration for generating a rectangular output voltage by means of a pulse-controlled inverter, 6
  • FIG. 37 is a schematic illustration for generating a sinusoidal output voltage by means of a pulse converter
  • 39 shows a circuit diagram of a further variant of an arrangement according to the invention with a three-phase inverter
  • Fig. 40 is a circuit diagram showing how a plurality of gas discharge lamps can be connected to the arrangement of Fig. 39, and
  • 41 shows a basic illustration of a three-phase transformer in a preferred design.
  • gas discharge lamps e.g. those for the generation of UV radiation or for the generation of IR radiation.
  • the following description refers to UV lamps, but always includes all gas discharge lamps.
  • the invention is preferably used in the case of mercury vapor discharge lamps. In these, the mercury is often doped with an additional element in order to obtain a specific frequency distribution of the radiated energy.
  • UV lamps such as those used in printers for drying printing inks, have considerable electrical power and are therefore operated from the AC network.
  • Apparent power S is defined as the product of the effective value u e ff of the voltage and the effective value i ⁇ ff of the current, i.e.
  • the active power P is defined as
  • the reactive power is defined as
  • a non-sinusoidal current usually flows in a UV lamp, since such a lamp has the character of a non-linear resistor. Due to its strong temperature dependence, it distorts voltage and current and therefore has the characteristics of a non-linear load.
  • a radiator does not have the character of an inductor or a capacitance, but that of a resistor, i.e. the zero crossings of current and voltage are the same, but because of the non-linear resistance of the radiator, a sinusoidal voltage also generates a non-sinusoidal current in it.
  • the distortion power is also a reactive power.
  • the distortion power S v arises solely from the harmonics of the non-sinusoidal quantities (current, voltage) on the radiator.
  • the reactive power which has to be supplied to it essentially consists of this distortion power S v .
  • This therefore essentially determines the power factor ⁇ , which is referred to in the UV industry as the lamp factor LF and which is between 0.7 and 0.9 in the arrangements customary today.
  • This lamp factor is sometimes - wrongly - also referred to as the cos ⁇ of the lamp. According to the inventors' knowledge, this relatively low lamp factor is a consequence of the fact that such radiators require a considerable distortion power S v which does not contribute to the active power P, so that the apparent power S is substantially greater than the active power P.
  • FIG. 1 shows an overview of a preferred arrangement 10 according to the invention.
  • This has a controllable rectifier 12, which is shown here as a fully controlled three-phase rectifier in a so-called B6 circuit, which is fed from a three-phase network with the phases R, S, T.
  • a fully controlled rectifier it is also possible to use a semi-controlled rectifier, as shown in FIG. 6, or a single-phase, fully or semi-controlled rectifier bridge circuit of known design for smaller powers.
  • the rectifier 12 can also be connected to the three-phase network R, S, T via a transformer (not shown).
  • a disadvantage of the use of rectifier circuits according to FIG. 1 or 6 is the type of network load, ie the current from the network to the rectifier 12 is not sinusoidal. This is because a current can only flow through the rectifier 12 if its output voltage is higher than the voltage across the capacitor which is connected to it. 9
  • a sinusoidal network load can be achieved by using a so-called switching power supply instead of the rectifier 12.
  • switching power supplies are e.g. sold by AEG under the name AC2000.
  • they allow the use of relatively small smoothing capacitors (see capacitors 20, 22 in FIG. 1 or capacitor 150 in FIG. 9). This is very advantageous if the output voltage of such a switching power supply is to be regulated, since this results in short dead times in the control circuit.
  • the use of a switching power supply represents a preferred solution. Since the components are known per se, a switching power supply is not shown separately.
  • a positive line 14 and a negative line 16 are connected to the output of the rectifier 12, which together form a direct current intermediate circuit 18, at which a variable direct voltage U is applied during operation, which is shown symbolically at 19. If the voltage on the three-phase network e.g. 400 V, the voltage U has a maximum value of 560 V. This voltage is smoothed by two series-connected capacitors 20, 22 of the same size, which form an artificial zero point 24 between them.
  • the capacitors 20, 22 have relatively high capacitance values of e.g. 1500 ⁇ F each, since they have to deliver high currents for the operation of a gas discharge lamp in the form of a UV lamp 26, which e.g. outputs a power of 3.2 kW at an operating voltage ("burning voltage") of 600 V. In addition, they may have to compensate for inductive loads (long connecting lines) via which the UV lamp 26 is supplied with current.
  • an inverter half bridge 30 is connected to the DC intermediate circuit 18. This has an upper IGBT transistor 32, which is connected to the positive line 14 with its collector and with its 1 0
  • An arrangement 42 is used to control the bases of the two IGBT transistors 32, 36.
  • This contains an ignition module for each of the two transistors 32, 36 (cf. FIG. 25), and also the necessary drive amplifiers for these ignition modules.
  • the frequency of the inverter 30, which will usually be in the range between 50 and 800 Hz, is predetermined by an oscillator 44. In the exemplary embodiment, it is approximately 400 Hz. For the reasons stated at the outset, it is more advantageous to use a higher frequency, since the temperature of the gas in the lamp can then be kept within a narrower temperature band, which, according to the inventors' knowledge, leads to the distortion power V s reduced, thereby improving the lamp factor LF.
  • IGBT e.g. the type FF150R12 KF22FN from EUPEC
  • the type must be selected according to the lamp wattage.
  • an ignition module e.g. (from Motorola) type MC33153D or MC33153P can be used.
  • a suitable circuit for such an ignition module is given by way of example in FIG. 25.
  • a transformer is required to generate this high voltage.
  • a stray field transformer 46 is used, as will be explained in more detail below with reference to FIGS. 2 and 3 or from FIG. 13.
  • the primary winding 48 of this transformer 46 is connected between the artificial zero point 24 and the node 34.
  • the voltage U / 2 is approximately at the artificial zero point 24. If the IGBT 32 is controlled by its ignition module, a current flows from the positive line 14 via the primary winding 48 to the artificial zero point 24, the capacitor 20 being partially discharged and 1 1 the capacitor 22 is charged, so that the potential at point 24 becomes somewhat more positive.
  • the transistor 32 is then blocked by its ignition module (in the arrangement 42), and after a predetermined time delay (closed period) of, for example, 100 ⁇ s, the transistor 36 is switched on, whereupon a current flows from point 24 via primary winding 48 to negative line 16 .
  • This current charges the capacitor 20 and discharges the capacitor 22, so that the potential at point 24 drops somewhat.
  • the potential at the point 24 changes only slightly, and an approximately rectangular primary voltage u p is obtained overall at the primary winding, as is shown schematically at 50 in FIG. 1.
  • This primary voltage generates in the secondary winding 52 of the stray field transformer 46 a secondary voltage u s , which is shown schematically at 54, for the state in which the UV lamp 26 has reached its operating voltage, for example 600 V. In this state the lamp has 26 reaches its operating temperature and delivers its full power. (At the start, there are different curve shapes and different voltages, since at the start the lamp 26 must first be ignited and heated, which may require a higher voltage. This also results in other current shapes.)
  • the voltage u s shown at 54 is an essentially rectangular voltage, but due to the properties of the stray field transformer 46 used, it has a somewhat delayed rise, for example in the manner of an e-function.
  • This voltage u s generates a lamp current i s in the lamp 26, which is shown schematically at 58. As already explained, this is the current after the operating temperature of the lamp has been reached. This current has an approximately E-shaped rise 60 at the beginning of a half-wave, and then an approximately linear rise up to a maximum 64 in the region 62, at which the current commutates (by switching the current direction in the primary winding 48) becomes.
  • the negative half wave has the same form as the positive half wave, cf. the representation at 58 and in the following Fig. 5b. 1 2
  • An arrangement 66 is used to control the controllable rectifier 12.
  • This contains the ignition modules for the phase control of the thyristors of the controllable rectifier 12, furthermore the drive amplifiers for these ignition modules, and an arrangement for synchronizing the bridge control with the frequency of the three-phase network R, S, T ( via connecting lines 68).
  • Controllable rectifiers of this type, their ignition modules and drive amplifiers are commercially available components. They are therefore not described in detail.
  • the voltage U at the intermediate circuit 18 can be controlled by means of a signal 69 which is fed to an input 70 of the arrangement 66.
  • the signal 69 is generated by a control or regulating arrangement 72, which in the simplest case can be designed as a setting potentiometer with which a signal 69 is generated by manual adjustment.
  • the arrangement 72 is a control device which can be constructed differently depending on the requirements. This is explained in more detail below.
  • the current i s on this secondary side is detected by a current transformer 74 on the secondary side of the transformer 46 and converted into an effective value i ef t for this current in an arrangement 76.
  • an arrangement is preferably used which delivers the true RMS value of this current, the so-called TRUE RMS.
  • TRUE RMS the true RMS value of this current
  • ICs which also provide a true RMS value for current forms that deviate significantly from the sinusoidal form.
  • An example of such an IC can be given: True RMS to DC Converter AD736 from Analog Devices.
  • the voltage u s on the secondary side of the transformer 46 is fed to an arrangement 80 by means of a voltage converter 78, which delivers the real effective value of this voltage at its output, ie u e t f .
  • An example of the arrangement 80 can be given: True RMS to DC converter AD736 from Analog Devices.
  • the effective values for current and voltage are fed to the arrangement 72 and processed further therein. Possibly.
  • the arrangement 42 can also control the arrangement 42, 1 3 the time during which the transistors 32, 36 are each conductive. Examples are given below.
  • Fig. 2 shows - as an example - a typical design of a stray field transformer 46.
  • This has an approximately rectangular magnetic core which is made up of individual parts, namely two straight longitudinal legs 90, 92, each of which has a coil of the primary winding 48 and a coil of the secondary winding 52 wears. As shown, these coils are each at a distance D from one another in order to enable the formation of a stray field and the installation of a stray field bridge between the coils of the primary winding 48 and the secondary winding 52.
  • the connections of the primary winding 48 are labeled 48a, 48b and the connections of the secondary winding 52 are labeled 52a, 52b.
  • FIG. 13 shows a second, preferred embodiment 46 'of a stray field transformer which, according to current knowledge, is particularly suitable for higher frequencies. This variant will be described below.
  • the parts 90, 92, 94, 96 are constructed from transformer sheet metal, preferably as so-called module elements.
  • Fig. 3 indicates 98 such sheets in the usual way. They typically have a thickness of 0.35 mm and are insulated from one another. When operating at 50 Hz, sheet thicknesses of 0.5 mm are also used
  • Module elements are constructed from individual sheets 98. These sheets are under 1 4 high mechanical pressure glued together and therefore form a compact block. This block is provided with a ground surface where another block adjoins it. For example, the yokes 94 and 96 are ground on their two longitudinal ends so that there is no annoying air gap at the transition to the adjacent leg 90 (left) or 92 (right), and the leg 90, 92 are at the relevant interfaces, for example surface 100 , also ground complementary.
  • the module elements are held close together by screw bolts (not shown) in order to keep the air gaps at the interfaces as small as possible, or to give them a defined size.
  • So-called stray field plates 102 are arranged within the distance D between the longitudinal legs 90, 92 in the manner shown. They are insulated at the ends by means of an insulating layer 104, 106 and thus clamped between the two longitudinal legs 90, 92. Their number determines the properties of the stray field transformer 46. This number is determined experimentally, corresponding to the values of the UV lamp 26.
  • the two coils of the primary winding 48 mainly generate a magnetic flux 108 through the longitudinal legs 90, 92 and the two yokes 94, 96.
  • This flux 108 causes the desired high ignition voltage at the two coils of the secondary winding 52 before ignition.
  • the lamp 26 When the lamp 26 ignites, it acts almost like a short circuit and the transformer 46 is loaded. This creates a very large magnetic excitation x wi in the primary winding 48, which counteracts the excitation I 2 x W 2 which is almost the same size in the secondary winding 52. Only the difference between these two excitations forms the common magnetic flux in the iron path. A considerable flow 110, driven by these large magnetic excitations, dodges the counter pressure and closes itself by bypassing the secondary coils 52 through the stray plates 102 1 5 weakened, so that the secondary voltage between the connections 52a, 52b becomes correspondingly smaller after ignition.
  • a UV lamp 26 When a UV lamp 26 is switched on, it is cold and it is first heated for a period of one or more minutes by the current flowing through it until it has reached its operating temperature. At this operating temperature, it then works with a voltage which is referred to as the operating voltage and which is usually substantially lower than the ignition voltage. The difference between ignition and burning voltage is a consequence of the construction of this special type of transformer.
  • FIGS. 4a to 4c show the case according to DE-UM 9 304 274 that the stray field transformer 46 is operated with a sinusoidal primary voltage u p , that is to say, for example, that a sinusoidal AC voltage of 400 V is connected between the connections 48a, 48b.
  • the representation according to FIGS. 4a to 4c thus corresponds to the prior art.
  • 4a and 4b are oscillograms.
  • Figure 4c is a schematic representation that is not based on measurements but on theoretical considerations.
  • the transformer 46 initially works in a similar manner to idling when the voltage rises, that is to say mainly the flux 108 and thus a voltage curve 1 14 on the secondary side, which corresponds approximately to the open circuit voltage of the transformer 46, that is to say the ignition voltage already explained.
  • the lamp 26 begins to conduct more, ie its resistance decreases, and the magnetic flux in the transformer 46 therefore increases greatly, so that from this point in time an increasing magnetic flux 110 flows through the stray plates 102 and the secondary voltage u s is therefore reduced.
  • This then essentially follows a sinusoidal voltage curve 118, the amplitude of which is substantially lower than the amplitude of the voltage curve 114.
  • the voltage curve 118 is also referred to as the operating voltage of the lamp. 1 6
  • region 120 the stray flux through the stray plates 102 increases continuously, and this is therefore a transition region that acts like a continuous current limitation for the lamp 26.
  • this current limitation in conjunction with the non-linear characteristic of the lamp 26, results in a curve of the lamp current which deviates very greatly from the shape of the voltage u s supplied. This can be seen in particular at the locations 122, 124, where the current i s indentations has, so that the lamp 26 its actual lamp power L (Fig. 4c) is substantially only between these points 122, 124 outputs, which greatly in Fig. 4c is shown schematically.
  • FIG. 4c also shows the amplitude L ma ma of the maximum lamp power and an amplitude L 70% of 70% of the maximum lamp power.
  • the ratio of the time T 0% , during which the lamp power L is at least 70%, to the total time T I0 tai of a half-wave is approximately 40%, ie the lamp 26 gives only for approximately 40% Total time from their peak performance.
  • the power decreases sharply after the maximum of the sinus, which presumably leads to a strong cooling of the lamp and a corresponding increase in its resistance.
  • the stray field transformer 46 when it supplies a UV lamp 26 with current, acts as a current limit for this lamp, specifically as a current limit with an extremely short response time, that is to say with extremely high operational reliability.
  • a stray field transformer 46 is therefore preferably used in the present invention, but other solutions are also possible, as will be explained below.
  • a stray field transformer also has the advantage of providing increased ignition voltage for lamp 26, but trials have shown that in many cases with an arrangement as shown in FIG. 1, the ignition voltage need not be much higher 1 7 as the operating voltage, so that in principle a "normal" transformer without stray field would also suffice, the secondary winding of which is connected in series with a choke, as shown below in FIG. 7 or FIG. 40.
  • the core of the transformer including the two windings 48, 52, can be designed correspondingly smaller, so that there is a smaller size than when operating at 50 Hz.
  • the invention can also be carried out with a lower or higher frequency than 400 Hz.
  • the transformer 46 Since, at a frequency of 400 Hz, higher magnetic reversal losses occur in the module elements 90, 92, 94, 96, it has proven to be advantageous to design the transformer 46 for a relatively low magnetic flux density of 0.3 to 0.5 T, while at Module elements, and operation at 50 Hz, a magnetic flux density in the range between 1, 2 and 1, 8 T is used, depending on the quality of the sheets 98, which are used in the module elements 90, 92, 94, 96. If the transformer 46 is forced cooled, e.g. a higher magnetic flux density is also possible by means of a blower, but in most cases a transformer is preferred in which no additional cooling is required. The design naturally also depends on the maximum permissible ambient temperature of the transformer and on the height above sea level at which the transformer 46 is operated. The corresponding rules for the design of stray field transformers are known to the person skilled in the art and therefore need not be set out here.
  • 5a to 5c show the same values for an arrangement according to the invention according to FIG. 1 as are shown for the prior art in FIGS. 4a to 4c.
  • the primary voltage u p of the transformer 46 has an essentially rectangular profile. This course is therefore not shown again in FIG. 5.
  • the transformer 46 does not transmit all the harmonics of a rectangular primary voltage, so that the secondary voltage u s has the shape 1 8 as shown in Fig. 5a.
  • a switchover (commutation) of the transistors 32, 36 takes place at the time t 1, ie the previously conductive transistor is blocked, and after a short switching pause (closed period) of, for example, 100 ⁇ s, the other transistor is controlled to be conductive.
  • this results in an approximately e-shaped rise in the secondary voltage u s , which reaches its positive maximum at about time t 2 and maintains it until time t 3 , at which a new commutation takes place in the inverter 30.
  • the secondary current i s changes between the times ti and t 2 , that is to say in the region 60 of FIG. 1, likewise in the manner of an e-function, and then increases approximately between the times t 2 and t 3 linearly, so that it reaches its maximum 64 at the time of commutation t 3 , at which the lamp 26 also reaches the maximum instantaneous value of its lamp power L.
  • the lamp power L decreases very quickly to zero in a region 61, which brings about a — necessary — reduction in the temperature of the gas in the lamp 26.
  • the lamp power then increases again rapidly in a region 63 and approximately after an e-function, and then increases approximately linearly in a region 65 to a maximum 67 where commutation takes place.
  • FIGS. 5a and b are oscillograms, but not FIG. 5c, which shows only a schematic representation, since such rapid courses are very difficult to measure.
  • the maximum lamp power is designated Lm ax , and a line L 70% with 70% of the maximum light output is also entered.
  • the ratio of the time period T 7 0 % to the time T to tai of a half-wave is approximately 65 to 75%, which results in better utilization of the lamp and a correspondingly better lamp factor LF.
  • the same lamp 26 with the invention enables a higher lamp wattage than with a prior art arrangement.
  • the lamp power can be varied over a wide range, 1 9 because the lowest output of a lamp is limited by the fact that it cools down too much due to a lack of energy supply and therefore goes out; if the maximum lamp power is increased, this automatically means an enlargement of the area in which the lamp power can be changed, because this extends this area upwards.
  • the lamp factor LF is defined according to equations (1) and (5) as
  • the lamp factor LF is generally between 0.7 and 0.9. In the invention, it can be increased up to a value of approximately 0.98, which is very advantageous for various reasons.
  • FIGS. 4a and 5a are not extremely dissimilar, and yet in FIG. 5b there is a completely different form of lamp current i s than in FIG. 4b, and a greatly improved lamp factor.
  • the temperature in the lamp presumably follows the form of the current with a very slight delay, since the gas in the lamp can store practically no energy and the energy storage capacity of the lamp bulb is also very low. Furthermore, according to the inventors' knowledge, it appears essential that the lamp 26 should always be operated in a narrow temperature range, i.e. the lamp temperature should fluctuate as little as possible, since then the resistance of the lamp fluctuates as little as possible and only slight transient changes in resistance occur.
  • the lamp power L increases approximately linearly in the region 62, ie this is in principle an unstable situation because the conductivity of the lamp 26 is getting better and better here. If the lamp 26 were constantly connected to a constant DC voltage, it would because of the constantly increasing 2 0
  • This part of the current curve (area 62) must therefore be interrupted at regular intervals by a short cooling phase, and this occurs during commutation, where the lamp current i s goes through zero and consequently the power supply to the lamp 26 is briefly interrupted.
  • Fig. 6 shows as a variant the use of a semi-controlled three-phase bridge rectifier 12 '.
  • diodes 13 In the lower bridge branches there are diodes 13, and in the upper bridge branches thyristors 15, which are controlled by the ignition pulse generators in the arrangement 66.
  • the remaining part of the arrangement corresponds to FIG. 1 and is therefore not shown. 6 has a simpler structure, but reacts less quickly to changes in the signal at input 70.
  • FIG. 7 shows a variant of the stray field transformer 46 of FIG. 1.
  • a standard transformer 130 (without a stray field bridge) is used here in its place, the lamp 26 being connected in series with a choke 132.
  • the choke 132 prevents an uncontrolled current rise in the lamp 26 and therefore acts as a current limitation.
  • the solution according to FIG. 1 with the stray field transformer 46 is preferred, however, since it is more flexible and requires fewer components.
  • Fig. 8 shows the connection of two UV lamps 26, 26 'to the same arrangement 10, that is to the same DC intermediate circuit 18.
  • Each lamp has its own stray field transformer 46 and 46', which is designed according to the needs of this lamp, and it has its own inverter 30 or 30 ', which can be constructed identically.
  • the parts of the inverter 30 ' are therefore labeled with the same reference numerals, but with a trailing apostrophe, e.g. IGBT transistor 32 '.
  • the primary winding of the transformer 46 is between the point 24 and the 2 1
  • the primary winding of transformer 46 ' is connected between point 24 and node 34', i.e. point 24 serves as an artificial zero point for both inverters 30 and 30 '.
  • the inverter 30 is controlled by the arrangement 42 and the oscillator 44, which specifies the frequency, e.g. 400 Hz.
  • the inverter 30 ' is controlled by an arrangement 42', and also by the oscillator 44.
  • the capacitance of the capacitors 20, 22, which together form a capacitor arrangement 136 must be adapted to the power requirements of the two lamps 26 and 26 ' .
  • FIG. 9 shows a section of an arrangement 10 in which a full-bridge circuit is used as the inverter 30 ".
  • This is constructed as a so-called H-bridge from four IGBT transistors, namely two upper transistors 140, 142 and two lower transistors 144, 146. Zu Swinging diodes 140 ', 142', 144 ', 146' are connected in parallel to these transistors.
  • the transistor 140 has its collector connected to the positive line 14 and its emitter to the connection 48a of the primary winding 48 of the stray field transformer 46, to which the collector of the transistor 144 is also connected, the emitter of which is connected to the negative line 16.
  • the transistor 142 is connected with its collector to the positive line 14 and with its emitter to the connection 48b of the primary winding 48, to which the collector of the transistor 146 is also connected, the emitter of which is connected to the negative line 16.
  • An arrangement 42 " which contains the ignition modules (cf. FIG. 25) for these transistors and which controls their clock, e.g. 400 Hz, received by an oscillator 44.
  • the capacitor arrangement 136 ′ here contains only a single capacitor 150, for example an electrolytic capacitor with 1000 ⁇ F.
  • the size of this capacitor 150 2 2 mainly depends on three factors: a) The structure of the rectifier 12. The harmonics of this rectifier must be effectively filtered out in order to obtain the most rectangular possible voltage on the primary winding 48 of the stray field transformer 46. For example, a switching power supply needs a smaller capacitor. b) The power of the lamp 26 and its operating frequency. A lamp 26 of high power needs a larger capacitor 150 than a lamp of low power, since the lamp current is largely supplied from the capacitor 150, particularly in the region 62 (FIG. 5b). c) The type of scheme.
  • the capacitor 150 must not become too large, since otherwise the dead time of the controller becomes too long, ie it then reacts too slowly. If the control is not performed via the DC link voltage U, but at the inverter, this consideration is irrelevant.
  • the arrangement 42 “first switches on the two transistors 140 and 146, for example, so that a current flows from the plus line 14 through the transistor 140 to the primary winding 48 and from there through the transistor 146 to the minus line 16. After a half-wave, that is at 400 Hz after 1150 ⁇ s, the current through the transistors 140, 146 is interrupted by corresponding signals of the arrangement 42 ′′.
  • the transistors 142 and 144 are switched on by the arrangement 42 "(" commutation "), so that now a current from the positive line 14 through the transistor 142, the primary winding 48, and the transistor 144 to Minus line 16 flows, which is also interrupted after 1150 ⁇ s, which is followed by a rest period of 100 ⁇ s, in which none of the transistors conducts, and the cycle described repeats itself continuously with the frequency of 400 Hz, for example.
  • FIG. 9 The remaining parts of the arrangement 10 are not shown in FIG. 9. For the sake of brevity, reference is made to FIG. 1.
  • each lamp has its own stray field transformer (not shown), and its primary winding is connected to the points 48a, 48b of the circuit according to FIG. 9, that is to say in parallel with the primary winding 48 of the transformer 46.
  • FIG. 10 shows a so-called current regulation for the lamp 26.
  • the same or equivalent parts as in the previous figures are designated with the same reference numerals as there and usually not described again.
  • the variants described, e.g. 6 to 9, can be used in Fig. 10 in the same manner.
  • the controllable rectifier 12 is supplied with current from an AC or three-phase network 11 and in turn feeds the DC intermediate circuit 18 to which the inverter 30 is connected, which is controlled by the arrangement 42 and the oscillator 44 in order to generate a rectangular AC voltage , e.g. at 400 Hz.
  • the lamp 26 is connected to the inverter 30 via the stray field transformer 46 and the current transformer 74.
  • the arrangement 76 converts the current detected by the current transformer 74 into a true rms current value i e ff (TRUE RMS), and this is fed to the controller 72, as is a current setpoint from the setpoint generator 84.
  • the difference between the Value i eff and the set current setpoint, and thus the controllable rectifier 12 and thus the DC voltage U on the intermediate circuit 18 is controlled via the control arrangement 66. So if the lamp current 2 4 becomes too high, the voltage U at the intermediate circuit 18 is reduced, and if the lamp current is too low, this voltage is increased. It is therefore an amplitude control.
  • 11 A and 11 B show the basic principle of amplitude control, which has proven to be very advantageous in the present case.
  • 11A shows the voltage u p on the primary side of the stray field transformer 46. It is a square-wave voltage with a frequency of, for example, 400 Hz, and a change in the voltage U at the intermediate circuit 18 causes a change in the amplitude of this voltage, as indicated by the arrows 154 indicated.
  • the voltage u p has the amplitude U / 2
  • in an inverter according to FIG. 9 it has the amplitude U.
  • the amplitude of the voltage u s on the secondary side of the stray field transformer 46 also changes, as shown in FIG. 11B, ie if the voltage U at the intermediate circuit 18 increases, the amplitude of u s also increases accordingly.
  • FIG. 12A shows (for the arrangement according to FIG. 1) measured values for a reduced primary voltage u p (with a frequency of 400 Hz), and FIG. 12B shows the corresponding current values for the lamp current i s , which also has a reduced amplitude.
  • the current curve according to FIG. 12B (with a low current i s ) can be compared with the current curve according to FIG. 5B, which shows a higher current.
  • the same favorable form of the current i s results, that is to say a rapid e-shaped increase 60 and subsequently a slower linear increase 62 up to the current maximum 64, but with a reduced amplitude of the current.
  • the spikes 156 of the lamp current visible in FIG. 12B result from the switching processes during the commutation of the current.
  • FIG. 13 shows a variant 46 "of the stray field transformer 46 of FIGS. 2 and 3.
  • This also has a rectangular iron core which is constructed in the manner shown from two legs 160, 164 and two yokes 162, 166.
  • the plates of these module parts have a thickness of 0.35 mm and are glued together under pressure, see Fig. 3.
  • These module parts are clamped together by screw bolts, not shown, so that small and.
  • screw bolts not shown, so that small and.
  • At the connection points between the individual module parts, for example at the parting line 168 result in precisely defined air gaps. (Of course, a build-up of the iron core from individual sheets is not excluded.)
  • the primary winding here has four coils 170, 172, 174, 176, and the secondary winding has two coils 178, 180, which are fastened in the middle of the two legs 160, 164.
  • the two lower primary coils 174, 176 are at a distance D1 from the secondary coils 178, 180, and at this distance D1 stray plates 182 are fastened between the legs 160, 164 with the interposition of two insulators 184, 186.
  • the two upper primary coils 170, 172 are at a distance D2 from the secondary coils 178, 180, and at this distance D2 stray plates 188 are fastened between the legs 160, 164 with the interposition of insulators 190, 192.
  • the number of scattering sheets 182, 188 is determined by tests, namely by operating the transformer 46 ′′ with the associated UV lamp 26, as already described in FIG. 2.
  • connections of the primary winding are designated 48a and 48b, the electrical connections V 2 6 of these coils from FIG. 13 result directly.
  • the connections of the secondary winding are designated 52a and 52b in FIG. 2, and their electrical connection also results from FIG. 13.
  • the upper primary coils 170, 172 When loaded, the upper primary coils 170, 172 create a leakage flux 192 through the upper leakage sheets 188, and the lower primary coils 174, 176 generate a leakage flux 190 through the lower leakage sheets 182, so that the voltage across the coils 178, 180 of the secondary winding is already on explained burning voltage drops.
  • the short-circuit voltage U k rises so much with increasing frequency that the power required to operate the lamp 26 can no longer be transmitted by the transformer 46. If the primary winding is divided into a larger number of coils 170, 172, 174, 176 according to FIG. 13, the short-circuit voltage U k is reduced, with the result that the required power can be transmitted by the stray field transformer 46 ′′.
  • Short-circuit voltage U k in transformer construction means the following: If the secondary winding in FIG. 13 is short-circuited between connections 52a, 52b, a short-circuit current flows there. 2 7
  • the magnetic flux density is also designed here to be about 0.3 ... 0.5 T if the transformer 46 "is operated, for example, at 400 Hz.
  • the explanations given in connection with FIGS. 4a to c apply analogously Way for the stray field transformer 46 "according to FIG. 13.
  • FIG. 14 shows the block diagram of a so-called power control, ie the actual value S of the apparent electrical power absorbed by the lamp 26 is calculated and regulated to a value S s (reference variable) which is specified on a setpoint generator 84.
  • the regulation takes place by changing the voltage U at the DC intermediate circuit 18, as already described in FIG. 1. Parts that are the same or have the same effect as in the previous figures are designated with the same reference symbols as there and are usually not described again.
  • the real effective value i e ff of the lamp current is determined by means of the IC 76, and the real effective value u e f of the voltage on the lamp 26 is determined by means of the IC 80, which have already been described in FIG. 1.
  • the apparent power S which is absorbed by the lamp 26 is then determined from these values at 198, according to the formula (1) already given at the beginning
  • the apparent power S is compared with a target value S s by the target value generator 84, and the difference between the apparent power and target value is amplified in a control amplifier 200 and fed to the arrangement 66, which controls the controllable rectifier 12 and thus the voltage U on the intermediate circuit 18 .
  • voltage regulation is also possible, in which case only the voltage u e ff on the lamp 26 is regulated to a desired value.
  • This variant is not shown, but in principle has the same structure as FIG. 10, except that it regulates the value u e ff. It can be useful for countries in which the mains voltage fluctuates strongly and you still want to keep the voltage at the lamp 26 as constant as possible so that the light output of the lamp 26 fluctuates as little as possible. The same result is obtained with a power control, but with more effort.
  • the arrangement according to FIG. 15 contains a microcontroller 204 with a processor 206 and two A / D converters 208, 210.
  • a microcontroller 204 with a processor 206 and two A / D converters 208, 210.
  • ACN-1 / 1V1.06n type from Wilke Technology is suitable. 2 9
  • the curve shape of the lamp current i s is digitized into n values during a period, as shown schematically in FIG. 18B, and these n values are stored in a field.
  • the A / D converter 210 also digitizes the curve shape u s of the voltage across the lamp 26 into n values, as shown in FIG. 18A, and these are also stored in the field. 16 shows this sequence in step S212 of the flow diagram shown there.
  • the label for step S212 reads: "Load n voltage values u n (t) and n current values i n (t) into a field during the period T.”
  • step S214 These values are then converted in processor 206 according to the formula in step S214 into the active power on lamp 26, specifically by digital integration over a period T, that is to say at 400 Hz over 2.5 ms.
  • a setpoint P s for the active power is digitally specified here via the encoder 84, and a comparison is made in step S216 to determine whether the active power is too small, correct or too large.
  • step S218 If the active power P is too low, the voltage U at the intermediate circuit 18 is increased by the control device 66 in step S218.
  • step S 220 If the active power P has the correct value, the voltage U is left unchanged in step S 220.
  • step S222 If the active power P is too high, the voltage U is reduced accordingly in step S222.
  • S224 denotes further steps (subroutine) which can be used if necessary and which are shown in FIG. 17. However, these steps are only an additional luxury for the regulation.
  • step S226 the effective current is calculated in step S226 according to the formula given there, and the effective voltage on the lamp 26 in step S228. 3 0
  • step S230 the apparent power S is calculated from these effective values, and in step S232 the lamp factor LF as the quotient of the active power P and the apparent power S, cf. Equation (5).
  • the lamp factor LF is digitally displayed to allow an accurate adjustment of the arrangement, e.g. the correct setting of the cooling for the lamp 26.
  • the cooling must be reduced with decreasing output of the lamp 26, the lamp factor LF being able to be kept at an optimal value by adapting the cooling exactly to the requirements of the lamp.
  • FIG. 19 shows a further variant of the invention.
  • the voltage U at the intermediate circuit 18 is kept constant during operation.
  • this voltage U must be slowly increased in this case, and therefore a so-called start-up control 240 is provided here, which controls the control unit 66 in such a way that the voltage, starting from zero, is increased as a function of time up to its operating value U.
  • the capacitors 20, 22 (FIG. 1) and the capacitor 150 (FIG. 9) act like a short circuit when switched on and could be destroyed by excessive charging currents.
  • the delayed rise is caused by the controller 72, which usually has an integrating behavior, so that its output signal 69 slowly rises.
  • signal 69 With manual control, in Fig. 1, starting from zero, signal 69 must be slowly increased to the desired value.
  • the output signal of the arrangement 72 (comparison of actual and setpoint value) is fed to a PWM controller 244 via a control amplifier 242.
  • the "voltage blocks" of the rectangular primary voltage u p are chopped into a plurality of shorter individual pulses 246.
  • FIG. 20A only two individual pulses 246 are shown per half-wave in order to keep the drawing clear, but in reality these are, for example, ten or twenty individual pulses 246 per half-wave. 3 1
  • the pulse duty factor of these pulses 246 is changed by the PWM control, that is to say the ratio of T / TG, which is 100% in FIG. 11A and only approximately 70% in FIG. 20A, with decreasing Duty cycle the pause T p between adjacent pulses 246 becomes longer, but the time TG remains essentially unchanged.
  • this variation of the duty cycle changes the amplitude A of the secondary voltage u s accordingly, that is, a high amplitude A is obtained with a large duty cycle, and a low amplitude with a small duty cycle, cf. arrows 250 in Fig. 20B. 19 thus acts similarly to an arrangement in which the voltage U at the intermediate circuit 18 is changed, and the power of the lamp 26 can also be varied within wide limits in this way.
  • the control or regulation of the intermediate circuit voltage U must be regarded as the better solution.
  • FIG. 21 shows a further variant of the invention, similar to the variant according to FIG. 19. Also in FIG. 21, the voltage U at the intermediate circuit 18 is kept constant during operation, and the output voltage of the inverter 30 is influenced by keeping its frequency constant (an additional variation of the frequency is also not excluded), but the length of the voltage block of a half wave at the output of the inverter 30 is varied. This is done with a so-called block controller 256, the function of which can be seen in FIGS. 22A and B. Parts which are the same or have the same effect as in the previous figures are designated with the same reference numerals as there, and are usually not described again.
  • 22A shows the primary voltage u p of the stray field transformer 46.
  • a short voltage block is labeled 258 and is highlighted in gray.
  • a medium length voltage block is designated 260, and the full output of inverter 30 is obtained with voltage block 262, which is the length of a full wave.
  • the short voltage block 258 corresponds to a low secondary voltage 3 2
  • the middle voltage block 260 a middle secondary voltage 260', and the long voltage block 262 the highest secondary voltage 262 '.
  • the disadvantage here is that the shape of the secondary voltage u s changes with decreasing voltage in the direction of a sine curve, which can lead to a poorer lamp factor LF.
  • a lowering of the secondary voltage u s by about 20% is generally sufficient for the lowest output of a lamp, the problems in practice are less serious than would be deduced from the - highly schematic - illustration according to FIG. 22.
  • FIG. 23 shows a further variant of the invention, which is constructed very similarly to the variant according to FIG. 21, and which, in contrast to FIG. 21, could be called inverse block control. Parts that are the same or have the same effect as in the previous figures are designated with the same reference symbols as there and are usually not described again.
  • the signal from the control amplifier 242 is fed to an arrangement 266, the function of which can be seen in FIGS. 24A and B.
  • FIG. 24A the shape of a rectangular half-wave of the primary voltage u p is made narrower or wider, that is to say the width of a voltage block is changed, in FIG. 24A a voltage block 268 is split into two blocks 272a, 272b from its center 270, so to speak. which are highlighted in gray in FIG. 24A.
  • the outer limits 274, 276 of a half wave thus remain unchanged, and with it the abrupt transition at 280 from a positive to a negative half wave of the primary voltage u p or vice versa, which is very favorable in order to obtain a good lamp factor LF.
  • 24B shows the shape of the secondary voltage u s .
  • 282 denotes the full voltage that results when the gap 284 between the two blocks 272a, 272b becomes zero, and with increasing size of this gap 284, the secondary voltage u s decreases in the direction of an arrow 286.
  • the main advantage of the arrangement according to FIG. 23 is that you have a favorable form of the secondary voltage u s and therefore also one for the 3 3
  • Lamp factor LF receives a favorable form of the lamp current i s (not shown in FIG. 24).
  • FIG. 25 shows an arrangement 290 for driving an IGBT transistor schematically indicated at 291, e.g. of transistor 32 or 36 of FIG. 1. Such an arrangement is required for each transistor 32 and 36, and therefore a second arrangement 290 'is shown schematically in FIG.
  • the arrangement 290 'then emits pulses which are in phase opposition to the pulses of the arrangement 290.
  • the output signals 292 of the oscillator 44 (eg 400 Hz) are inverted by a NAND gate 296 and fed to the upper terminal 298 of a changeover switch 300, the lower terminal 302 of which is connected to the output of a NAND gate 304 and the input of the arrangement 290 "
  • the outputs of the NAND gate 296 are fed to both inputs of the NAND gate 304.
  • the gate 304 thus functions as an inverter.
  • a signal is fed to a NAND gate 306, which is push-pull (inverse) to the signal 292. If the switch 300 is in the lower position, as shown, the NAND gate 306 receives a signal which is in synchronism with the signal 292.
  • assembly 290 may receive the common mode signal, and in assembly 290 'switch 300 must then be set up to receive a push-pull signal. In this way, only one of the transistors 32, 36 of FIG. 1 can always be conductive. If both transistors 32, 36 in FIG. 1 were to be conductive at the same time, a short circuit between lines 14 and 16 of the intermediate circuit 18 would clearly appear.
  • An IC 308 which serves as a monostable multivibrator, is used to generate them 3 4 whose input 310 the output signal of the NAND gate 306 is supplied.
  • the duration of the closed season is determined by a potentiometer 312 and one
  • Capacitor 314. Preferred values for the components are:
  • Capacitor 314 ... 10 nF.
  • the time constant of the potentiometer 312 and the capacitor 314 determine the above-mentioned gentle period of approximately 100 ⁇ s, which can be set by means of the potentiometer 312.
  • the monostable multivibrator 308 is triggered on each negative edge of the signal at input 310 (change of signal from high to low). As a result, the voltage U3 16 at its output 316 briefly goes to zero during the gentle period 317 (FIG. 26B).
  • This voltage U 3 i 6 is fed to two series-connected NAND gates 318, 320, which together act as an AND gate, and this AND gate 318, 320 is also fed the signal 292 (at the output of the NAND gate 304) 26C, so that these signals are combined to form a signal U 322 , the pulses of which - compared to the signal 292 - are somewhat shortened at the beginning. This causes the desired delayed switch-on of the IGBT transistor in question.
  • the signal U 322 is fed to the upper input 327 of a NAND gate 328, the lower input 330 of which can either be supplied with a constant voltage Vcc (+ 5 V) or a PWM signal from the arrangement 244 (FIG. 19). This can be set with a changeover switch 332.
  • the NAND gate 328 is followed by a NAND gate 334, which together form an AND gate, the output signal of which is fed to an npn transistor 336, which is connected in series with the LED 338 of an optocoupler 340.
  • LED 338 outputs a light signal to an optotransistor 342, the collector of which is connected to a positive line 344 (+ 15 V).
  • Its emitter is connected to the bases of an npn transistor 346 and a pnp transistor 348 (both Darlington transistors), likewise via a resistor 350 with a negative line 352 (- 15 V), to which the resistor 354 connects 3 5
  • the collector of the transistor 348 is connected, the emitter of which is connected via a node 356 to the emitter of the transistor 346, the collector of which is connected to the positive line 344 via a resistor 358.
  • the gate G of an IGBT transistor is connected to node 356, e.g. of transistor 32 or 36 of FIG. 1.
  • the emitter of this transistor is connected to ground 360, that is to the potential 0 V.
  • gate G momentarily receives a high positive current of e.g. 5 A, which turns on this transistor, and if, conversely, the transistor 348 conducts, the gate G briefly receives a strong negative current of e.g. also 5 A, which turns this transistor off.
  • the center tap 372 is connected to the positive line 14.
  • Terminal 374 of winding 48 ' is connected to the anode of a thyristor 378 and the cathode of a diode 380, the anode of which is connected to the negative line 16.
  • the right connection 376 of the winding 48 ' is connected to the anode of a thyristor 382 and the cathode of a diode 384, the anode of which is connected to the negative line 16.
  • the cathodes of the thyristors 378, 382 are connected to one another and to the negative line 16 via an inductor (choke) 386.
  • the control electrodes of thyristors 378, 382 receive corresponding control pulses from a pulse generator 42 which is controlled by an oscillator 44, e.g. at 600 Hz.
  • thyristor 378 when the thyristor 378 is turned on, a current flows from the center tap 372 to the left terminal 374 of the primary winding 48 '.
  • thyristor 382 When thyristor 382 subsequently becomes conductive, thyristor 378 is blocked via a capacitor 388 between the anodes of thyristors 378, 382 and a current flows from center tap 372 to right connection 376 of primary winding 48 '.
  • alternating current pulses flow through it, which induce a corresponding secondary voltage in the secondary winding 52.
  • FIG. 28 shows a variant of FIG. 9. Parts which are the same or have the same effect as in FIG. 9 are designated with the same reference numerals as there and are usually not described again.
  • the primary winding 448 of a stray field transformer 446 lies here between points 48a and 48b of the full bridge circuit of the inverter 30 ".
  • a gas discharge lamp in the form of a mercury vapor UV lamp 26 is connected to its secondary winding 452.
  • the voltage at the secondary winding 452 is denoted by u s , and the current through the secondary winding 452 with i s Current and voltage can be measured as shown in Fig. 1.
  • the voltage across the primary winding 448 is labeled u p .
  • a control arrangement 442 which can control these transistors with a variable frequency f.
  • This frequency f is supplied by a controllable oscillator 444 which can supply a frequency in the frequency range f 1 to f2, for example between 400 and 1400 Hz, or between 300 and 1500 Hz.
  • the pulses which control the transistors of the inverter 30 " are composed in the manner of PWM signals from individual pulses with a higher PWM frequency f3, for example with a frequency f3 in the range from 12 to 14 kHz.
  • the inverter 30 " is also referred to as a pulse inverter.
  • pulses 462 which are used to control the transistors 140, 146. These are composed of a plurality of individual pulses 464, between which there are pulse pauses. These pulse pauses can be of different sizes, e.g. to generate a sinusoidal voltage us or a more rectangular voltage us. This is explained below with reference to FIGS. 36 and 37.
  • a rather rectangular voltage cf. FIG. 36
  • a sinusoidal voltage cf. FIG. 37
  • a sinusoidal voltage can also be used in the context of the present invention, as a result of which the lamp factor then becomes somewhat lower. (In practice, an exactly rectangular voltage is not obtained, but rather a trapezoidal voltage, see FIG. 32A.)
  • pulses 466 which are used to control transistors 142, 144 and which are composed in the same way of individual PWM pulses.
  • the individual PWM pulses are generated in a PWM part 443 of the control unit 442.
  • Your duty cycle TV (see FIG. 36, below) is preferably a function of the voltage U at the DC intermediate circuit 14, 16. This voltage is supplied as the actual value Uj S t to the PWM part 443, which also receives a setpoint voltage from a setpoint generator 445 Usoii is fed. If the voltage U drops, the duty cycle TV is increased accordingly, and if the voltage U rises, it is decreased, so that the amplitude Z (FIG. 36) of the voltage u p , which of the
  • Primary winding 448 is supplied, is not or not significantly influenced by fluctuations in the intermediate circuit voltage U. 3 8
  • a pulse 462 has the (variable) length T / 2. It is followed by a pulse 466, likewise with the length T / 2, then another pulse 462, etc., as is familiar to the person skilled in the art in this field.
  • a control device 460 is used to control the oscillator 444, e.g. a potentiometer that can be used to adjust the frequency of oscillator 444.
  • the oscillator 444 can be controlled via an input 470 by means of an analog or digital signal. For example, can be switched directly between the frequencies f 1 and f2 by means of a switch 472.
  • the transformer 446 is dimensioned in such a way that it supplies the lamp 26 with its maximum power, or with the maximum desired power, at the lower frequency f1, that is to say usually with 100% of the possible lamp power N
  • _ emitted by the lamp 26 can thus be in a range of e.g. 100% to e.g. 15% can be adjusted continuously, with the light output being very stable in all areas. It is of great advantage that the light output can be precisely adjusted in this way.
  • UV lamp 26 (mercury vapor discharge lamp), which was measured using an arrangement according to the invention.
  • this lamp is supplied with its nominal power of 3620 VA, i.e. 100%.
  • this power has dropped to 1830 VA, i.e. to 50.5%.
  • At 1100 Hz it fell to 1100 VA, i.e. to 30.4%, and at 1400 Hz to 680 VA, i.e. to 18.8%. So you get a range of 100% to 20% of the lamp power with very stable operation of the lamp.
  • the frequency see Fig. 32A
  • the power of the lamp 26 can e.g. be switched back and forth between 100% and 18.8%.
  • the voltage across lamp 26 is determined by the transformation ratio of transformer 446 and therefore changes little when the 3 9
  • _ is changed in this way. Rather, essentially only the current i s through the lamp 26 changes.
  • the lamp 26 can also be operated stably at the frequency f2 with 18.8% of its light output, and that within a very short time - by switching from f1 to f2 or vice versa - the light output can be switched between 100% and 18.8% or vice versa is possible. This proves to be very advantageous in industrial application and in many cases leads to high savings in electrical energy.
  • FIG. 29 shows the voltage u s on the secondary winding 452 when the transformer 446 is operated at idle at a frequency f 1 of 400 Hz, that is to say without a lamp 26.
  • the time T / 2 in this case is 1250 ⁇ s.
  • the maximum amplitude of the voltage u s here is +1080 V or -1100 V. It corresponds to the ignition voltage of the lamp 26, which is higher than the so-called operating voltage of this lamp.
  • the voltage at the output of the transformer 446 drops to the "operating voltage", the maximum amplitude of which is approximately 640 V here.
  • the half-waves 478, 480 are each composed of approximately 16 PWM pulses 482. Since no current i s flows during idling, these pulses 482 are not filtered by the inductance of the secondary winding 452, which is advantageous for the ignition, since this results in a large number of ignition pulses with a high du / dt, which the lamp 26 immediately after switching on and ignite safely.
  • FIG. 30 shows an example of a preferred way of dimensioning the transformer 446 in such a way that the desired large setting range for the light output of the lamp 26 is obtained.
  • transformers can be calculated in many ways, which never lead to exactly identical results.
  • a computer program 490 is used as an example, the data of which are given below.
  • the desired core material eg "sheet 1”.
  • the primary voltage u p This is the effective value of the primary voltage, which essentially corresponds to the maximum amplitude Z (FIG. 36) for a rectangular primary voltage up. If a sine voltage is used, the effective value of the sine voltage is used in the usual way.
  • the frequency f 1 that is the lower frequency at which the transformer 446 is to deliver the full power.
  • the desired peak value Bmax of the magnetic flux density B This value depends on the cooling of the transformer, since at higher frequencies the iron losses increase sharply and the transformer 446 must therefore be sufficiently cooled to avoid inadmissible temperatures.
  • a lower Bmax means lower iron losses.
  • Bmax at 400 Hz will usually be between 0.3 and 0.6 T.
  • the magnetic flux density Bmax automatically decreases with increasing frequency during operation, which results in a corresponding reduction in the power transmitted by the transformer 446. If Bmax is selected too high, this leads to a very strong heating of the transformer iron, which is taken into account by the program mentioned.
  • the program calculates a suitable cross section AF ⁇ of the transformer core at 506, and the winding data of the transformer at 508.
  • the program 490 used in the example below is "Dry Transformer Program 1 / M. mid-section For Transformers with a Power of 4 kVA or Higher"
  • This program allows the quick calculation of dry-type transformers, but in any case the special expertise of a specialist in transformer construction is very useful in order to obtain optimal results. It should also be noted that the manufacturers of transformer sheets offer a calculation service and that they have special calculation documents for their own sheets and module cores.
  • Dry transformers are usually sufficient for the performance of UV lamps, but the use of transformers with oil cooling is of course not excluded.
  • a single-phase dry transformer is assumed as example 1, with the following data:
  • Insulation class F cos ⁇ 1, 0
  • Core type 1 Rectangular core, as shown in Fig. 13
  • Sheet type 3 module cores, as shown in FIG. 13
  • Sheet 1 Type M111-35N
  • This transformer is designed for a mercury vapor discharge lamp with an ignition voltage of 750 V and an operating voltage of 544 V.
  • the "Craciun" program then calculates an iron cross-section of 35.91 cm2 at 506. Since one works in transformer construction with standardized values for the core cross section and the transformer should be tightly dimensioned, one can use the next smaller standardized module core in this case, the 4 2 has a cross section of 30.2 cm ⁇ .
  • Such module cores are manufactured, for example, by
  • the program calculates a total of 184 turns for the primary winding at 508 and 388 turns for the secondary winding.
  • the primary winding is divided between the four coils 170, 172, 174, 176 (FIG. 13), each of which therefore receives 46 turns.
  • the secondary winding is divided between the two coils 178, 180 (FIG. 13), each of which receives 194 turns.
  • the exact setting of the number of stray plates 182, 188 (FIG. 13) is carried out when the transformer is loaded with its nominal load, that is, best with the UV lamp 26 which it is intended to operate.
  • the operating voltage e.g. here 544 V
  • the iron weight of the calculated transformer is 16 kg, the copper weight is 6.7 kg. Compared to a transformer for the same power at 50 Hz, the size and weight are greatly reduced.
  • a transformer 446 dimensioned in this way produces a power of 5.28 kVA on the secondary side at 400 Hz.
  • This power decreases with increasing frequency because the magnetic flux density B in the transformer core decreases with increasing frequency if the primary voltage u p is kept constant.
  • the voltage on the secondary side remains essentially constant since it does not depend on the magnetic flux density, but on the transmission ratio of the primary winding 448 to the secondary winding 452.
  • the voltage on the secondary side increases with increasing frequency something off.
  • a standard transformer (without a stray field device) to which the lamp 26 is connected via a series choke can also be used in FIG. 28.
  • a series choke 608, 610, 612 are shown in series with the lamps 26 ', 26 "and 26'", respectively. It also specifies how such a series choke is appropriately dimensioned.
  • the upper frequency be determined so that the lamp 26 consumes only a small amount of power - while still operating safely - and that the lower frequency is set so that the lamp at least approximates its maximum power.
  • the lamp power can be switched from the maximum to a very low power by changing the frequency f within a very short time, or vice versa.
  • Mains voltage fluctuations can be corrected in a converter according to the invention by means of the controller 443 in that the pulse duty factor of the PWM pulses 464 is automatically increased as the mains voltage decreases. Conversely, if the mains voltage becomes too high, the width of the pulses 464 is reduced by the controller 443, so that mains voltage fluctuations can be corrected extremely quickly.
  • lamp 26 is turned on.
  • the amplitude A of the ignition voltage in this lamp is, for example, approximately 820 V. Since the lamp 26 initially acts like a very low resistor after switching on, a high current i s flows with an amplitude of approximately 40 A according to FIG. 31 B, and the Voltage u s drops to a low value of, for example, 120 V, ie a large part of the magnetic flux flows through the stray plates 182, 188 (FIG. 13), and large losses occur in the stray field.
  • the transformer 446 behaves here like an ideal inductance, which is why the secondary current i s becomes triangular.
  • the lamp 26 After about two minutes, the lamp 26 has reached its operating temperature, and 4 4 the current i s then changes into the desired shape, as shown in FIG. 5b for the case in which a rectangular primary voltage is used. It should be pointed out that the ignition of the lamp takes place very quickly in an arrangement according to FIG. 28, since a high du / dt is generated by the converter, which causes a practically instantaneous ignition. As a result, the operating temperature is reached very quickly.
  • FIG. 32 shows the transition from a higher frequency (eg 700 Hz) to a low frequency f1 (eg 400 Hz).
  • FIG. 32A shows the secondary voltage u s, and it is seen that the frequency is switched from f2 to f1 within a very short period of time, and that the voltage U S has a trapezoidal shape.
  • the secondary current i s is only very low and has an amplitude below 5 A. After switching to the lower frequency, this amplitude rises to values above 10 A, and the light output of the lamp 26 thus increases very much quickly to a high value, e.g. 100%.
  • 33 and 34 show the secondary voltage at a frequency of 400 Hz 4 5 u s (each above) and the secondary current i s (each below) for different powers.
  • a variable resistor 530 (FIG. 28) was connected to the secondary winding 452, that is to say no mercury vapor discharge lamp 26.
  • the PWM pulses 482 are still clearly recognizable, i.e. the inductance of the transformer 446 brings about a large reduction in the amplitude of these pulses compared to FIG. 29 (idling), but not their suppression. 33A, the maximum amplitudes of the secondary voltage are +776 V and -784 V.
  • the effective value of the secondary voltage is 570 V and the effective value of the secondary current is 7.75 A, corresponding to an apparent power of 4.42 kVA.
  • the maximum amplitude of the voltage us in FIG. 34 is +740 V or - 760 V.
  • the transformer 446 can be designed for the lower frequency f 1 in such a way that the lamp 26 is operated there at full load and that the lamp power decreases with increasing frequency. 4 6
  • the voltage u s on the lamp 26 is determined by the transformation ratio of the transformer 446 and changes only slightly when the frequency is changed. Rather, only the current through lamp 26 essentially changes.
  • FIG. 36 shows what the PWM pulses 464 look like if the primary voltage up of the transformer 446 is to look approximately rectangular. 28 serves to control the inverter 30 ".
  • Each half-wave is a plurality of individual pulses 464, all with the same length t1 and with the same duty cycle TV, the definition of which is given in FIG. 36 below
  • Such an inverter 30 " is also referred to as a pulse inverter because it is driven by a plurality of pulses during each half-wave.
  • the pulse duty factor TV is reduced, ie all the pulses 446 become somewhat narrower, so that the amplitude Z of the voltage u p also remains unchanged.
  • FIG. 37 shows the pulse shape for a sinusoidal primary voltage u p , which leads to a lower lamp factor LF than a square-wave voltage, but can also be used in connection with FIG. 28.
  • the pulses 464 ' also have the same frequency here, for example 14 kHz, but they have different lengths within a half wave.
  • the outer pulses 550, 551 are namely narrow and the middle pulses 553 are wide, ie the middle pulses have a high duty cycle TV, and this decreases towards the outside. This is also called a "sinus evaluation".
  • the amplitude X of the primary voltage u p 4 7 can be increased by proportionally increasing the pulse duty factor of all pulses 464 ', for example by a factor of 0.05.
  • the voltage form according to FIG. 36 is preferred since it gives a better lamp factor.
  • a UV lamp 26 is located above the belt 560 and is supplied with energy by an arrangement 566.
  • the arrangement 566 corresponds to the arrangement according to FIG. 28.
  • Lacquered objects 570 are transported on the belt 560 and are to be dried by the lamp 26 by means of UV radiation.
  • a light barrier 572, 574 serves to switch the lamp 26 to full power when an object 570 is found on the belt 560.
  • the lamp 26 is switched back to a reduced power, e.g. of 15%, switched.
  • the invention provides enormous opportunities for saving energy.
  • FIG. 39 shows an inverter analogous to FIG. 28, but with a three-phase inverter 30 ′′ ', the three outputs of which are denoted by U, V and W.
  • the same or equivalent parts as in FIG. 28 are denoted by the same reference numerals as there and usually not described again.
  • the inverter 30 '" has the form of a known three-phase full-bridge circuit, which compared to FIG. 28 has a third branch with two 4 contains 8 additional transistors 141, 143, the connection point of which is designated 48c and to which freewheeling diodes 141 ', 143' are connected in anti-parallel fashion.
  • the transistors 141, 143 are controlled by the control arrangement 442.
  • the control pulses S are composed of individual pulses 464 with a frequency of, for example, 14 kHz, ie the inverter 30 '"is a pulse inverter.
  • the transformer 40 shows the connection of a three-phase transformer 600 to the inverter 30 '".
  • the transformer has a primary winding 602, which is connected here in a triangle, and this is connected to the outputs U, V, W of the inverter 30'".
  • the secondary winding 604 of the transformer 600 is here, for example, in a zigzag configuration.
  • This has the advantage that a secondary load, which lies between only one output and the star point labeled 606 (single-phase load), is distributed over two cores of the transformer 600.
  • This is due to the fact that the secondary winding on each core of the transformer 600 is divided into two equivalent parts, cf. Fig. 41, and that two parts of different cores are switched against each other.
  • a single-phase load on the secondary side is then always distributed over two cores, and a current can then always flow on the primary side, which eliminates the flooding of the secondary winding on each of these two cores.
  • the transformer 600 is the switching group Dz, here DzO.
  • a transformer with a star connection on the secondary side would be vector group Dy.
  • the switching groups Yy or Yz are e.g. defined in Bödefeld sequence, electrical machines, Springer-Verlag, Vienna, 1952, page 69.
  • the outputs of the secondary winding 604 are labeled u, v and w.
  • the three-phase transformer 600 is designed for the maximum power of the three radiators 26 ', 26 ", 26'", an example of which is given below. If the frequency f rises, the magnetic flux density in the transformer 600 and thus the power transmitted by it decrease, and consequently the power emitted by the radiators decreases with increasing frequency f, but the voltage at the radiators only decreases slightly.
  • two radiators can also be connected in parallel, so that one can also operate six radiators with the arrangement according to Fig. 40.
  • the series chokes 608, 610, 612 serve to limit the current through the radiators 26 to their nominal current.
  • each of the emitters 26 ', 26 “, 26'” has an output of 3.2 kW, with an ignition voltage of 750 V and an operating voltage of 544 V.
  • the transformer 600 a standard transformer (i.e. no stray field transformer), is designed for the ignition voltage, i.e. 750 V.
  • the chokes 608, 610, 612 should therefore each have an inductance of 12 mH in this example. 5 0
  • N (I x L) / (Bmax x A F ⁇ ) ... (12)
  • Sheet 4 (sheet of type V330)
  • Primary winding 124 turns per leg of the transformer, i.e. here 62 turns per coil
  • Secondary winding 272 turns per leg, so here 136 turns per coil
  • FIG. 41 shows an example of a design of a three-phase transformer 600 of this type. It has an iron core 620 with three legs 622, 624, 626, which are connected at the top by a cross yoke 628 and at the bottom by a cross yoke 630. Brackets 632 are attached below by means of screws 634. At the top are the electrical connections U, V, W for the primary winding 602, the connections u, v, w for the secondary winding 604, and the star point 606.
  • the leg 622 carries a primary coil 602a and 602b at the top and bottom and a secondary coil 604a in between.
  • the leg 624 carries a primary coil 602c and 602d at the top and bottom, and a secondary coil 604b between them.
  • the leg 626 carries a primary coil 602e and 602f at the top and bottom and a secondary coil 604c in between.
  • the two primary coils 602 of each leg here have 62 turns each, that is a total of 124 turns.
  • the secondary coils 604 are implemented here in a zigzag connection, i.e. each secondary coil 604 consists of two separate windings with 136 turns each, which are connected according to FIG. 40.
  • the zigzag circuit has proven to be very advantageous, particularly in the case of radiators with a high ignition voltage, because it avoids an excessive increase in the ignition voltage in the radiator, which ignites last. With a normal secondary star connection (y), this radiator may experience a sharp rise in the ignition voltage.
  • the invention is suitable for all types of gas discharge lamps, but finds preferred application in mercury vapor discharge lamps with their high energy requirements.

Abstract

L'invention concerne un procédé permettant de faire fonctionner une lampe à décharge en atmosphère gazeuse, alimentée par intermittence avec des impulsions de courant via un transformateur, notamment un transformateur à champ de dispersion. Selon ce procédé, il se produit entre des impulsions de courant successives une commutation du courant. Après une commutation, une impulsion de courant augmente rapidement, par rapport à une valeur absolue, d'abord approximativement selon une fonction exponentielle puis elle reste, approximativement jusqu'à la prochaine commutation, à une valeur qui permette à la lampe de diffuser une puissance émise.
PCT/EP1999/002556 1998-04-18 1999-04-16 Procede permettant de faire fonctionner des lampes a decharge en atmosphere gazeuse, et configuration destinee a l'application d'un tel procede WO1999055124A1 (fr)

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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP2599207B1 (fr) * 2010-07-29 2020-03-11 Xylem Water Solutions Herford GmbH Convertisseur à deux commutateurs

Families Citing this family (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
DE10312549B3 (de) 2003-03-21 2004-08-26 Hüttinger Elektronik Gmbh + Co. Kg Gasentladungsprozess-Spannungsversorgungseinheit
DE102006007754A1 (de) * 2006-02-20 2007-08-23 Patent-Treuhand-Gesellschaft für elektrische Glühlampen mbH Schaltungsanordnung und Verfahren zum Betreiben einer Hochdruckentladungslampe

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GB2030388A (en) * 1978-09-05 1980-04-02 Thorn Electrical Ind Ltd Lamp drive circuits for cine film projectors or cameras
US4240009A (en) * 1978-02-27 1980-12-16 Paul Jon D Electronic ballast
EP0104264A1 (fr) * 1982-09-24 1984-04-04 White Castle System, Inc. Source de courant réglable pour lampes à décharge de gaz et pour lampes analogues
US4667132A (en) * 1986-03-03 1987-05-19 Dianalog Systems, Inc. Electronic transformer system for neon lamps
US4933605A (en) * 1987-06-12 1990-06-12 Etta Industries, Inc. Fluorescent dimming ballast utilizing a resonant sine wave power converter
EP0394966A2 (fr) * 1989-04-25 1990-10-31 Matsushita Electric Works, Ltd. Alimentation
EP0505947A2 (fr) * 1991-03-29 1992-09-30 Kabushiki Kaisha Sanyo Denki Seisakusho Transformateur pour lampes à décharge dans un gaz
DE4219958C1 (en) * 1992-06-18 1993-06-24 Trilux-Lenze Gmbh + Co Kg, 5760 Arnsberg, De Ballast circuit for discharge lamp - uses phase gate control to short out electrodes for interval in each half cycle, depending on brightness
EP0583841A2 (fr) * 1992-08-20 1994-02-23 Koninklijke Philips Electronics N.V. Alimentation de puissance à découpage
US5349270A (en) * 1991-09-04 1994-09-20 Patent-Treuhand-Gesellschaft F. Elektrische Gluehlampen Mbh Transformerless fluorescent lamp operating circuit, particularly for a compact fluorescent lamp, with phase-shifted inverter control
EP0633711A1 (fr) * 1992-03-25 1995-01-11 Toto Ltd. Regulateur de tension de lampe a decharge et appareil d'eclairage en couleur modulable mettant en uvre ce regulateur
US5384516A (en) * 1991-11-06 1995-01-24 Hitachi, Ltd. Information processing apparatus including a control circuit for controlling a liquid crystal display illumination based on whether illuminatio power is being supplied from an AC power source or from a battery
US5412288A (en) * 1993-12-15 1995-05-02 General Electric Company Amalgam support in an electrodeless fluorescent lamp
US5534755A (en) * 1990-08-17 1996-07-09 Hughes Aircraft Company System for providing a constant current to a fluorescent tube
DE19639873A1 (de) * 1995-10-05 1997-06-12 Int Rectifier Corp MOS-Gate-Treiberschaltung
WO1997021264A1 (fr) * 1995-12-05 1997-06-12 Swedeponic Holding Ab Alimentation electrique pour charges non lineaires

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US4240009A (en) * 1978-02-27 1980-12-16 Paul Jon D Electronic ballast
US4187450A (en) * 1978-03-09 1980-02-05 General Electric Company High frequency ballast transformer
GB2030388A (en) * 1978-09-05 1980-04-02 Thorn Electrical Ind Ltd Lamp drive circuits for cine film projectors or cameras
EP0104264A1 (fr) * 1982-09-24 1984-04-04 White Castle System, Inc. Source de courant réglable pour lampes à décharge de gaz et pour lampes analogues
US4667132A (en) * 1986-03-03 1987-05-19 Dianalog Systems, Inc. Electronic transformer system for neon lamps
US4933605A (en) * 1987-06-12 1990-06-12 Etta Industries, Inc. Fluorescent dimming ballast utilizing a resonant sine wave power converter
EP0394966A2 (fr) * 1989-04-25 1990-10-31 Matsushita Electric Works, Ltd. Alimentation
US5534755A (en) * 1990-08-17 1996-07-09 Hughes Aircraft Company System for providing a constant current to a fluorescent tube
EP0505947A2 (fr) * 1991-03-29 1992-09-30 Kabushiki Kaisha Sanyo Denki Seisakusho Transformateur pour lampes à décharge dans un gaz
US5349270A (en) * 1991-09-04 1994-09-20 Patent-Treuhand-Gesellschaft F. Elektrische Gluehlampen Mbh Transformerless fluorescent lamp operating circuit, particularly for a compact fluorescent lamp, with phase-shifted inverter control
US5384516A (en) * 1991-11-06 1995-01-24 Hitachi, Ltd. Information processing apparatus including a control circuit for controlling a liquid crystal display illumination based on whether illuminatio power is being supplied from an AC power source or from a battery
EP0633711A1 (fr) * 1992-03-25 1995-01-11 Toto Ltd. Regulateur de tension de lampe a decharge et appareil d'eclairage en couleur modulable mettant en uvre ce regulateur
DE4219958C1 (en) * 1992-06-18 1993-06-24 Trilux-Lenze Gmbh + Co Kg, 5760 Arnsberg, De Ballast circuit for discharge lamp - uses phase gate control to short out electrodes for interval in each half cycle, depending on brightness
EP0583841A2 (fr) * 1992-08-20 1994-02-23 Koninklijke Philips Electronics N.V. Alimentation de puissance à découpage
US5412288A (en) * 1993-12-15 1995-05-02 General Electric Company Amalgam support in an electrodeless fluorescent lamp
DE19639873A1 (de) * 1995-10-05 1997-06-12 Int Rectifier Corp MOS-Gate-Treiberschaltung
WO1997021264A1 (fr) * 1995-12-05 1997-06-12 Swedeponic Holding Ab Alimentation electrique pour charges non lineaires

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP2599207B1 (fr) * 2010-07-29 2020-03-11 Xylem Water Solutions Herford GmbH Convertisseur à deux commutateurs

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DE19917180A1 (de) 1999-10-21

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