US7190596B2 - Resonant converter with phase controlled switching - Google Patents

Resonant converter with phase controlled switching Download PDF

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US7190596B2
US7190596B2 US10/514,173 US51417304A US7190596B2 US 7190596 B2 US7190596 B2 US 7190596B2 US 51417304 A US51417304 A US 51417304A US 7190596 B2 US7190596 B2 US 7190596B2
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switch
current
voltage
circuit
switches
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US20050226010A1 (en
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Thomas Dürbaum
Georg Sauerländer
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TP Vision Holding BV
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Koninklijke Philips Electronics NV
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/083—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the ignition at the zero crossing of the voltage or the current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H02M3/1588—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load comprising at least one synchronous rectifier element
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
    • H05B41/282—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices
    • H05B41/2825—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices by means of a bridge converter in the final stage
    • H05B41/2828—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices by means of a bridge converter in the final stage using control circuits for the switching elements
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
    • H05B41/282—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices
    • H05B41/285—Arrangements for protecting lamps or circuits against abnormal operating conditions
    • H05B41/2851—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the circuit against abnormal operating conditions
    • H05B41/2856—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the circuit against abnormal operating conditions against internal abnormal circuit conditions
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B20/00—Energy efficient lighting technologies, e.g. halogen lamps or gas discharge lamps
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the invention relates to a circuit arrangement for a converter with switches for chopping a DC voltage U 1 into a chopped DC voltage U 3 , comprising control means for controlling the switch-on times of the switches, in which switch-on times of the switches alternate with each other and are separated from each other by dead-time phases, and circuit elements comprising a resonant circuit having at least one capacitor and at least one coil for converting the chopped DC voltage U 3 into an output voltage U 2 .
  • the invention also relates to a method of switching the switches in such a circuit arrangement.
  • Converters with resonant circuit elements are generally used for supplying a load connected to its output with a DC voltage or a DC current. They may be used in versatile ways and are particularly used for operating gas discharge lamps, display screens, audio apparatuses, televisions, video recorders or in the automobile technique. Resonant converters may be formed as AC/AC, DC/AC, AC/DC or DC/DC converters.
  • a DC voltage U 1 is chopped into a chopped DC voltage U 3 by means of a bridge circuit or half bridge circuit consisting of switches.
  • the chopped DC voltage is applied to circuit means having at least one inductive and one capacitive resonant circuit element, i.e. with inductive and capacitive reactance components, such that an AC current flows in the circuit means in the case of operation proximate to the resonance frequency, which DC current is approximately sinusoidal, for example, in circuit means having exactly one inductive and exactly one capacitive resonant circuit element.
  • This AC current is then rectified and smoothed to an output voltage U 2 as a power supply voltage for a load connected to the converter.
  • the load is operated with an AC voltage, thus without any rectifiers, (for example, induction heating).
  • Resonant converters can be operated at high switching frequencies so that, in comparison with the possible power supply, apparatuses having a small volume and a light weight can be built.
  • ZVS operation Zinc Voltage Switching
  • MOSFETs Metal-oxide-semiconductor-semiconductor-semiconductor-semiconductor-semiconductor-semiconductor-semiconductor-semiconductor-semiconductor-semiconductor-semiconductor-semiconductor-semiconductor-s.
  • dead-time phases should be realized in which all switches of the relevant half bridge are switched off (i.e. in the non-conducting state).
  • the dead-time phases should be adapted to the area of operation of the converter.
  • Known control ICs such as, for example, the STR-Z4000 series of the firm of Allegro-Sanken or the controller IC MC 34067 of Motorola do not provide the possibility of automatic adaptation and can therefore ensure ZVS operation only within limited areas. Outside these areas, they can no longer ensure a reliable ZVS operation.
  • mismatching the dead-time phases and a resultant elimination of the ZVS operation the switching losses are increased, which may lead to destruction of the switches in extreme cases.
  • the switching of the switches at a capacitive load is to be particularly avoided.
  • a converter circuit arrangement of the type described in the opening paragraph is known from EP 0 430 358 A1.
  • the phase difference between the voltage applied to the circuitry and the current flowing in the circuit arrangement is indirectly monitored by monitoring the current flowing in the circuitry. In this mode of determining the converter load, it is detrimental that the phase difference determination is elaborate from a circuit technical point of view and that the measurements are subject to losses.
  • a circuit arrangement is known from DE 199 25 490 in which the voltage at one of the switches and possibly also the voltage gradient dI/dt is measured for determining the type of converter load in a dead-time phase before switching on, and is compared with a threshold value so as to control the dead times between the switch-on times of the switches.
  • This has the drawback that the operation of the converter at a capacitive load is only determined at the switch-on instant at the end of the dead time and thus not until after the load is already capacitive. In such a case, the switching cycle must be interrupted and re-initiated, which may cause disturbances.
  • control means comprise a comparator which compares a value dependent on the current I flowing in the resonant circuit with a threshold value, and in that the circuit means control the switching of the switches in dependence upon the comparison result.
  • the object is also solved in that means for determining the current gradient dI/dt of the current I flowing in the resonant circuit are provided, in that the control means comprise a comparator which compares a value dependent on the current gradient dI/dt with a threshold value, and the circuit means control the switching of the switches in dependence upon the comparison result.
  • a value dependent on the current I flowing in the resonant circuit is determined and compared with a threshold value during the switch-on phase of a switch;
  • the switches can be alternately switched off until the current I has reached the threshold value.
  • the object is also solved by means of a method in which:
  • a value dependent on the current gradient dI/dt of the current I flowing in the resonant circuit is determined and compared with a threshold value during the switch-on phase of a switch;
  • the switches can be alternately switched off until the current gradient has reached the threshold value.
  • An essential fundamental aspect of the invention is that, with the determination of the current I flowing in the resonant circuit and/or the current gradient dI/dt, it is possible at any instant to determine a value for the energy in the resonant circuit or the charge in the inductive mode of operation, and thus to determine at which instant a switch should be switched off at the latest in order that the switch-on process for the subsequent switch is not performed in the capacitive mode of operation.
  • the energy/charge remaining in the inductive mode of operation must be large enough when switching off the switches so as to ensure that the relevant subsequent switch is switched on again in the ZVS operation.
  • the circuit arrangement according to the invention provides the particular possibility of checking the switch-on release of a subsequent switch already before the start of the dead time and hence before the instantaneously active switch is switched off, so that the reliability of operation of the converter is significantly enhanced.
  • control means supply a signal at the latest for switching off the active (switched-on) switch, as soon as they have received a signal from the comparator indicating that the given value for the energy/charge remaining in the inductive mode has reached the threshold value, or the subsequent switch is prevented from being switched-on when the comparator has supplied a signal indicating that the value for the energy/charge remaining in the inductive mode of operation has fallen below the threshold value.
  • a switch is usually not switched on directly at the instant when the switch receives a switch-on signal, but when the switching process has been terminated after a given time delay.
  • the method may be optionally implemented in such a way that a switch can still be switched on when the current I has reached the first threshold value but has not yet exceeded or fallen below this value, or in such a way that the switch can no longer be switched on when the current I has reached the first threshold value.
  • the means for measuring the current I and/or the current gradient dI/dt may be arranged in the resonant circuit, at which a measuring point is sufficient. The means should then be suitable for processing both positive and negative measuring values and for forming values for negative measuring values.
  • the means for measuring the current I and/or the current gradient dI/dt may also be provided on the switches of a half bridge or on the switch pairs of a full bridge circuit. In this case, two measuring points are required, for which the means for measuring the current I should then be able to process only positive signals.
  • a resistor or a current transformer as well as current-sense FETs can be used in both cases as means for measuring the current I and/or the current gradient dI/dt.
  • the current I and/or the current gradient dI/dt can be comparatively easily tapped via its drain source voltage in a further preferred embodiment of the invention.
  • switch-off of the active switch can be triggered in a preferred embodiment of the method at the latest when the threshold value has been reached so that the switch-on process for the subsequent switch is timely initiated before the energy/charge available for recharging the capacitances falls below the at least required energy/charge in the inductive mode of operation.
  • a switch-on of the subsequent switch can be prevented after the value has fallen below the threshold value, and in each case, a switch-on of a switch can be prevented in the capacitive mode of operation.
  • the method according to the invention for switching the switches in a circuit arrangement is further preferably implemented in such a way that the threshold value is adjusted in dependence upon the input voltage U 1 , which is raised with an increasing input voltage U 1 . Since the required energy/charge for charging or discharging the parasitic and possibly external capacitance in the resonant circuit increases with an increasing input voltage U 1 , it will thus be possible to adapt the threshold value to the load condition.
  • FIG. 1 is a block diagram of a circuit arrangement including a resonant converter
  • FIG. 2 shows the circuit structure of a resonant converter according to the invention
  • FIG. 3 a shows time variations for an inductive load
  • FIG. 3 b shows time variations for a capacitive load
  • FIG. 3 c is a diagram for the remaining residual charge as a measure of the remaining energy in the case of an inductive load
  • FIG. 4 is a block diagram of a control circuit arrangement for controlling the switches
  • FIG. 5 is a block diagram of a control circuit
  • FIG. 6 shows a transfer function as a function of the frequency for a constant load resistance.
  • FIG. 1 shows a load-resonant converter, here a power supply unit, with a circuit block 1 for converting a DC input voltage U 1 into an output voltage U 2 , here a DC voltage, which is used for supplying a load represented by a circuit block 3 .
  • the input voltage U 1 is generated in the conventional manner for power supply units, for example, by rectifying an AC voltage of an AC voltage mains 4 by means of a circuit block 2 .
  • FIG. 2 shows in a more detailed way the essential elements of a converter as shown in FIG. 1 .
  • the DC input voltage U 1 is present at a half bridge of series-arranged switches S 1 and S 2 , which chop the DC voltage U 1 .
  • the switches S 1 and S 2 are MOSFET transistors in this case, which comprise body diodes D 1 and D 2 each being arranged anti-parallel to the corresponding switches S 1 and S 2 .
  • a capacitance Cp at which a chopped DC voltage U 3 decreases during operation of the converter 1 is arranged parallel to the switch S 2 .
  • the capacitance Cp need not necessarily be an external component but it may be exclusively the output capacitance C iss of the MOSFET transistors, which are present anyway.
  • the chopped DC voltage U 3 is applied to a circuit configuration 6 , which comprises resonant circuit elements and generates a DC output voltage U 2 .
  • the circuit configuration 6 comprises a capacitance Cr and an inductance Lr, arranged in series, as resonant circuit elements.
  • a rectifier arrangement 7 is arranged in the direction of the converter output, which rectifier arrangement rectifies a current I flowing through the resonant circuit elements Cr and Lr and, as usual, applies it to a smoothing capacitance C arranged at the output, from which smoothing capacitance the DC output voltage U 2 can be tapped.
  • the DC output voltage U 2 is present at a load R, which is shown as an ohmic resistor in this case.
  • the converter 1 may, however, also be used for supplying an AC voltage instead of a DC voltage. In such a case, rectification by a rectifier arrangement and a smoothing capacitor is not required and the output voltage would be equal to the AC voltage decreasing at the rectifier arrangement 7 in the embodiment shown in FIG. 2 .
  • a control unit 5 for controlling switching-on and switching-off of the switches S 1 , S 2 is provided.
  • an ohmic resistor W 1 , W 2 Arranged in series with each of the switches S 1 , S 2 is an ohmic resistor W 1 , W 2 from which the decreasing voltage U W1 or U W2 is tapped by the control unit 5 .
  • an ohmic resistor arranged in the resonant circuit may also be used.
  • current transformers or current-sense FETs for determining the current I flowing during a dead-time phase and/or the current gradient dI/dt can be used, whose signals are then tapped accordingly by the control unit 5 .
  • the upper one of the three diagrams shown in FIG. 3 a represents the difference
  • the control voltages U G1 and U G2 serving as control signals for controlling the switches S 1 and S 2 represent corresponding gate voltages of the MOSFET transistors. There is a dead-time phase, denoted by T tot , whenever the difference between the values of the control voltages is zero.
  • T on (S 1 ) When the switch S 1 is transferred to the switched-on state by applying a suitable control voltage U G1 to the control input of the switch, the time intervals denoted by T on (S 1 ) are present. In these time intervals, the control voltage U G2 is zero so that the switch S 2 is switched off. The time intervals in which the switch S 2 is switched on and the switch S 1 is switched off are denoted by T on (S 2 ). During these time intervals, a control voltage U G2 , which is different from zero and causes the switch S 2 to switch on is applied to the control input of the switch S 2 . Within these time intervals, the control voltage U G1 is zero.
  • the central diagram in FIG. 3 a shows the variation with respect to time of the current flowing through the resonant circuit elements C r and L r .
  • the lower diagram in FIG. 3 a shows the variation with respect to time of the voltage U 3 present at the parasitic capacitance C p .
  • the time axes of the three diagrams with the time t are plotted on the same scale.
  • the switch S 2 is finally switched off so that no current can flow through this switch any longer.
  • the current I further flowing due to the energy stored in the inductance L now causes the capacitance C p to be charged from the instant t 1 so that the voltage U 3 increases.
  • the value U 3 has finally reached the value of the DC input voltage U 1 so that the diode D 1 starts conducting. From this instant, a switch-on of the switch S 1 below a switching voltage U S1 of about 0 volt (ZVS at the diode forward voltage) is ensured.
  • a time interval T on (S 1 ) with the switch S 1 switched on and the switch S 2 switched off is thereby initiated.
  • the voltage U 3 has reached the value of zero so that, from this instant, the diode D 2 starts conducting and the switch S 2 can be switched on below a switching voltage US 2 of about 0 volt (at the diode forward voltage), which actually happens at the instant t 9 a short time after a corresponding control voltage U N2 has been applied.
  • a time interval T on (S 2 ) starts from this instant at which the switch S 2 is switched on and the switch S 1 is switched off.
  • dead-time phase T tot between the instants t 0 and t 4 and between the instants t 5 and t 9 , during which dead-time phase both the control voltage U G1 and the control voltage UG 2 are zero so that control voltages operating as switch-off control signals are provided.
  • the dead-time phases T tot are adjusted in such a way that ZVS operation is possible.
  • the shaded areas represent a measure of the available charge for recharging the capacitance C p . In the case shown in FIG. 3 a , the available charge is present to a sufficient extent.
  • the state of operation shown by means of the variations with respect to time in FIG. 3 a represents, for example, an inductive load, i.e. the current I lags with respect to the first harmonic of the voltage U 3 .
  • ZVS operation of the converter 1 is possible in so far as sufficient inductively stored energy is available for the transfer.
  • FIG. 3 b shows, by way of example, corresponding variations with respect to time for a capacitive load.
  • the current I leads with respect to the first harmonic of the voltage U 3 .
  • ZVS operation of the converter 1 is no longer possible.
  • the switch S 2 is switched off.
  • the current I is then positive so that a gradual charging of the capacitance C p up to the voltage U 1 (as in the case shown in FIG. 3 a between the instants t 1 and t 2 ) is not possible due to the current I which is constantly driven on by the energy stored in the inductance L r , but the current continues to flow instead through the diode D 2 .
  • the voltage U 3 is abruptly raised from zero to the value U 1 at the instant t 4 at which the switch S 1 is switched on, i.e. the full voltage having the value of U 1 is still present at this switch S 1 when it is switched on.
  • the switch S 2 is not switched on in the voltageless state in the capacitive load, because the voltage U 3 still has the value U 1 at the instant t 9 at which the switch S 2 is switched on, which value is abruptly decreased to zero.
  • the capacitive load high switching losses are produced (and corresponding large values for the product of the current I and the switch voltages U S1 and U S2 at the instants t 4 and t 9 , respectively) in the switches S 1 and S 2 which are here formed as MOSFET transistors, which may even lead to a destruction of the switches.
  • switching is certainly avoided in the case of a capacitive load.
  • FIG. 3 c shows diagrammatically a section of the upper diagram of FIG. 3 a , representing the dead time after the switch S 1 has been switched off.
  • the shaded area is a comparatively good measure of the energy/charge available for ZVS operation.
  • FIG. 4 shows a configuration of an embodiment according to the invention for the control unit 5 . It comprises a half bridge control 11 , two control circuits 12 , 12 ′ and a controller 13 .
  • the control unit 5 has its own control circuit 12 , 12 ′ for each switch S 1 , S 2 , in which the voltage U W1 decreasing at the resistor W 1 is present at the input of the control circuit 12 and the voltage U W2 decreasing at the resistor W 2 is present at the input of the control circuit 12 ′.
  • the control circuits 12 and 12 ′ make the signals “ZVS S2 possible” and “S 1 off allowed” or “ZVS S 1 possible” and “S 2 off allowed” available at their outputs so as to inform the half bridge control whether there is sufficient energy/charge for zero voltage switching of switches S 2 and S 1 , respectively, and whether the switches S 2 and S 1 can be switched off.
  • the control circuits 12 , 12 ′ thus operate as protective circuits, which ensure a reliable ZVS operation, particularly in the normal mode of operation, after which the load resonant converter has started up.
  • this protective circuit should be deactivated, or its signals should not be taken into account by the half bridge control 11 in order that this does not disturb a start-up of the load resonant converter.
  • the controller 13 In dependence upon the voltage U 2 present at its input 1 , the controller 13 generates control signals for the required frequency and the duty cycle at which the switches S 1 , S 2 should be switched. These signals are present at the outputs of the controller 13 .
  • the half bridge circuit 11 In dependence upon the signals of the control circuits 12 , 12 ′, the frequency signal and/or the duty cycle signal of the controller 13 , as well as upon a regulating value for the instantaneous dead time T tot , which are present at the inputs of the half bridge circuit 11 , the half bridge circuit 11 generates the control voltages U G1 and U B2 present at its outputs for the purpose of switching the switches S 1 and S 2 .
  • the control unit 5 with the control circuits 12 and 12 ′ may be realized in a single IC together with the controller 13 and the half bridge circuit 11 , as is shown. It is particularly also possible to realize the control circuits 12 and 12 ′ by means of a single control circuit and then doubly utilizing control circuit parts by multiplexing the voltages U W1 and U W2 . Similarly as the half bridge circuit 11 and the controller 13 , the control circuits 12 and 12 ′ may also be realized by means of separate ICs.
  • FIG. 5 shows the fundamental structure of a preferred embodiment of the control circuit 12 used for controlling the switch S 1 in a block diagram, with a functional block 21 , 22 , an estimation device 23 , a threshold matching device 24 , a comparator 25 and a circuit block 26 .
  • the functional block 21 , 22 incorporates a combined measuring and evaluating device which determines the current I flowing through the switch S 1 from the voltage U W1 tapped from the resistor W 1 and passes on a signal, equivalent to the current I, to an estimation device 23 . Additionally, the low-pass filtered current gradient dI/dt can be determined from the voltage U W1 tapped from the resistor W 1 and a signal equivalent thereto can be passed on to the estimation device 23 .
  • the estimation device 23 determines the energy/charge from the input values, which energy/charge is available for recharging the parasitic switch capacitances as well as C p .
  • the threshold matching circuit 24 generates a signal, which corresponds to the minimal energy/minimal charge required for recharging the parasitic switch capacitances as well as the capacitance C p . To this end, it adapts a nominal value at its input 1 to the input voltage U 1 , in which the threshold value is raised with an increasing input voltage U 1 because the required energy/charge for charging or discharging the parasitic capacitance in the resonant circuit increases with an increasing input voltage U 1 and is decreased accordingly when the input voltage U 1 decreases.
  • the threshold value can be adjusted in dependence upon the switching frequency of the switches S 1 , S 2 and/or in dependence upon the load present at the converter. During operation, the threshold value can thus be adapted to the working point of the converter 1 .
  • the comparator 25 checks whether the value for the energy/charge determined by the estimation device 23 is larger than the threshold value predetermined by the threshold matching circuit 24 so that a zero voltage switch-on of the next switch is possible, in which it generates a logic “one” (corresponds to “ZVS S 2 possible”) in this case.
  • the input signals which are proportional to the current I and possibly the current gradient dI/dt are applied to the circuit block 26 which performs a validity check and checks whether the switch-off of the instantaneously switched-on switch is principally allowed, and generates a corresponding output signal.
  • criteria for validity check may be, for example:
  • Switching S 1 off I>0, dI/dt ⁇ 0, T on >T min , A>A min or a sub-combination of these criteria (in a corresponding manner, the criteria for switching off S 2 in a circuit block of a control circuit 12 ′ are: I ⁇ 0, dI/dt>0, T on >T min , A>A min or a sub-combination of these criteria).
  • Both the absolute value of the current I and the absolute value of the current I in connection with the current gradient dI/dt (and also the current gradient d/dt I(t) when knowing the transfer function of the resonant circuit or its elements, and estimation of the current I correlating with the current gradient d/dt (I(t) by an estimation device) are a measure of the available energy/charge for causing the voltage U 3 to increase via charging of the parasitic switch capacitances including C p to the input voltage U 1 , which is necessary for switching on the switch S 1 in the ZVS operation, and they are also a measure of the available energy/charge for causing the voltage U 3 to decrease by complete discharging of the parasitic capacitance C p to the value of zero which is necessary for switching on the switch S 2 in the ZVS operation.
  • the threshold value predetermines a limit at which the available residual energy/residual charge is still reliably sufficient to ensure a switch-over of the switches S 1 , S 2 in the ZVS operation.
  • FIG. 6 shows a transfer function A(f) showing the variation of the quotient U 2 /U 3 in dependence upon the frequency f.
  • the transfer function A(f) has its maximum.
  • the capacitive load is provided.
  • Frequencies higher than f r correspond to states of converter operation with in inductive converter load. Accordingly, the converter is to be operated at frequencies f above the resonance frequency f r . It will be evident from FIG.
  • the capacitive mode of operation (range I) should also be avoided because the control mechanisms customarily used for controlling the converter output voltage U 2 no longer react.
  • the value of A(f) decreases with a decreasing frequency in the range 1 so that there is a positive feedback, preventing control of the output voltage U 2 , instead of a negative feedback as in range II (increasing values of A(f) with a decreasing frequency f).

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)
  • Inverter Devices (AREA)
  • Magnetic Resonance Imaging Apparatus (AREA)
US10/514,173 2002-05-15 2003-05-09 Resonant converter with phase controlled switching Expired - Fee Related US7190596B2 (en)

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DE10221450.6 2002-05-15
DE10221450A DE10221450A1 (de) 2002-05-15 2002-05-15 Schaltungsanordnung für einen resonanten Konverter und Verfahren zu dessen Betrieb
PCT/IB2003/001814 WO2003098790A1 (en) 2002-05-15 2003-05-09 Circuit arrangement for a resonant converter and method of operating said converter

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US7190596B2 true US7190596B2 (en) 2007-03-13

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EP (1) EP1506613B1 (ja)
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US20080278984A1 (en) * 2007-05-07 2008-11-13 Harman International Industries, Incorporated Automatic zero voltage switching mode controller
WO2010125066A1 (en) * 2009-04-28 2010-11-04 St-Ericsson Sa (St-Ericsson Ltd) Cross current minimisation
CN108736727A (zh) * 2017-04-14 2018-11-02 台达电子工业股份有限公司 电源转换器及其控制方法
US10763743B1 (en) * 2019-03-06 2020-09-01 Infineon Technologies Ag Analog predictive dead-time
US11539231B1 (en) * 2016-09-23 2022-12-27 Apple Inc. Method and system for single stage battery charging

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JP4957180B2 (ja) * 2006-10-25 2012-06-20 サンケン電気株式会社 電力変換装置及びその制御方法
CN101409971A (zh) * 2007-10-08 2009-04-15 奥斯兰姆有限公司 双重峰值电流控制的电路和方法
FI121561B (fi) 2009-06-30 2010-12-31 Helvar Oy Ab Elektronisen liitäntälaitteen toimintojen säätäminen ja mittaaminen
EP2385617A1 (de) 2010-05-06 2011-11-09 Brusa Elektronik AG Gleichstromsteller mit Steuerung
CN106793219B (zh) * 2016-12-20 2023-04-07 赫高餐饮设备（苏州）有限公司 基于延迟时间以实现电路zvs的方法及系统
CN111385924B (zh) * 2018-12-29 2022-03-22 佛山市顺德区美的电热电器制造有限公司 电磁加热器具及其控制方法、装置

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CN108736727B (zh) * 2017-04-14 2020-02-21 台达电子工业股份有限公司 电源转换器及其控制方法
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ATE339799T1 (de) 2006-10-15
US20050226010A1 (en) 2005-10-13
EP1506613B1 (en) 2006-09-13
JP2005526478A (ja) 2005-09-02
EP1506613A1 (en) 2005-02-16
DE10221450A1 (de) 2003-11-27
JP4376775B2 (ja) 2009-12-02
DE60308359T2 (de) 2007-09-20
DE60308359D1 (de) 2006-10-26
WO2003098790A1 (en) 2003-11-27

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