US6956727B1 - High side current monitor with extended voltage range - Google Patents

High side current monitor with extended voltage range Download PDF

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Publication number
US6956727B1
US6956727B1 US10/762,647 US76264704A US6956727B1 US 6956727 B1 US6956727 B1 US 6956727B1 US 76264704 A US76264704 A US 76264704A US 6956727 B1 US6956727 B1 US 6956727B1
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current
terminal
transistor
resistor
voltage
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US10/762,647
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English (en)
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A. Paul Brokaw
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Analog Devices Inc
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Analog Devices Inc
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Assigned to ANALOG DEVICES, INC. reassignment ANALOG DEVICES, INC. ASSIGNMENT OF ASSIGNORS INTEREST (SEE DOCUMENT FOR DETAILS). Assignors: BROKAW, A. PAUL
Priority to DE602004008734T priority patent/DE602004008734T2/de
Priority to EP04257815A priority patent/EP1557679B1/de
Priority to AT04257815T priority patent/ATE372520T1/de
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Publication of US6956727B1 publication Critical patent/US6956727B1/en
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    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R19/00—Arrangements for measuring currents or voltages or for indicating presence or sign thereof
    • G01R19/0023—Measuring currents or voltages from sources with high internal resistance by means of measuring circuits with high input impedance, e.g. OP-amplifiers
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R1/00—Details of instruments or arrangements of the types included in groups G01R5/00 - G01R13/00 and G01R31/00
    • G01R1/20—Modifications of basic electric elements for use in electric measuring instruments; Structural combinations of such elements with such instruments
    • G01R1/203—Resistors used for electric measuring, e.g. decade resistors standards, resistors for comparators, series resistors, shunts
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R19/00—Arrangements for measuring currents or voltages or for indicating presence or sign thereof
    • G01R19/0092—Arrangements for measuring currents or voltages or for indicating presence or sign thereof measuring current only

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  • This invention relates to the field of current monitors, and particularly to “high side” current monitors.
  • a high side current monitor is designed to measure the signal current through a sensing element connected in series with a circuit's high side (as opposed to its return side).
  • a shunt voltage proportional to the signal current is developed across the sensing element—typically a small resistor.
  • the current monitor measures the differential voltage across the sensing element, and produces a ground or common-referred output that varies with the sensed current.
  • a conventional high side current monitor is shown in FIG. 1 .
  • a sensing element 10 here a resistor having a resistance R s , is connected in series with a signal 12 having a voltage V 1 and carries a current of interest I sense to a load 14 ; R s is typically on the order of 0.1 ⁇ .
  • An operational amplifier A 1 is connected across the sensing element, with its inverting input connected to the sensing element's load side, and its non-inverting input connected to the sensing element's high side via a resistor 15 having a resistance R 1 .
  • a feedback transistor Q 0 here a NPN, has its base connected to the output of A 1 , its collector connected to the junction of R 1 and A 1 's non-inverting input, and its emitter providing an output I out .
  • I out is delivered to an output resistor 16 having a resistance R out to produce an output voltage V out .
  • I sense develops a shunt voltage V sense across R s ; A 1 responds by causing Q 0 to conduct a current through R 1 necessary to equalize A 1 's inverting and non-inverting inputs.
  • This current (I out ) is proportional to the voltage (V sense ) across—and thus to the current (I sense ) through—sensing element 10 .
  • output voltage V out I out R out , it is also proportional to current of interest I sense .
  • the differential voltage applied to A 1 can have a large common mode potential.
  • An op amp IC has an associated breakdown voltage determined by its fabrication process, which limits its common mode input range—which in turn limits the signals with which the current monitor of FIG. 1 can be safely used.
  • a high side current monitor circuit is presented which overcomes the problems noted above.
  • the present high side current monitor circuit includes components of the monitor described above: a sensing element is connected between high and load side terminals, carries a current of interest I sense , and develops a shunt voltage V sense between the terminals in response to I sense .
  • An op amp's non-inverting input is coupled to the load side terminal, and a resistor is connected between the high side terminal and the amplifier's inverting input.
  • a feedback transistor is connected to the op amp's output and conducts an output current I out through the resistor to a current output node necessary to make the voltages at the amp's inverting and non-inverting inputs equal—such that I out is proportional to I sense .
  • the present current monitor circuit may also include an output load resistor connected between a second node and ground, and a second transistor coupled between the current output node and the second node and connected to conduct I out to the output load resistor such that a ground-referred voltage proportional to V sense is developed at the second node.
  • the op amp and feedback transistor are preferably contained within an integrated circuit (IC) package, while the second transistor is preferably external to the IC and fabricated with a high voltage process. When so arranged, the external transistor stands off most of the common mode voltage, thereby reducing the voltage across the IC to less than the breakdown voltage associated with the IC's fabrication process. This permits measurement of shunt voltages having common mode voltages in excess of the breakdown voltage.
  • the discrete transistor can be a P-type field-effect transistor (FET) or a PNP bipolar. If the latter, the invention preferably includes a base current recycling circuit which corrects for errors that would otherwise arise due to the external transistor's base current.
  • FET field-effect transistor
  • PNP bipolar P-type field-effect transistor
  • the current monitor circuit may be configured as a “dual-use” IC, which can be used either with or without a discrete external transistor.
  • the monitor circuit is further arranged such that it can be powered with a limited fraction of the common mode voltage when used with an external transistor, and is self-biased when used without an external transistor.
  • FIG. 1 is a schematic diagram of a known high side current monitor.
  • FIG. 2 is a schematic diagram of a high side current monitor per the present invention.
  • FIG. 3 is a schematic diagram of another embodiment of a high side current monitor per the present invention.
  • FIG. 4A is a schematic diagram of a dual-use IC implementation of the present high side current monitor, in one of its two applications.
  • FIG. 4B is a schematic diagram of a dual-use IC implementation of the present high side current monitor, in the other of its two applications.
  • FIG. 5 is a schematic diagram of a bandgap shunt regulator as might be used with the present high side current monitor.
  • FIG. 2 One embodiment of a high side current monitor circuit in accordance with the present invention is shown in FIG. 2 .
  • the circuit comprises a sensing element 10 , typically a resistor having a resistance R s (though other devices having a known impedance could also be used), connected between high side and load side terminals 20 and 22 and in series with a signal 12 having a voltage V 1 .
  • R s carries a current I sense to a load 14 .
  • Op amp A 1 is connected across sensing element 10 , with its non-inverting input coupled to load side terminal 22 and its inverting input connected to high side terminal 20 via a resistor 15 having a resistance R 1 .
  • a 1 may be powered between V 1 and a local circuit common point (COM).
  • COM local circuit common point
  • the present monitor circuit can include a voltage limiter 23 , which, when coupled to ground via a resistance R lim , enables A 1 to be powered from a voltage much lower than the signal common mode voltage.
  • a feedback transistor Q 1 here a PNP bipolar (though a FET could also be used), has its base connected to the output of A 1 , its emitter connected to the junction ( 24 ) of R 1 and A 1 's non-inverting input, and its collector providing an output current I out .
  • I sense develops a shunt voltage V sense across R s ;
  • a 1 responds by causing Q 1 to conduct current I out through R 1 necessary to equalize A 1 's inverting and non-inverting inputs, such that I out is proportional to V sense and I sense .
  • the differential voltage applied to A 1 can have a large common mode potential.
  • An op amp IC has an associated breakdown voltage determined by its fabrication process (referred to herein as the “process breakdown voltage”), which limits its common mode input range—which in turn limits the signals with which the current monitor circuit can be safely used.
  • the invention overcomes this limitation with the addition of a transistor Q 2 , which is connected to conduct output current I out to output resistor 16 , the other side of which is connected to ground.
  • the voltage developed across resistor 16 is the circuit's output voltage V Out .
  • Components A 1 , Q 1 and R 1 are preferably housed within an IC 26 , and Q 2 is preferably a discrete transistor external to IC 26 .
  • the IC portion 26 can be biased so that most of the voltage between V 1 and ground is stood off by transistor Q 2 , instead of being mostly across Q 1 and A 1 as in the prior art.
  • Q 2 is preferably made with a high voltage process, so that the monitor circuit can tolerate a V sense having a common mode voltage in excess of the process breakdown voltage.
  • Voltage limiter 23 limits the voltage which can be made to appear across A 1 , and Q 2 stands off most of the remaining common mode voltage—thereby enabling the measurement of a small shunt voltage (V sense ) having a common mode voltage in excess of the process breakdown voltage.
  • Transistor Q 2 is preferably a P-type FET (as shown in FIG. 2 ), or a PNP bipolar (described below and shown in FIG. 3 ). If a PFET is used, its gate should be connected to circuit common point COM, and the monitor circuit must be arranged such that the voltage between V 1 and COM is sufficient to allow A 1 to drive Q 1 and for the collector of Q 1 to drive Q 2 to the gate-source voltage needed for Q 2 to conduct I out to R out .
  • the discrete external transistor is a PNP bipolar transistor
  • the magnitude of I out conducted to output resistor 16 will be reduced by the PNP's base current, resulting in an error in V out .
  • the invention preferably includes a base current recycling circuit to reduce or eliminate this error.
  • FIG. 3 A preferred arrangement is shown in FIG. 3 .
  • the discrete external transistor is a PNP bipolar transistor Q 3 .
  • Q 3 's base is connected to a simple current mirror, made from transistor Q 4 (diode-connected) and Q 5 , each of which is referenced to A 1 's circuit common point (COM).
  • a resistor 30 having a resistance R 2 approximately equal to R 1 is interposed between load side terminal 22 and A 1 's non-inverting input, and the collector of Q 5 is connected to the junction 32 of R 2 and A 1 .
  • Q 3 In operation, Q 3 's base current I base biases Q 4 so that its base voltage biases Q 5 such that Q 5 conducts a current I Q5 nearly equal to I base (assuming a 1:1 mirror).
  • Current I Q5 is conducted through resistance R 2 .
  • the component of voltage that results from this displaces A 1 's non-inverting input by a small amount.
  • a 1 responds by driving the base of Q 1 to force a similar displacement across R 1 . This adds an increment of current to the signal current in R 1 , thereby increasing I out .
  • the increment of current added to the signal current should closely approximate the base current I base , thereby correcting for the error that would otherwise arise due to Q 3 's base current.
  • COM is the most negative of all the terminals, so that no more than 5 volts difference can appear anywhere inside the IC.
  • the present current monitor circuit is suitably configured as a “dual-use” IC, which can be used when the common mode potential of V sense is greater than or less than the process breakdown voltage.
  • IC 40 is shown employed in its two uses in FIGS. 4A and 4B .
  • IC 40 is connected to high side terminal 20 and load side terminal 22 via I/O pins VP and VSENS, respectively, which are also connected to resistors 15 and 30 .
  • the common mode potential of V sense is expected to be less than the process breakdown voltage. In this case, there is no need for an external discrete transistor to stand off the voltage across A 1 , so output current I out is connected directly to output resistor 16 (via an I/O pin IOUT) to generate VOUT.
  • IC 40 preferably includes circuitry which allows it to be self-biased when the common mode potential of V sense is within the IC's safe operating range; one possible embodiment of such circuitry is shown in FIG. 4A .
  • a voltage limiter 23 connected between V 1 and COM provides A 1 's operating voltage;
  • a 1 's operating current is set with a current I bias generated with a bias circuit 44 .
  • Bias circuit 44 preferably comprises a PNP transistor Q 6 having its emitter connected to COM and voltage limiter 23 , and a PNP transistor Q 7 having its base and emitter connected in common with the base and emitter of Q 6 .
  • the collector of Q 6 is connected to the collector of an NPN transistor Q 8 at a node 45 , and the collector and base of Q 7 are connected to the emitter of a PNP transistor Q 9 .
  • the base of Q 9 is connected to node 45 , and Q 9 's collector is connected to the collector of a diode-connected NPN transistor Q 10 having its base connected in common with the base of Q 8 and its emitter connected to Q 8 's emitter via an emitter degeneration resistor 46 .
  • a resistor 47 is connected between Q 7 's base and Q 9 's collector, and a resistor 48 is connected between the emitter and base of Q 6 /Q 7 .
  • the bias circuit is connected to an I/O pin BIAS at Q 10 's emitter.
  • resistor 46 in the emitter circuit of Q 8 makes the emitter voltages of Q 8 and Q 10 different.
  • Q 8 is made larger than Q 10 (8 times larger in FIG. 4A ), so that in the absence of resistor 48 , the Wilson current mirror composed of Q 6 , Q 7 , and Q 9 could force Q 10 to operate at the same current as Q 8 .
  • Q 8 and Q 10 would operate at a fixed current density ratio of 8, causing a voltage (kT/q)ln8 ⁇ 54 mV at room temperature to appear across resistor 46 .
  • This known temperature proportional voltage determines the magnitude of the equal currents in inverse proportion to resistor 46 .
  • bias current I bias would be the sum of the equal proportional-to-absolute-temperature (PTAT) currents in Q 8 and Q 10 .
  • PTAT proportional-to-absolute-temperature
  • Adding resistor 48 provides an additional component of current in Q 9 since it must drive resistor 48 until the voltage across it makes the base-emitter voltage for Q 6 .
  • This current complements the PTAT current with a complementary-to-absolute-temperature (CTAT) current so that the total bias current is nearly temperature-invariant.
  • CTAT complementary-to-absolute-temperature
  • Bias circuit 44 should be arranged to provide an I bias having a magnitude sufficient to operate the IC, but no more than the voltage limiter's maximum allowable current.
  • the voltage limiter is preferably arranged to stabilize A 1 's operating voltage near the minimum that it needs.
  • FIG. 4B illustrates the use of the IC when the common mode potential of V sense is greater than the process breakdown voltage.
  • an external discrete transistor Q 3 stands off most of the common mode voltage, thereby reducing the voltage across A 1 to that portion of the common mode voltage permitted by voltage limiter 23 and making the voltage across the IC less than the process breakdown voltage.
  • the COM and BIAS pins are connected together, thereby disabling bias circuit 44 , and a resistor 52 is connected between BIAS/COM and ground which provides I bias to IC 40 .
  • Resistor 52 is selected to provide an I bias current within the range described above.
  • Voltage limiter 23 limits the voltage across A 1 and the other IC circuits to protect them, but must allow enough voltage for A 1 to operate, and for the added voltage required due to the drop across Q 4 and the forward-biased Q 3 base-emitter voltage.
  • the present current monitor may be used as a stand-alone IC for use in low voltage applications, or in an extended voltage application by adding an external discrete transistor (Q 2 /Q 3 ) and a resistor ( 52 ).
  • the invention further enables the IC to be fabricated using a basic low voltage process.
  • bias circuit implementation shown in FIGS. 4A and 4B is preferred, many other circuits could be employed to establish appropriate operating points for IC 40 .
  • voltage limiter 23 could be implemented in many different ways. For example, a zener diode (as shown in FIGS. 2–4 ) or an avalanche breakdown diode could be used. Another possibility is to use an electronic bandgap shunt regulator; one possible embodiment of such a regulator is shown in FIG. 5 .

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  • Physics & Mathematics (AREA)
  • General Physics & Mathematics (AREA)
  • Measurement Of Current Or Voltage (AREA)
  • Other Investigation Or Analysis Of Materials By Electrical Means (AREA)
  • Measurement Of Radiation (AREA)
  • Analysing Materials By The Use Of Radiation (AREA)
  • Control Of Electrical Variables (AREA)
US10/762,647 2004-01-21 2004-01-21 High side current monitor with extended voltage range Expired - Lifetime US6956727B1 (en)

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US10/762,647 US6956727B1 (en) 2004-01-21 2004-01-21 High side current monitor with extended voltage range
DE602004008734T DE602004008734T2 (de) 2004-01-21 2004-12-15 Hochspannungsstromdetektor
EP04257815A EP1557679B1 (de) 2004-01-21 2004-12-15 Hochspannungsstromdetektor
AT04257815T ATE372520T1 (de) 2004-01-21 2004-12-15 Hochspannungsstromdetektor

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Cited By (30)

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US20050269657A1 (en) * 2004-06-03 2005-12-08 Timothy Dupuis On chip transformer isolator
US20050272378A1 (en) * 2004-06-03 2005-12-08 Timothy Dupuis Spread spectrum isolator
US20070126493A1 (en) * 2005-12-06 2007-06-07 Supertex, Inc. Low power high side current monitor which operates at high voltages and method therefor
US7376212B2 (en) 2004-06-03 2008-05-20 Silicon Laboratories Inc. RF isolator with differential input/output
US20080143266A1 (en) * 2006-12-18 2008-06-19 Microsemi Corp. - Analog Mixed Signal Group Ltd. Voltage Range Extender Mechanism
US7421028B2 (en) 2004-06-03 2008-09-02 Silicon Laboratories Inc. Transformer isolator for digital power supply
US20080267301A1 (en) * 2004-06-03 2008-10-30 Silicon Laboratories Inc. Bidirectional multiplexed rf isolator
US20090017773A1 (en) * 2004-06-03 2009-01-15 Silicon Laboratories Inc. Capacitive isolator
US20090027125A1 (en) * 2007-07-24 2009-01-29 Analog Devices, Inc. Common mode rejection calibration method for difference amplifiers
US7737871B2 (en) 2004-06-03 2010-06-15 Silicon Laboratories Inc. MCU with integrated voltage isolator to provide a galvanic isolation between input and output
US7738568B2 (en) 2004-06-03 2010-06-15 Silicon Laboratories Inc. Multiplexed RF isolator
US7821428B2 (en) 2004-06-03 2010-10-26 Silicon Laboratories Inc. MCU with integrated voltage isolator and integrated galvanically isolated asynchronous serial data link
US7902627B2 (en) 2004-06-03 2011-03-08 Silicon Laboratories Inc. Capacitive isolation circuitry with improved common mode detector
US20110095793A1 (en) * 2009-10-22 2011-04-28 Oki Semiconductor Co., Ltd. Bias potential generating circuit
WO2012039735A1 (en) * 2010-09-21 2012-03-29 Sendyne Corp. High-accuracy low-power current sensor with large dynamic range
US8198951B2 (en) 2004-06-03 2012-06-12 Silicon Laboratories Inc. Capacitive isolation circuitry
US8441325B2 (en) 2004-06-03 2013-05-14 Silicon Laboratories Inc. Isolator with complementary configurable memory
US8451032B2 (en) 2010-12-22 2013-05-28 Silicon Laboratories Inc. Capacitive isolator with schmitt trigger
JP2014021090A (ja) * 2012-07-24 2014-02-03 Panasonic Corp 電流検出回路及び電流検出回路を用いた超音波診断装置
US8902005B2 (en) 2012-09-25 2014-12-02 Analog Devices, Inc. Apparatus and method for wide common mode difference
US9052343B2 (en) 2011-03-01 2015-06-09 Sendyne Corporation Current sensor
US9264002B2 (en) 2014-02-19 2016-02-16 Analog Devices Global Apparatus and methods for improving common mode rejection ratio
US9496835B2 (en) 2014-12-15 2016-11-15 Semiconductor Components Industries, Llc Current sense amplifer with extended common mode input voltage range
US9729140B2 (en) 2014-03-05 2017-08-08 Analog Devices, Inc. Circuits with floating bias
US9960741B2 (en) 2016-06-27 2018-05-01 Dialog Semiconductor (Uk) Limited High frequency common mode rejection technique for large dynamic common mode signals
US10290608B2 (en) 2016-09-13 2019-05-14 Allegro Microsystems, Llc Signal isolator having bidirectional diagnostic signal exchange
US20210124386A1 (en) * 2019-10-24 2021-04-29 Nxp Usa, Inc. Voltage reference generation with compensation for temperature variation
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US20220228929A1 (en) * 2021-01-20 2022-07-21 Kioxia Corporation Semiconductor integrated circuit

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Cited By (46)

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US8198951B2 (en) 2004-06-03 2012-06-12 Silicon Laboratories Inc. Capacitive isolation circuitry
US7902627B2 (en) 2004-06-03 2011-03-08 Silicon Laboratories Inc. Capacitive isolation circuitry with improved common mode detector
US20080267301A1 (en) * 2004-06-03 2008-10-30 Silicon Laboratories Inc. Bidirectional multiplexed rf isolator
US7302247B2 (en) 2004-06-03 2007-11-27 Silicon Laboratories Inc. Spread spectrum isolator
US20050269657A1 (en) * 2004-06-03 2005-12-08 Timothy Dupuis On chip transformer isolator
US7376212B2 (en) 2004-06-03 2008-05-20 Silicon Laboratories Inc. RF isolator with differential input/output
US8441325B2 (en) 2004-06-03 2013-05-14 Silicon Laboratories Inc. Isolator with complementary configurable memory
US8049573B2 (en) 2004-06-03 2011-11-01 Silicon Laboratories Inc. Bidirectional multiplexed RF isolator
US8064872B2 (en) 2004-06-03 2011-11-22 Silicon Laboratories Inc. On chip transformer isolator
US20050272378A1 (en) * 2004-06-03 2005-12-08 Timothy Dupuis Spread spectrum isolator
US7421028B2 (en) 2004-06-03 2008-09-02 Silicon Laboratories Inc. Transformer isolator for digital power supply
US8169108B2 (en) 2004-06-03 2012-05-01 Silicon Laboratories Inc. Capacitive isolator
US7650130B2 (en) 2004-06-03 2010-01-19 Silicon Laboratories Inc. Spread spectrum isolator
US7737871B2 (en) 2004-06-03 2010-06-15 Silicon Laboratories Inc. MCU with integrated voltage isolator to provide a galvanic isolation between input and output
US7738568B2 (en) 2004-06-03 2010-06-15 Silicon Laboratories Inc. Multiplexed RF isolator
US7821428B2 (en) 2004-06-03 2010-10-26 Silicon Laboratories Inc. MCU with integrated voltage isolator and integrated galvanically isolated asynchronous serial data link
US7856219B2 (en) 2004-06-03 2010-12-21 Silicon Laboratories Inc. Transformer coils for providing voltage isolation
US20090017773A1 (en) * 2004-06-03 2009-01-15 Silicon Laboratories Inc. Capacitive isolator
US7336122B2 (en) * 2005-12-06 2008-02-26 Supertex, Inc. Low power high side current monitor which operates at high voltages and method therefor
US20070126493A1 (en) * 2005-12-06 2007-06-07 Supertex, Inc. Low power high side current monitor which operates at high voltages and method therefor
US20080143266A1 (en) * 2006-12-18 2008-06-19 Microsemi Corp. - Analog Mixed Signal Group Ltd. Voltage Range Extender Mechanism
US7928662B2 (en) 2006-12-18 2011-04-19 Microsemi Corp.—Analog Mixed Signal Group Ltd. Voltage range extender mechanism
US7570114B2 (en) 2007-07-24 2009-08-04 Analog Devices, Inc. Common mode rejection calibration method for difference amplifiers
US20090027125A1 (en) * 2007-07-24 2009-01-29 Analog Devices, Inc. Common mode rejection calibration method for difference amplifiers
US20110095793A1 (en) * 2009-10-22 2011-04-28 Oki Semiconductor Co., Ltd. Bias potential generating circuit
US8432194B2 (en) * 2009-10-22 2013-04-30 Oki Semiconductor Co., Ltd. Bias potential generating circuit
WO2012039735A1 (en) * 2010-09-21 2012-03-29 Sendyne Corp. High-accuracy low-power current sensor with large dynamic range
US8264216B2 (en) 2010-09-21 2012-09-11 Sendyne Corp. High-accuracy low-power current sensor with large dynamic range
US8451032B2 (en) 2010-12-22 2013-05-28 Silicon Laboratories Inc. Capacitive isolator with schmitt trigger
US9052343B2 (en) 2011-03-01 2015-06-09 Sendyne Corporation Current sensor
JP2014021090A (ja) * 2012-07-24 2014-02-03 Panasonic Corp 電流検出回路及び電流検出回路を用いた超音波診断装置
US8902005B2 (en) 2012-09-25 2014-12-02 Analog Devices, Inc. Apparatus and method for wide common mode difference
US9264002B2 (en) 2014-02-19 2016-02-16 Analog Devices Global Apparatus and methods for improving common mode rejection ratio
US9729140B2 (en) 2014-03-05 2017-08-08 Analog Devices, Inc. Circuits with floating bias
US9503039B2 (en) 2014-12-15 2016-11-22 Semiconductor Components Industries, Llc Trimming method for current sense amplifiers
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EP1557679A2 (de) 2005-07-27
DE602004008734D1 (de) 2007-10-18
EP1557679B1 (de) 2007-09-05
EP1557679A3 (de) 2005-11-02
DE602004008734T2 (de) 2008-06-12
ATE372520T1 (de) 2007-09-15

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