TWI393399B - Quadrature imbalance mitigation using unbiased training sequences - Google Patents

Description

Orthogonal non-equilibrium mitigation using unbiased calibration sequences

SUMMARY OF THE INVENTION The present invention relates to communication channel estimation, and more particularly to systems and methods for using quadrature modulation unbiased calibration sequences in the adjustment of receiver channel estimates.

This patent application is the patent application No. 11/684,566 (Attorney Docket No. 060395) filed on March 9, 2007 and entitled "QUADRATURE MODULATION ROTATING TRAINING SEQUENCE" Part of the continuation of the application, the status of which is hereby incorporated by reference in its entirety in its entirety in its entirety in its entirety in its entirety in its entirety in its entirety in its entirety.

Figure 1 is a schematic block diagram of a conventional receiver front end (prior art). A conventional wireless communication receiver includes an antenna that converts a radiation signal into a conductive signal. After some initial filtering, the conduction signal is amplified. Assuming a sufficient power level is given, the carrier frequency of the signal can be converted by mixing the signal with a local oscillator signal. Since the received signal is quadrature modulated, the signal is demodulated by separating it before combining the I and Q paths. After frequency conversion, the analog signal can be converted to a digital signal by a analog-to-digital converter (ADC) for baseband processing. The process can include a Fast Fourier Transform (FFT).

There are a large number of errors that can be introduced into the receiver to adversely affect channel estimation and recovery of the intended signal. Errors can be introduced from mixers, filters, and passive components such as capacitors. These errors can cause the I path and the Q path in it. The imbalance between the two deteriorates. In an effort to estimate the channel and thus eliminate some of these errors, the communication system may utilize a message format that includes a calibration sequence that may be a repeating or predetermined data symbol. For example, by utilizing an orthogonal frequency division multiplexing (OFDM) system, the same IQ constellation point can be repeatedly transmitted for each subcarrier.

In an effort to preserve the power of portable battery operating devices, some OFDM systems use only a single modulation symbol for tuning. For example, a unique direction (eg, an I path) in the constellation is excited without exciting the other direction (eg, the Q path). One-way tuning of the same type can also be used with pilot tones. It should be noted that scrambling a single modulation channel (eg, I channel) with a ±1 symbol value does not rotate the constellation point and does not provide an excitation for the orthogonal channel.

The power save tuning sequence mentioned above results in a bias channel estimate when quadrature path imbalance occurs, which is more popular in large bandwidth systems. A bias channel estimate can align the IQ constellation in one direction (i.e., the I path) but provide orthogonal unbalance in the orthogonal direction. Preferably, any non-equilibrium can be equally distributed between the two channels.

2 is a schematic diagram illustrating orthogonal non-equilibrium (prior art) on the receiver side. Although not shown, the transmitter side imbalance is similar. Assume that the Q path is a reference. The impact waveform is cos(wt+ θ ), where θ is the channel phase. The Q path is downconverted with -sin( wt ). The I path is downconverted by (1+2 ε )cos( wt +2Δ φ ). 2 △ φ and 2 ε are hard unbalanced, which are respectively a phase error and an amplitude error. The low pass filters H _I and H _{Q of} each path are different. These filters introduce additional amplitude and phase distortion. However, these additional distortions move within 2Δ φ and 2 ε . It should be noted that these two filters are practical and affect +w and -w in exactly the same way.

Suppose these errors are small:

The first component cos( wt ) on the right hand side of the equation is a slightly scaled ideal I path. The second component -2 Δ φ .sin( wt ) is a small leak from the Q path. After down-conversion of the impulse waveform: in the I path: (1+2ε)cos( θ )+2 ε .sin( θ ).

In the Q path: sin( θ ).

These errors lead to misunderstanding of symbol positioning in the quadrature modulation constellation, which in turn leads to incorrect demodulation of the data.

Wireless communication receivers are prone to errors due to lack of tolerance in hardware components associated with mixers, amplifiers, and filters. In a quadrature demodulation transformer, such errors can also result in an imbalance between the I path and the Q path, resulting in incorrect processed data.

A calibration signal can be used to calibrate the receiver channel. However, a calibration signal that cannot simultaneously excite both the I path and the Q path cannot solve the imbalance problem between the two paths.

Accordingly, a method for transmitting a non-biased communication tuning sequence is provided. The method produces an unbiased calibration sequence in a quadrature modulation transmitter. The unbiased calibration sequence represents an equal cumulative power uniformly distributed in the complex plane. More specifically, via an in-phase with a cumulative power (I) The modulation path is used to transmit the calibration information in the time domain. The calibration information in the time domain is transmitted via an orthogonal (Q) modulation path having a cumulative power equal to the modulation power of the I.

In one aspect, the unbiased calibration sequence is generated as a signal pair comprising a composite value reference signal (p) having a frequency of +f and a composite image signal (p _m ) having a frequency of -f . This method makes the product (p.p _m ) zero.

A method for calculating an unbiased channel estimate is also provided. The method accepts an unbiased alignment sequence in a quadrature demodulation receiver. The unbiased calibration sequence includes a predetermined reference signal (p) that represents an evenly distributed power uniformly distributed in the complex plane. The method processes the unbiased calibration sequence and produces a processed symbol (y) representative of the complex plane information in the unbiased calibration sequence. The processed symbol (y) is conjugated to the corresponding reference signal (p ^* ) and an unbiased channel estimate (h _u ) is obtained.

Additional details of the above described methods, systems, and variations of such systems and methods for generating an unbiased calibration sequence and calculating an unbiased channel estimate are provided below.

Various embodiments will now be described with reference to the drawings. In the following description, numerous specific details are set forth However, it is apparent that such embodiments may be practiced without such specific details. In other instances, well-known structures and devices are shown in block diagrams in order to illustrate the embodiments.

As used in this application, the terms "processor", "processing device", "component", "module", "system" and similar terms are intended to relate to a computer-related reality. A body, which can be a hard body, a tough body, a combination of a hardware and a soft body, a soft body, or a soft body in execution. For example, a component can be, but is not limited to, a process running on a processor, a process, a processor, an object, an executable file, an execution thread, a program, and/or a computer . For example, an application running on a computing device and the computing device can be a component. One or more components can reside within a process and/or execution thread, and a component can be limited to a computer and/or distributed between two or more computers. In addition, such components can be executed from a variety of computer readable media having various data structures stored thereon. The components can be communicated by the local and/or remote process, for example based on a signal having one or more data packets (eg, data from a component, the component and a local system, a distributed system) Another component interacts and/or interacts with other systems via signals across a network such as the Internet.

Various embodiments are provided below by a system that can include a large number of components, modules, and the like. It is to be understood and appreciated that the various systems may include additional components, modules, etc. and/or may not include all of the components, modules, etc. discussed in connection with the drawings. You can also use one of these methods to combine.

The various illustrative logic blocks, modules, and circuits that have been described herein can be constructed or implemented by means of a general purpose processor, a digital signal processor (DSP), a dedicated integrated circuit (ASIC), and a programmable gate. An array (FPGA) or other programmable logic device, discrete gate or transistor logic circuit, discrete hardware component, or any combination thereof designed to implement the functions described above. A general purpose processor can be a microprocessor, or the processor It can also be any conventional processor, controller, microcontroller or state machine. A processor can also be constructed as a combination of computing devices, for example, a combination of a DSP and a microprocessor, a plurality of microprocessors, a combination of one or more microprocessors and a DSP core, or any other This type of configuration.

The methods or algorithms described in connection with the embodiments disclosed herein may be embodied directly in hardware, in a software module executed by a processor, or in a combination of both. A software module can reside in: RAM memory, flash memory, ROM memory, EPROM memory, EEPROM memory, scratchpad, hard disk, a removable disk, a CD-ROM Or any other form of storage medium known in the art. A storage medium can be coupled to the processor such that the processor can read information from the storage medium and write information to the storage medium. Alternatively, the storage medium can be integrated into the processor. The processor and the storage medium can reside in an ASIC. The ASIC can reside in the node or elsewhere. Alternatively, the processor and the storage medium can reside as discrete components in a node, or at other locations on the access network.

FIG. 3 is a schematic block diagram of an exemplary data transmission system 300. A baseband processor 302 has an input on line 304 for accepting digital information from a medium access control (MAC) level. In one aspect, baseband processor 302 includes an encoder 306 having an input on line 304 for accepting digital (MAC) information and a line 308 for use in the frequency domain. The output of the encoded digital information is supplied. An interleaver 310 can be used to interleave the encoded digital information to provide interleaved information in the frequency domain on line 312. Interleaver 310 is a device that converts a single high speed input signal into a plurality of parallel lower rate streams, each of which The lower rate stream is associated with a particular subcarrier. An inverse fast Fourier transform (IFFT) 314 accepts information in the frequency domain, performs an IFFT operation on the input information, and supplies a digital time domain signal on line 316. A digital-to-analog converter 318 converts the digital signal on line 316 into an analog fundamental signal on line 320. As explained in more detail below, a transmitter 322 modulates the baseband signal and supplies a modulated carrier signal as an output on a line 324. It should be noted that alternative circuit configurations capable of performing the same functions as described above will be well known to those skilled in the art. Although not explicitly shown, a receiver system will consist of a set of similar components to reversely process information received from a transmitter.

4 is a schematic block diagram of a system or apparatus for transmitting a non-deviation communication tuning sequence. System 400 includes a transmitter or transmission component 402 having an input on line 404 for accepting digital information. For example, this information can be supplied from the MAC level. The transmitter 402 has an output on the line 406 for supplying a quadrature modulation unbiased calibration sequence, the quadrature modulation unbiased calibration sequence representing an evenly distributed uniform distribution in a complex plane power.

Transmitter 402 can include a transmitter subsystem 407, such as a radio frequency (RF) transmitter subsystem that uses an antenna 408 to communicate via an air or vacuum medium. However, it should be appreciated that the present invention is applicable to any communication medium (e.g., wireless, wireline, optical) capable of carrying orthogonally modulated information. Transmitter subsystem 407 includes an in-phase (I) modulation path 410, or a means for generating I-modulated tuning information having a cumulative power in the time domain. Transmitter subsystem 407 also includes an orthogonal (Q) modulation path 412, or one for A component of the Q modulation calibration information having a cumulative power equal to the power of the I modulated path is generated in the time domain. The I path information on line 404a is upconverted at mixer 414 by carrier fc, and the Q path information on line 404b is upconverted at mixer 416 by a phase shifted version of the carrier (fc + 90 ∘). I path 410 and Q path 412 are summed at combiner 418 and supplied to line 420. In some aspects, the signal is amplified at amplifier 422 and supplied to antenna 408 on line 406, wherein the unbiased calibration sequence is radiated. The I path and the Q path are alternatively referred to as an I channel and a Q channel. A non-deviation tuning sequence may also be referred to as a rotational tuning signal, an orthogonal balance tuning sequence, a balanced tuning sequence, a balanced tuning sequence, or an unbiased calibration signal.

For example, the unbiased calibration sequence can be sent initially via the I modulation path 410, and then the calibration information is sent via the Q modulation path 412. That is, the calibration signal can include information, such as a symbol transmitted only via the I modulation path or a series of repeated symbols, followed by transmission of a symbol or series of repeated symbols transmitted only via the Q modulation path. Alternatively, the calibration information can be sent initially via the Q modulation path and then the path is modulated via the I. In the case where a single symbol is instead transmitted through the I path and the Q path, the transmitter transmits a rotational tuning signal. For example, the first symbol can always be (1, 0), the second symbol can always be (0, 1), the third symbol is (-1, 0), and the fourth symbol is (0, -1). .

However, it is not necessary to replace the symbol transmission by the I and Q modulation paths as described above to obtain symbol rotation. For example, the transmitter can simultaneously transmit the calibration information through the I and Q modulation paths, and combine the I and Q modulated signals. number.

The unbiased calibration sequence of the rotation type mentioned above initially (only) transmits the calibration signal via the I modulation path, and can be energized by the I modulation path without energizing the Q modulation path to realise. Then, after transmitting the calibration information via the I modulation path, the transmitter transmits a calibration signal via the Q modulation path by enabling the Q modulation path. The tuned symbols can also be rotated by supplying symbols each having I and Q components, as is conventionally associated with quadrature modulation.

Typically, transmitter 402 also transmits orthogonally modulated (unscheduled) communication material. The unbiased calibration sequence is used to create an unbiased channel estimate by a receiver (not shown) that permits more accurate restoration of unscheduled communication material. In one aspect, the orthogonally modulated communication data is transmitted after transmitting the unbiased calibration sequence. In another aspect, the unbiased calibration sequence and the communication data are simultaneously transmitted in the form of a pilot signal. The system is not limited to any particular temporal relationship between the calibration signal and the orthogonally modulated communication material.

To be unbiased, the symbol values associated with any particular subcarrier can be periodically changed. The simplest component that evenly distributes information in the complex plane when there is an even number of symbols per message is rotated 90 turns per cycle. As used herein, a message is a group of symbols in a predetermined format. A message has a duration of a number of symbol periods. One or more symbols can be transmitted per symbol period. Some messages contain a preamble before the body of the message. For example, a message can be formed as a long packet containing a number of OFDM symbols. Each OFDM symbol contains a number of subcarriers. In some aspects, the message preamble contains an unbiased calibration sequence. Yu Qi In his aspect, the unbiased calibration sequence is a sequence of pilot signals transmitted simultaneously with unscheduled communication data.

If an odd number of symbols are used in the calibration sequence of a message, it is not always beneficial to rotate the phase of the symbol by 90 degrees per cycle. For a sequence of 3 symbols, a 60 ∘ or 120 ∘ rotation can be used to evenly distribute the symbol | _[r1] values in the complex plane. For 5 symbols, you can use a 180/5∘ or 360/5∘ rotation. If the number of symbols in a tuning sequence is a prime number, a combined solution can be used. For example, if there are a total of 7 symbols in a message, then a 90" rotation can be used for the first 4 symbols and a 120" (or 60" rotation for the last 3 symbols. In another aspect, the unbiased calibration sequence can be averaged for more than one message. For example, if a message contains 3 calibration symbols, the combination of 2 messages contains 6 symbols. In the context of a 6-symbol calibration signal, a 90-turn rotation can be used between symbols.

Since the power system corresponds to a measure of the square of a complex symbol value, the power associated with the symbol vector at an angle θ in the composite space can also be considered as the power at (θ + 180). Therefore, the cumulative power at a 60-degree angle is the same as the power at 240 。. Alternatively, the power associated with the sign at angle θ can be summed with the power at the angle (θ + 180). By summing the powers at angles θ and (θ+180), the complex plane spans only 180 在 when considered from the corresponding power. For this reason, when the unbiased calibration sequence consists of only 2 orthogonal symbols or 3 symbols separated by 60 ,, the equal accumulation of one power is uniformly distributed in the complex plane.

FIG. 5A is a diagram showing an unbiased calibration sequence represented by both time domain and frequency domain. In one aspect, the transmitter generates a signal pair comprising a composite value reference signal (p) having a frequency of +f and a composite value image signal (p _m ) having a frequency of -f, wherein the product (p.p _m ) is zero. For example, at time i=1, the product (p ₁ .p _1m )=0. As mentioned above, p and p _m have a composite value of amplitude and phase components. In another aspect, the transmitter generates i occurrences of the reference signal (p) and the image signal (p _m ) and causes the sum of the products (p _i · p _im ) to be zero. Or, the sum of (p _i ·p _im ) = 0, where i = 1 to N. It should be noted that the "dot" between the p _i and p _im symbols is intended to represent a conventional multiplication between scalar numbers.

Similarly, when the transmitter produces the reference signal and the image signal i occurrences, the signal pair values p and p _m may, but need not, vary for each occurrence. For example, the transmitter can generate the sum of the products (p _i .p _im ) to zero by representing the p as a composite value that remains constant for each occurrence. To represent p _m , the transmitter can generate information as a composite value that is rotated 180 每次 per occurrence. However, there are almost an infinite number of other ways to make the product (p _i .p _im ) zero.

In another aspect, the transmitter produces i occurrences of the reference signal (p) and the image signal (p _m ), and the product of each occurrence (p _i · p _im ). The emitter causes the pairs to appear in pairs and the sum of the products from each pair is zero.

For example, one or more messages may have a time sequence with N pilot tones for a given subcarrier f and N pilot tones for the mirror subcarrier -f. As mentioned above in the discussion of Figure 5A, to create an unbiased calibration sequence using this pilot tone, the general solution is to have a sum of (p _i ·p _im ) = 0, where i = 1 to N. For a particular solution, the pilots of i = 1 and 2 are tuned in pairs. Therefore, p ₁ ·p _1m +p ₂ ·p _2m =0. Similarly, the pilot tones of i=3 and 4 can be paired as follows: p ₃ · p _3m + p ₄ . p _4m =0. This pair can continue to i=N. If each pair has a sum of zero, the sum is also zero, that is, the sum p _i ·p _im =0. Pairing will simplify the problem of zero. Instead of searching for N pilots to verify the sum of p _i · p _im = 0, it is sufficient to make 2 pairs of pilots zero.

As explained above, a simple example of creating an unbiased calibration sequence involves rotating the symbol 90 degrees in the time domain or in the frequency domain, maintaining the sign of the reference signal at +f, but inverting the sign of the image signal to -f . Both examples use 2 pairs of tones and satisfy the equation p ₁ ·p _1m +p ₂ ·p _2m =0.

In other words, the unbiased calibration sequence may include: time 1: p ₁ is +f and p _1m is -f; time 2: p ₂ is +f and p _2m is -f; time 3: p ₃ is +f and p _3m is -f; and time 4: p ₄ is +f and p _4m is -f.

The unbiased calibration sequence can be obtained by averaging. The principle of the unbiased calibration sequence indicates that the pilot must satisfy: p ₁ . p _1m +p ₂ . p _2m +p ₃ ·p _3m +p ₄ ·p _4m =0.

As a variant, the unbiased calibration sequence can be organized as follows: p ₁ · p _1m + p ₂ · p _2m =0 and p ₃ · p _3m + p ₄ · p _4m =0.

Figures 5B and 5C are graphical representations of the equal accumulation of power evenly distributed across a complex plane. The complex plane can be used to represent real axis (R) and imaginary axis (I) information. The circle represents the boundary of equal power or energy, and a normalized value is 1. In FIG. 5B, the unbiased calibration sequence is formed by three symbols: a first symbol (A) at 0 ;, a second symbol (B) at 120 ;, and a The third symbol (C) at 240 。. When the first symbol (A) is maintained at 0 、, the second symbol (B') is at 60 、, and the third symbol (C') is at 120 ,, an identical power distribution is obtained. The power associated with each symbol is one.

In Figure 5C, the unbiased calibration sequence is formed by 5 symbols: 2 symbols at 0∘, each having a power of 0.5, such that the cumulative power is 1; a symbol at 90∘, which has a Power 1; a symbol at 180 , having a power of 1; and a symbol at 270 , having a power of 1.

As used herein, the above "equal accumulation of power" may be an accumulation of exactly equals in the direction of each complex plane, as in many cases, a zero-error error may be transmitted and received with a zero error correction sequence. That is, the calibration sequence is 100% deviation. Or, as mentioned above, p _i . The sum of p _im = 0. In a worst case analysis, L pilot symbols are averaged, each having an equal cumulative power as follows: |p _i ·p _im sum |=|p _i | ² sum = L.

If L is 100%, and if the sum of |p _i ·p _im |=L/4, the (equal cumulative power) error is 25%. An unbiased calibration sequence with a 25% error will still produce excellent results. If L/2 (a 50% error) is used, the IQ interference with the channel estimate is still reduced by 6 dB, and good results are obtained.

FIG. 6 is a diagram showing an unbiased calibration sequence enabled as a pilot tone sequence in the time domain. In a plurality of symbol periods, the transmitter can generate an unbiased alignment sequence by supplying P pilot symbols per symbol period. Each pulse in the illustration represents a symbol. In the plurality of symbol periods, The transmitter generates (N-P) orthogonally modulated communication data symbols per symbol period and simultaneously supplies N symbols per symbol period. Many communication systems, such as IEEE 802.11 and UWB, use pilot tones for channel tuning purposes.

FIG. 7 is a diagram showing an unbiased calibration sequence enabled as a preamble before unscheduled communication material. The transmitter generates quadrature modulated communication data and supplies the unbiased calibration sequence for a first plurality of symbol periods (eg, at times 1 to 4), followed by a second plurality of symbol periods (eg, Orthogonally modulated communication data is supplied at time 5 to N). Again, the pulses in the illustration represent symbols.

For example, an ultra-wideband (UWB) system utilizes six symbols transmitted prior to transmission of communication data or a beacon signal. Therefore, 3 consecutive symbols can be generated on the I modulation path, and then 3 consecutive symbols are generated on the Q modulation path. Using this procedure, it is only necessary to briefly activate the Q channel for 3 symbols before the Q channel returns to sleep. However, there are many other combinations of symbols that can be used to generate an unbiased tuning sequence.

By looking at Figure 5B or 5C, it can be seen that the transmitter produces a time series of complex plane symbols having equal cumulative power in a plurality of directions (in the complex plane). As used herein, "direction" refers to the sum of vectors at each angle θ and (θ + 180). For example, the power associated with the symbol at 0∘ is cumulative with the power from the symbol at 180∘, since 0∘ and 180∘ are in the same direction. As a result of this relationship, the symbol time series in the unbiased calibration sequence has an accumulated power associated with real axis information in the time domain, and an equal accumulated power associated with the virtual axis information in the time domain, such as emission. The device is supplied in a plurality of symbol periods. In another aspect, the unbiased calibration sequence representing the equal cumulative power uniformly distributed in the complex plane can be expressed as a time series having i complex symbols (a) in the time domain, as follows: a _i (k) The sum of a _i (k) = 0; where k is the number of samples per symbol period. It should be noted that the "dot" between the a _i and a _i symbols is intended to represent a conventional multiplication between scalar numbers.

Since the symbol a _i is usually a subcarrier having a periodic waveform, there is no specific value for a. That is, a _i varies with time and can be represented as a _i (t). However, assuming that T is normalized to 1, if t samples are obtained, the symbol can be expressed as a _i (kT) or a _i (k). For the time domain system, the summation of k disappears. With only one sample per symbol, the symbol and the sample become the same, and the equation can be written as: the sum of a _i · a _i = 0.

To illustrate by means of a simple 2-symbol orthogonal unbiased calibration sequence, if the first symbol (i = 1) has a 0-turn angle, an equal amount of power must exist at a 180-degree angle to satisfy such formula. Similarly, if the second symbol is at 90 ,, the equivalent amount of power must be present at a 270 corner. Other more complex instances may require summing the symbols for i indexes to get a final result of zero.

Alternatively, it is considered that the sum of the formulas a _i · a _i is the fact that the power is always the same regardless of the magnitude of the angle in the case where the projection is performed in either of the complex planes and the power has been calculated. The power of the direction φ is: |Re a _i (-jφ)| ² sum = 0.5 | a _i | ² sum + 0.5 Re (-2jφ) a _i · a _i sum = 0.

If and only if a _i . When the sum of a _i is 0, this power is constant for all φ.

It can be seen that the frequency domain formula (the sum of p _i · p _im = 0) is equal to a _i . The sum of a _i = 0. The time domain signals corresponding to p _i and p _im are: a _i = p _i exp(j2πft) + p _im exp(-j2πft); because p _{i is} modulated +f and p _{im is} modulated by -f.

In a symbol i, the integer part of a _i · a _i over time is: the integer part of a _i · a _i = {p _i · p _i exp(j4πft) + p _im · p _im exp(-j4πft) + p _i The integer part of p _im } = p _i · p _im ; because exp(j4πft) is rotated several times and disappears when integrated into a symbol.

Therefore, a _i · a _i accumulated in one symbol is equal to p _i · p _im .

If all the symbols are added: the sum of the integer parts of a _i · a _{i =} the sum of p _i · p _im = 0.

FIG. 8 is a diagram showing an unbiased calibration sequence enabled by averaging symbols for a plurality of messages. A symbol (or more than one, not shown) is generated in the first symbol period in a first message. After the first message, a symbol is generated in the second symbol period in the second message. More generally, a tuning information symbol is generated in a plurality of messages. The transmitter generates the unbiased calibration sequence by creating equal power accumulated over the plurality of messages in a plurality of complex plane directions. Although a similar pre-coded calibration sequence is shown in Figure 7, the same type of analysis can be applied to the pilot-type unbiased calibration sequence.

FIG. 9 is a schematic block diagram showing a processing apparatus for transmitting a non-deviation communication tuning sequence. Processing device 900 includes a transmitter module 902 for receiving digital information on line 904 and on line 906 Supply a quadrature modulation unbiased calibration sequence. The unbiased calibration sequence represents an equal accumulation of power evenly distributed in the complex plane. The functionality associated with processing device 900 is similar to the transmitters set forth above in Figures 3-8 and will not be repeated here for the sake of brevity.

Figure 10 is a schematic block diagram of a system for calculating an unbiased channel estimate. System 1000 includes a quadrature demodulation receiver or receiving component 1002 having an input on line 1004 to accept an unbiased calibration sequence. As with the transmitter of FIG. 4, the receiver 1002 can be an RF device that is coupled to an antenna 1005 to receive radiation information. However, the receiver may alternatively receive the unbiased calibration sequence via a wired or optical medium (not shown). The unbiased calibration sequence includes a predetermined reference signal (p) that represents an equal cumulative power uniformly distributed in a complex plane, as defined above.

Receiver 1002 produces a processed symbol (y) on line 1006 to represent the complex plane information in the unbiased calibration sequence and sends it to multiplier 1008. Since the p value is predetermined, the multiplier 1008 can multiply each processed symbol (y) by the (predetermined) conjugate (p ^* ) of the corresponding reference signal and supply an unbiased channel at the output on line 1010. Estimated value (h _u ). For example, the conjugate information can be stored in memory 1012 and supplied to multiplier 1008 on line 1014.

In one aspect, receiver 1002 accepts an unbiased alignment sequence having a plurality of simultaneously accepted predetermined reference signals (p _n ). For example, the receiver can accept a message with P pilot symbols (per symbol period), see Figure 6. The receiver 1002 generates a plurality of processed symbols (y _n ) from the corresponding plurality of reference signals, multiplying each processed symbol by its corresponding reference signal conjugate to obtain a plurality of channel estimation values (h _un ), and for n Each value is averaged to estimate the channel estimate (h _un ). By using the example of Figure 6, P unbiased channel estimates are obtained. Methods for determining channel estimates are well known in the art. However, the receiver of the present invention is capable of utilizing predetermined data to calculate an extremely accurate unbiased type of channel estimate.

In another aspect, the receiver subsystem 1016 has an in-phase (I) demodulation variable path 1018, or a means for accepting I demodulation tuning information having a cumulative power in the time domain. A quadrature (Q) demodulation variable path 1020 or a means for accepting Q demodulation tuned information in the time domain has a cumulative power equal to the I modulated path power.

Comparing Figure 10 with Figure 6, receiver 1002 accepts an unbiased calibration sequence having a time series of _n predetermined reference signals ( _pn ). Receiver 1002 generates a time series of n processed symbols (y _n ) from a time sequence of reference signals and multiplies each processed symbol in the time series by its corresponding reference signal conjugate. In Figure 6, P processed symbols (y) are generated per symbol period. Receiver 1002 obtains a time series having n channel estimates (h _un ) and averages the n channel estimates.

In one aspect, the receiver 1002 accepts the unbiased calibration sequence as a signal pair comprising a composite value reference signal (p) having a frequency of +f and a composite image signal having a frequency of -f. (p _m ), where the product (p.p _m ) is zero, see Figure 5. Further, the receiver can accept the unbiased calibration sequence as the i-time occurrence of the reference signal (p) and the image signal (p _m ), wherein the sum of the products (p _i ·p _im ) is zero. In one variation, the receiver 1002 is the i-th occurrence of the image signal and the reference signal, the signal value varies with p _m and p each occurrence. In another variation, the receiver accepts the unbiased calibration sequence as the i-time occurrence of the reference signal (p) and the image signal (p _m ), and produces a product for each occurrence (p _i ·p _im ). The receiver causes each occurrence to be paired and produces a processed symbol by zeroing the sum of the products from each pair. For example, the receiver can accept a signal pair such that the sum of the products (p _i · p _im ) is zero, as follows. The information is accepted as a representative p as a composite value that is constant for each occurrence. The information is accepted as a composite value of 180 每次 per occurrence to represent p _m .

Comparing Figures 10 and 6, in one aspect, the receiver accepts the unbiased alignment sequence with P pilot symbols per symbol period and obtains P unbiased pilot channel estimates over a plurality of symbol periods. . The receiver simultaneously accepts (N-P) orthogonally modulated communication data symbols in each symbol period to produce a processed symbol (y _c ) for the communication material in each symbol period. That is, (N-P) processed symbols are generated. The receiver extrapolates the channel estimate derived from the unbiased pilot channel estimate for each processed symbol (y _c ) and multiplies each processed symbol by the extrapolated channel estimate to derive a transmitted symbol (x ). The symbol x is an unknown symbol value transmitted as a communication data. Extrapolation of channel estimates for data channels based on unbiased channel estimates of adjacent pilot channels will be familiar to those skilled in the art.

Comparing Figures 10 and 7, receiver 1002 accepts quadrature modulated communication data during the symbol period after accepting the unbiased calibration sequence. The receiver generates a processed symbol (y _c ) for each communication data symbol and multiplies each processed symbol by the unbiased channel estimate to derive a transmitted symbol (x).

As mentioned above in the explanation of the transmitted unbiased calibration sequence, receiving The device accepts a time series of complex plane symbols having equal cumulative power in a plurality of directions in the complex plane (as defined above). Similarly, the time series of the unbiased calibration sequence symbols have an accumulated power associated with real axis information in the time domain and an equal accumulated power associated with the virtual axis information in the time domain.

In another aspect, the unbiased calibration sequence accepted by the receiver can be expressed as a time series having i complex symbols (a) in the time domain, as follows: a _i (k). The sum of a _i (k) = 0; where k is the number of samples per symbol period.

Comparing Figures 10 and 8, the receiver can accept the unbiased calibration sequence as a symbol in a plurality of messages having an equal power accumulated in the plurality of messages in a plurality of complex plane directions.

FIG. 11 is a schematic block diagram of a processing apparatus for calculating an unbiased channel estimate. The processing device 1100 includes a quadrature demodulation receiving module 1102 having an input on the line 1104 to receive an unbiased calibration sequence having a uniform distribution in the complex plane. A predetermined reference signal (p) of equal cumulative power. Receiver module 1102 generates processed symbols (y) representing complex plane information in an unbiased calibration sequence supplied to line 1106. A multiply module 1108 multiplies the processed symbol (y) by the conjugate (p ^* ) of the corresponding reference signal and supplies an unbiased channel estimate (h _u ) at the output on line 1110. Many of the features of processing device 1100 are shared with the receiver of Figure 10 and will not be repeated here for the sake of brevity.

Whether it is a calibration sequence enabled in a preamble or as a pilot signal The calibration sequence has the following similarities: the information content of the transmitted data is typically predetermined or "preferred" data that permits receiver calibration and channel measurement. When receiving communication (non-scheduled) data, there are 3 unknowns: the data itself, the channel, and the noise. The receiver cannot be calibrated for noise because the noise changes randomly. Channels are typically measured in association with delay and multipath. For relatively short time periods, errors due to multipath can be measured when using predetermined data (such as tuning or pilot signals). Once the channel is known, this measurement can be used to remove errors in the received communication (unscheduled) material. Therefore, some systems supply a calibration signal to measure a channel before data decoding begins.

However, for example, the channel may change as the transmitter or receiver moves in space or clock shifts. As a result, many systems continue to send more "known" and "unknown" data to track slow changes in the channel.

Although not specifically shown, the transmitter of FIG. 3 and the receiver of FIG. 10 can be combined to form a transceiver. In fact, the transmitter and receiver of the transceiver can share components such as an antenna, a baseband processor, and a MAC level circuit. The explanations set forth above are intended to illustrate a transceiver that transmits an unbiased calibration sequence and calculates an unbiased channel estimate based on receiving an unbiased calibration sequence from other transceivers in a network of devices.

Function Description

Modern data rate communication systems transmit signals on two discrete channel-in-phase channels and quadrature phase channels (I and Q). The two channels form a 2D constellation in a complex plane. QPSK and QAM are examples of constellations. The I and Q channels can be carried by RF hardware, which may not be due to changes in the RF components Better balance, which leads to IQ imbalance. In the growing number of shared direct conversion systems, this imbalance problem is getting worse. IQ unbalance distorts the constellation and causes an intersection between the I and Q channels: the signal interferes with itself. Increasing the transmission power does not help because the self-generated interference increases as the signal power increases. The signal-to-noise ratio (SINR) reaches an upper limit that sets a limit on the highest data rate achievable with a given RF hardware. To increase data rates, a costly solution uses more chic and more expensive hardware. A potentially cost-effective solution is to estimate and compensate IQ imbalance in a digital manner. The concept of digital estimation and compensation algorithms has previously been enhanced in this technology. However, such solutions tend to be expensive because they do not rely on a particular type of tuning sequence. These solutions generally only consider the imbalance of one side (typically at the receiver).

An example focusing on orthogonal frequency division multiplexing (OFDM) is given below, and attention is paid to studying end-to-end (from transmitter to receiver) unbalanced time domain systems. Furthermore, in OFDM, the non-equilibrium model is modeled as a function of frequency and takes into account variations in the frequency response of the filter.

Two enhancements are provided herein: a zero cost, which eliminates interference from channel estimates by utilizing a non-biased calibration sequence. A sufficient gain is achieved because errors in channel estimates are generally more detrimental to performance than errors in the data itself. If more gain is needed, a second, relatively low cost enhancement will compensate for data distortion.

An IQ imbalance model is provided below. An analysis is provided to show how the conventional channel imbalance can be mitigated by using the conventional channel estimates using the unbiased calibration sequence. Then, provide a direct extension to calculate the IQ imbalance parameter, assuming the The algorithm is valid. By using the estimated parameters, a simple compensation algorithm is provided to mitigate data distortion. Simulation results for WiMedia's UWB and suggestions for modifying the standard are also given.

The IQ non-equilibrium model IQ imbalance occurs when the power (amplitude) balance or orthogonality (phase) between the in-phase (I) and quadrature phase (Q) channels is not maintained. Therefore, the IQ non-equilibrium system consists of an amplitude non-equilibrium 2ε and a phase unbalanced 2△ To characterize.

The time domain signal transmits and receives a complex symbol x via the I and Q channels. In an ideal noise-free channel, the symbol x is completely received. However, when IQ imbalance occurs, it may receive a noise or distortion version.

y=αx+βx ^* , (1) where:

The non-equilibrium modeled complex, where α 1 and β 0. The nonlinear model (1) is linearized via the following vector form:

B is an unbalanced matrix. The second line is obsolete because of the copy of the first line. However, it gives an input and output of the same size and type so that the unbalanced blocks at the transmitter and receiver can be connected in series, as described below. The unbalanced matrix at the transmitter is defined by B _t and it is defined by B _r at the receiver.

Single tap channel Now consider a single tap channel suitable for OFDM. A single tap channel h system with a suitable matrix form

When there is an imbalance between the transmitter and the receiver, and when the Gaussian (AWGN) noise n, the vector form N = ( nn ^* ) ^T is in an average state, the received signal is expressed as a series of linear blocks.

The overall result is IQ unbalanced and channel combination to create a total channel h', plus an undesired distortion or interference characterized by a total unbalanced parameter β'. The total unbalanced parameter β' changes as the channel changes and may need to be regularly estimated.

Then, consider the condition in which the symbol x is limited to a predetermined (1D) axis instead of crossing the entire complex plane. For example, the axis may be associated with BPSK modulation, real axis, imaginary axis, or any axis between the two. In this case, x ^* = kx can be written, where k is a complex constant (one rotation), and

If x is limited to a unique axis, the IQ imbalance disappears and becomes an integer part of the total channel response.

Frequency Domain Signals Although the previous model is applicable to time domain signals, it is now considered that the signal x of interest is given at a frequency f in the frequency domain. In the time domain, this signal is carried by a composite tone xe ^j2πft . By substituting the term in equation (1), the following equation αxe ^j2πft + βx ^* e - ^{j2πft is obtained} . (7)

In OFDM, interference created by IQ unbalance is not shown at the same frequency f, but at the image frequency -f, and vice versa. Signals transmitted at -f create interference with frequency +f. If the signal x _m is a signal transmitted at the frequency -f, where the index m identifies an amount at the image frequency -f, then the following equation α _m x _m e ^-j2πft +β _m x _m ^* e is obtained at the frequency -f ^J2πft . (8)

It has been generalized using one of these time domain equations. The IQ non-equilibrium parameters α and β are here a frequency function. This will model a non-equilibrium due to different low pass (fundamental) or band pass (IF) filters in the system. The I and Q paths cannot have identical filters, and therefore the imbalance will vary with frequency. In time domain systems, such non-equilibrium exists but is extremely expensive to compensate. An equalizer and an extension of the model are needed to handle different convolutions on different channels. Therefore, in the time domain, the overall or average unbalance is used. The frequency domain system is able to utilize a common equalizer structure and model the non-equilibrium based on each frequency.

If the outputs of equations ( 7 ) and ( 8 ) are combined per subcarrier, then the following equation Y = (αx + β _m x _m ^* ) e ^j2πft y _m = (α _m x _m + βx ^* )e - ^{j2πft is observed} . (9)

By ignoring the subcarriers (automatically processed by the FFT), one of the linear model functions at +f and -f can be written as

In the frequency domain model, the second line is no longer obsolete. In a scenario, the model handles a pair of image frequencies. A single tap channel h with a frequency f and a h _{m with a} frequency of -f are modeled by the following matrix

AWGN noise n with frequency f and noise n _{m with} frequency -f form noise vector . End-to-end model system

h', h _m ' is the total channel tap, and β', β _m ' is the total unbalanced parameter. The unbalanced parameters change as the channel changes and may need to be estimated regularly.

Since IQ imbalances produce interference entirely from the image frequency, two interesting scenarios are worth noting. If no signal is transmitted at the image frequency, or the channel is in a recession, no interference is created. On the other hand, if the signal or channel is strong, the interference may be strong. Therefore, in OFDM, the effect of IQ imbalance is more problematic.

The custom channel estimate shows how to solve half of the problem without cost by using a non-deviation tuning sequence before examining the compensation algorithm. A non-biased calibration sequence completely eliminates interference from channel estimates, resulting in significant performance improvements. In fact, errors in channel estimates are often more unfavourable than errors in the data, since channel estimates may create a bias in the constellation.

The pilot tone is used to excite the model ( 12 ). At frequency + f, the transmission of the pilot p, and the frequency -f, the transmission of the pilot p _m. Before the generality is assumed, assuming that the pilot has a unit specification (the channel carries the effective power), the estimated channel value at the frequency f is obtained by derotating with p ^* .

By averaging several channel observations, the noise is automatically reduced (for clarity, the noise is eliminated). Regarding the term β' _m p _m ^* p ^* , many OFDM systems (eg, UWB of WiMedia) use a calibration sequence that is only a repeat symbol. Therefore, this item does not decline with averaging. The distortion of a set of +1 or -1 applied to the entire OFDM symbol does not help, which was due to the p ^* and p _m ^* does not change in the time between the two symbols is inverted. Instead, the following is achieved: after accumulating a large number of observations, the sum of the following products is made zero: Σ _i p _i p _im =0. (14)

Typically, the calibration sequence consists of an even number of symbols and is sufficient to ensure that each pair is added to zero p ₁ p _1m + p ₂ p _2m =0. (15)

An example of a simple sequence that satisfies this condition is given in Table 1 . Such a calibration sequence is labeled as an unbiased calibration sequence because, on the one hand, an unbiased channel estimate is produced, and on the other hand, the calibration signal equally spans the I and Q dimensions of the complex plane in the time domain. For example, a non-deviation tuning sequence is not only concentrated along the real axis.

As an experiment, considering the unit halfway between p _i and p _im specification complex scalar ^{_{a i = p i e jθ =}} p im e -jθ. In the time domain, the pilot is added up to 2 a _i cos (2 _II ft + θ). In the time domain and a predetermined OFDM symbol, the two mirrored pilots span a unique direction determined by the composite constant a _i . If L symbols are transmitted, the total (or average, or cumulative) power system in one direction φ is Σ _i | ^R a _i exp(-jφ)| ² =0.5L+0.5 ^R exp(-jφ)Σa _i a _i . This power is constant in any direction φ if and only if Σ _i a _i a _i ≡Σ _i p _i p _im =0. Achieve equal crossing of the complex plane.

IQ Unbalance Estimation After estimating the total channel h', consider the estimated value of the total unbalanced parameter β _m '. Careful analysis of equation ( 12 ) explains that this parameter can be obtained in much the same way as the estimated channel estimates. That is, β _m ' can be handled similarly to a "channel" carrying the pilot p _m ^* . Thus, by derotating with p _m , an estimate of the imbalance can be obtained. The condition for the unbalanced unbiased estimate is consistent with equation ( 14 ).

In summary, by using an unbiased calibration sequence and two custom channel estimates, a good estimate of end-to-end channel and unbalanced parameters is obtained (Table 2 ).

Smoothing of Adjacent Subcarriers In addition to averaging adjacent OFDM symbols, channel estimates within one symbol can also be smoothed for adjacent subcarriers. In OFDM, the cyclic preamble is designed to be shorter, and the channel of the tone-by-tone modulation is assumed to change slowly. Similarly, the filter in the RF chain should have a short time response and its frequency response will slowly change, ie the IQ imbalance will slowly change across the subcarriers. The same channel smoothing technique can be used to smooth and improve the unbalanced parameter estimates. By utilizing the unbiased calibration sequence, there is no interaction between the channel estimate and the unbalanced estimate. Each estimate can be smoothed independently.

If an estimate is made using a unique OFDM symbol, it is impossible to find an unbiased alignment sequence that satisfies equation ( 14 ). In this case, a nearly unbiased calibration sequence can be obtained by applying the summation of equation ( 14 ) to a group of two or more adjacent subcarriers. Smoothing then automatically cancels all or part of the interference from the image frequency. One solution is to rotate the pilot by 90 毗邻 on adjacent subcarriers (moving in the mirror direction at positive and negative frequencies).

The best estimator The use of the unbiased calibration sequence and the above-mentioned conventional channel estimate results is a minimum variance (LS) estimator. Among all LS estimators, the Minimum Mean Square Error (MMSE) sensor displays a valid value.

The minimum variance estimator L transmissions X _i , L noise terms N _i and L observation values Y _i may be serially connected to the 2×L matrix respectively x = ( X ₁ X ₂ ... X _L ) N = ( N ₁ N ₃ ... N _L ) y = ( Y ₁ Y ₂ ... Y _L ). (16)

Then, equation ( 12 ) becomes y = H' x + N . (17)

H' is unknown. LS estimator

When the condition ( 14 ) is satisfied, it is easy to verify the diagonal of the xx ^H system (the intersection term disappears). It is proportional to a unit matrix because the pilots are normalized to unit specifications. then

It is precisely the four conventional channel estimates that are de-rotated by p _i ^* , p _im , p _im ^* and p _i , respectively, as explained in the previous section. Two estimates are obtained for the frequency f and two estimates are obtained for the image frequency -f.

Best Estimator The unbiased calibration sequence and the estimated channel estimate are an LS estimator. But any estimator Also an LS estimator. The following shows the use of unbiased alignment sequence results in a very good estimator. The model ( 17 ) can be viewed as an unknown information H' sent over 2 consecutive transmissions for 2 vectors (columns of x ) in one L dimension space. To I x _j, N _j and y _j are labeled x, y and N columns of j, where . Models ( 12 ) and ( 17 ) can be written as y ₁ =h 'x ₁ +β ' _m x ₂ + N ₁ y ₂ =β 'x ₁ +h ' _m x ₂ + N ₂ . (20)

There are 2 transmissions, and each transmission involves 2 vectors x ₁ , x ₂ , and each of these vectors carries the complex amplitude information to be estimated. The LS estimator includes a projection onto each vector and a parallel projection to other vectors to cancel the interference. A very good result is obtained when the two vectors are orthogonal (i.e., when the point ( 14 ) is zero). The unbiased calibration sequence is defined by a calibration sequence that verifies this condition. Other sequences use non-orthogonal vectors and suffer from a loss of the performance function of the angle between the vectors x ₁ and x ₂ . Many OFDM systems currently use a calibration sequence of a range of differences, where x ₁ , x _{2 are} collinear, and it is not possible to correctly estimate 4 entries in H ' . These training sequence tends to estimate the channel h 'and h' more noise version of _m.

To calculate the mean square error (MSE), estimate the error . This is a 2 x 2 matrix, which is 4 error values. Each value can be isolated by multiplying the left and right ends by a combination of vectors (1 0) ^T and (0 1) ^T . Assuming that NN ^H is a unit matrix, or more generally a diagonal matrix with elements σ ² and σ _m ² , it can be displayed and The MsE is the first and second diagonal elements of σ ² ( xx ^H ) ^-1 , respectively. And for and In terms of MSE, the first and second diagonal elements of σ _m ² ( xx ^H ) ^-1 are respectively.

The total MSE is 2(σ ² +σ _m ² )tr( xx ^H ) ^-1 . Now, the problem is to find the minimized x of tr( xx ^H ) ^-1 under the constraint that the total pilot power is constant (ie tr( xx ^H )=2L). Using eigenvalue decomposition, the problem can be written to minimize Σ1/λ _j under the condition that Σλ _j is constant. This problem is solved by means of a Lagrangian multiplier and is usually optimal when all eigenvalues are equal. This means Proportional to a unit matrix.

The total MSE has been minimized and the resulting MSE is σ ² /L or σ _m ² /L per element. But this MSE per element may be the best available, even with a unique vector transmission. This MSE cannot be improved for a 2-vector transmission, and thus the MSE per element has been minimized. The unbiased calibration sequence plus the custom channel estimator is the MMSE for all LS estimators.

IQ Unbalance Compensation If the gain from the undistorted channel estimate is insufficient, the IQ imbalance parameter (as previously stated) can be estimated and used to compensate for data distortion. H' is estimated in model ( 12 ), Y = H'X + N. Now the focus returns to the unknown data X. The model is identical to any of the 2 tap channels with cross-correlation. Any channel equalization algorithm can be applied. A simple equalization algorithm for QAM and recession channels for universal bit interleaved coding is provided.

One of the points associated with the zero-forcing (ZF) method H' ^-1 Y=X+H' ^-1 N enhances the noise when the mirrored channel is weak, unless statistics are performed for complex color noise. This solution utilizes ZF, but only when the mirrored channel is not weak. In equation (12), x _m is replaced by the value of x _m to obtain the following formula: among them Noise enhancement. It should be noted that the second-order unbalanced term β' ^* β _m '<<h'h _m ' ^{* is} assumed. When this approximation is invalid, consider the corrected channel It is accompanied by an accurate estimate of the channel and unbalanced parameters.

Basically, ZF technology consists of the following formula

By subtracting the image frequency (β _m '/h _m ') y _m from the received signal y, a simple channel model without IQ non-equilibrium is obtained. The rest of the decoding chain does not change.

As long as the noise enhancement is weaker than the original interference from IQ unbalance, ie |n'| ² <|β _m 'x _m ^* | ² , this solution works well. If not, the original y is used instead of the unbalanced correction z. There is no need to estimate n' to make a decision. A robust averaging improvement can be selected. Therefore, consider the expected value

When the signal-to-noise ratio SNR _{m of the} image frequency is greater than 1, the non-equilibrium correction term z is used. Otherwise, the original signal y is maintained. Since the channel and unbalance estimation are not accurate, it is safer to use a larger SNR, for example SNR _m > 2 would be good for WiMedia UWB. It should be noted that the SNR _m can generally be obtained from the total SNR via the formula SNR _m =|h _m '| ² SNR.

Table 3 summarizes the ZF algorithm with noise enhancement avoidance.

Simulation Results Figure 12 illustrates the performance achieved by applying the above algorithm to the WiMEdia UWB standard. The highest data rate of 480 Mbps is simulated in the IEEE 802.15.3 channel model CM2 (approximately 4 meters indoor pico environment). Avoid shadow techniques and band hopping. IQ non-equilibrium is constant and the amplitude is equal to 2ε=10% (0.8 dB) and the phase is . Provide the same amount of unbalance at the transmitter and receiver. This figure shows the packet error rate (PER) as a function of Eb/No. Performance is quickly degraded without any form of compensation. Table 4 lists the losses of various algorithms for the ideal situation.

The end-to-end IQ unbalance and channel are combined to form a total 2 x 2 channel matrix. Use the unbiased calibration sequence to achieve a significant cost-free gain. The unbiased calibration sequence automatically cancels end-to-end self-interference from channel estimation. In addition, this tuning sequence is ideally used to estimate the IQ non-equilibrium parameters, and a simple algorithm is presented to compensate for data distortion: a zero-forcing method with noise enhancement avoidance.

In particular, WiMedia UWB benefits from the enhancements described below: a conventional deviation adjustment sequence consisting of six symbols specifically transmitted on the I channel can be divided into two halves to create an unbiased sequence. The first 3 symbols are transmitted on the I channel, and the last 3 symbols are transmitted on the Q channel. By equally spanning the complex plane, an unbiased calibration sequence with large gain is created for high data rates. For reverse compatibility purposes, this scheme can be reversed for high data rate mode and transmitted via beacons, or the calibration sequence type can be blindly detected.

In OFDMA (eg, WiMAX), subcarriers f and -f can be assigned to different users. If power control drives a user to a high power level, considerable interference can occur. Therefore, it is a good idea to locate the pilots of different users on the mirrored subcarriers. The pilot should meet the unbiased calibration sequence then. Each user automatically benefits without any additional capabilities. The pilot should jump to a different position while maintaining the mirror position.

The time domain formula can be extended to code division multiple access (CDMA), where a one-band equalizer is combined with several single tap channels. The unbiased calibration sequence automatically improves the channel estimate per tap. A simple unbiased calibration sequence for CDMA involves rotating the complex symbol 90 turns frequently.

Figure 13 is a flow chart illustrating a method for transmitting a non-deviation communication tuning sequence. Although the method is illustrated in the figures as a series of numbered steps for clarity, this numbering does not necessarily specify the order of the steps. It will be appreciated that some of these steps may be skipped, performed in parallel, or there may be no need to maintain a strict sequence when implementing some of the steps. The method begins in step 1300.

Step 1302 generates an unbiased calibration sequence in a quadrature modulation transmitter, wherein the unbiased calibration sequence represents an equal cumulative power uniformly distributed in a complex plane, as defined above. Step 1304 transmits the unbiased calibration sequence. The terms "produce," "export," and "multiply" refer to processes that may be enabled by the use of machine readable software instructions, hardware, or a combination of software and hardware.

In one aspect, generating an unbiased calibration sequence in step 1302 includes sub-steps. Step 1302a generates tuning information transmitted in the time domain via an in-phase (I) modulation path having a cumulative power. Step 1302b generates calibration information transmitted in the time domain via an orthogonal (Q) modulation path having a cumulative power equal to the I modulated path power.

In another aspect, generating the unbiased calibration sequence in step 1302 includes the following sub-steps. Step 1302c generates a signal, which signal comprises a frequency to f + of the complex value reference signal (p) and a frequency of the composite value of the image signal -f (p _m). Step 1302d causes the product (p·p _m ) to be zero.

For example, the i-time occurrence of the reference signal (p) and the image signal (p _m ) can be generated and the sum of the products (p.p _m ) is zero. The generation of the i occurrence of the reference signal and the image signal may include generating signal pair values p and p _m that vary for each occurrence. In one aspect, the sum of the products (p _i .p _im ) can be zero by generating the information as a composite value (representing p) that is constant for each occurrence. To represent p _m , the information can be generated as a composite value that is rotated 180 每次 per occurrence.

As another example, it may generate a reference signal (p) and mirror signal (p _m) of the i-th occurrence, and generates a product (p _i · p _m) for each occurrence. These can then be made paired and the sum of the products from each pair appearing to be zero.

In one aspect, generating the unbiased alignment sequence in step 1302 includes generating P pilot symbols per symbol period in a plurality of symbol periods. Then, step 1303 generates (N-P) orthogonally modulated communication data symbols per symbol period. Transmitting the unbiased calibration sequence in step 1304 includes transmitting N symbols simultaneously per symbol period in a plurality of symbol periods.

In another aspect, step 1303 produces orthogonally modulated communication data. Step 1304 transmits the unbiased calibration sequence in a first plurality of symbol periods, and then transmits the quadrature modulated communication data in a second plurality of symbol periods.

In a different aspect, step 1302 produces a time series of complex plane symbols having equal cumulative power in a plurality of directions in the complex plane. That is, the symbol time series has an accumulated power associated with real axis information in the time domain, and an equal accumulated power associated with the virtual axis information in the time domain. Step 1304 then transmits the symbol time series in a plurality of symbol periods. In another aspect, step 1302 transmits the unbiased alignment sequence expressed as a time series of i complex symbols (a) in the time domain, as follows: a _i (k)·a _i (k) And =0; where k is the number of symbols per symbol period. In one aspect, step 1302 generates a symbol in a plurality of messages having an equal power accumulated in the plurality of messages in a plurality of complex plane directions.

The above flow diagram can also be interpreted as an expression of a machine readable medium having stored therein instructions for transmitting an unbiased communication tuning sequence. The instructions for transmitting a rotational tuning signal should correspond to steps 1300 through 1304, as explained above.

Figure 14 is a flow chart illustrating a method for calculating an unbiased channel estimate. The method begins in step 1400. Step 1402 accepts a non-deviation tuning sequence in a quadrature demodulation receiver having a predetermined reference signal (p) representative of an evenly distributed power uniformly distributed in the complex plane. Step 1404 processes the unbiased calibration sequence to generate a processed symbol (y) representative of the complex plane information in the unbiased calibration sequence. Step 1406 multiplies the processed symbol (y) by the conjugate (p ^* ) of the corresponding reference signal. Step 1408 obtains an unbiased channel estimate (h _u ).

In one aspect, accepting the unbiased calibration sequence in step 1402 includes accepting an unbiased alignment sequence having a plurality of simultaneously accepted predetermined reference signals ( _pn ). Generating the processed symbol (y) in step 1404 includes generating a plurality of processed symbols (y _n ) from the corresponding plurality of reference signals. Multiplying the processed symbol (y) by the conjugate (p ^* ) of the reference signal in step 1406 includes multiplying each processed symbol by its corresponding reference signal conjugate. Then, step 1408 obtains a channel estimate by obtaining a plurality of channel estimates (h _un ) and averages the channel estimate (h _un ) for each value of n.

In another aspect, step 1402 accepts the offset calibration sequence by accepting calibration information in the time domain via an in-phase (I) modulation path having a cumulative power, and via one having an equalization (as above) The quadrature (Q) modulation path of the cumulative power of the I modulated path power is subjected to tuning information in the time domain.

In a different aspect, step 1402 accepts an unbiased calibration sequence having n time series of predetermined reference signals (p _n ) having an accumulated power associated with real axis information in the time domain, and having An equal number of accumulated powers associated with the virtual axis information in the time domain. Step 1404 generates a time series of n processed symbols (y _n ) from a time sequence of reference signals. Step 1406 multiplies each processed symbol in the time series by its corresponding reference signal conjugate. Then, obtaining the channel estimate in step 1408 includes: obtaining a time series having n channel estimates (h _un ); and averaging the n channel estimates.

In one aspect, step 1402 accepts the unbiased calibration sequence as a signal pair comprising a composite value reference signal (p) having a frequency of +f and a composite value image signal having a frequency of -f ( p _m ), where the product (p.p _m ) is zero. For example, i occurrences of the reference signal (p) and the image signal (p _m ) may be generated, wherein the sum of the products (p _i ·p _m ) is zero. Further, the signal pair values p and p _m vary for each occurrence. In another variation, the sum of the products (p _i ·p _im ) is zero by the following steps: the information is accepted as a composite value (representing p) that is constant for each occurrence; and the information is taken as A composite value of 180 旋转 (representing p _m ) is accepted for acceptance each time it appears.

As another example, an acceptable reference signal (p) and mirror signal (p _m) of occurrences of i, for each occurrence, and generates a product (p _i .p _m). Then, the pairs are made paired and the sum of the products from each pair is zero.

In one aspect, step 1402 accepts the unbiased calibration sequence as P pilot symbols per symbol period in a plurality of symbol periods, and step 1408 obtains P unbiased pilot channel estimates. Step 1403 accepts (N-P) orthogonally modulated communication data symbols simultaneously in each symbol period. Step 1405 generates a processed symbol (y _c ) for the communication material in each symbol period. Since step 1410 is such an unbiased channel estimate for each pilot symbol derived by processing the value (y _c) extrapolated channel estimate. Each step 1412 the processed symbol (y _c) multiplying this value by the extrapolation of the channel estimates, to derive the transmitted symbols by a (x).

In another aspect, step 1403, after accepting the unbiased calibration sequence, accepts orthogonally modulated communication data during the symbol period. Step 1405 generates a processed symbol (y _c ) for each communication data symbol, and step 1414 multiplies each processed symbol by the unbiased channel estimate to derive a transmitted symbol (x).

In a different aspect, step 1402 accepts a time series of complex planes having equal cumulative power in a plurality of directions in the complex plane. Alternatively, the unbiased calibration sequence can be expressed as a time series having i complex symbols (a) in the time domain, as follows: sum of a _i (k)·a _i (k) = 0; where k is per The number of symbols in the symbol period.

In one aspect, accepting the unbiased calibration sequence in step 1402 includes accepting symbols in a plurality of messages having equal power accumulated in the plurality of messages in a plurality of complex plane directions.

The above flow diagram can also be interpreted as a representation of a machine readable medium having instructions stored thereon for calculating an unbiased channel estimate. The instructions for calculating the unbiased channel estimate should correspond to steps 1400-1414, as explained above.

Systems, methods, apparatus, and processors have been provided herein for the transmission and reception of quadrature modulated unbiased calibration sequences in a communication device, and the calculation of unbiased channel estimates. Examples of specific communication protocols and formats have been given to illustrate the invention. However, the invention is not limited to these examples. Other variations and embodiments of the invention will be apparent to those skilled in the art.

300‧‧‧Transmission system

302‧‧‧Baseband processor

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Figure 1 is a schematic block diagram of a conventional receiver front end (prior art).

Figure 2 is a schematic diagram illustrating orthogonal non-equilibrium (prior art) on the receiver side.

FIG. 3 is a schematic block diagram showing an exemplary data transmission system.

4 is a schematic block diagram of a system or apparatus for transmitting a non-deviation communication tuning sequence.

FIG. 5A is a diagram showing an unbiased calibration sequence represented by both time domain and frequency domain.

Figures 5B and 5C are graphical representations of the equal accumulation of power evenly distributed across a complex plane.

FIG. 6 is a diagram showing an unbiased calibration sequence enabled as a pilot tone sequence in the time domain.

FIG. 7 is a diagram showing an unbiased calibration sequence of a preamble enabled as a non-predetermined communication material.

FIG. 8 is a diagram showing an unbiased calibration sequence enabled by averaging symbols by means of a plurality of messages.

FIG. 9 is a schematic block diagram showing a processing apparatus for transmitting a non-deviation communication tuning sequence.

Figure 10 is a schematic block diagram of a system for calculating an unbiased channel estimate.

FIG. 11 is a schematic block diagram of a processing apparatus for calculating an unbiased channel estimate.

Figure 12 illustrates the performance achieved by applying the above algorithm to the WiMedia UWB standard.

Figure 13 is a flow chart illustrating a method for transmitting a non-deviation communication tuning sequence.

Figure 14 is a flow chart illustrating a method for calculating an unbiased channel estimate.

300‧‧‧Transmission system

302‧‧‧Baseband processor

306‧‧‧Encoder

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310‧‧‧Interlacer

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314‧‧‧IFFT

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318‧‧‧DAC

322‧‧‧transmitter

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Claims (49)

A method for transmitting a calibration sequence, the method comprising: generating a calibration sequence in an orthogonal modulation transmitter, the calibration sequence representing a plurality of at least three symbols, each symbol representing a composite value, The composite value has an in-phase component value and an orthogonal component value, wherein a sum of squares of the equivalent phase component values is equal to a sum of squares of the orthogonal component values; and the calibration sequence is transmitted. The method of claim 1, wherein transmitting the calibration sequence comprises transmitting: adjusting power in a domain including one of the equivalent phase component values of an in-phase (I) modulation path having a cumulative power, the cumulative power Equal to the sum of the squares of the equivalent phase component values; and the calibration power in the time domain including the orthogonal component values having one of the cumulative powers of the orthogonal (Q) modulation path, the cumulative power is equal to The sum of the squares of the orthogonal component values. The method of claim 1, wherein generating the calibration sequence comprises generating a first calibration sequence comprising a plurality of at least three reference symbols, each symbol representing a reference composite value, and generating comprising a plurality of at least three corresponding mirror images a second calibration sequence, each corresponding mirror symbol representing a corresponding mirror composite value, wherein transmitting the calibration sequence comprises the first calibration sequence with a transmission frequency of +f and the second transmission frequency of -f The sequence is adjusted, and the product of each of the reference composite values and each corresponding mirror composite value is zero. The method of claim 1, wherein generating the calibration sequence comprises generating a first calibration sequence comprising one of a plurality of at least three reference symbols, each symbol Representing a reference composite value, and generating a second calibration sequence comprising a plurality of at least three corresponding mirror symbols, each corresponding mirror symbol representing a corresponding mirror composite value, wherein transmitting the calibration sequence comprises transmitting a frequency of +f The first calibration sequence and the transmission frequency are the second calibration sequence of -f, and wherein the product sum of the reference composite values and the corresponding mirror composite values is zero. The method of claim 4, wherein each reference symbol is different from the adjacent reference symbol. The method of claim 4, wherein each reference symbol is the same as the adjacent reference symbol. The method of claim 1, further comprising: generating the quadrature modulated communication data; and transmitting the orthogonally modulated communication data after transmitting the calibration sequence. A method for calculating a channel estimate, the method comprising: receiving a calibration sequence in a quadrature demodulation receiver, the calibration sequence representing a plurality of at least three symbols, each symbol representing a composite value The composite value has an in-phase component value and an orthogonal component value, wherein a sum of squares of the equivalent phase component values is equal to a sum of squares of the orthogonal component values; and based on the received calibration sequence to obtain a Channel estimate. The method of claim 8, wherein receiving the calibration sequence comprises receiving: adjusting power in a domain including one of the equivalent phase component values of an in-phase (I) modulation path having an accumulated power Equal to the sum of the squares of the equivalent phase components; and The cumulative power is equal to the sum of the squares of the orthogonal component values via the calibration information in the time domain including the orthogonal component values of one of the cumulative powers of the quadrature (Q) modulation path. The method of claim 8, wherein receiving the calibration sequence comprises receiving a first calibration sequence comprising a plurality of at least three reference symbols having a frequency of +f, each symbol representing a reference composite value, and receiving comprises a plurality of At least three second tuning sequences of one of the corresponding mirror symbols having a frequency of -f, each corresponding mirror symbol representing a corresponding mirror composite value, wherein the product of each reference composite value and each corresponding mirror composite value is zero. The method of claim 8, wherein receiving the calibration sequence comprises receiving a first calibration sequence comprising a plurality of at least three reference symbols having a frequency of +f, each symbol representing a reference composite value, and receiving comprises a plurality of At least three second tuning sequences of one of the corresponding mirror symbols having a frequency of -f, each corresponding mirror symbol representing a corresponding mirror composite value, wherein a product sum of the reference composite values and the corresponding mirror composite values is zero. The method of claim 11, wherein each reference symbol is different from the adjacent reference symbol. The method of claim 11, wherein each reference symbol is the same as the adjacent reference symbol. The method of claim 8, further comprising: receiving the quadrature modulated communication data symbols; and deriving the transmitted symbols for each of the communication data symbols based on the channel estimate. The method of claim 8, further comprising: After receiving the calibration sequence, the quadrature modulated communication data symbols are received; one of the processed symbols for each communication data symbol is generated; and each processed symbol is multiplied by the channel estimate to derive a transmitted symbol. A system for transmitting a calibration sequence, the system comprising: a processor configured to generate a calibration sequence, the calibration sequence representing a plurality of at least three symbols, each symbol representing a composite value, The composite value has an in-phase component value and an orthogonal component value, wherein a sum of squares of the equivalent phase component values is equal to a sum of squares of the orthogonal component values; and a transmitter configured to transmit The calibration sequence. The system of claim 16, wherein the transmitter is configured to transmit: the calibration power in the time domain of one of the in-phase (I) modulation paths having one of the cumulative powers, the cumulative power being equal to the equivalent phase component value The sum of squares; and the calibration information in the time domain via a quadrature (Q) modulation path having a cumulative power equal to the sum of the squares of the orthogonal component values. The system of claim 16, wherein the processor generates the calibration sequence by generating a first calibration sequence comprising one of a plurality of at least three reference symbols, each symbol representing a reference composite value, and generating comprising a plurality of a second calibration sequence of at least three corresponding mirror symbols, each corresponding mirror symbol representing a corresponding mirror composite value, wherein the transmitter transmits the calibration sequence by the first calibration sequence with a transmission frequency of +f And transmission frequency The second tuning sequence of -f, and wherein the product of each reference composite value and each corresponding mirror composite value is zero. The system of claim 16, wherein the processor generates the calibration sequence by generating a first calibration sequence comprising one of a plurality of at least three reference symbols, each symbol representing a reference composite value, and generating comprising a plurality of a second calibration sequence of at least three corresponding mirror symbols, each corresponding mirror symbol representing a corresponding mirror composite value, wherein the transmitter transmits the calibration sequence by the first calibration sequence with a transmission frequency of +f And the second calibration sequence having a transmission frequency of -f, and wherein the product sum of the reference composite values and the corresponding mirror composite values is zero. A system as claimed in claim 19, wherein each reference symbol is different from the adjacent reference symbol. A system as claimed in claim 19, wherein each reference symbol is identical to an adjacent reference symbol. The system of claim 16, wherein the processor is further configured to generate quadrature modulated communication data and wherein the transmitter is further configured to transmit the quadrature modulated communication material. A system for calculating a channel estimate, the system comprising: a quadrature demodulation receiver configured to receive a calibration sequence, the calibration sequence representing a plurality of at least three symbols, each symbol Representing a composite value having an in-phase component value and an orthogonal component value, wherein a sum of squares of the equivalent phase component values is equal to a sum of squares of the orthogonal component values; and a processor Configuring to obtain based on the received calibration sequence The channel estimate. A system as claimed in claim 23, wherein the receiver is configured to receive the calibration sequence by receiving: the calibration information in the domain via one of the in-phase (I) modulation paths having a cumulative power, the accumulation The power is equal to the sum of the squares of the equivalent phase component values; and the calibration power is equal to the orthogonal component values via the calibration information in the time domain having a quadrature (Q) modulation path of a cumulative power sum of square. The system of claim 23, wherein the receiver is configured to receive the calibration sequence by receiving a first calibration sequence comprising a plurality of at least three reference symbols having a frequency of +f, each symbol representing a Referring to the composite value, and receiving a second calibration sequence comprising a plurality of at least three corresponding mirror symbols having a frequency of -f, each corresponding mirror symbol representing a corresponding mirror composite value, wherein each reference composite value corresponds to each The product of the mirrored composite value is zero. The system of claim 23, wherein the receiver is configured to receive the calibration sequence by receiving a first calibration sequence comprising a plurality of at least three reference symbols having a frequency of +f, each symbol representing a Referring to the composite value, and receiving a plurality of second correction sequences including at least three corresponding mirror symbols having a frequency of -f, each corresponding mirror symbol representing a corresponding mirror composite value, wherein the reference composite values correspond to the plurality The product of the mirrored composite values is zero. The system of claim 26, wherein each reference symbol is different from the adjacent reference symbol. A system as claimed in claim 26, wherein each reference symbol is identical to an adjacent reference symbol. The system of claim 23, wherein the receiver is further configured to receive the quadrature modulated communication data symbols and wherein the processor is further configured to derive for each communication data symbol based on the channel estimate One of the transmitted symbols. The system of claim 23, wherein the receiver is configured to receive the quadrature modulated communication data symbols after receiving the calibration sequence, and wherein the processor is configured to generate for each communication material One of the symbols is processed with a symbol and each processed symbol is multiplied by the channel estimate to derive a transmitted symbol. A machine readable medium having stored thereon instructions for transmitting a communication adjustment sequence, the instructions comprising: generating a calibration sequence in a quadrature modulation transmitter, the calibration sequence representing a plurality of at least three a symbol, each symbol representing a composite value having an in-phase component value and an orthogonal component value, wherein a sum of squares of the equivalent phase component values is equal to a sum of squares of the orthogonal component values; The tuning sequence is transmitted. A machine readable medium having stored thereon instructions for calculating a channel estimate, the instructions comprising: receiving a calibration sequence in a quadrature demodulation receiver, the calibration sequence representing a plurality of at least three a symbol, each symbol representing a composite value having an in-phase component value and an orthogonal component value, wherein The sum of the squares of the in-phase component values is equal to the sum of the squares of the orthogonal component values; and based on the received tuning sequence to obtain a channel estimate. An apparatus for transmitting a communication tuning sequence, the apparatus comprising: generating means for generating a tuning sequence, the tuning sequence representing a plurality of at least three symbols, each symbol representing a composite value, the composite The value has an in-phase component value and an orthogonal component value, wherein the sum of squares of the equivalent phase component values is equal to the sum of squares of the orthogonal component values; and a transmission component for transmitting the calibration sequence. The apparatus of claim 33, wherein the transmission component comprises: a transmission component for transmitting calibration information in a time domain in a phase having one of the in-phase (I) modulation paths having a cumulative power equal to the equivalent phase a sum of squares of component values; and a transmission component for transmitting calibration information in the time domain via a quadrature (Q) modulation path having a cumulative power equal to the orthogonal components The sum of the squares of the values. The apparatus of claim 33, wherein the generating means comprises means for generating a first tuning sequence comprising one of a plurality of at least three reference symbols, each symbol representing a reference composite value, and generating means for Generating a second tuning sequence comprising a plurality of at least three corresponding mirror symbols, each corresponding mirror symbol representing a corresponding mirror composite value, wherein the transmission component includes a component for transmitting the first tone of the frequency +f And a transmission sequence for transmitting the second calibration sequence with a frequency of -f, and wherein the product of each reference composite value and each corresponding mirror composite value is zero. The apparatus of claim 33, wherein the generating means comprises means for generating a first tuning sequence comprising one of a plurality of at least three reference symbols, each symbol representing a reference composite value, and generating means for Generating a second tuning sequence comprising a plurality of at least three corresponding mirror symbols, each corresponding mirror symbol representing a corresponding mirror composite value, wherein the transmission component includes a component for transmitting the first tone of the frequency +f And a transmission sequence for transmitting the second calibration sequence having a frequency of -f, and wherein a product sum of the reference composite values and the corresponding mirror composite values is zero. The apparatus of claim 36, wherein each reference symbol is different from an adjacent reference symbol. The apparatus of claim 36, wherein each reference symbol is the same as the adjacent reference symbol. The apparatus of claim 33, wherein the generating component comprises means for generating orthogonally modulated communication material, and wherein the transmitting component comprises means for transmitting the orthogonal after transmitting the tuning sequence Modulated communication data. An apparatus for calculating a channel estimate, the apparatus comprising: a receiving component for receiving a calibration sequence, the calibration sequence representing a plurality of at least three symbols, each symbol representing a composite value, the composite value having An in-phase component value and an orthogonal component value, wherein a sum of squares of the equivalent phase component values is equal to a sum of squares of the orthogonal component values; and an obtaining component for selecting based on the received calibration sequence Get a channel estimate. The apparatus of claim 40, wherein the receiving means comprises: means for receiving calibration information via an in-phase (I) modulation path in a time domain having a cumulative power equal to the equivalent phase group a sum of squares of the values; and means for receiving calibration information via one of the orthogonal (Q) modulation paths in the time domain having a cumulative power equal to the orthogonal component values sum of square. The apparatus of claim 40, wherein the receiving component comprises means for receiving a first tuning sequence comprising a plurality of at least three reference symbols having a frequency of +f, each symbol representing a reference composite value, and the symbol The receiving component includes a component for receiving a second tuning sequence including a plurality of at least three corresponding mirror symbols having a frequency of -f, each corresponding mirror symbol representing a corresponding mirror composite value, wherein each reference composite value is The product of each corresponding mirror composite value is zero. The apparatus of claim 40, wherein the receiving component comprises means for receiving a first tuning sequence comprising a plurality of at least three reference symbols having a frequency of +f, each symbol representing a reference composite value, and the symbol The receiving component includes a component for receiving a second tuning sequence including a plurality of at least three corresponding mirror symbols having a frequency of -f, each corresponding mirror symbol representing a corresponding mirror composite value, wherein the reference composite values are The product sum of the corresponding mirror composite values is zero. The apparatus of claim 43, wherein each reference symbol is different from the adjacent reference symbol. The apparatus of claim 43, wherein each reference symbol and adjacent reference symbol the same. The apparatus of claim 40, wherein the receiving component comprises means for receiving orthogonally tuned communication data symbols, and wherein the obtaining means includes means for deriving for each based on the channel estimate One of the communication data symbols is transmitted by a symbol. The apparatus of claim 40, wherein the receiving means includes means for receiving the orthogonally modulated communication data symbols, and the obtaining means includes means for generating a processed symbol for each of the communication data symbols And includes means for multiplying each processed symbol by the channel estimate to derive a transmitted symbol. A processing device for transmitting a calibration sequence, the processing device comprising: a generation module configured to generate a calibration sequence, the calibration sequence representing a plurality of at least three symbols, each symbol representing a A composite value having an in-phase component value and an orthogonal component value, wherein the sum of squares of the equivalent phase component values is equal to the sum of squares of the orthogonal component values. A processing apparatus for calculating a channel estimate, the processing apparatus comprising: a receiver module configured to receive a calibration sequence, the calibration sequence representing a plurality of at least three symbols, each symbol representing a composite value having an in-phase component value and an orthogonal component value, wherein a sum of squares of the equivalent phase component values is equal to a sum of squares of the orthogonal component values; and a obtaining module A channel estimate is obtained based on the received calibration sequence.