JPWO2020208785A1 - Motor drive, electric blower, vacuum cleaner and hand dryer - Google Patents

Abstract

An inverter (11) that outputs an AC voltage to the single-phase motor (12), a control unit (25) that controls the AC voltage output by the inverter (11), and a current that detects the current flowing through the single-phase motor (12). The detection unit (22) is provided, and the control unit (25) is calculated by the product of a sinusoidal wave having a phase based on a change in the polarity of an AC voltage and a current detected by the current detection unit (22). The rotation speed of the single-phase motor (12) is controlled according to the physical quantity.

Description

The present invention relates to a motor drive device for driving a single-phase motor, an electric blower, an electric vacuum cleaner, and a hand dryer.

Conventionally, there are various types of motors such as a DC motor with a brush, an induction motor, and a PM (Permanent Magnet) motor, and the number of phases of the motor also includes types such as single-phase and three-phase. Among these various motors, the single-phase PM motor has a brushless structure that does not use a brush, which is a mechanical structure, as compared with a DC motor with a brush, so that brush wear does not occur. Due to this feature, the single-phase PM motor can ensure a long life and high reliability. Further, the single-phase PM motor is a highly efficient motor because a secondary current does not flow in the rotor as compared with an induction motor.

The single-phase PM motor has the following advantages as compared with the three-phase PM motor having different numbers of phases.
(1) In the case of a three-phase PM motor, a three-phase inverter is required, whereas in a single-phase PM motor, a single-phase inverter may be used.
(2) When a full-bridge inverter generally used as a three-phase inverter is used, six switching elements are required, whereas in the case of a single-phase PM motor, four switching elements are required even if a full-bridge inverter is used. Can be configured with.
(3) Due to the features of (1) and (2), the single-phase PM motor can be made smaller than the three-phase PM motor.
Patent Document 1 discloses a technique relating to a drive system for a single-phase PM motor.

Japanese Unexamined Patent Publication No. 2012-13378

However, Patent Document 1 includes "a control means for controlling the energization amount to the electric blower, and estimates and estimates the air volume from the relationship of" energization amount-current-air volume "preliminarily obtained by an experiment or the like. When the air volume is within the first predetermined range, as the estimated air volume decreases, the energization amount to the electric blower is controlled to decrease, and the energization amount is the first predetermined range. In the air volume region within the range, the degree of vacuum in the dust collecting chamber is controlled to be substantially constant and to be a value set in advance by an experiment or the like. " That is, in Patent Document 1, the air volume by the electric blower is determined by the work amount of the electric blower, and is determined by the active power from the viewpoint of the electric energy in the electric blower.

In the technique of Patent Document 1, the energization amount is controlled according to the estimated air volume, but the control is not performed from the viewpoint of active power and ineffective power. Further, in the technique of Patent Document 1, although the apparent power is controlled only by the amount of energization, it is not possible to individually control the required active power. Therefore, the technique of Patent Document 1 has a problem that the current flowing through the electric motor becomes larger than the maximum efficiency point and the efficiency deteriorates.

The present invention has been made in view of the above, and an object of the present invention is to obtain a motor drive device for driving a single-phase motor by controlling active power and reactive power.

In order to solve the above-mentioned problems and achieve the object, the motor drive device according to the present invention includes an inverter that outputs an AC voltage to a single-phase motor, a control unit that controls the AC voltage output by the inverter, and a single-phase motor. It is provided with a current detection unit that detects the current flowing through the motor. The control unit controls the rotation speed of the single-phase motor according to the physical quantity calculated by the product of the sine wave based on the phase based on the change in the polarity of the AC voltage and the current detected by the current detection unit.

The motor drive device according to the present invention has the effect of being able to drive a single-phase motor by controlling active power and reactive power.

The figure which shows the structural example of the motor drive system which includes the motor drive device which concerns on Embodiment 1. The figure which shows the example of the circuit configuration of the inverter shown in FIG. A block diagram showing a functional part that generates a PWM signal among the functional parts of the control unit shown in FIG. A block diagram showing a configuration example of the carrier comparison unit shown in FIG. A time chart showing an example of waveforms of a main part in the carrier comparison part shown in FIG. A block diagram showing another configuration example of the carrier comparison unit shown in FIG. A time chart showing an example of waveforms of a main part in the carrier comparison part shown in FIG. The figure which shows the phase difference of the voltage applied to a single-phase motor from the inverter which concerns on Embodiment 1 and the current flowing through a single-phase motor on the rotating coordinate. The figure which shows the state which added the auxiliary winding to the single-phase motor which concerns on Embodiment 1. The figure which shows the voltage, the current, the active current and the reactive current when the current phase difference is 0 degree in the motor drive apparatus which concerns on Embodiment 1. The figure which shows the voltage, the current, the active current and the reactive current when the current phase difference is 30 degrees in the motor drive apparatus which concerns on Embodiment 1. The figure which shows the voltage, the current, the active current and the reactive current when the current phase difference is 60 degrees in the motor drive apparatus which concerns on Embodiment 1. The figure which shows the 1st configuration example of the control part which concerns on Embodiment 1. The figure which shows the 2nd structural example of the control part which concerns on Embodiment 1. The figure which shows the relationship of voltage command, active voltage, invalid voltage and phase in the motor drive device which concerns on Embodiment 1. The figure which shows the structural example of the control cycle determination part shown by FIG. 13 and FIG. The figure which shows the relationship between the control cycle and the trigonometric function of the control part which concerns on Embodiment 1. The figure which shows the 3rd structural example of the control part which concerns on Embodiment 1. The figure which shows the example of the change of efficiency when the reactive current is increased or decreased when the active power is set to a constant value in the control part which concerns on Embodiment 1. The figure which shows the relationship between the motor work and the copper loss in the motor drive device which concerns on Embodiment 1. The figure which shows the state of the change of the rotation speed, the active power and the ineffective power when the sealing is detected in the motor drive device which concerns on Embodiment 1. The figure which shows the 4th structural example of the control part which concerns on Embodiment 1. A flowchart showing the operation of the control unit according to the first embodiment. The figure which shows an example of the hardware composition which realizes the control part provided in the motor drive device which concerns on Embodiment 1. The figure which shows the structural example of the electric blower provided with the motor drive device which concerns on Embodiment 2. The figure which shows the structural example of the electric vacuum cleaner provided with the electric blower which concerns on Embodiment 2. The figure which shows the structural example of the hand dryer provided with the electric blower which concerns on Embodiment 2.

Hereinafter, the motor drive device, the electric blower, the vacuum cleaner, and the hand dryer according to the embodiment of the present invention will be described in detail with reference to the drawings. The present invention is not limited to this embodiment. Further, in the following, the description will be simply referred to as "connection" without distinguishing between an electrical connection and a physical connection.

Embodiment 1.
FIG. 1 is a diagram showing a configuration example of a motor drive system 1 including a motor drive device 2 according to the first embodiment of the present invention. The motor drive system 1 shown in FIG. 1 includes a single-phase motor 12, a motor drive device 2, a battery 10, a voltage sensor 20, a position sensor 21, and a switch 102.

The motor drive device 2 supplies AC power to the single-phase motor 12 to drive the single-phase motor 12. The battery 10 is a DC power source that supplies DC power to the motor drive device 2. The voltage sensor 20 detects the _{DC voltage Vdc} output from the battery 10 to the motor drive device 2. The position sensor 21 detects the rotor rotation position, which is the rotation position of the rotor 12a built in the single-phase motor 12.

The single-phase motor 12 is used as a rotary electric machine for rotating an electric blower (not shown). The single-phase motor 12 and the electric blower are mounted on devices such as a vacuum cleaner and a hand dryer.

In the present embodiment, the voltage sensor 20 _{detects the DC voltage V dc} , but the detection target of the voltage sensor 20 is not limited to the _{DC voltage V dc output from the battery 10.} The detection target of the voltage sensor 20 may be the inverter output voltage, which is the output voltage of the motor drive device 2. The "inverter output voltage" is synonymous with the "motor applied voltage" described later.

The motor drive device 2 includes an inverter 11, a control unit 25, and a drive signal generation unit 32. The inverter 11 is connected to the single-phase motor 12 and outputs an AC voltage to the single-phase motor 12. A capacitor (not shown) may be inserted between the battery 10 and the inverter 11 for voltage stabilization. The control unit 25 controls the AC voltage output by the inverter 11.

Further, the motor drive device 2 includes a current detection unit 22 for detecting the current flowing through the single-phase motor 12, that is, the motor current. The current detection unit 22 may be arranged anywhere as long as the current flowing through the single-phase motor 12 can be detected. In the motor drive device 2, the current detection unit 22 may be arranged in series with the wiring of the single-phase motor 12, may be arranged in series with the switching element of the inverter 11, or may be arranged in series with the switching element of the inverter 11. Alternatively, it may be placed on the ground line. Further, regarding the current detection method in the current detection unit 22, a method of calculating the current value from Ohm's law by inserting a resistor having a known resistance value and detecting the voltage value, a detection method using a transformer, and a Hall effect are used. The detection method used may be mentioned, but any method may be used as long as the current can be detected. In this embodiment, a method of inserting a transformer current sensor in series with the wiring of the single-phase motor 12 will be described. Further, although the inverter 11 is assumed to be a single-phase inverter, it may be any as long as it can drive the single-phase motor 12.

The control unit 25, the DC voltage V _dc detected by the voltage sensor _20, current I _m detected by the current detection unit _22, the protection signal, the position sensor signal 21a is a rotation position detection signal outputted from the position sensor 21 And the command value output from the switch 102 is input. Examples of the command value include an effective current command value Ip ^* due to torque, a rotation speed command value ω ^{*, and the} like. The control unit 25 generates PWM (Pulse Width Modulation) signals Q1, Q2, Q3, and Q4 based on the direct current voltage V _{dc, the position sensor signal 21a, and the command value.}

The switch 102 is, for example, a physical switch, and is a switch for switching between strong operation and weak operation at hand, which is used for vacuum cleaners and the like. There are various types of the switch 102, such as a push button type and a toggle type, but any shape can be used as long as the user's request can be transmitted to the control unit 25. Further, the switch 102 is not limited to the physical switch, and may be a software process in the case of a configuration in which the command value is automatically switched according to the usage time and the state.

The drive signal generation unit 32 generates drive signals S1, S2, S3, S4 for driving the switching element of the inverter 11 based on the PWM signals Q1, Q2, Q3, Q4 output from the control unit 25. The position sensor signal 21a is a binary digital signal that changes according to the direction of the magnetic flux generated by the rotor 12a. In the motor drive system 1, the position sensor 21 and the position sensor signal 21a are unnecessary when the control unit 25 estimates the position of the rotor 12a using the current value of the single-phase motor 12 and drives the rotor 12a without a position sensor. May be.

The drive signal generation unit 32 converts the PWM signals Q1, Q2, Q3, Q4 output from the control unit 25 into drive signals S1, S2, S3, S4 for driving the inverter 11 and outputs the PWM signals to the inverter 11. do. The drive signal generation unit 32 may have a structure built in the inverter 11 or may be integrated with the control unit 25, and is shown as an example in FIG.

An example of the single-phase motor 12 is a brushless motor. When the single-phase motor 12 is a brushless motor, a plurality of permanent magnets (not shown) are arranged in the circumferential direction on the rotor 12a of the single-phase motor 12. These plurality of permanent magnets are arranged so that the magnetizing directions are alternately reversed in the circumferential direction, and form a plurality of magnetic poles of the rotor 12a. A winding (not shown) is wound around the stator 12b of the single-phase motor 12. An alternating current flows through the winding. The current flowing through the winding of the single-phase motor 12 is appropriately referred to as "motor current". In the present embodiment, the number of magnetic poles of the rotor 12a is assumed to be four poles, but the number of magnetic poles of the rotor 12a may be other than four poles.

FIG. 2 is a diagram showing an example of the circuit configuration of the inverter 11 shown in FIG. The inverter 11 has a plurality of bridge-connected switching elements 51, 52, 53, 54. The switching elements 51 and 52 form the first leg 5A. In the first leg 5A, the switching element 51 and the switching element 52 are connected in series. The switching elements 53 and 54 form the second leg 5B. In the second leg 5B, the switching element 53 and the switching element 54 are connected in series.

The switching elements 51 and 53 are located on the high potential side, and the switching elements 52 and 54 are located on the low potential side. In an inverter circuit, the high potential side is generally referred to as an "upper arm" and the low potential side is referred to as a "lower arm". In the following description, the switching element 51 of the first leg 5A may be referred to as an "upper arm first element", and the switching element 53 of the second leg 5B may be referred to as an "upper arm second element". Further, the switching element 52 of the first leg 5A may be referred to as a "lower arm first element", and the switching element 54 of the second leg 5B may be referred to as a "lower arm second element".

The connection point 6A between the switching element 51 and the switching element 52 and the connection point 6B between the switching element 53 and the switching element 54 form an AC end in the bridge circuit. A single-phase motor 12 and a current detection unit 22 for detecting the current flowing through the single-phase motor 12 are connected between the connection point 6A and the connection point 6B. As described above, the current detection unit 22 may be inserted anywhere as long as the current flowing through the single-phase motor 12 can be detected, and the method for detecting the current in the current detection unit 22 is not limited.

For each of the plurality of switching elements 51, 52, 53, 54, a MOSFET (Metal-Oxide-Semiconductor Field-Effective Transistor), which is a metal oxide film semiconductor field effect transistor, is used. MOSFET is an example of FET (Field-Effective Transistor).

The switching element 51 is formed with a body diode 51a connected in parallel between the drain and the source of the switching element 51. The switching element 52 is formed with a body diode 52a connected in parallel between the drain and the source of the switching element 52. The switching element 53 is formed with a body diode 53a connected in parallel between the drain and the source of the switching element 53. The switching element 54 is formed with a body diode 54a connected in parallel between the drain and the source of the switching element 54. Each of the plurality of body diodes 51a, 52a, 53a, 54a is a parasitic diode formed inside the MOSFET and is used as a freewheeling diode. A freewheeling diode may be connected separately, or an IGBT (Insulated Gate Bipolar Transistor) may be used instead of the MOSFET.

The plurality of switching elements 51, 52, 53, 54 are not limited to MOSFETs formed of silicon-based materials, and may be MOSFETs formed of silicon carbide, gallium nitride-based materials, or wide bandgap semiconductors such as diamond. At least one of the plurality of switching elements 51, 52, 53, 54 may be formed of a wide bandgap semiconductor.

In general, wide bandgap semiconductors have higher withstand voltage and heat resistance than silicon semiconductors. Therefore, by using the wide bandgap semiconductors for the plurality of switching elements 51, 52, 53, 54, the withstand voltage resistance and the allowable current density of the switching elements are increased, and the semiconductor module incorporating the switching elements can be miniaturized. In addition, since the wide bandgap semiconductor has high heat resistance, it is possible to reduce the size of the heat dissipation part for dissipating the heat generated by the semiconductor module, and the heat dissipation structure for dissipating the heat generated by the semiconductor module is simple. It is possible to change.

FIG. 3 is a block diagram showing a functional part that generates PWM signals Q1, Q2, Q3, and Q4 among the functional parts of the control unit 25 shown in FIG. In FIG. 3, a _{voltage phase θ v} used when generating a _{voltage command V ref,} which will be described later, is input to the carrier comparison unit 38. Further, the carrier generated by the carrier generation unit 33, the DC voltage V _dc, ^{and the voltage command V *} , which is the amplitude value of the voltage command V _ref , are input to the carrier comparison unit 38. The carrier comparison unit 38 generates PWM signals Q1, Q2, Q3, and Q4 based on the carrier, voltage phase θ _v , DC voltage V _dc, and voltage command V ^*.

FIG. 4 is a block diagram showing a configuration example of the carrier comparison unit 38 shown in FIG. FIG. 4 shows a detailed configuration of the carrier comparison unit 38A and the carrier generation unit 33, which are examples of the carrier comparison unit 38. In FIG. 4, the carrier frequency f _C [Hz], which is the frequency of the carrier, is set in the carrier generation unit 33. At the tip of the arrow of the carrier frequency f _C [Hz], as an example of the carrier waveform, a triangular wave carrier that moves up and down between "0" and "1" is shown. The PWM control of the inverter 11 includes synchronous PWM control and asynchronous PWM control. In the case of synchronous PWM control, it is necessary to synchronize the carrier with _{the voltage phase θ v.} On the other hand, in the case of asynchronous PWM control, it is not necessary to synchronize the carrier with the _{voltage phase θ v.}

As shown in FIG. 4, the carrier comparison unit 38A has an absolute value calculation unit 38a, a division unit 38b, a multiplication unit 38c, a multiplication unit 38d, a multiplication unit 38f, an addition unit 38e, a comparison unit 38g, a comparison unit 38h, and an output inversion. It has a unit 38i and an output inversion unit 38j.

The absolute value calculation unit 38a calculates the absolute value | V ^* | of ^{the voltage command V *.} The division unit 38b divides the absolute value | V ^* | by the DC voltage V _dc detected by the voltage sensor 20. In the configuration shown in FIG. 4, the output of the division unit 38b is the modulation factor. The battery voltage, which is the output voltage of the battery 10, fluctuates as the current continues to flow. Therefore, in the carrier comparison unit 38A, the value ^{of the modulation factor can be adjusted by dividing the absolute value | V *} | by the DC voltage V _dc , and the motor applied voltage can be prevented from decreasing due to the decrease in the battery voltage.

The multiplication unit 38c calculates a sine value of the _{voltage phase θ v.} The multiplication unit 38c multiplies _{the sine value of the voltage phase θ v} by the modulation factor which is the output of the division unit 38b. _{The multiplication unit 38d multiplies the voltage command V ref} , which is the output of the multiplication unit 38c, by “1/2”. The addition unit 38e adds "1/2" to the output of the multiplication unit 38d. The multiplication unit 38f multiplies the output of the addition unit 38e by "-1". The output of the addition unit 38e is input to the comparison unit 38g as _{a positive voltage command V ref1} for driving the two switching elements 51, 53 of the upper arm among the plurality of switching elements 51, 52, 53, 54. .. The output of the multiplication unit 38f is input to the comparison unit 38h as _{a negative voltage command V ref2} for driving the two switching elements 52 and 54 of the lower arm.

The comparison unit 38g compares the positive voltage command V _{ref 1} with the amplitude of the carrier. The output of the output inversion unit 38i in which the output of the comparison unit 38g is inverted becomes the PWM signal Q1 to the switching element 51, and the output of the comparison unit 38g becomes the PWM signal Q2 to the switching element 52. Similarly, the comparison unit 38h compares the negative voltage command V _{ref 2} with the carrier amplitude. The output of the output inversion unit 38j, which is the inverted output of the comparison unit 38h, is the PWM signal Q3 to the switching element 53, and the output of the comparison unit 38h is the PWM signal Q4 to the switching element 54. The output inversion unit 38i does not turn on the switching element 51 and the switching element 52 at the same time. The output inversion unit 38j does not turn on the switching element 53 and the switching element 54 at the same time.

FIG. 5 is a time chart showing a waveform example of a main part in the carrier comparison unit 38A shown in FIG. _{In FIG. 5, the waveform of the positive voltage command V ref1} output from the addition unit 38e, the waveform of the negative voltage command V _ref2 output from the multiplication unit 38f, and the waveforms of the PWM signals Q1, Q2, Q3, and Q4 are shown. And the waveform of the inverter output voltage are shown.

PWM signal Q1 is "high (High)" when "low (Low)" next when the positive voltage command _{V ref1} is greater than the carrier, the positive voltage command _{V ref1} is smaller than the carrier. The PWM signal Q2 is an inverted signal of the PWM signal Q1. PWM signal Q3 is "high (High)" when "low (Low)" becomes when negative voltage instruction _{V ref2} is larger than the carrier, the negative-side voltage instruction _{V ref2} smaller than the carrier. The PWM signal Q4 is an inverted signal of the PWM signal Q3. As described above, the circuit shown in FIG. 4 is configured with "Low Active", but even if each signal is configured with "High Active" having opposite values. good.

As shown in FIG. 5, the waveform of the inverter output voltage shows a voltage pulse due to the difference voltage between the PWM signal Q1 and the PWM signal Q4 and a voltage pulse due to the difference voltage between the PWM signal Q3 and the PWM signal Q2. These voltage pulses are applied from the inverter 11 to the single-phase motor 12 as the motor applied voltage.

Bipolar modulation and unipolar modulation are known as modulation methods used by the carrier comparison unit 38A to generate PWM signals Q1, Q2, Q3, and Q4. Bipolar modulation is a modulation method that outputs a voltage pulse that changes with a positive or negative potential for each cycle of the voltage command V _ref. Unipolar modulation is a modulation method that outputs a voltage pulse that changes at three potentials in each cycle of the voltage command V _ref , that is, a voltage pulse that changes between a positive potential, a negative potential, and a zero potential. The waveform shown in FIG. 5 is due to unipolar modulation. In the motor drive device 2 of the present embodiment, any modulation method may be used. In applications where it is necessary to control the motor current waveform to a more sinusoidal wave, it is preferable to adopt unipolar modulation having a lower harmonic content than bipolar modulation.

Further, the waveform shown in FIG. 5 shows the switching elements 51 and 52 constituting the first leg 5A and the switching elements 53 and 54 constituting the second leg 5B in the section of the half cycle T / 2 of the _{voltage command V ref.} It is obtained by a method of switching operation of the four switching elements of. This method is called "both-sided PWM" because the switching operation is performed by both the positive side voltage command V _ref1 and the negative side voltage command V _ref2. On the other hand, _{in one half cycle of one cycle T of the voltage command V ref} , the switching operation of the switching elements 51 and 52 is suspended, and in the other half cycle of the one cycle T of the _{voltage command V ref, the switching operation is suspended.} There is also a method of suspending the switching operation of the switching elements 53 and 54. This method is called "one-sided PWM". Hereinafter, "one-sided PWM" will be described.

FIG. 6 is a block diagram showing another configuration example of the carrier comparison unit 38 shown in FIG. FIG. 6 shows an example of a PWM signal generation circuit by the above-mentioned “one-sided PWM”, and specifically, a detailed configuration of a carrier comparison unit 38B and a carrier generation unit 33, which are examples of the carrier comparison unit 38, is shown. Has been done. The configuration of the carrier generation unit 33 shown in FIG. 6 is the same as or equivalent to that shown in FIG. Further, in the configuration of the carrier comparison unit 38B shown in FIG. 6, the same or equivalent components as the carrier comparison unit 38A shown in FIG. 4 are designated by the same reference numerals.

As shown in FIG. 6, the carrier comparison unit 38B has an absolute value calculation unit 38a, a division unit 38b, a multiplication unit 38c, a multiplication unit 38k, an addition unit 38m, an addition unit 38n, a comparison unit 38g, a comparison unit 38h, and an output inversion. It has a unit 38i and an output inversion unit 38j.

The absolute value calculation unit 38a calculates the absolute value | V ^* | of ^{the voltage command V *.} The division unit 38b divides the absolute value | V ^* | by the DC voltage V _dc detected by the voltage sensor 20. Even in the configuration of FIG. 6, the output of the division unit 38b is the modulation factor.

The multiplication unit 38c calculates a sine value of the _{voltage phase θ v.} The multiplication unit 38c multiplies _{the sine value of the voltage phase θ v} by the modulation factor which is the output of the division unit 38b. _{The multiplication unit 38k multiplies the voltage command V ref} , which is the output of the multiplication unit 38c, by “-1”. The addition unit 38m adds “1” to the _{voltage command V ref} which is the output of the multiplication unit 38c. The addition unit 38n adds "1" to the output of the multiplication unit 38k, that is, the inverted output of the _{voltage command V ref.} The output of the addition unit 38m is input to the comparison unit 38g as _{the first voltage command V ref3} for driving the two switching elements 51, 53 of the upper arm among the plurality of switching elements 51, 52, 53, 54. .. The output of the addition unit 38n is input to the comparison unit 38h as a _{second voltage command V ref 4} for driving the two switching elements 52 and 54 of the lower arm.

The comparison unit 38g compares the first voltage command V _{ref 3} with the amplitude of the carrier. The output of the output inversion unit 38i in which the output of the comparison unit 38g is inverted becomes the PWM signal Q1 to the switching element 51, and the output of the comparison unit 38g becomes the PWM signal Q2 to the switching element 52. Similarly, the comparison unit 38h compares the second voltage command V _{ref 4} with the amplitude of the carrier. The output of the output inversion unit 38j, which is the inverted output of the comparison unit 38h, is the PWM signal Q3 to the switching element 53, and the output of the comparison unit 38h is the PWM signal Q4 to the switching element 54. The output inversion unit 38i does not turn on the switching element 51 and the switching element 52 at the same time. The output inversion unit 38j does not turn on the switching element 53 and the switching element 54 at the same time.

FIG. 7 is a time chart showing a waveform example of a main part in the carrier comparison unit 38B shown in FIG. _{FIG. 7 shows} the waveform of the first voltage command V ref3 output from the adder 38m, the waveform of the second voltage command V _ref4 output from the adder 38n, and the waveforms of the PWM signals Q1, Q2, Q3, and Q4. And the waveform of the voltage applied to the motor are shown. In FIG. 7, for convenience, _{the waveform portion of the first voltage command V ref3} whose amplitude value is larger than the peak value of the carrier and the second voltage command V _{ref 4} whose amplitude value is larger than the peak value of the carrier. The corrugated portion is represented by a flat straight line.

PWM signal Q1 is "low (Low)" next when the first voltage command _{V ref3} is larger than the carrier, the first voltage command _{V ref3} is "high (High)" when less than the carrier. The PWM signal Q2 is an inverted signal of the PWM signal Q1. PWM signal Q3, the second voltage command _{V ref4} is "high (High)" when "low (Low)" becomes when larger than the carrier, the second voltage command _{V ref4} smaller than the carrier. The PWM signal Q4 is an inverted signal of the PWM signal Q3. As described above, the circuit shown in FIG. 6 is configured with "Low Active", but even if each signal is configured with "High Active" having opposite values. good.

As shown in FIG. 7, the waveform of the inverter output voltage shows a voltage pulse due to the difference voltage between the PWM signal Q1 and the PWM signal Q4 and a voltage pulse due to the difference voltage between the PWM signal Q3 and the PWM signal Q2. These voltage pulses are applied from the inverter 11 to the single-phase motor 12 as the motor applied voltage.

In the waveforms shown in FIG. 7, in one half cycle of one cycle T of the voltage command V _ref, the switching operation of the switching elements 51 and 52 are at rest, the other of the one period T of the voltage command V _ref In the half cycle, the switching operation of the switching elements 53 and 54 is suspended.

Further, as shown in FIG. 7, the waveform of the inverter output voltage is unipolar modulation that changes at three potentials in each cycle _{of the voltage command V ref.} As described above, bipolar modulation may be used instead of unipolar modulation, but in applications where it is necessary to control the motor current waveform to a more sinusoidal wave, it is preferable to adopt unipolar modulation.

The carrier generation unit 33 generates a triangular wave of peaks and valleys from a desired frequency fc as a carrier used for generating these PWM signals Q1, Q2, Q3, and Q4. The frequency fc is variable according to the rotation speed of the single-phase motor 12, and the carrier generation unit 33 increases the frequency fc of the carrier as the rotation speed of the single-phase motor 12 increases. It is desirable that the carrier generation unit 33 generate carriers at a frequency fc that is at least twice the rotation speed of the single-phase motor 12. The carrier generation unit 33 may generate a carrier, such as a sawtooth wave, instead of a triangular wave.

Figure 8 is a diagram showing the phase difference of the current I _m flowing through the voltage V _m and the single-phase motor 12 is applied from the inverter 11 according to the first embodiment to the single-phase motor 12 on the rotating coordinates. Voltage V _m and current I _m can be expressed by the following equation for an alternating respectively (1) and (2).

V _m = V _{m_amp} sinθ _v ... (1)
_{I m = I m_amp sin (θ} v + α) ... (2)

Here, V _{m_amp} is the voltage waveform amplitude, and _{Im_amp} is the current waveform amplitude. Further, theta _v is the voltage phase, i.e. _{θ v = ω e t, ω} e is the electrical angular velocity, i.e. ω _e = 2πf _{e, f e} is the electrical angular frequency. Moreover, alpha is a current phase difference of the current I _m with respect to the voltage V _m. The voltage phase θ _v is a phase that rotates at a certain electric angular frequency _{fe from 0 to 360 degrees.} The voltage phase θ _v is a phase based on a change in the polarity of the _{voltage V m,} that is, the AC voltage applied to the single-phase motor 12. The current phase difference α represents the delay of the current waveform with respect to the voltage waveform, and is a constant in the steady state. The following equations multiplying sin [theta _v and cos [theta] _v of the voltage _{V m} and the phase current _{I m} (3) and (4) is obtained. Equation (3) is a physical quantity calculated by the product of the sine wave, the current I _m that is detected by the current detection unit 22 according to the phase and base changes in the polarity of the AC voltage. Further, Equation (4) is a cosine wave by a phase in which the base point change in polarity of the AC voltage, which is a physical quantity calculated by the product of the current I _m that is detected by the current detection unit 22.

From equation (3) and (4), the second term and the third term can be confirmed to contain the frequency component of the 2 [Theta] _v. Further, the first term is a constant including cosα or sinα, and is a DC component not including a frequency component. α is because it is delayed component of the current I _m with respect to the voltage V _m, cos [alpha] represents the power factor. Generally, the active current and reactive current that contribute to the rotation of the motor are represented by the following equations (5) and (6).

Active current I _p = I _{m_rms} cos α… (5)
Reactive current I _q = I _{m_rms} sin α… (6)

Here, _{Im_rms} is an effective value of the current waveform, that is, _{Im_rms} = _{Im_amp} / (√2). Note that √2 indicates the square root of 2. The equation (5) and (6), the first term of equation (3) is 1 / √2 times the effective current of the current I _m flowing through the single-phase motor 12, the first term of equation (4) is indicates that a 1 / √2 times the reactive current of the current I _m flowing through the single-phase motor 12. Therefore, the control unit 25, the calculation result of equation (3) and (4) to remove the component of twice the frequency 2 [Theta] _v of the fundamental wave by a signal processing such as a low-pass filter, the effective current I to √2 times the value Treat as _p and invalid current I _q. The voltage phase θ _v is known because it is the phase of the voltage V _m. Control unit 25, a current sensor a current I _m, and calculates the equations (3) and (4) by estimating acquired from the current detecting unit 22 such as a shunt resistor, or an observer device and the like.

The above-mentioned low-pass filter can reduce the frequency twice that of the fundamental wave from the equations (3) and (4), and as long as the DC component can be extracted, there are no restrictions on the form and coefficient of the filter. Since the frequency to be reduced increases as the rotation speed of the single-phase motor 12 increases, a low-pass filter having a variable band to be reduced with respect to the rotation speed of the single-phase motor 12 is designed. If possible, a low-pass filter that has a small delay in the signal output from the low-pass filter and can attenuate the band other than the DC component as much as possible is ideal.

The method for calculating the active current I _p and the reactive current I _q is not limited to the above. The control unit 25 calculates, for example, a virtual current having a phase difference of 90 degrees with respect to the detected current. The control unit 25 can calculate the _{effective current Ip} as a constant by integrating the _{voltage phase waveform sin θ v} into the waveform obtained by adding the detected current and the virtual current. In the control unit 25, as a method for calculating a virtual current is to use a differentiation result or the integration result of the current I _m. Since the differential of the alternating current advances the alternating current phase by 90 degrees and the integral of the alternating current delays the alternating current phase by 90 degrees, the control unit 25 adjusts the coefficient as shown in the following equation (7) to 90 degrees with the same amplitude. It is possible to calculate a virtual current with a phase difference. As described above, a θ _{v =} ω _{e t,} the voltage phase theta _v is a variable varies with time t.

That is, the control unit 25 calculates the first and second currents having a phase difference from the first current is the current I _m that is detected by the current detection unit 22, the first current and the second The rotation speed of the single-phase motor 12 is controlled by using an electric current. In addition to this, the control unit 25 may _{store the detected current Im} for each cycle and calculate a waveform having a phase difference of 90 degrees from the waveform saved in the next cycle. Further, in the motor drive device 2, by adding an auxiliary winding having a phase difference of 90 degrees physically to the single-phase motor 12, it is possible for the current detection unit 22 to detect a current having a phase difference. .. FIG. 9 is a diagram showing a state in which the auxiliary winding 101 is added to the single-phase motor 12 according to the first embodiment. FIG. 9 shows a state in which an auxiliary winding 101 having a phase difference of 90 degrees is added to the main winding 100 in the single-phase motor 12. In this case, the current detection unit 22 detects the second current having a phase difference relative to the first current and the first current is the current I _m. The control unit 25 controls the rotation speed of the single-phase motor 12 by using the first current and the second current.

As can be seen from the equations (5) and (6), when α = 0, that is, when the power factor cos α is large, the active current I _p becomes large and the reactive current I _q becomes small. This is a state where the current I _m is delayed 0 to the phase of the voltage V _m, it is an expression that delay increases the proportion of reactive current I _q as the larger (5) and (6), further drawing It can be confirmed from 10 to FIG. FIG. 10 is a diagram showing _{a voltage V m} , a current _Im , an active current I _p, and a reactive current I _q when the current phase difference is 0 degrees in the motor drive device 2 according to the first embodiment. FIG. 11 is a diagram showing _{a voltage V m} , a current _Im , an active current I _p, and a reactive current I _q when the current phase difference is 30 degrees in the motor drive device 2 according to the first embodiment. FIG. 12 is a diagram showing _{a voltage V m} , a current _Im , an active current I _p, and a reactive current I _q when the current phase difference is 60 degrees in the motor drive device 2 according to the first embodiment. As shown in FIGS. 10 to 12, the larger the current phase difference α, the smaller the _{effective current Ip.}

The active power P is calculated by multiplying _{the active current I p} obtained from the equation (3) by the effective value V _{m_rms} = V _{m_amp} / √2 of the voltage V _m. The following equation (8) is an equation meaning active power and is generally shown.

Active power P = V _{m_rms} I _{m_rms cosα} = (V _{m_amp} / √2) (I _{m_amp} / √2) cosα… (8)

Assuming that the parasitic impedance between the inverter 11 and the single-phase motor 12 is small and does not affect the control, the active power P calculated by the equation (8) and the inside of the single-phase motor 12 are implemented. The motor work W becomes equal. For the sake of simplicity, copper loss, iron loss and mechanical loss are omitted as minute values.

Active power P = motor power W + copper loss + iron loss + mechanical loss _{≒ e m I m + R m} I m 2 ... (9)

Here, _{e m} is the motor induced voltage _{e m = P n ω m φ} f sinω m t, P n is the number of pole pairs, omega _m is the rotation speed, phi _f is the flux linkage, R _m is the internal resistance of the single-phase motor 12. From equation (8), to increase the active power P, it is possible to increase the work e _m I _m of a single-phase motor 12. At this time, an increase in the current I _m denotes an increase in the torque of the single-phase motor 12, the increase in the motor induced voltage e _m is meant an increase in the rotational speed omega _m of the single-phase motor 12. The control that realizes the increase / decrease of the active power P and the active power P is defined as PQ control.

A specific configuration of the control unit 25 that realizes PQ control using the calculated active current I _p and reactive current I _{q will be described.} FIG. 13 is a diagram showing a first configuration example of the control unit 25 according to the first embodiment. Further, FIG. 14 is a diagram showing a second configuration example of the control unit 25 according to the first embodiment.

FIG. 13 is a configuration for the purpose of speed control, and the command value input to the control unit 25 is the rotation speed command value ω ^* . In FIG. 13, the calculator 45 calculates the effective current command value _Ip ^* required to achieve the desired rotation speed. Detection current signal processing unit 44, the signal processing with respect to the current I _m that is detected by the current detection unit 22, for example, from an analog signal into a digital signal. Control unit 25 multiplies sinθ against current _{I m} after the signal processing, to calculate the actual effective current _{I p} by performing filtering with LPF (Low Pass Filter). Further, the subtractor 71 subtracts the actual active current _{Ip from the active current command value Ip} ^* calculated by the arithmetic unit 45, and outputs the difference to the PI (Proportional Industrial) controller 73. The PI controller 73 calculates the _{effective voltage V p} ^* required to realize the ^{rotation speed command value ω *} , which is the command value. The calculator 46 calculates the reactive current command value I _q ^* required to achieve the desired rotation speed. Control unit 25 multiplies cosθ with respect to the current _{I m} after the signal processing, to calculate the actual reactive current _{I q} by performing a filtering process in LPF. The subtractor 72 subtracts the actual reactive current _{Iq from the reactive current command value Iq} ^* calculated by the arithmetic unit 46, and outputs the difference to the PI controller 74. The PI controller 74 calculates the _{invalid voltage V q} ^* required to realize the ^{rotation speed command value ω *} , which is the command value.

FIG. 15 is a diagram showing the relationship between the ^{voltage command V *} , the effective voltage V _p , the invalid voltage V _q, and the phase θ _{angle in} the motor drive device 2 according to the first embodiment. As shown in FIG. 15, the inverse PQ conversion unit 42 performs inverse PQ conversion using the active voltage V _{p calculated by the PI controller 73 and the invalid voltage V q} calculated by the PI controller 74, as shown below. ^{The voltage command V *} and the voltage phase θ _v are calculated from the equations (10) to (12) of.

V ^* = √ (V _p ² + V _q ² )… (10)
θ _angle = tan ^-1 (V _p / V _q )… (11)
θ _v = θ _e ^ + θ _angle + θ _lead … (12)

Here, θ _angle is the phase of the desired active voltage V _p and invalid voltage V _{q , θ e} ^ is the reference phase, that is, P _n ∫ω ^* dt, and θ _lead is the advance angle for correcting the delay due to arithmetic processing. It is a directive. The notation of "^" in "θ _e ^" should be that the symbol of "^" should be added to the upper part of the character of _{"θ e", but that notation is not possible.} Therefore, in the present specification, except for the formula to be inserted in the image, the character "^" is added after the summary character. As shown in FIG. 13, the control unit 25, the theta _{e ^} is used to multiply the sinθ or cosθ with respect to the current I _m after the signal processing, be calculated using a rotational speed command value omega ^* Can be done. In the control unit 25, the carrier comparison unit 38 inputs the PWM signals Q1, Q2, Q3, and Q4 to the single-phase motor 12 in order to apply the ^{calculated voltage command V *} and the AC voltage V ^* _{sin θ v} by the voltage phase θ _{v to the single-phase motor 12.} It is generated and controls the switching elements 51, 52, 53, 54 of the inverter 11. From the above, the voltage waveform amplitude V _{m_amp} and the voltage command V ^* described above are equal to each other.

The above-mentioned reference phase θ _e ^ is a reference voltage phase when it is assumed that the single-phase motor 12 follows the target rotation speed command value ω ^{* without stepping out.} In the control unit 25, when the power is not controlled, the carrier comparison unit 38 generates PWM signals Q1, Q2, Q3, Q4 from the voltage waveform V ^* sinθ _{e ^ with reference to the reference phase θ e ^, and the inverter 11} The switching elements 51, 52, 53, 54 are controlled. On the other hand, the control unit 25 can control the ratio of the active power P and the active power Q by adding _{the θ angle} to the reference phase θ _{e ^.}

The advance angle command θ _lead described above is inserted in the _{voltage phase θ v} for the purpose of correcting the delay due to the arithmetic processing of the control unit 25. Control unit 25, the actual calculation process performs calculation for each certain control period T _s. In the motor drive system 1, since the single-phase motor 12 is rotated by the time spent in the arithmetic processing of each component of the control unit 25 shown in FIG. 13, it is necessary to correct the delay by this amount. For example, since the control unit 25 _{realizes arithmetic processing for the control cycle T s} and insertion of a value into the next component by one arithmetic (N _step × 1), the following equation (13) is used. The advance command θ _lead is calculated from, and the voltage phase θ _v is corrected.

θ _lead = N _step × ω ^* × T _s … (13)

Step number _{N step} of arithmetic process, a filter other than the arithmetic processing of the PI controller 73 and 74, specifically, by the LPF shown in FIG. 13 delays the detected processed like the various delays of the current _{I m} Factors are assumed. Further, the influence of delay due to processing as a single-phase motor 12 is the difference of the rotation period and the control period T _s faster rotation is reduced increases. In the present embodiment, the correction is performed with the configuration as described in the above equation (13), but the correction method is not limited. Thus, the control unit 25 changes the control period T _s in accordance with the rotational speed of the single-phase motor 12, the phase of the base point change in polarity of the AC voltage, the advance angle command value for correcting a delay in operation Is added.

FIG. 14 has a torque control configuration in which the effective current command value I _p ^* is used as the command value, and the rotation speed ω is random. The control unit 25 maintains the rotation of the single-phase motor 12 by controlling the _{effective current Ip that contributes to the torque.} The control unit 25 acquires the rotation speed ω by using a speed estimator, a position sensor, or the like. FIG. 14 _{shows an example in which the speed estimator 43 estimates the rotation speed ω from the current Im,} and the rotation speed ω estimated by the speed estimator 43 is defined as the rotation speed ω ^. As shown in FIG. 14, the control unit 25, the theta _{e ^} is used to multiply the sinθ or cosθ with respect to the current I _m after the signal processing can be calculated using a rotational speed omega ^ .. The method of controlling the _{active current I p} and the reactive current I _q is the same except that the control target is different from the configuration of FIG. 13.

As described above, the control unit 25 removes twice the frequency of the fundamental wave by using a filter for the calculation results of the equations (3) and (4), and the active current I _p and the reactive current I _q. Only the DC component corresponding to this is extracted. However, when the rotation speed of the single-phase motor 12 is low, the frequency twice the fundamental wave cannot be completely removed by a general filter. Therefore, the calculated active current I _p and reactive current I _q vibrate. The PI controllers 73 and 74 output a voltage command that vibrates at a frequency twice that of the fundamental wave in order to eliminate this vibration. As a result, the PI controllers 73 and 74 themselves vibrate the _{active current I p} and the reactive current I _{q. If the vibrations of the active current I p} and the reactive current I _q are fed back by positive feedback due to the delay due to the current detection, the filter, etc., the PI controllers 73 and 74 may diverge. Therefore, preventing the divergence by delaying the control period T _s. That is, the control unit 25, when detecting the vibration of the physical quantity calculated by the product of the detected current due to the phase in which the base point change in polarity of the AC voltage sine wave and the current detecting section 22, the control period T _s Slow down.

FIG. 16 is a diagram showing a configuration example of the control cycle determination unit 48 shown in FIGS. 13 and 14. As shown in FIG. 16, the control cycle determination unit 48 detects the calculated _{amplitudes of the active current I p} and the reactive current I _q by the amplitude detection unit 105, and determines the _{control cycle T s} from the value having a large amplitude by the selector 108. do. Table of the amplitude and the control period, previously obtained the control period T _s of the control does not collapse experiments in advance and the like.

Here, the formulas for calculating the active current I _p and the reactive current I _{q are reprinted and expanded.} As described above, ω _e is the electric angular velocity and _fe is the electric angular frequency.

_{I m × sinθ v = I m} × sin (ω e t) = I m × sin (2πf e t) ... (14)
_{I m × cosθ v = I m} × cos (ω e t) = I m × cos (2πf e t) ... (15)

As the rotation speed of the single-phase motor 12 increases, the electric angular frequency _{fe of} ^{the applied voltage command V *} also increases. As shown in FIG. 17, when the control period _{T s} is not fast enough, distorted waveform of trigonometric functions sin (2πf _e t) and cos (2πf _e t), the control unit 25, the calculated active current _{I p} and The calculation accuracy of the reactive current _Iq deteriorates, and in the worst case, control failure may occur. Figure 17 is a diagram showing the relationship between the control period _{T s} and trigonometric functions sin (2πf _e t) of the control unit 25 according to the first embodiment. In FIG. 17, at the electric angular frequency f ₁ at low speed, the control cycle T _s 1 was sufficient calculation accuracy, but at the electric angular frequency f ₂ at high speed, the control cycle T _s 1 became the electric angular frequency f ₂ . Since it is relatively slower than that, the calculation accuracy deteriorates. Therefore, the control unit 25 maintains the calculation accuracy by performing the calculation in the _{control cycle T s} 2 which is faster than the _{control cycle T s 1.} Thus, the control unit 25, by accelerating the control period T _s in accordance with the rotational speed of the single-phase motor 12, to ensure the calculation accuracy.

Specifically, the control unit 25, the control cycle determining unit 48 shown in FIG. 16 to determine the control period T _s. As described above, it was necessary to slow down the control cycle T _{s from the vibrations of the active current I p} and the reactive current I _{q during low-speed rotation.} This time, at the time of high speed rotation, it is necessary to increase the control period T _s in accordance with the increase in the rotational speed. Therefore, the control cycle determining section 48, the end with the selector 109, to determine the control period T _s to be used. The selector 109, to determine the control period T _s, the current rotational speed, i.e., referring to the rotation speed omega ^ shown in the rotational speed command value omega ^* or 14 shown in FIG. 13. The selector 109 selects the output of the control period calculating unit 106 according to the amplitude at low speed, the high speed when it selects the output of the control period calculating unit 107 according to the rotation speed is used as control period T _s.

Here, there is a limit to increase the control period T _s. During particularly high-speed rotation exceeding 50,000 rpm, the control period T _s as well deterioration of the operational accuracy due roughness, operation result will be a single-phase motor 12 is rotated in until it is reflected on the motor control, the controlled object There is a problem that the situation changes. As described above, the control unit 25 corrects such a problem by using the _{advance command θ lead.}

FIG. 18 is a diagram showing a third configuration example of the control unit 25 according to the first embodiment. FIG. 18 is a specific configuration of the control unit 25 that realizes the PQ control using the virtual current described above. Here, the control target is the rotation speed command value ω ^* as in FIG. 13, but the control target may be the _{effective current command value I p} ^* that contributes to the torque shown in FIG. When the control target is the effective current command value I _p ^* , the control unit 25 needs to estimate or detect the rotation speed ω ^ as shown in FIG. _{The control unit 25 obtains} an effective current I _p using a virtual current Im_virtual having a phase difference of 90 degrees. Control unit 25 is different from the configuration shown in FIGS. 13 and 14, have added differentiator 47 for performing differential processing with respect to the current I _m. The control unit 25, arithmetic unit for multiplying the cosθ with respect to the current _{I m} after differentiation calculation for multiplying the sinθ against current _{I m} after differentiation, current after sinθ multiplying _{I m} and cosθ after multiplication It has added adder for adding an adder for adding the current I _m after differentiation, and the current I _m after differential current I _m and sinθ after multiplication after cosθ multiplication. The control unit 25 obtains a result equivalent to that of the equation (5) by dividing the result calculated by the following equation (16) by √2 for coefficient adjustment. Although detailed description is omitted, the reactive current _Iq can also be obtained by the control unit 25 by the same calculation. Since the active current I _p and the reactive current I _q calculated using the virtual current do not include the vibration component having a period twice that of the fundamental wave, the control unit 25 can calculate without using the LPF. be.

The PI controllers 73 and 74 used in FIGS. 13, 14 and 18 are examples, and if the controller is sufficient to realize the command value, a P (Proportional) controller or a PID (Proportional Integral) is used. Differential) controller or the like may be used.

_{Relationship of effective current command value Ip} ^* with respect to the rotation speed used in the arithmetic unit 45 of FIGS. 13 and 18, and invalid current command with respect to the rotation speed used in the arithmetic unit 46 of FIGS. 13, 14 and 18. Regarding the relationship between the values I _q ^* , a method of calculating a required value by an experiment in advance or a method of calculating a required value from a motor parameter can be mentioned. In general, _{increasing the active current Ip} increases the work of the motor, so it is required to increase the rotation speed of the motor to be controlled. On the other hand, in general, the reactive current _Iq does not directly contribute to the work of the motor, and therefore needs to be controlled to be 0. FIG. 19 is a diagram showing an example of a change in efficiency _{when the reactive current Iq} is increased or decreased when the active power P is set to a constant value in the control unit 25 according to the first embodiment. FIG. 19 shows a state in which a constant active power P is supplied in a state where the rotation speed of the single-phase motor 12 is known. At time t1, _{the command value of the reactive current Iq} is set to 0. As the reactive current _Iq decreases, the current flowing through the motor winding decreases. As a result, in the motor driving system 1, as shown in FIG. 20 reduces the proportion of R _m I _m ² which has been consumed part of the active power P, increasing the proportion of active power P according to e _m I _m , The rotation speed increases. FIG. 20 is a diagram showing the relationship between the motor work W and the copper loss in the motor drive device 2 according to the first embodiment. Further, it can be confirmed from FIG. 19 that the motor efficiency is also increased because almost all of the active power P is consumed as the motor work W.

Generally, in the high-speed rotation region of a motor, there is a problem that the difference between the induced voltage and the voltage applied by the inverter becomes small due to the increase in the induced voltage due to the rotation speed, and the current cannot flow through the motor winding. be. Further, since the frequency of the applied AC voltage becomes high, there is a problem that the current delay due to the inductance component of the motor winding is large, the motor torque becomes small, and the rotation speed cannot be increased. In order to solve these problems, in the present embodiment, the control unit 25 advances the voltage phase by increasing the reactive power that does not directly contribute to the rotation as PQ control. The control unit 25 allows a current to flow in the motor winding by advancing the voltage phase beyond the induced voltage. Further, the control unit 25 can suppress a decrease in the motor torque by correcting the delayed current phase. _{Therefore, the reactive current command value I q} ^{* with} respect to the rotation speed of the arithmetic unit 46 shown in FIGS. 13, 14 and 18 increases at high speed rotation. This is particularly effective in electric vacuum cleaners, hand dryers, etc., which require the motor to operate at a high speed of 50,000 rpm or more.

Here, when controlling a motor that uses a wing as a load, such as a vacuum cleaner, using torque as a command value, it is conceivable that the suction port is closed and the motor is sealed. FIG. 21 is a diagram showing changes in the rotational speed ω, the active power P, and the active power Q when the motor drive device 2 according to the first embodiment detects sealing. In a vacuum cleaner or the like, the load on the wings decreases during sealing, so if a certain torque command value is continuously output, the rotation speed ω increases as shown in FIG. 21 from time t1 to time t2. At this time, the control unit 25 avoids step-out of the motor by increasing the invalid power command in accordance with the increase in the rotation speed of the single-phase motor 12. As shown at time t2 in FIG. 21, the control unit 25 detects that the suction port is sealed when the rotation speed of the rotation speed ω reaches the maximum rotation speed guaranteed by the single-phase motor 12. The maximum rotation speed may be referred to as a first rotation speed. When the rotation speed reaches the specified first rotation speed, the control unit 25 calculates by the product of the sine wave based on the phase based on the change in the polarity of the AC voltage and the current detected by the current detection unit 22. It is controlled to reduce the physical quantity to be generated. When the control unit 25 detects that the suction port is sealed, the control unit 25 lowers the command values of the active power P and the ineffective power Q, thereby lowering the rotation speed ω. In the example of FIG. 21, the control unit 25 completely stops the rotation of the single-phase motor 12 for safety, although the single-phase motor 12 maintains a constant rotation speed ω after the time t3. May be good.

FIG. 22 is a diagram showing a fourth configuration example of the control unit 25 according to the first embodiment. FIG. 22 shows a part of the configuration of the control unit 25 that reduces the rotational speed ω of the single-phase motor 12 by detecting the sealing. The control unit 25 holds in advance a table of _{the effective current command value Ip} ^* at the time of sealing in the command value selection unit 104, in addition to the command value by the switch 102. As for the effective current command value I _p ^*, it is desirable that the rotation speed ω can be reduced or stopped without the single-phase motor 12 stepping out under a load in a sealed state, and the value is obtained in advance by actual measurement or the like. When the current rotation speed ω reaches the maximum rotation speed, the seal detection unit 103 notifies the command value selection unit 104 that the current rotation speed ω has been reached. The rotation speed ω may be an estimated value or an actually measured value. The command value selection unit 104 outputs the command value from the switch 102 as it is when the notification is not acquired from the seal detection unit 103, and when the notification is acquired from the seal detection unit 103, the command held in the table in advance. Output the value. The command value selection unit 104 outputs the command value from the switch 102 again when the sealing is released or the switch 102 is pressed again.

The operation of the control unit 25 included in the motor drive device 2 will be described with reference to a flowchart. FIG. 23 is a flowchart showing the operation of the control unit 25 according to the first embodiment. Control unit 25 obtains the DC voltage _{V dc} detected by the voltage sensor 20, current _{I m} detected by the current detection unit 22, information such as the command value from the switch 102 (step ST1). The control unit 25 calculates a physical quantity according to the phase based on the change in the polarity of the AC voltage applied from the inverter 11 to the single-phase motor 12 (step ST2). Specifically, the control unit 25 calculates a sine wave by a phase in which the base point change in polarity of the AC voltage, by multiplying the current I _m that is detected by the current detection unit 22, the effective current I _p as a physical quantity do. Or, the control unit 25, and the cosine wave by a phase in which the base point change in polarity of the AC voltage, by multiplying the current I _m that is detected by the current detection unit 22 calculates a reactive current I _q as the physical quantity. The control unit 25 controls the rotation speed of the single-phase motor 12 according to the calculated physical quantity (step ST3). Specifically, the control unit 25 generates PWM signals Q1, Q2, Q3, Q4 according to the calculated physical quantity, and controls the switching elements 51, 52, 53, 54 of the inverter 11.

Next, the hardware configuration of the control unit 25 included in the motor drive device 2 will be described. FIG. 24 is a diagram showing an example of a hardware configuration that realizes the control unit 25 included in the motor drive device 2 according to the first embodiment. The control unit 25 is realized by the processor 201 and the memory 202.

The processor 201 is a CPU (Central Processing Unit, central processing unit, processing unit, arithmetic unit, microprocessor, microprocessor, processor, DSP (Digital Signal Processor)), or system LSI (Large Scale Integration). The memory 202 includes RAM (Random Access Memory), ROM (Read Only Memory), flash memory, EPROM (Erasable Programmable Read Only Memory), EEPROM (registered trademark) (Registered Trademark) (Electrically volatile Memory), and EEPROM (Registered Trademark). A semiconductor memory can be exemplified. Further, the memory 202 is not limited to these, and may be a magnetic disk, an optical disk, a compact disk, a mini disk, or a DVD (Digital Versaille Disc).

As described above, according to the present embodiment, in the motor drive device 2, the control unit 25 has a sine wave with a phase based on a change in the polarity of the AC voltage and a current I detected by the current detection unit 22. The rotation speed of the single-phase motor 12 is controlled according to the physical quantity calculated by the product with _m. The control unit 25, the physical quantity calculated by the product of the cosine wave and the current I _m by base and the phase of the change in polarity of the AC voltage is controlled so as to increase in proportion to the rotational speed. As a result, the motor drive device 2 can control the active power and the reactive power to realize effective power control and rotation speed control of the single-phase motor 12.

Embodiment 2.
In the second embodiment, an application example of the motor drive device 2 described in the first embodiment will be described.

FIG. 25 is a diagram showing a configuration example of an electric blower 64 including the motor drive device 2 according to the second embodiment. The electric blower 64 includes the motor drive device 2 described in the first embodiment, and the propeller 69 is attached to the single-phase motor 12 driven by the motor drive device 2. The electric blower 64 has a structure in which the motor driving device 2 rotates the single-phase motor 12 to send out or suck the wind.

FIG. 26 is a diagram showing a configuration example of an electric vacuum cleaner 61 including the electric blower 64 according to the second embodiment. The vacuum cleaner 61 includes a battery 67 corresponding to the battery 10 shown in FIG. 1, a motor driving device 2 shown in FIG. 1, and an electric blower 64 driven by a single-phase motor 12 shown in FIG. Further, the vacuum cleaner 61 includes a dust collecting chamber 65, a sensor 68, a suction port 63, an extension pipe 62, and an operation unit 66.

The user who uses the vacuum cleaner 61 has an operation unit 66 and operates the vacuum cleaner 61. The motor drive device 2 of the vacuum cleaner 61 drives the electric blower 64 using the battery 67 as a power source. By driving the electric blower 64, dust is sucked from the suction port 63. The sucked dust is collected in the dust collecting chamber 65 via the extension pipe 62.

When outputting the voltage based on the voltage command to the single-phase motor 12, the control unit 25 suspends the switching operation between the upper arm first element and the lower arm first element in one half cycle of the voltage command cycle. In the other half cycle of the voltage command cycle, the switching operation between the upper arm second element and the lower arm second element is suspended. As a result, an increase in switching loss is suppressed, and an efficient vacuum cleaner 61 can be realized.

Further, in the vacuum cleaner 61 according to the embodiment, the switching elements 51, 52, 53, 54 of the inverter 11 are formed of wide bandgap semiconductors, so that the heat dissipation parts can be simplified to reduce the size and weight. can.

FIG. 27 is a diagram showing a configuration example of a hand dryer 90 including the electric blower 64 according to the second embodiment. The hand dryer 90 includes a casing 91, a hand detection sensor 92, a water receiving unit 93, a drain container 94, a cover 96, a sensor 97, an intake port 98, and an electric blower 64. Here, the sensor 97 is either a gyro sensor or a motion sensor. In the hand dryer 90, when a hand is inserted into the hand insertion portion 99 at the upper part of the water receiving portion 93, water is blown off by the blown air by the electric blower 64, and the blown water is collected by the water receiving portion 93. After that, it is stored in the drain container 94.

Similar to the vacuum cleaner 61 shown in FIG. 26, the hand dryer 90 is a product in which the motor rotation speed fluctuates from 0 [rpm] to more than 100,000 [rpm]. Therefore, also in the hand dryer 90, the control method according to the above-described embodiment is suitable, and the same effect as that of the vacuum cleaner 61 can be obtained.

As described above, in the present embodiment, the configuration example in which the motor drive device 2 is applied to the vacuum cleaner 61 and the hand dryer 90 has been described, but the motor drive device 2 is applied to an electric device on which a motor is mounted. can do. Electrical equipment equipped with motors includes incinerators, crushers, dryers, dust collectors, printing machines, cleaning machines, confectionery machines, tea making machines, woodworking machines, plastic extruders, cardboard machines, packaging machines, hot air generators, and OA. Equipment, electric blowers, etc. The electric blower is a blowing means for transporting an object, sucking dust, or for general blowing and exhausting.

The configuration shown in the above-described embodiment shows an example of the content of the present invention, can be combined with another known technique, and is one of the configurations as long as it does not deviate from the gist of the present invention. It is also possible to omit or change the part.

1 motor drive system, 2 motor drive, 5A 1st leg, 5B 2nd leg, 6A, 6B connection points, 10,67 batteries, 11 inverters, 12 single-phase motors, 12a rotors, 12b stators, 20 voltage sensors, 21 Position sensor, 21a position sensor signal, 22 current detection unit, 25 control unit, 32 drive signal generation unit, 33 carrier generation unit, 38, 38A, 38B carrier comparison unit, 38a absolute value calculation unit, 38b division unit, 38c, 38d , 38f, 38k multiplication unit, 38e, 38m, 38n addition unit, 38g, 38h comparison unit, 38i, 38j output inversion unit, 42 inverse PQ conversion unit, 44 detection current signal processing unit, 45, 46 arithmetic unit, 47 differential unit. , 48 Control cycle determination unit, 51, 52, 53, 54 switching element, 51a, 52a, 53a, 54a body diode, 61 electric vacuum cleaner, 62 extension tube, 63 suction port, 64 electric blower, 65 dust collector, 66 operation unit, 68,97 sensor, 69 propeller, 71,72 subtractor, 73,74 PI controller, 90 hand dryer, 91 casing, 92 hand detection sensor, 93 water receiver, 94 drain container, 96 cover, 98 Intake port, 99 manual insertion part, 100 main winding, 101 auxiliary winding, 102 switch, 103 sealed detection part, 104 command value selection part, 105 amplitude detection part, 106, 107 control cycle calculation part, 108, 109 selector.

In order to solve the above-mentioned problems and achieve the object, the motor drive device according to the present invention includes an inverter that outputs an AC voltage to a single-phase motor, a control unit that controls the AC voltage output by the inverter, and a single-phase motor. It is provided with a current detection unit that detects the current flowing through the motor. The control unit, so that the rotational speed of the single-phase motor according to the average value by the filter processing of the physical quantity calculated by the product of the sine wave and the current with a polar phase of a base point of change of the AC voltage is controlled Control the AC voltage.

Claims (12)

An inverter that outputs AC voltage to a single-phase motor,
A control unit that controls the AC voltage output by the inverter,
A current detector that detects the current flowing through the single-phase motor,
Equipped with
The control unit determines the rotation speed of the single-phase motor according to a physical quantity calculated by the product of a sine wave having a phase based on a change in the polarity of the AC voltage and the current detected by the current detection unit. Motor drive to control. The motor drive device according to claim 1, wherein the control unit controls the physical quantity calculated by the product of the cosine wave due to the phase and the current to increase in proportion to the rotation speed. The control unit controls to reduce the physical quantity calculated by the product of the sine wave due to the phase and the current when the rotation speed reaches the specified first rotation speed. The motor drive device described in. The motor drive device according to any one of claims 1 to 3, wherein the control unit changes a control cycle according to the rotation speed and adds an advance command value for correcting a delay in calculation to the phase. The control unit calculates a second current having a phase difference from the first current from the first current, which is the current, and uses the first current and the second current to drive the single-phase motor. The motor drive device according to any one of claims 1 to 4, which controls the rotation speed of the motor. The single-phase motor has an auxiliary winding having a phase difference with respect to the main winding.
The current detecting unit detects a first current, which is the current, and a second current having a phase difference with respect to the first current.
The motor drive device according to any one of claims 1 to 4, wherein the control unit controls the rotation speed of the single-phase motor using the first current and the second current. The motor drive device according to any one of claims 1 to 6, wherein the control unit delays a control cycle when a vibration of a physical quantity calculated by a product of a sine wave due to the phase and the current is detected. The motor drive device according to any one of claims 1 to 7, wherein at least one of the plurality of switching elements included in the inverter is formed of a wide bandgap semiconductor. The motor drive device according to claim 8, wherein the wide bandgap semiconductor is silicon carbide, gallium nitride, or diamond. An electric blower comprising the motor driving device according to any one of claims 1 to 9. A vacuum cleaner comprising the electric blower according to claim 10. The hand dryer provided with the electric blower according to claim 10.

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