JP6896032B2 - Motor drive, vacuum cleaner and hand dryer - Google Patents

Description

The present invention relates to a motor drive device for driving a single-phase permanent magnet synchronous motor (hereinafter, appropriately referred to as a "single-phase PM motor" (Permanent Magnet Motor)), and an electric vacuum cleaner and a hand dryer using the single-phase PM motor. Regarding.

There are various types of motors such as brushed DC motors, induction motors and PM motors, and there are also types such as single-phase and three-phase motors. Among these various motors, the single-phase PM motor has a "brushless" structure that does not use a brush, which is a mechanical structure, as compared with a DC motor with a brush, so that brush wear does not occur. Due to this feature, the single-phase PM motor can ensure a long life and high reliability.

Further, the single-phase PM motor is a highly efficient motor because a secondary current does not flow through the rotor as compared with an induction motor.

Further, the single-phase PM motor has the following advantages as compared with the three-phase PM motor having different numbers of phases.
(1) In the case of a three-phase PM motor, a three-phase inverter is required, whereas in a single-phase PM motor, a single-phase inverter may be used.
(2) When a full-bridge inverter generally used as a three-phase inverter is used, six switching elements are required, whereas in the case of a single-phase PM motor, four switching elements are required even if a full-bridge inverter is used. Can be configured with.
(3) Due to the features of (1) and (2), the single-phase PM motor can be made smaller than the three-phase PM motor.

In addition, as prior documents disclosing the technology relating to the drive system of a single-phase PM motor, for example, there are Patent Document 1 and Non-Patent Document 1 described below.

Japanese Unexamined Patent Publication No. 2012-13378

Institute of Electrical Engineers of Japan D, Vol. 121, No. 10, 2001 "Control method for single-phase system-connected inverter using Hilbert transform"

According to the above-mentioned Patent Document 1, "the air volume is estimated and estimated from the relationship of" energization amount-current-air volume "which has a control means for controlling the energization amount to the electric blower and is obtained in advance by an experiment or the like. When the air volume is within the first predetermined range, as the estimated air volume decreases, the energization amount to the electric blower is controlled to decrease, and the energization amount is the first predetermined range. In the air volume region within the range, the degree of vacuum in the dust collecting chamber is controlled to be substantially constant and to a value set in advance by an experiment or the like. " That is, in Patent Document 1, the air volume by the electric blower is determined by the work amount of the electric blower, and is determined by the active power from the viewpoint of the electric energy in the electric blower.

As described above, in Patent Document 1, the energization amount is controlled according to the estimated air volume, but the control is not performed from the viewpoint of active power and ineffective power. Moreover, although the apparent power is controlled only by the amount of energization, the required active power cannot be controlled individually. Therefore, the technique of Patent Document 1 has a problem that the current flowing through the electric motor becomes larger than the maximum efficiency point and the efficiency deteriorates.

Further, Non-Patent Document 1 describes one method of current control and power control of a single-phase system interconnection inverter. According to this method, the system current is converted into an instantaneous current vector on the complex plane by the Hilbert transform, and the system voltage is converted into the instantaneous voltage vector on the complex plane by the Hilbert transform. Then, the instantaneous current vector is subjected to coordinate conversion to the d-axis and the q-axis using the phase of the instantaneous voltage vector. Further, a method of controlling single-phase instantaneous power by controlling each of the dq-converted d-axis system current and the q-axis system current is described.

Further, in the method of Non-Patent Document 1, the handling of single-phase instantaneous power is facilitated by converting the system current as a DC value, but the application is limited to the single-phase system interconnection inverter. No study has been made on application to a single-phase PM motor as in the present invention.

The present invention has been made in view of the above, and provides a motor drive device, a vacuum cleaner, and a hand dryer that enable effective single-phase instantaneous power control and rotation speed control in a single-phase PM motor. The purpose is to do.

In order to solve the above-mentioned problems and achieve the object, the motor drive device according to the present invention is a motor drive device that drives a single-phase PM motor. The motor drive device includes a plurality of switching elements, a single-phase inverter that applies an AC voltage to the single-phase PM motor, and a motor current detector that outputs a motor current detection signal according to the motor current flowing through the single-phase PM motor. It also includes an inverter control unit that receives a motor current detection signal as an input and outputs a drive signal of a switching element. The inverter control unit controls the active power according to the rotation speed of the single-phase PM motor.

According to the present invention, it is possible to realize effective single-phase instantaneous power control and rotation speed control in a single-phase PM motor.

The figure which shows the structure of the motor drive device in Embodiment 1. The figure which shows the relationship between the rotor rotation position and the position detection signal in Embodiment 1. The block diagram which shows the structure of the inverter control part in Embodiment 1. The figure which shows the relationship between the position detection signal and the motor rotation speed estimated value in Embodiment 1. The figure which shows the relationship between the motor rotation speed estimated value and the rotor rotation position estimated value in Embodiment 1. The figure which shows the vector relation in the dq coordinate system of the instantaneous vector of a motor current and the instantaneous vector of an inverter output voltage in Embodiment 1. A time chart provided for explaining the operation of the switching element drive signal generation unit according to the first embodiment. The figure which shows an example of the structure of the electric vacuum cleaner as an application example of the motor drive device in Embodiment 1. The figure which shows an example of the structure of the hand dryer as another application example of the motor drive device in Embodiment 1. The figure which shows the relationship between a rotor rotation position and a position detection signal, and a rotor magnet magnetic flux in Embodiment 2. Vector diagram with rotor magnet magnetic flux added to the vector diagram shown in FIG. In the vector diagram shown in FIG. 11, the vector diagram when the d-axis is set on the instantaneous vector of the inverter output voltage. The block diagram which shows the structure of the inverter control part in Embodiment 2. A block diagram showing an example of a hardware configuration related to the inverter control unit of the first embodiment and the second embodiment. A block diagram showing another example of the hardware configuration related to the inverter control unit of the first embodiment and the second embodiment.

Hereinafter, the motor drive device, the vacuum cleaner, and the hand dryer according to the embodiment of the present invention will be described in detail with reference to the drawings. The present invention is not limited to the following embodiments.

Embodiment 1.
FIG. 1 is a diagram showing a configuration of a motor drive device according to the first embodiment. The motor drive device 100 in the first embodiment is a motor drive device for driving a load including a single-phase PM motor 3, and is a DC power supply 1, a single-phase inverter 2, an inverter control unit 4, a motor current detection unit 5, and a DC. It is configured to include a power supply voltage detection unit 6 and a rotor position detection unit 7. Examples of the load including the single-phase PM motor 3 include a vacuum cleaner equipped with an electric blower and a hand dryer.

The DC power supply 1 supplies DC power to the single-phase inverter 2. The single-phase inverter 2 includes diodes 221 to 224 connected in antiparallel to each of the switching elements 211 to 214 and the switching elements 211 to 214, and applies an AC voltage to the single-phase PM motor 3. The inverter control unit 4 outputs drive signals S1 to S4 to the switching elements 211 to 214 of the single-phase inverter 2. Rotor position detector 7 outputs the position detection signal S_ _rotation is a signal corresponding to the rotor rotational position theta _m is the rotational position of the single-phase PM motor 3 of the rotor 3a to the inverter control unit 4. Motor current detector 5 outputs a signal corresponding to the motor current I _m flowing through the single-phase PM motor 3 to the inverter control unit 4. The DC power supply voltage detection unit 6 detects the DC voltage V _dc , which is the voltage of the DC power supply 1. Drive signals S1~S4 the rotor rotational position theta _m and pulse width modulation are generated based on the motor current _{I m:} is (Pulse Width Modulation hereinafter referred to as "PWM") signal. By driving the switching elements 211 to 214 of the single-phase inverter 2 with the drive signals S1 to S4 which are PWM signals, an arbitrary voltage can be applied to the single-phase PM motor 3.

The DC power supply 1 may be a DC power supply that rectifies and smoothes the AC voltage from the AC power supply with a diode bridge or the like to generate a DC voltage, and there is no problem even if a DC power supply such as a solar cell or a battery is used. Further, the switching elements 211 to 214 of the single-phase inverter 2 include an IGBT (Insulated Gate Bipolar Transistor), a MOS-FET (Metal Oxide Semiconductor-Field Effect Transistor), a thyristor, and a GTO (Gate Turn-off). There is no problem even if a switching element is used. Further, there is no problem even if a semiconductor material called a wide bandgap semiconductor represented by SiC, GaN or the like is used as well as Si which is the mainstream semiconductor material for the switching element as described above.

_{The rotor position detection unit 7 generates a position detection signal S_rotation corresponding to the rotor rotation position θ m} of the motor as shown in FIG. 2, and outputs the position detection signal S_rotation to the inverter control unit 4. In the case of FIG. 2, it is assumed that a magnetic sensor such as a hall sensor is used _{to output a position detection signal S_rotation} , which is a pulsed voltage signal corresponding _{to the rotor rotation position θ m} , and 0 ≦ θ _m. In <π, S _{_rotation} = “High level”, and in π ≦ θ _m <2π, S _{_rotation} = “Low level” is described as an example. However, there is no problem even if a position detection sensor such as an encoder or a resolver is used instead of the magnetic sensor.

FIG. 3 is a block diagram showing the configuration of the inverter control unit 4 according to the first embodiment. The inverter control unit 4 has a first current control unit 411 that controls the d-axis current Id, _{a second current control unit 412 that controls the q-axis current I q,} and a rotation speed control that controls the estimated motor rotation speed ωm ^. Part 42, the first coordinate conversion unit 431 that performs coordinate conversion from the polar coordinate system to the coordinate system on the d-axis and the q-axis (hereinafter referred to as "dq-axis coordinate system"), coordinate conversion from the dq-axis coordinate system to the polar coordinate system. The second coordinate conversion unit 432 that detects the rotor rotation position estimated value θ _m ^ and the motor rotation speed estimated value ω _{m ^ according to the position detection signal S_rotation} , the motor position and rotation speed detection unit 44, and the rotation speed command. The rotation speed command value generator 45 that generates the value ω *, the switching element drive signal generator 46 that generates the drive signals S1 to S4 from _{the inverter output voltage command value V m *, and the q-axis current command value I q} * are generated. q-axis current command value generating unit 47, and the motor current I _m is configured with a Hilbert transform unit 48 for complex by Hilbert conversion, each part will be described details below. It should be noted that the notation "^" in the "θ _{m ^"} and "ω _{m ^",} and is a representation of the ^"→" in the "I ^→" and "V ^→" later, if the original "θ" or The "^" symbol should be added to the upper part of the "ω" character, and the " ^→ " symbol should be added to the upper part of the "I" and "V" characters, but that notation is not possible. Therefore, in this specification, except for the mathematical expression part to be inserted in the image, the character "^" or " ^→ " is added after the corresponding character or character string.

First, the detailed operation of the motor position and the rotation speed detection unit 44 will be described. As described above, the rotor position detection unit 7 _{generates the position detection signal S_rotation} as shown in FIG. 2 and outputs it to the motor position and rotation speed detection unit 44. FIG. 4 is a diagram showing the relationship between the _{position detection signal S_rotation} and the motor rotation speed estimated value ω _{m ^ in the first embodiment.} The motor position and rotation speed detection unit 44 can detect the _{motor rotation speed estimated value ω m} ^ by the calculation formula shown in the following equation (1) using the period T _{_rotation of the position detection signal S _rotation.} In the first embodiment, the pole logarithm P of the motor is described as P = 1, but of course P ≠ 1. However, when P ≠ 1, there is a relationship of ω _e = P × ω _{m between the electric angle rotation speed ω e and the motor rotation speed ω m} which is the mechanical angle rotation speed.

FIG. 5 is a diagram showing the relationship between the motor rotation speed estimated value ω _m ^ and the rotor rotation position estimated value θ _{m ^ in the first embodiment.} As shown in FIG. 5 and the following equation (2), the rotor rotation position estimated value θ _m ^ can be calculated by integrating the motor rotation speed estimated value ω _{m ^.} However, in the example of FIG. 5, _{the description assumes a discrete control system in the control cycle Tct} , and the rotor rotation position estimated value at the control timing n is described as θ _m ^ [n].

From the above, by using the equations (1) and (2), the motor rotation speed estimated value ω _m ^ and the rotor rotation position estimated value θ _m ^ can be calculated _{from the position detection signal S_rotation.} The above-described method of calculating the motor rotation speed estimation value omega _m ^ and the rotor rotational position estimate theta _m ^ than the period T _{_rotation} position detection signal S _{_rotation} is only an example, it is employed other methodological issues It goes without saying that there is no such thing.

Next, with respect to the Hilbert conversion unit 48 for complex the motor current I _m by a first coordinate conversion unit 431 and the Hilbert transform of performing coordinate conversion to the dq-axis from representation in the single-phase AC will be described with reference to FIG. 6 .. FIG. 6 is a diagram showing a vector relationship in the dq coordinate system of ^{the instantaneous vector I →} of the motor current and the instantaneous vector V ^{→ of the inverter output voltage.}

As also described in Non-Patent Document 1, when expressed the motor current I _m in a complex vector, therefore expression is regarded as the analysis function, the real and imaginary parts of the vector are connected by the relationship of the Hilbert transform to each other .. Although it directed to a current in the non-patent document 1, a single phase system, as in the present invention is directed to a motor current I _m. By combining the imaginary part _{I imaginal} satisfying the relation of the Hilbert transform to the motor current _{I m,} the instantaneous vector I ^→ the motor current _{I m,} can be detected by the following equation (3).

In the above equation, there is a relationship of _{I imaginal} = H [I _{real ] between I real} and I _imaginal. Further, H [] represents the Hilbert transform process, and has the characteristic represented by the following equation (4).

The first coordinate conversion unit 431 performs coordinate conversion from the single-phase alternating current to the dq coordinate system according to the above equation (4), and the d-axis current I _d and the q-axis current I _{q represented by the following equation (5).} Is calculated.

In the above equation, θ _V is the phase of the inverter output voltage (hereinafter referred to as “inverter output voltage phase”), and the value calculated by the second coordinate conversion unit 432 is used. Further, in the above equation, φ represents the power factor angle.

According to the example of FIG. 6, the first coordinate conversion unit 431 is performing coordinate conversion of the instantaneous vector I ^→ motor current I _m on the inverter output voltage phase theta _V.

Next, the equation (5) is expanded from the relationship shown in FIG. First, from FIG. 6, I _real and I _imaginal can be expressed by the following equation (6).

In the above formula, I _{M_rms,} the effective value of the motor current I _m (hereinafter, appropriately referred to as "motor current effective value") it represents.

By substituting the above equation (6) into the above equation (5) and expanding the equation, the following equation (7) can be obtained.

It can be seen that both of the above equations (7) _{do not include the inverter output voltage phase θ V} , and the d-axis current I _d and the q-axis current I _q are converted as DC quantities.

Then, when the instantaneous value i _m of the instantaneous value v _{m (t)} and the motor current I _m of the inverter output voltage _(t) is defined as the following equation (8), the instantaneous single-phase AC power P _m The relationship between the d-axis current I _d and the q-axis current I _q with respect to the value (hereinafter referred to as “single-phase AC power instantaneous value”) P _{m (t) will be described.}

The first equation in the above equation (8) is an equation representing an instantaneous value of the inverter output voltage (hereinafter referred to as "inverter output single-phase voltage instantaneous value") v _m (t), and the second equation is a motor current _Im. the instantaneous value (hereinafter referred to as "motor current instantaneous value") is an expression representing a i m _(t). Further, V _{m_rms} is an effective value of the inverter output single-phase voltage instantaneous value v _{m (t).} In FIG. 6, it is assumed that the capacitive load motor current instantaneous value i _{m (t)} is the phase lead with respect to the inverter output single-phase voltage instantaneous value v _{m (t),} the inverter output single phase voltage instantaneous value v _{m (t)} with respect to the motor current instantaneous value i _{m (t)} may be a delay inductive load is the phase. If a motor current instantaneous value i _{m (t)} is delayed phase, so that the power factor angle φ takes a negative value.

The single-phase AC power instantaneous value P _m (t) is expressed by the following equation (9).

When the above equation (9) is expanded by the addition theorem, it is expressed by the following equation (10).

The above equation (10) is an equation expressing instantaneous power, in particular, the first term represents the active power component of instantaneous power (hereinafter referred to as "instantaneous active power value"), and the second term is the reactive power component of instantaneous power (hereinafter referred to as "instantaneous active power value"). Hereinafter referred to as "instantaneous value of ineffective power"). Here, by substituting the equation (7) into the equation (10), the following equation (11) is obtained.

As is clear from comparing the above equation (11) with the above equation (10), the d-axis current Id is included in the equation representing the active power, and the q-axis current Iq is included in the equation representing the ineffective instantaneous power. Therefore, it can be said that the active power can be controlled by the d-axis current Id and the invalid instantaneous power can be controlled by the q-axis current Iq.

Here, it is known that in PQ conversion, which is a general method for controlling active power and ineffective power in single-phase power, pulsation occurs in the value after PQ conversion. On the other hand, according to the method of the present invention, as shown in the above equation (7), the d-axis current Id and the q-axis current Iq are converted as DC quantities, so that controllability can be improved. .. The details of improving the controllability will be described later.

Up to this point, in the explanation of the first coordinate transforming unit 431 and the Hilbert transforming unit 48, the formula has been expanded after defining as shown in FIG. 6, but such a definition has been made for convenience of explanation. It does not necessarily have to be the definition described.

Next, a second coordinate conversion unit 432 that performs coordinate conversion from the dq coordinate system to the polar coordinate system, that is, the single-phase alternating current coordinate system will be described. The second coordinate conversion unit 432 uses the d-axis voltage command value V _d * and the q-axis voltage command value V _q *, and is an AC voltage inverter output voltage command value V based on the following equation (12). Convert to _{m *.}

The above equation (12) is _{an example of the coordinate conversion equation to the inverter output voltage command value V m} *, and the content of the equation (12) changes depending on the definition in the first coordinate conversion unit 431 described above. ..

_{The inverter output voltage phase θ V} used by the first coordinate conversion unit 431 corresponds to the phase angle in cos in the above equation (12). That is, the inverter output voltage phase θ _V can be expressed by the following equation (13).

Next, the first current control unit 411 and the second current control unit 412 will be described. The first current control unit 411 is a feedback controller that controls the above-mentioned d-axis current I _d to match the d-axis current command value I _d *, and the second current control unit 412 is the above-mentioned q. This is a feedback controller that controls the shaft current I _q so that it matches the q-axis current command value I _{q *.} Both the first current control unit 411 and the second current control unit 412 can employ, for example, a PID control system having _{a transfer function GPID (s) as shown in the following equation (14).} .. (14) In the equation, _{K p} is a proportional gain, _{K I} is an integral gain, _{K d} is the differential gain, s represents the Laplace operator. Needless to say, the points of PID control, the point of feedback control, and the like are examples of the control methods given as explanatory examples in the present specification.

Next, the rotation speed control unit 42 will be described. The rotation speed control unit 42 is a feedback controller that controls the motor rotation speed estimated value ω ^ to match the rotation speed command value ω *, and is combined with the first current control unit 411 or the second current control unit 412. Similar PID control and the like can be adopted. Needless to say, the point of PID control and the point of feedback control are examples of the control method like the first current control unit 411 and the second current control unit 412.

Next, the switching element drive signal generation unit 46 will be described. FIG. 7 is a time chart provided for explaining the operation of the switching element drive signal generation unit 46. In the upper part of the figure, the thin line shows the carrier waveform, and the thick line shows the waveform of the inverter output voltage command value V _m *. In the example of FIG. 7, with respect to the carrier period _{T c,} and setting the control period _{T cnt} to 1/2 of carrier period _{T c.} Further, in the lower part of FIG. 7, the waveforms of the drive signals S1 to S4 for driving the switching elements 211 to 214 are shown.

Assuming a discrete control system with a control cycle T _ct , the inverter output voltage command value V _m * [n] is changed discretely. For example, when focusing on the control timing n, the high level and the low level of the drive signals S1 to S4 are determined by the magnitude relationship between the _{inverter output voltage command value V m * [n] at the control timing n and the carrier.} At this time, S1 and S4, S2 and S3 are the same signal, and S2 and S3 are waveforms inverted with respect to S1 and S4. Here, since the switching element has a delay time peculiar to the switching element such as a rise time and a fall time, a short-circuit prevention time generally called a dead time is often provided.

Although the dead time is described as 0 in FIG. 7, there is no problem even if the dead time ≠ 0. The method for generating the drive signals S1 to S4 in FIG. 7 is merely an example, and any method may be used as long as it is a method for generating a PWM signal.

Next, the rotation speed command value generation unit 45 will be described. FIG. 8 is a diagram showing an example of the configuration of a vacuum cleaner as an application example of the motor drive device according to the first embodiment. In FIG. 8, the vacuum cleaner 8 includes a DC power source 1 such as a battery, an electric blower 81 driven by the single-phase PM motor 3 described above, a dust collecting chamber 82, a sensor 83, a suction port body 84, and an extension pipe. It is configured to include 85 and an operation unit 86.

The vacuum cleaner 8 drives the single-phase PM motor 3 using the DC power supply 1 as a power source, sucks the dust from the suction port 84, and sucks the dust into the dust collecting chamber 82 through the extension pipe 85. At the time of use, the operation unit 86 is held to operate the vacuum cleaner 8.

The operation unit 86 is provided with an operation switch 86a for adjusting the suction amount of the vacuum cleaner 8. The user of the vacuum cleaner 8 operates the operation switch 86a to set the operating state of the vacuum cleaner 8 to strong operation, weak operation, or the like. The operation unit 86 assigns the rotation speed setting value ω ** according to the set operating state to the rotation speed command value generation unit 45 (see FIG. 3). The rotation speed setting value ω ** is input to the rotation speed command value generation unit 45, and the rotation speed command value ω * is output from the rotation speed command value generation unit 45. The method of setting the rotation speed setting value ω ** by the operation unit 86 is an example in the present embodiment, and other methods may be used. For example, a method of automatically setting the rotation speed setting value ω ** according to the sensor 83 may be used, or any method may be used.

FIG. 9 is a diagram showing an example of the configuration of a hand dryer as another application example of the motor drive device according to the first embodiment. In FIG. 9, the hand dryer 90 includes a casing 91, a hand detection sensor 92, a water receiving portion 93, a drain container 94, a cover 96, a sensor 97, and an intake port 98. An electric blower (not shown) driven by the motor drive device of the first embodiment is provided in the casing 91. The hand dryer 90 has a structure in which water is blown off by blowing air from an electric blower by inserting a hand into the hand insertion portion 99 above the water receiving portion 93, and water is stored from the water receiving portion 93 into the drain container 94. ing. The sensor 97 is either a gyro sensor or a motion sensor, and the control system of the inverter control unit 4 may be configured so as to automatically set the rotation speed setting value ω ** according to the sensor 97.

Next, the q-axis current command value generation unit 47 will be described. The q-axis current command value generation unit 47 outputs the _{q-axis current command value I q *.} The q-axis current I _q is an operation amount for controlling the reactive power as described above. Reactive power is power that does not contribute to the actual amount of work. However, when the reactive power increases, the motor current _Im increases, so that the efficiency deteriorates. Therefore, normally, the q-axis current command value I _q * = 0 is set. However, when a control method such as field weakening is used in combination, the negative power may be controlled to a value other than zero according to the increase in the number of rotations. In such a case, the q-axis current command value I _q * may be held by the q-axis current command value generation unit 47 as table data.

With the above configuration, the active power can be controlled by the _{d-axis current I d} so that the rotation speed estimated value ω ^ matches the rotation speed command value ω *. _{Specifically, the d-axis current Id} is controlled so that the active power increases when the rotation speed command value ω * increases, and the active power decreases when the rotation speed command value ω * decreases. The d-axis current _Id is controlled so as to do so. At the same time, the reactive power _{can be controlled by the q-axis current Iq} , and the power factor at the time of driving the motor can be controlled. Thus, for example, by reactive power is controlled to be zero, it is possible to suppress only the amount that only relevant motor current I _m in effective power. As a result, the motor current is controlled to the minimum, the copper loss of the motor including the loss in the winding resistance, the conduction loss of the inverter including the on-resistance or the loss due to the on-voltage in the switching element, and the on / off of the switching element. Since switching loss, which is a loss, can be suppressed, high efficiency of applied products to which a motor drive device is applied can be realized.

So far, mode - have been described a method of controlling the active power by performing coordinate transformation to the dq-axis coordinate system parsing functions motor current I _m - applying a Hilbert transform to the data current I _m, Hilbert transformed mode. Next, the effect of the method according to the present embodiment will be described in comparison with the control using the PQ conversion, which is one of the conventional methods.

First, the equations representing the PQ conversion in single-phase alternating current are defined by the following equations (15-1) and (15-2).

In the above (15-1) and (15-2) below, _{I P} is the P-axis current, _{I Q} represents the Q-axis current. Further, in the equations (15-1) and (15-2), _{the coordinate transformation is performed using the rotor rotation position estimated value θ m} ^, but this is merely an example and is not limited thereto.

As described above, the single-phase AC power instantaneous value P _m (t) is expressed by the equation (9). Therefore, when the equations (15-1) and (15-2) are substituted into the equation (9), the following ( 16) Equation is obtained.

According to the above (16), an active power can be controlled by the P-axis current I _P, to say that the reactive power by the Q-axis current I _Q is controllable can. However, representative of the d-axis current _{I d} and the q-axis current _{I q} and (7), is compared with the representative of the P-axis current _{I P} and Q-axis current _{I Q} (15-1) and (15-2) below The meaning of each formula is very different.

More particularly, (15-1) and (15-2), the P-axis current _{I P} and Q-axis current _{I Q} shown in equation includes the inverter output voltage phase theta _V is an AC component in the trigonometric It has been. Thus, when the current control is performed such as the first current control section 411 and the second current control section 412, since the P-axis current I _P and Q-axis current I _Q contains AC component, the frequency characteristic It is necessary to design in consideration. In particular, when the motor rotating at a high speed such as a vacuum cleaner of the controlled object, since the frequency range of the P-axis current I _P and Q-axis current I _Q is wide, it is necessary to control design considering the characteristics of a wide band, control Design becomes difficult.

On the other hand, in the d-axis current I _d and the q-axis current I _q shown in Eq. (7), the argument of the trigonometric function is the power factor angle φ that does not include the AC component, and the d-axis current I _d and the q-axis current I _q is the amount of DC. Therefore, in the control method according to the present embodiment, it is not necessary to consider the frequency characteristic, which is a problem of control by PQ conversion, and control that does not depend on the rotation speed range of the motor to be controlled becomes possible. ..

In the first embodiment, a method of generating an instantaneous vector of the motor current by Hilbert transform and converting the instantaneous vector into a current component on the dq coordinate axis on the inverter output voltage phase has been described, but a method similar to the Hilbert transform has been described. Alternatively, an instantaneous vector of the motor current may be generated by a different method. The important point is that the AC component contained in the motor current after the coordinate conversion should be smaller than the DC component, and the motor current after the coordinate conversion can be regarded as a DC amount.

If the motor current after coordinate conversion can be treated as a DC amount, it is not necessary to consider the frequency characteristics when designing the current controller, and the control construction can be easily realized.

Further, as described above, since the required active power can be controlled according to the motor rotation speed, highly efficient control can be realized. In particular, when a load device that rotates at high speed such as the vacuum cleaner shown in FIG. 8 is driven by the built-in battery, the driving time by the battery can be extended. Furthermore, the capacity of the battery can be reduced by taking advantage of the high-efficiency drive, so that the load device can be miniaturized.

Embodiment 2.
In the first embodiment, the rotation speed control and the power control for the single-phase PM motor have been described. In the second embodiment, the configuration of the motor control device shown in FIG. 1 is the same, and the additional operation of the inverter control unit 4 will be described. Before explaining the additional operation in the inverter control unit 4, the vector diagram shown in FIG. 6 will be described again.

The amount of magnetic flux of the rotor magnet of the single-phase PM motor (hereinafter referred to as "rotor magnet magnetic flux") is defined as Φ _rotor . Here, in the second embodiment, it is assumed that _{the relationship between the rotor rotation position θ m} and the rotor magnet magnetic flux Φ _rotor is defined as shown in FIG. The relationship of FIG. 10 can be expressed by the following equation (17) when the rotor magnet magnetic flux Φ _rotor is expressed as a function of the rotor rotation position estimated value θ _{m ^.}

In the above equation (17), it is assumed that _{θ m} = θ _{m ^.} Further, Φ _{rotor_max} is the maximum value of the rotor magnet magnetic flux Φ _rotor.

When the rotor magnet magnetic flux Φ _rotor represented by the equation (17) is added to the vector diagram of FIG. 6, FIG. 11 is obtained. FIG. 11 is a vector diagram in the case where the d-axis current I _d and the q-axis current I _{q are defined by the equation (5) as in the first embodiment.}

Here, (5) the d-axis current I _d and a q-axis current I _q as defined in formula, the definitions of setting the d-axis instantaneous vector V ^→ on the inverter output voltage. Next, a case where the d-axis is set on the _{rotor magnet magnetic flux Φ rotor} as shown in FIG. 12 will be examined. In this case, the d-axis current I _d and the q-axis current I _q are redefined by Eq. (18).

From FIG. 12, the d-axis current I _d is parallel to the rotor magnet magnetic flux Φ _rotor , and the q-axis current I _q is orthogonal to the rotor magnet magnetic flux Φ rotor. Therefore, the d-axis current I _d can be regarded as an exciting component current, and the q-axis current I _q can be regarded as a torque component current. As a result, torque control becomes possible by manipulating the q-axis current I _q.

Here, shown again the instantaneous value v _{m (t)} and representing the instantaneous value i _{m (t)} of the motor current (8) of the inverter output voltage as defined in the first embodiment as follows.

In FIG. 12, the inverter output voltage phase θ _V is represented by the following equation (20).

In the above equation (20), θ _VV ^{is the advance angle of the instantaneous vector V →} of the inverter output voltage with respect to the d-axis (hereinafter referred to as “inverter output voltage advance angle”), and is represented by the following equation (21). ..

_{Substituting Eqs.} (6) and (21) into Eq. (19), the following (22) _{shows the relationship between the d-axis currents I d} and the q-axis currents I _q and the I _real and I image. The formula is obtained.

From the equation (22), since the inverter output voltage advance angle θ _VV and the power factor angle φ can be regarded as constant _{in the steady state, the d-axis current I d} and the q-axis current I _{q also in the vector diagram of FIG.} Can be regarded as a DC quantity. Therefore, the same control configuration as in the first embodiment can be realized.

FIG. 13 is a block diagram showing the configuration of the inverter control unit 4 according to the second embodiment. In FIG. 13, the same or equivalent parts as those of the configuration of the first embodiment shown in FIG. 3 are designated by the same reference numerals. The differences from the first embodiment will be described below.

First, with respect to the first coordinate conversion unit 431, as described above, the definition of the dq axis for coordinate conversion is different. Therefore, in the first coordinate conversion unit 431 of the second embodiment, the rotor rotation position estimated value _{is input to θ m} ^, and the coordinate conversion is performed by the equation (18). That is, according to the example of FIG. 12, the first coordinate conversion unit 431 is performing coordinate conversion of the instantaneous vector I ^→ motor current I _m on the rotor rotational position theta _m.

Further, the second current control unit 412 controls the q-axis current I _q , but as described above, unlike the first embodiment, it can be regarded as a torque current component, so that the second current control unit 412 can be regarded as a torque current component. It is set as a q-axis current control unit as a minor loop for rotation speed control.

The above is the difference between the first embodiment and the second embodiment. Since the first current control unit 411, the rotation speed control unit 42, the motor position and rotation speed detection unit 44, and the rotation speed command value generation unit 45 are the same as those in the first embodiment, detailed description thereof will be omitted. ..

With the above configuration, the rotation speed control of the single-phase PM motor can be realized as in the first embodiment, and the same effect as that of the first embodiment can be obtained.

Finally, the hardware configuration when the various control units according to the first and second embodiments are realized by software will be described with reference to FIG. The various control units referred to here are the first current control unit 411, the second current control unit 412, the rotation speed control unit 42, the first coordinate conversion unit 431, and the second in the inverter control unit 4. Coordinate conversion unit 432, motor position and rotation speed detection unit 44, rotation speed command value generation unit 45, switching element drive signal generation unit 46, q-axis current command value generation unit 47, Hilbert conversion unit 48, and the like correspond to the above.

When the functions of the various control means described above are realized by software, as shown in FIG. 14, a CPU (Central Processing Unit) 200 that performs calculations and a memory in which a program read by the CPU 200 is stored. The configuration may include 202 and an interface 204 for inputting / outputting signals. The CPU 200 may be referred to as an arithmetic unit, a microprocessor, a microcomputer, a processor, a DSP (Digital Signal Processor), or the like. The memory 202 is, for example, a non-volatile or volatile semiconductor memory such as a RAM (Random Access Memory), a ROM (Read Only Memory), a flash memory, an EPROM (Erasable Program ROM), or an EPROM (Electrically EPROM). Applies to.

Specifically, the memory 202 stores programs that execute the functions of various control means. The CPU 200 executes various arithmetic processes described in the present embodiment by exchanging and exchanging necessary information via the interface 204.

The CPU 200 and the memory 202 shown in FIG. 14 may be replaced with the processing circuit 203 as shown in FIG. The processing circuit 203 includes, for example, a single circuit, a composite circuit, a programmed processor, a parallel programmed processor, an ASIC (Application Specific Integrated Circuit), an FPGA (Field-Programmable Gate Array), or a combination thereof. Applicable.

The configuration shown in the above-described embodiment shows an example of the content of the present invention, can be combined with another known technique, and is configured without departing from the gist of the present invention. It is also possible to omit or change a part of.

1 DC power supply, 2 single-phase inverter, 3 single-phase PM motor, 3a rotor, 4 inverter control unit, 5 motor current detector, 6 DC power supply voltage detector, 7 rotor position detector, 8 electric vacuum cleaner, 42 rotations Control unit, 44 motor position and rotation speed detection unit, 45 rotation speed command value generation unit, 46 switching element drive signal generation unit, 47 q-axis current command value generation unit, 48 hill belt conversion unit, 81 electric blower, 82 dust collection chamber , 83 sensor, 84 suction port, 85 extension tube, 86 operation unit, 86a operation switch, 90 hand dryer, 91 casing, 92 hand detection sensor, 93 water receiver, 94 drain container, 96 cover, 97 sensor, 98 intake Port, 99 manual insertion part, 100 motor drive, 200 CPU, 202 memory, 203 processing circuit, 204 interface, 211-214 switching element, 221-224 diode, 411 first current control part, 412 second current control 431 First coordinate conversion unit, 432 Second coordinate conversion unit.

Claims (9)

A motor drive device that drives a single-phase permanent magnet synchronous motor.
A single-phase inverter equipped with a plurality of switching elements and applying an AC voltage to the single-phase permanent magnet synchronous motor.
A motor current detector that outputs a motor current detection signal according to the motor current flowing through the single-phase permanent magnet synchronous motor, and a motor current detector.
An inverter control unit that receives the motor current detection signal as an input and outputs a drive signal of the switching element.
With
The inverter control unit is a motor drive device that controls active power and ineffective power according to the rotation speed of the single-phase permanent magnet synchronous motor. The motor drive device according to claim 1, wherein the single-phase inverter is controlled so that the q-axis current obtained from the motor current increases when the rotation speed of the motor exceeds the first value. The motor drive device according to claim 1 or 2, wherein the inverter control unit generates an instantaneous current vector by converting the motor current into a Hilbert transform. The motor drive device according to claim 3, wherein the inverter control unit includes a coordinate conversion unit that performs coordinate conversion of the instantaneous current vector on the output voltage phase of the single-phase inverter. The motor drive device according to claim 3, wherein the inverter control unit includes a coordinate conversion unit that converts the instantaneous current vector into coordinates on the rotor rotation position of the single-phase permanent magnet synchronous motor. The motor drive device according to any one of claims 1 to 5, wherein the inverter control unit includes a current control unit that controls a motor current after coordinate conversion. The inverter control unit includes a rotation speed control unit that controls the rotation speed of the single-phase permanent magnet synchronous motor according to a command value, and uses the output of the rotation speed control unit as a current command value in the current control unit. 6. The motor drive device according to 6. An electric vacuum cleaner equipped with the motor drive device according to any one of claims 1 to 7. A hand dryer equipped with the motor drive device according to any one of claims 1 to 7.

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