JP7050951B2 - Load drive device, refrigeration cycle device and air conditioner - Google Patents

Description

The present invention relates to a load drive device that supplies AC power to a motor to drive a load, a refrigerating cycle device provided with the load driving device, and an air conditioner provided with the refrigerating cycle device.

When the drive target of the load drive device is, for example, a compressor, the load torque fluctuates in one cycle or a plurality of cycles of the rotation cycle of the motor. At this time, the current supplied to the motor of the compressor also pulsates according to the fluctuation of the load torque. When the current supplied to the motor pulsates, the operating efficiency of the compressor, which is a load, decreases.

The following Patent Document 1 discloses a technique for controlling the pulsating component of the q-axis current command value to be zero by integral control. The q-axis current command value is a torque current command value. The pulsating component of the torque current command value is an AC component included in the torque current command value.

By controlling the pulsating component of the torque current command value to be zero, the difference between the output torque of the motor and the load torque of the compressor becomes small, and the speed fluctuation of the motor is suppressed. It is considered that this will improve the operating efficiency of the compressor.

Japanese Unexamined Patent Publication No. 2016-082636

However, in the technique described in Patent Document 1, it takes time to adjust the control gain value required for integral control, so that it is assumed that a desired control gain value cannot be obtained in the control process. As a result, there is a problem that the pulsating component of the torque current command value cannot be completely reduced and highly efficient drive control cannot be reliably performed.

The present invention has been made in view of the above, and an object of the present invention is to obtain a load drive device capable of reliably performing highly efficient drive control.

In order to solve the above-mentioned problems and achieve the object, the present invention is a load drive device that supplies AC power to a motor to drive a load. The load drive device includes a rectifier circuit that rectifies an AC voltage and converts it into a DC voltage, an inverter that converts DC power output from the rectifier circuit into AC power, and a control device that controls the inverter. The control device is a speed estimator that calculates a speed estimate that is an estimated value of the rotational speed of the motor, and a speed controller that generates a current command value based on the speed deviation that is the deviation between the speed command value and the speed estimated value. And a mechanical angle phase calculator that calculates the mechanical angle phase indicating the rotation position of the motor based on the estimated speed value. The control device includes a filter that removes a specific frequency component from the fluctuation component due to the load fluctuation included in the speed deviation or the current command value output from the speed controller. The controller includes a current controller that calculates a voltage command value, which is a command value of a voltage applied to a motor, based on a current command value from which a specific frequency component has been removed and a motor current flowing through the motor. The specific frequency component is a frequency component determined by the load, and is a frequency component that is a natural number of times the mechanical angular frequency in which the rotation speed of the motor is expressed by the rotation angle per unit time.

According to the load drive device according to the present invention, there is an effect that highly efficient drive control can be reliably performed.

A circuit diagram showing a configuration example of the load drive device according to the first embodiment. Partially enlarged view of the inverter shown in FIG. A block diagram showing a configuration example of a control device according to the first embodiment. A block diagram showing a configuration example of the voltage command calculation unit shown in FIG. A first waveform diagram provided for explaining the effect of the first embodiment. Characteristic diagram used for explaining the effect of the first embodiment A second waveform diagram for explaining the effect of the first embodiment. A third waveform diagram for explaining the effect of the first embodiment. A block diagram showing a configuration example when the filter shown in FIG. 4 is a notch filter. A block diagram showing an example of a hardware configuration that realizes the function of the control device according to the first embodiment. A block diagram showing another example of a hardware configuration that realizes the function of the control device according to the first embodiment. The figure which shows the structural example of the refrigerating cycle apparatus which concerns on Embodiment 2.

The load drive device, the refrigeration cycle device, and the air conditioner according to the embodiment of the present invention will be described in detail with reference to the accompanying drawings. The present invention is not limited to the following embodiments.

Embodiment 1.
First, the configuration and function of the load drive device according to the first embodiment will be described with reference to FIGS. 1 and 2. FIG. 1 is a circuit diagram showing a configuration example of the load drive device 300 according to the first embodiment. FIG. 2 is a partially enlarged view of the inverter 30 shown in FIG.

As shown in FIG. 1, the load drive device 300 according to the first embodiment temporarily converts the AC voltage output from the AC power supply 1 which is a single-phase power supply into a DC voltage, and again inside the load drive device 300. It is configured to convert to an AC voltage and drive the motor 7 mounted on the load. The motor 7 can be applied as a driving motor for a compressor provided in a refrigeration cycle device. An example of the motor 7 is a three-phase permanent magnet synchronous motor.

As shown in FIG. 1, the load drive device 300 includes a reactor 2, a rectifier circuit 3, a smoothing capacitor 5, a voltage detector 10, an inverter 30, a current detector 40, and a control device 100. Be prepared. The rectifier circuit 3 and the inverter 30 are electrically connected by the DC buses 12a and 12b. The smoothing capacitor 5 is connected between the DC bus 12a on the high potential side and the DC bus 12b on the low potential side.

The rectifier circuit 3 includes four diodes D1, D2, D3, and D4. The four diodes D1 to D4 are bridge-connected to form a diode bridge circuit. The rectifier circuit 3 receives the AC voltage output from the AC power supply 1 via the reactor 2. The rectifier circuit 3 rectifies the AC voltage and converts it into a DC voltage. AC voltage and DC voltage may be paraphrased as "AC power" and "DC power", respectively.

The rectifier circuit 3 shown in FIG. 1 has a configuration in which four diodes D1, D2, D3, and D4 are bridge-connected, and this configuration is adapted to the AC power supply 1 which is a single-phase power supply. When the AC power supply 1 is a three-phase power supply, the rectifier circuit 3 is also configured to correspond to the three-phase power supply. Specifically, the configuration is such that six diodes are bridge-connected.

The output voltage of the rectifier circuit 3 is applied to both ends of the smoothing capacitor 5. The smoothing capacitor 5 smoothes the output voltage of the rectifier circuit 3. The smoothing capacitor 5 is connected to the DC bus 12a and 12b, and the voltage smoothed by the smoothing capacitor 5 is appropriately referred to as a "bus voltage".

The voltage detector 10 detects the bus voltage. In the case of the configuration of FIG. 1, the bus voltage is also the input voltage to the inverter 30. The detected value Vdc of the bus voltage detected by the voltage detector 10 is input to the control device 100.

The current detector 40 detects the bus current. In the case of the configuration of FIG. 1, the bus current is the current input to the inverter 30. The bus current may be referred to as "input current of inverter 30" or simply "input current". The detection value Idc of the bus current detected by the current detector 40 is input to the control device 100.

An example of the current detector 40 is a shunt resistor. When the current detector 40 is a shunt resistance, the detection value Idc of the bus current is an analog signal. The analog signal is converted into a digital signal by an analog digital converter (not shown) inside the control device 100. The same applies to the detected value Vdc of the bus voltage. That is, when the detected value Vdc of the bus voltage is an analog signal, it is converted into a digital signal inside the control device 100.

A bus voltage is applied to the inverter 30. The inverter 30 drives the motor 7 by converting the DC power output from the rectifying circuit 3 into AC power and supplying the converted AC power to the motor 7 which is a load.

As shown in FIG. 2, the inverter 30 includes a main circuit 310 and a drive circuit 350.

The main circuit 310 includes a leg 310A in which an upper arm switching element UP and a lower arm switching element UN are connected in series, a leg 310B in which an upper arm switching element VP and a lower arm switching element VN are connected in series, and an upper leg 310B. A leg 310C in which an arm switching element WP and a lower arm switching element WN are connected in series is provided. The legs 310A, 310B and 310C are connected in parallel to each other.

FIG. 2 illustrates a case where the upper arm switching elements UP, VP, WP and the lower arm switching elements UN, VN, WN are isolated gate bipolar transistors (IGBTs). Instead of this IGBT, a metal oxide semiconductor field effect transistor (Metal Oxide Semiconductor Field Effect Transistor: MOSFET) may be used.

The upper arm switching element UP includes a transistor 311 and a diode 312 connected in antiparallel to the transistor 311. The antiparallel means that the anode side of the diode is connected to the first terminal corresponding to the emitter of the IGBT, and the cathode side of the diode is connected to the second terminal corresponding to the collector of the IGBT. Other upper arm switching elements VP, WP, and lower arm switching elements UN, VN, WN are also connected in the same manner.

When the transistors 311 of the upper arm switching elements UP, VP, WP and the lower arm switching elements UN, VN, WN are MOSFETs, the diode 312 connected in antiparallel has a parasitic diode inside the MOSFET itself. It can be used. Parasitic diodes are also called body diodes. By using a parasitic diode, the number of parts can be reduced and the cost can be reduced because individual diodes connected in antiparallel are not required.

When the transistors 311 of the upper arm switching elements UP, VP, WP and the lower arm switching elements UN, VN, WN are MOSFETs, at least one of the transistors 311 is formed of a wide bandgap semiconductor. You may. Examples of the wide bandgap semiconductor include silicon carbide (SiC), gallium nitride (GaN), gallium oxide (Ga ₂ O ₃ ), and diamond.

In general, wide bandgap semiconductors have higher withstand voltage and heat resistance than silicon semiconductors. Therefore, if a MOSFET formed of a wide bandgap semiconductor is used for at least one of each transistor 311, the effects of withstand voltage and heat resistance can be enjoyed.

Further, FIG. 2 has a configuration including three legs in which the upper arm switching element and the lower arm switching element are connected in series, but the configuration is not limited to this. The number of legs may be four or more. Further, the circuit configurations of FIGS. 1 and 2 are adapted to the motor 7 which is a three-phase motor. When the load is a multi-phase motor, the main circuit 310 is also configured to correspond to the multi-phase motor. When the load is either a three-phase motor or a multi-phase motor, one leg may be composed of a plurality of pairs of upper and lower arm switching elements.

Further, the inverter 30 has output lines 331, 332, 333. The output line 331 is drawn from the connection point 321 between the upper arm switching element UP and the lower arm switching element UN. The output line 332 is drawn from the connection point 322 between the upper arm switching element VP and the lower arm switching element VN. The output line 333 is drawn from the connection point 323 between the upper arm switching element WP and the lower arm switching element WN. In the inverter 30, the connection points 3211, 322, and 323 form an AC terminal.

The output line 331 is connected to the first phase (for example, U phase) of the motor 7. The output line 332 is connected to the second phase (for example, V phase) of the motor 7. The output line 333 is connected to a third phase (for example, W phase) of the motor 7.

The control device 100 generates pulse width modulation (PWM) signals Sm1 to Sm6 for controlling the operation of the inverter 30 based on the detection value Vdc of the bus voltage and the detection value Idc of the input current. ..

The PWM signals Sm1 to Sm6 are output corresponding to each of the six upper arm switching elements UP, VP, WP and the lower arm switching elements UN, VN, WN of the main circuit 310.

The PWM signals Sm1 to Sm6 are input to the drive circuit 350 of the inverter 30. The drive circuit 350 generates drive signals Sr1 to Sr6 based on the PWM signals Sm1 to Sm6. Each of the upper arm switching elements UP, VP, WP and the lower arm switching elements UN, VN, WN is controlled to be turned on or off by each of the drive signals Sr1 to Sr6. As a result, a voltage at which at least one of the frequency and the voltage value is controlled, that is, a frequency-variable or voltage-variable voltage is applied to the motor 7, and a load including the motor 7 is driven.

The PWM signals Sm1 to Sm6 are signals at a voltage level required to control a logic circuit. An example of a voltage level is 0-5V. The PWM signals Sm1 to Sm6 use the ground potential of the control device 100 as a reference potential. On the other hand, the drive signals Sr1 to Sr6 are voltage level signals necessary for controlling the on operation and the off operation of the upper arm switching elements UP, VP, WP and the lower arm switching elements UN, VN, WN. Examples of voltage levels are -15V to + 15V. The drive signals Sr1 to Sr6 use the potential of the terminal on the low potential side of the corresponding switching element as a reference potential. In the example of FIG. 1, the emitter terminal of the IGBT is the reference potential.

Next, the configuration and function of the control device 100 according to the first embodiment will be described with reference to FIG. FIG. 3 is a block diagram showing a configuration example of the control device 100 according to the first embodiment. As shown in FIG. 3, the control device 100 includes an operation control unit 102 and an inverter control unit 110.

The operation control unit 102 receives the command information Qe from the outside, and generates the speed command value ω * and the stop signal St based on the command information Qe. The stop signal St is a signal for stopping the operation of the inverter 30. The speed command value ω * is a signal for generating a voltage command value, which is a command value of the voltage applied to the motor 7. Both the speed command value ω * and the stop signal St are input to the inverter control unit 110.

When the control device 100 controls the air conditioner, the command information Qe is, for example, the temperature detected by the temperature sensor. Other examples of the command information Qe include information indicating a set temperature instructed from the remote controller, operation mode selection information, operation start and operation end instruction information, and the like.

The inverter control unit 110 includes a current restoration unit 111, a coordinate conversion unit 112, a voltage command calculation unit 115, a coordinate conversion unit 116, a PWM signal generation unit 117, an electric angle phase calculation unit 118, and an excitation current command control. A unit 119 is provided.

The current restoration unit 111 restores the motor currents iu, iv, and iwa based on the detected value Idc of the bus current detected by the current detector 40. The motor currents iu, iv, and iwa are currents flowing through each phase of the motor 7, that is, currents of each phase of the motor 7. The current restoration unit 111 restores the motor currents iu, iv, and iwa by sampling the detected value Idc of the bus current at a timing determined based on the signal from the PWM signal generation unit 117. In the following description, the motor current iu may be referred to as "U-phase current", the motor current iv may be referred to as "V-phase current", and the motor current if may be referred to as "W-phase current".

The coordinate conversion unit 112 is a conversion unit that converts the current value of the UVW phase into the current value of the γ-δ axis. More specifically, the coordinate conversion unit 112 uses the electric angle phase θ generated by the electric angle phase calculation unit 118, which will be described later, to γ-axis the motor currents iu, iv, and iwa restored by the current restoration unit 111. Convert to current iγ and δ-axis current iδ. The γ-axis current iγ is an exciting current component, and the δ-axis current iδ is a torque current component.

The exciting current command control unit 119 calculates a γ-axis current command value iγ * suitable for driving the motor 7 with high efficiency based on the δ-axis current iδ. More specifically, the exciting current command control unit 119 obtains a current phase angle at which the output torque is equal to or higher than the set value or becomes the maximum value based on the δ-axis current iδ, and the γ-axis current is based on the obtained current phase angle. Calculate the command value iγ *. The γ-axis current command value iγ * is an excitation current command value. The γ-axis current command value iγ * may be calculated based on the current phase angle at which the motor current is equal to or less than the set value or becomes the minimum value.

Further, FIG. 3 illustrates a configuration in which the γ-axis current command value iγ * is obtained based on the δ-axis current iδ, but the configuration is not limited to this configuration. Instead of the δ-axis current iδ, the γ-axis current command value iγ * may be calculated based on the γ-axis current iγ and the velocity command value ω *.

The voltage command calculation unit 115 sets the γ-axis voltage command value Vγ * and the δ-axis voltage command based on the γ-axis current command value iγ *, the γ-axis current iγ, the δ-axis current iδ, and the speed command value ω *. Calculate the value Vδ * and the estimated velocity value ωest. The γ-axis voltage command value Vγ * is an excitation voltage command value, and the δ-axis voltage command value Vδ * is a torque voltage command value. The speed estimation value ωest represents an estimated value of the speed of one cycle for controlling the inverter 30 by an electric angle. The detailed configuration of the voltage command calculation unit 115 will be described later.

The electric angle phase calculation unit 118 integrates the velocity estimation value ωest to generate the electric angle phase θ used inside the inverter control unit 110.

The coordinate conversion unit 116 is a conversion unit that converts the voltage command value of the γ-δ axis into the voltage command value of the UVW phase, that is, the voltage command value of the three-phase coordinate system. More specifically, the coordinate conversion unit 116 sets the γ-axis voltage command value Vγ * and the δ-axis voltage command value Vδ * into a three-phase voltage command value which is a voltage command value of the three-phase coordinate system using the electric angle phase θ. Convert to the values Vu *, Vv *, Vw *.

The PWM signal generation unit 117 generates PWM signals Sm1 to Sm6 based on the three-phase voltage command values Vu *, Vv *, and Vw *.

When the above-mentioned stop signal St is generated by the operation control unit 102, the generated stop signal St is given to the PWM signal generation unit 117. When the PWM signal generation unit 117 receives the stop signal St, the PWM signal generation unit 117 stops the output of the PWM signals Sm1 to Sm6. As a result, the switching element of the inverter 30 stops the switching operation.

In the example of FIG. 3, the motor currents iu, iv, and iwa are restored from the detected value Idc of the input current of the inverter 30, but the configuration is not limited to this. Current detectors may be provided at at least two of the output lines 331, 332, and 333 of the inverter 30, and the phase current may be directly detected by these current detectors. In the case of this configuration, the current restoration unit 111 can be omitted in the inverter control unit 110.

Next, the configuration of the voltage command calculation unit 115 will be described. FIG. 4 is a block diagram showing a configuration example of the voltage command calculation unit 115 shown in FIG. As shown in FIG. 4, the voltage command calculation unit 115 includes a speed estimator 131, an adder / subtractor 132, 136, 137, 140, 141, a mechanical angle phase calculator 133, a filter 134, a speed controller 135, and a γ-axis current control. It includes a device 138, a δ-axis current controller 139, a first non-interfering controller 142, and a second non-interfering controller 143. In each adder / subtractor, the addition or subtraction operation is performed according to the plus (+) or minus (−) sign attached to the side.

The speed estimator 131 calculates the speed estimation value ωest based on the γ-axis current iγ, the δ-axis current iδ, the γ-axis voltage command value Vγ *, and the δ-axis voltage command value Vδ *. The speed estimated value ωest is an estimated value corresponding to the rotation speed of the motor 7. The speed of the motor 7 changes due to load fluctuations. Therefore, the speed estimator 131 estimates the rotation speed of the motor 7 that changes due to the load fluctuation, and outputs the speed estimation value ωest corresponding to the estimated rotation speed. The estimated speed estimation value ωest is used in the calculation of the voltage command value described later.

The speed estimated value ωest is an estimated value used inside the control device 100. Inside the control device 100, the change of the electric angle from 0 to 2π is treated as one cycle corresponding to the control cycle of the inverter 30. Therefore, even if the estimated value of the rotational speed of the motor 7 is the same, the value of the speed estimated value ωest differs depending on the pole logarithm P of the motor 7. For example, if the pole logarithm P is 1, the estimated value of the rotational speed of the motor 7 and the estimated speed value ωest match. Further, when the pole logarithm P is 2, the value of the speed estimation value ωest is twice the estimated value of the rotation speed of the motor 7.

The speed estimation value ωest calculated by the speed estimator 131 is input to the addition / subtractor 132. The adder / subtractor 132 calculates the velocity deviation Δω, which is the deviation between the velocity command value ω * and the velocity estimated value ωest.

The mechanical angle phase calculator 133 includes an integrator 133a and a multiplier 133b. The velocity estimation value ωest becomes the value of the phase angle expressed by the electric angle by being integrated by the integrator 133a. Further, the output of the integrator 133a becomes the value of the phase angle expressed by the mechanical angle by multiplying by the multiplier 133b by the reciprocal of the pole logarithm P of the motor 7. In this way, the mechanical angle phase calculator 133 calculates the mechanical angle phase θme indicating the rotational position of the motor 7 based on the speed estimation value ωest.

The filter 134 removes the load fluctuation component included in the velocity deviation Δω based on the mechanical angle phase θme. The term "removal" as used herein includes the concept of "reduction" in which a part of the load fluctuation component is removed. Hereinafter, the term "removal" is used in the present specification including the concept of "reduction".

As the filter 134, a low-pass filter or a notch filter can be used. The output from which the load fluctuation component is removed from the velocity deviation Δω is input to the velocity controller 135 as a new velocity deviation Δωf. When distinguishing between the velocity deviation Δω and the velocity deviation Δωf, the velocity deviation Δω may be referred to as a “first velocity deviation” and the velocity deviation Δωf may be referred to as a “second velocity deviation”. The function and configuration of the filter 134 will be described later.

The speed controller 135 calculates the δ-axis current command value iδ * based on the speed deviation Δωf. The δ-axis current command value iδ * is the command value of the δ-axis current iδ at which the velocity deviation Δω becomes zero, in other words, the command value of the δ-axis current iδ for matching the speed command value ω * with the velocity estimated value ωest. Is. An example of the speed controller 135 is a Proportional-Integral (PI) controller.

Although the filter 134 is provided in the front stage of the speed controller 135 in FIG. 4, the filter 134 may be provided in the rear stage of the speed controller 135. When the filter 134 is provided after the speed controller 135, the load fluctuation component included in the δ-axis current command value iδ * for matching the speed command value ω * and the speed estimation value ωest is removed.

The adder / subtractor 136 calculates the deviation between the γ-axis current command value iγ * and the γ-axis current iγ. The γ-axis current controller 138 is composed of, for example, a PI controller, and operates so as to converge the deviation between the γ-axis current command value iγ * and the γ-axis current iγ to zero.

The adder / subtractor 137 calculates the deviation between the δ-axis current command value iδ * and the δ-axis current iδ. The δ-axis current controller 139 is composed of, for example, a PI controller, and operates so as to converge the deviation between the δ-axis current command value iδ * and the δ-axis current iδ to zero.

The first non-interfering controller 142 includes a multiplier 142a. The first non-interference controller 142 calculates the compensation value Vγff * of the γ-axis voltage command based on the δ-axis current command value iδ * and the velocity estimation value ωest. The compensation value Vγff * of the γ-axis voltage command is a compensation value for suppressing mutual interference with the γ-axis due to the δ-axis current command value iδ *. As shown in FIG. 4, the compensation value Vγff * of the γ-axis voltage command is calculated by multiplying the δ-axis current command value iδ * by the δ-axis inductance Lδ of the motor 7 and the speed estimation value ωest.

When the first non-interference controller 142 does not exist, the first γ-axis voltage command value Vγfb * output from the γ-axis current controller 138 is the γ-axis voltage command value Vγ output from the voltage command calculation unit 115. *. On the other hand, when the first non-interference controller 142 is present, the addition / subtractor 140 compensates for the γ-axis voltage command which is the output of the first non-interference controller 142 from the first γ-axis voltage command value Vγfb *. The value Vγff * is subtracted. Then, the second voltage command value (Vγfb * −Vγff *), which is the output of the adder / subtractor 140, is set to the γ-axis voltage command value Vγ * output from the voltage command calculation unit 115.

The second non-interfering controller 143 includes multipliers 143a, 143c and adder / subtractor 143b. The second non-interference controller 143 calculates the compensation value Vδff * of the δ-axis voltage command based on the γ-axis current command value iγ * and the velocity estimation value ωest. The compensation value Vδff * of the δ-axis voltage command is a compensation value for suppressing mutual interference with the δ-axis due to the γ-axis current command value iγ *. As shown in FIG. 4, in the multiplier 143a, the γ-axis current command value iγ * and the γ-axis inductance Lγ of the motor 7 are multiplied. In the adder / subtractor 143b, the magnetic flux chain crossover vector φf of the motor 7 is added to the output of the multiplier 143a. In the multiplier 143c, the output of the adder / subtractor 143b is multiplied by the velocity estimate ωest. Then, the output of the multiplier 143c is input to the adder / subtractor 141 as the compensation value Vδff * of the δ-axis voltage command.

When the second non-interference controller 143 does not exist, the first δ-axis voltage command value Vδfb * output from the δ-axis current controller 139 is the δ-axis voltage command value Vδ output from the voltage command calculation unit 115. *. On the other hand, when the second non-interference controller 143 is present, the addition / subtractor 141 compensates for the δ-axis voltage command which is the output of the second non-interference controller 143 from the first δ-axis voltage command value Vδfb *. The value Vδff * is subtracted. Then, the second voltage command value (Vδfb * −Vδff *), which is the output of the adder / subtractor 140, is set to the δ-axis voltage command value Vδ * output from the voltage command calculation unit 115.

Next, in the voltage command calculation unit 115 in the first embodiment, the significance of providing the filter 134 in the front stage or the rear stage of the speed controller 135 will be described. In the explanation, it is assumed that the load is a compressor.

When the load is a compressor, in the motor 7, the load fluctuation generated by the compressor structure may occur n times as many times as one cycle (n is a natural number) in one cycle in which the motor 7 rotates once. Are known. When the compressor is, for example, a single rotary compressor and a scroll compressor, n = 1. Further, when the compressor is, for example, a twin rotary compressor, n = 2.

When there is a load fluctuation, the speed estimation value ωest fluctuates according to the load fluctuation. Therefore, in the absence of the filter 134, the δ-axis current command value iδ * controlled based on the velocity deviation Δω contains a frequency component n times the mechanical angular frequency of the motor 7. The mechanical angular frequency of the motor 7 referred to here is the rotation speed of the motor 7 expressed in rotation angles (radians) per unit time (1 second). If this fluctuation component is included in the δ-axis current command value iδ *, the phase current of each phase of the motor 7 varies as described later, so that highly efficient operation cannot be performed.

On the other hand, if the filter 134 is controlled to remove a specific frequency component from the velocity deviation Δω or the δ-axis current command value iδ *, the influence of the load fluctuation is suppressed and the value of the δ-axis current command value iδ * is set. It can be made smaller. The "specific frequency component" referred to here is a frequency component n times the mechanical angular frequency in which the rotation speed of the motor 7 is represented by the rotation angle per unit time. “N” is a natural number described above. If the value of the δ-axis current command value iδ * can be reduced, the variation in the current of each phase of the motor 7 can be reduced, and highly efficient operation becomes possible.

Next, the effect of improving the operating efficiency depending on the presence or absence of the filter 134 will be described with reference to the drawings of FIGS. 5 to 8. FIG. 5 is a first waveform diagram for explaining the effect of the first embodiment. FIG. 6 is a characteristic diagram for explaining the effect of the first embodiment. FIG. 7 is a second waveform diagram for explaining the effect of the first embodiment. FIG. 8 is a third waveform diagram for explaining the effect of the first embodiment.

First, "vibration suppression control", which is a premise of the waveform of FIG. 5, will be described.

In the refrigeration cycle apparatus, in order to reduce the vibration of the motor 7, the speed fluctuation of the motor 7 is controlled to be small, in other words, the speed deviation Δω is controlled to be small. When the speed fluctuation of the motor 7 becomes small, the vibration becomes small. Therefore, the control for reducing the velocity deviation Δω is generally called “vibration suppression control”.

The speed deviation Δω can be expressed by the following equation (1) using the output torque τm of the motor 7, the load torque τL, and the inertia J of the motor 7.

As shown in the above equation (1), if the output torque τm is smaller than the load torque τL, the rotation speed of the motor 7 is smaller than the command value. On the contrary, if the output torque τm is larger than the load torque τL, the rotation speed of the motor 7 becomes larger than the command value.

FIG. 5 shows various waveform examples when vibration suppression control is performed. More specifically, in the upper part of FIG. 5, the rotational speed of the motor 7 (hereinafter, simply referred to as “rotational speed”) ωm is shown by a fine solid line, and the estimated speed value ωest is shown by a thick solid line. The command value ω * is shown by the broken line. In the middle upper part of FIG. 5, the U-phase current iu is shown by a fine solid line, the V-phase current iv is shown by a thick solid line, and the W-phase current iwa is shown by a broken line. In the lower middle part of FIG. 5, the δ-axis current command value iδ * is shown by a fine solid line, and the δ-axis current iδ is shown by a thick solid line. In the lower part of FIG. 5, the output torque τm is shown by a solid line, and the load torque τL is shown by a broken line.

According to the lower part of FIG. 5, it is shown that the output torque τm follows the fluctuation of the load torque τL. In the vibration suppression control, the output torque τm shown in the lower part of FIG. 5 is output to the motor 7, so that the amplitude of the phase current of the motor 7 becomes large. Further, in the vibration suppression control, as shown in the middle upper part of FIG. 5, the peak value of each phase current is large, the peak value of each phase current has a large time change, and the peak value of each phase current is large. Is scattered from phase to phase. Therefore, in the vibration suppression control, it cannot be expected to perform highly efficient operation control.

Next, FIG. 6 will be described. FIG. 6 shows the relationship between the attenuation factor of the overall value and the time constant of the filter when the high efficiency control is performed. The characteristics shown in FIG. 6 are examples when a low-pass filter is used for the filter 134 shown in FIG. Further, the characteristics shown in FIG. 6 are an example when the load is a single rotary compressor.

The horizontal axis of FIG. 6 shows a multiple of the time constant of the low-pass filter (hereinafter referred to as “filter time constant”) determined based on the mechanical angular frequency of the motor 7. For example, a multiple of 10 on the horizontal axis means that the filter time constant is set to 10 times the reciprocal of the mechanical angular frequency of the motor 7.

Further, the vertical axis of FIG. 6 shows the attenuation rate of the overall value of the motor current. Here, the overall value is the total value of the frequency components up to a certain frequency band after frequency analysis of the motor current. Specifically, FIG. 6 shows the attenuation rate of the overall value from 0 Hz to 1000 Hz. It should be noted that a small overall value is equivalent to a large attenuation rate of the overall value. Further, the larger the attenuation rate of the overall value, the smaller the variation in the phase current of the motor 7, so that more efficient operation becomes possible.

Further, the damping rate of the overall value shown in FIG. 6 is based on the overall value at the time of vibration suppression control shown in FIG. That is, the overall value at the time of vibration suppression control shown in FIG. 5 is set to the damping rate “0%” in FIG. Further, FIG. 6 shows that, for example, the attenuation factor when the time constant of the low-pass filter is set to 10 times the mechanical angular frequency is about 29%. This means that there is a 29% improvement effect on the overall value at the time of vibration suppression control shown in FIG.

Further, FIG. 6 shows two examples when the rotation speed of the motor 7 is 37 rps and when the rotation speed is 80 rps. According to these examples, it can be seen that the difference in the rotational speed of the motor 7 does not significantly affect the characteristics with respect to the multiple of the filter time constant.

Further, in FIG. 6, increasing the multiple of the filter time constant means that the cutoff frequency of the filter moves to the low frequency side and the δ-axis current command value iδ * approaches the DC value. As the filter time constant is increased, the attenuation rate of the overall value saturates at a value of about 33%, as shown in FIG. From this, the following can be said.

(1) If the filter time constant is set so that the frequency component n times the mechanical angular frequency of the motor 7 is sufficiently attenuated, the variable frequency component of the load fluctuation is removed from the current command.
(2) In the control in which the δ-axis current command value iδ * is only brought close to the DC value by using a low-pass filter, the limit of the attenuation rate of the overall value is about 33%.

In any case, if the filter time constant is set to 20 times or more the reciprocal of the mechanical angular frequency of the motor 7, the waveform of FIG. 6 has an improvement effect of 33%, which is considered to be the limit value of the attenuation rate of the overall value. It shows that it can be done. Considering the actual condition of control, it is not always necessary to obtain an improvement effect of 33%. For example, it is sufficient if an effect of 10% to 15% reduction, that is, an effect of 28% or more can be obtained. Is. In FIG. 6, when the multiple of the filter time constant is 10, the attenuation factor is about 29%. Therefore, if the multiple of the filter time constant is 10 or more, it can be said that it is suitable for the high efficiency control referred to in the present embodiment.

Further, FIGS. 7 and 8 show an example of a waveform similar to that in FIG. 5 when high-efficiency control is performed using a low-pass filter in which the filter time constant shown in FIG. 6 is set to 25 times. However, in FIG. 7, as the information of the angular velocity used in the first non-interference controller 142 and the second non-interference controller 143 shown in FIG. 4, the velocity command value ω * is used instead of the velocity estimation value ωest. ing. On the other hand, in FIG. 8, the information on the angular velocity used in the first non-interfering controller 142 and the second non-interfering controller 143 shown in FIG. 4 uses the estimated velocity value ωest as shown in FIG. There is. In Patent Document 1, the velocity command value ω * is used as the information of the angular velocity used in the non-interference controller.

As shown in the lower part of FIG. 5, it is a feature of vibration suppression control that the output torque τm follows the fluctuation of the load torque τL. On the other hand, in the high-efficiency control shown in FIG. 8, as shown in the middle and lower parts of FIG. 8, the δ-axis current iδ follows the δ-axis current command value iδ *, and the δ-axis current command value iδ * It is controlled so as to be substantially in agreement with the δ-axis current iδ.

Further, as shown in the lower part of FIG. 8, the output torque τe does not follow the load torque τL and becomes a substantially constant value. As a result, as shown in the middle upper part of FIG. 8, the peak value of each phase current becomes a constant value, and the variation of the peak value of each phase current for each phase is extremely small as compared with FIG. Although only the calculation result is shown, the attenuation rate of the overall value in the case of the waveform example shown in FIG. 8 is 43.1%.

In contrast to FIG. 8, in FIG. 7, as shown in the middle and lower stages, the δ-axis current command value iδ * does not follow the δ-axis current command value iδ *, and the δ-axis current command value iδ * and the δ-axis current iδ are one. I haven't done it. Further, as shown in the lower part of FIG. 7, the output torque τe contains more components n times the mechanical angular frequency. As a result, as shown in the middle upper part of FIG. 7, the peak value of each phase current is not constant, and the improvement effect by the high efficiency control is smaller than that in the case of FIG.

As shown in FIG. 8, when the peak value of each phase current is controlled to be constant, the output torque τe is a direct current value that does not include an AC component. Therefore, it can be said that the more the output torque τe is a DC value that does not include an AC component, the more efficient the control is. However, since the output torque is also a DC value, the control is such that the load torque and the output torque do not match, and the speed fluctuation becomes large.

Next, the configuration and intent when the filter 134 shown in FIG. 5 is used as a notch filter will be described with reference to FIG. FIG. 9 is a block diagram showing a configuration example when the filter 134 shown in FIG. 4 is a notch filter.

When the filter 134 is a notch filter, a value n times the mechanical angular frequency is set as the filter center frequency. As a result, the filter center frequency is adjusted to the fluctuation frequency component due to load fluctuation, and the fluctuation component due to load fluctuation is removed from the current command value.

In the case of a notch filter, a filter having a narrow removal band is generally used. Therefore, only the band of the fluctuation component due to the load fluctuation is attenuated to a low level from the current command value. Therefore, it is desirable to configure the filter center frequency to be variable according to the motor speed. If the filter center frequency is configured to be variable according to the motor speed, it is possible to appropriately remove the variable component that changes according to the set speed.

The filter 134, which is a notch filter, can be configured as shown in FIG. 9, for example. As shown in FIG. 9, the filter 134 includes a multiplier 161, a sine value calculator 162,166, a cosine value calculator 163,167, a low-pass filter 164,165, and an addition / subtractor 168,169.

The velocity deviation Δω, which is the output of the adder / subtractor 132, and the mechanical angle phase θme, which is the output of the mechanical angle phase calculator 133, are input to the filter 134.

The multiplier 161 outputs a value of n × θme obtained by multiplying the machine angle phase θme by n to the sine and cosine value calculator 162 and the cosine value calculator 163.

In the sine and cosine value calculator 162, a calculation process of Δω × sin (n × θme) is performed. The output of the sine and cosine calculator 162 is passed through a low-pass filter 164 having a time constant of Tf. A DC amount can be obtained by passing through the low-pass filter 164. This DC amount is multiplied by the value of sin (n × θme) by the sine and cosine value calculator 166, and is output to the adder / subtractor 168 as a DC amount in the imaginary axis direction on the complex plane.

Further, in the cosine value calculator 163, a calculation process of Δω × cos (n × θme) is performed. The output of the cosine value calculator 163 is passed through a low-pass filter 165 having a time constant Tf. A DC amount can be obtained by passing through the low-pass filter 165. This DC amount is multiplied by the value of cos (n × θme) by the cosine value calculator 167 and output to the addition / subtractor 168 as a DC amount in the real axis direction on the complex plane.

In the adder / subtractor 168, the DC amount in the imaginary axis direction on the complex plane and the DC amount in the real axis direction on the complex plane are added. As a result, the output of the adder / subtractor 168 becomes an alternating current amount Δωnf due to the n times component of the mechanical angular frequency. Then, in the adder / subtractor 169, the deviation between the velocity deviation Δω and the alternating current amount Δωnf is calculated. The output of the adder / subtractor 169 is an output obtained by removing the alternating current amount Δωnf caused by the n times component of the mechanical angular frequency from the velocity deviation Δω, and is input to the velocity controller 135 as a second velocity deviation Δωf. Subsequent operations are as described above.

Next, the hardware configuration for realizing the function of the control device 100 in the first embodiment will be described with reference to the drawings of FIGS. 10 and 11. FIG. 10 is a block diagram showing an example of a hardware configuration that realizes the function of the control device 100 according to the first embodiment. FIG. 11 is a block diagram showing another example of the hardware configuration that realizes the function of the control device 100 according to the first embodiment.

In order to realize a part or all of the functions of the control device 100, as shown in FIG. 10, the processor 200 that performs the calculation, the memory 202 that stores the program read by the processor 200, and the input / output of the signal are performed. It can be configured to include the interface 204.

The processor 200 may be an arithmetic unit such as an arithmetic unit, a microprocessor, a microcomputer, a CPU (Central Processing Unit), or a DSP (Digital Signal Processor). Further, the memory 202 includes a non-volatile or volatile semiconductor memory such as a RAM (Random Access Memory), a ROM (Read Only Memory), a flash memory, an EPROM (Erasable Program ROM), and an EPROM (registered trademark) (Electrically EPROM). Examples thereof include magnetic discs, flexible discs, optical discs, compact discs, mini discs, and DVDs (Digital Versailles Disc).

The memory 202 stores a program that executes the function of the control device 100. The processor 200 can realize the function of the control device 100 by exchanging necessary information via the interface 204 and executing the program stored in the memory 202 by the processor 200. The calculation result by the processor 200 can be stored in the memory 202.

Further, the processor 200 and the memory 202 shown in FIG. 10 may be replaced with the processing circuit 203 as shown in FIG. The processing circuit 203 corresponds to a single circuit, a composite circuit, an ASIC (Application Specific Integrated Circuit), an FPGA (Field-Programmable Gate Array), or a combination thereof.

It should be noted that some processing in the control device 100 may be performed by the processing circuit 203, and processing not performed by the processing circuit 203 may be performed by the processor 200 and the memory 202.

As described above, according to the first embodiment, the filter provided in the front stage of the speed controller is a deviation between the speed command value and the speed estimation value based on the mechanical angle phase indicating the rotation position of the motor. The fluctuation component due to the load fluctuation included in the first velocity deviation is removed. The speed controller generates a current command value based on the second speed deviation output from the filter. The current controller calculates a voltage command value, which is a command value of the voltage applied to the motor, based on the current command value from which a specific frequency component is removed and the motor current. The specific frequency component is a frequency component that is a natural number multiple of the mechanical angular frequency. This makes it possible to reduce the pulsating component of the current command value in the process in which the control device controls the inverter that drives the motor. This makes it possible to reliably carry out highly efficient drive control for the motor.

The filter may be provided after the speed controller. In this configuration, the speed controller generates a current command value based on the speed deviation, which is the deviation between the speed command value and the speed estimate value. The filter removes the fluctuation component due to the load fluctuation included in the current command value based on the mechanical angle phase indicating the rotation position of the motor. The current controller calculates a voltage command value, which is a command value of the voltage applied to the motor, based on the current command value from which the fluctuation component due to the load fluctuation is removed and the motor current. Also with this configuration, the pulsating component of the current command value can be reduced, and highly efficient drive control for the motor can be reliably performed.

In the conventional current control, the pulsating component of the torque current command value is controlled to be zero by integral control. However, in the integral control, a phase delay occurs, and the control may break down depending on the value of the integral gain. On the other hand, in the method of the first embodiment, the torque current command value is set to the front stage or the rear stage of the speed controller by using a notch filter without performing the integral control so that the pulsating component of the torque current command value becomes zero. I try to get close to a certain value. As a result, the occurrence of a phenomenon that a desired control gain value cannot be obtained in the control process can be reliably avoided.

Further, in the conventional non-interference control, the speed command value ω * is used for the information of the angular velocity used in the non-interference controller. When the angular velocity used in the non-interfering controller is set to the speed command value ω *, the non-interfering controller may not work effectively and the control for following the actual current to the current command value may be delayed. On the other hand, in the method of the first embodiment, since the speed estimation value ωest is used for the information of the angular velocity used in the non-interference controller, it is possible to quickly control to make the actual current follow the current command value. .. This enables more efficient drive control while speeding up control.

Embodiment 2.
FIG. 12 is a diagram showing a configuration example of the refrigeration cycle apparatus 500 according to the second embodiment. The refrigeration cycle device 500 according to the second embodiment includes the load drive device 300 described in the first embodiment. The refrigerating cycle device 500 according to the second embodiment can be applied to products including a refrigerating cycle such as an air conditioner, a refrigerator, a freezer, and a heat pump water heater. In FIG. 12, the components having the same functions as those of the first embodiment are designated by the same reference numerals as those of the first embodiment.

In the refrigeration cycle device 500, the compressor 501 having a built-in motor 7 in the first embodiment, the four-way valve 502, the indoor heat exchanger 506, the expansion valve 508, and the outdoor heat exchanger 510 are connected via a refrigerant pipe 512. Is attached.

Inside the compressor 501, a compression mechanism 504 for compressing the refrigerant and a motor 7 for operating the compression mechanism 504 are provided.

The refrigeration cycle device 500 can perform a heating operation or a cooling operation by switching the four-way valve 502. The compression mechanism 504 is driven by a motor 7 that is controlled at a variable speed.

During the heating operation, as shown by the solid arrow, the refrigerant is pressurized by the compression mechanism 504 and sent out, and passes through the four-way valve 502, the indoor heat exchanger 506, the expansion valve 508, the outdoor heat exchanger 510, and the four-way valve 502. Return to the compression mechanism 504.

During the cooling operation, as shown by the broken line arrow, the refrigerant is pressurized by the compression mechanism 504 and sent out, and passes through the four-way valve 502, the outdoor heat exchanger 510, the expansion valve 508, the indoor heat exchanger 506, and the four-way valve 502. Return to the compression mechanism 504.

During the heating operation, the indoor heat exchanger 506 acts as a condenser to release heat, and the outdoor heat exchanger 510 acts as an evaporator to absorb heat. During the cooling operation, the outdoor heat exchanger 510 acts as a condenser to release heat, and the indoor heat exchanger 506 acts as an evaporator to absorb heat. The expansion valve 508 depressurizes the refrigerant and expands it.

The configuration shown in the above embodiments is an example of the content of the present invention, can be combined with another known technique, and is configured without departing from the gist of the present invention. It is also possible to omit or change a part of.

1 AC power supply, 2 reactor, 3 rectifier circuit, 5 smoothing capacitor, 7 motor, 10 voltage detector, 12a, 12b DC bus, 30 inverter, 40 current detector, 100 controller, 102 operation control unit, 110 inverter control unit , 111 Current restoration unit, 112,116 Coordinate conversion unit, 115 Voltage command calculation unit, 117 PWM signal generation unit, 118 Electric angle phase calculation unit, 119 Excitation current command control unit, 131 Speed estimator, 132,136,137, 140, 141,143b, 168,169 Add / subtractor, 133 Mechanical angle phase calculator, 133a Inverter, 133b, 142a, 143a, 143c, 161 Multiplier, 134 Filter, 135 Speed controller, 138 γ-axis current controller, 139 δ-axis current controller, 142 first non-interfering controller, 143 second non-interfering controller, 162,166 sine value calculator, 163,167 chord value calculator, 164,165 low pass filter, 200 processor, 202 memory, 203 processing circuit, 204 interface, 300 load drive, 310 main circuit, 310A, 310B, 310C leg, 311 transistor, 312 diode, 3211,322,323 connection point, 331,332,333 output line, 350 drive Circuit, 500 refrigeration cycle device, 501 compressor, 502 four-way valve, 504 compression mechanism, 506 indoor heat exchanger, 508 expansion valve, 510 outdoor heat exchanger, 512 refrigerant piping, D1, D2, D3, D4 diode, UP, VP, WP upper arm switching element, UN, VN, WN lower arm switching element.

Claims (9)

A load drive device that supplies AC power to a motor to drive a load.
A rectifier circuit that rectifies AC voltage and converts it to DC voltage,
An inverter that converts DC power output from the rectifier circuit into AC power ,
A control device for controlling the inverter is provided.
The control device is
A speed estimator that calculates a speed estimate, which is an estimate of the rotational speed of the motor,
A speed controller that generates a current command value based on the speed deviation, which is the deviation between the speed command value and the speed estimation value.
A mechanical angle phase calculator that calculates the mechanical angle phase indicating the rotational position of the motor based on the speed estimation value, and
A filter that removes a specific frequency component from the fluctuation component due to the load fluctuation included in the speed deviation or the current command value output from the speed controller.
A current controller that calculates a voltage command value, which is a command value of a voltage applied to the motor, based on a current command value from which the specific frequency component has been removed and a motor current flowing through the motor.
Equipped with
The specific frequency component is a frequency component determined by the load, and is a frequency component that is a natural several times the mechanical angular frequency in which the rotation speed of the motor is expressed by the rotation angle per unit time.
The time constant of the filter is set to be 10 times or more the reciprocal of the mechanical angular frequency.
Load drive device. A load drive device that supplies AC power to a motor to drive a load.
A rectifier circuit that rectifies AC voltage and converts it to DC voltage,
An inverter that converts DC power output from the rectifier circuit into AC power,
A control device for controlling the inverter is provided.
The control device is
A speed estimator that calculates a speed estimate, which is an estimate of the rotational speed of the motor,
A speed controller that generates a current command value based on the speed deviation, which is the deviation between the speed command value and the speed estimation value.
A mechanical angle phase calculator that calculates the mechanical angle phase indicating the rotational position of the motor based on the speed estimation value, and
A filter that removes a specific frequency component from the fluctuation component due to the load fluctuation included in the speed deviation or the current command value output from the speed controller.
A current controller that calculates a voltage command value, which is a command value of a voltage applied to the motor, based on a current command value from which the specific frequency component has been removed and a motor current flowing through the motor.
A non-interference controller that calculates the compensation value of the voltage command value for suppressing mutual interference between the exciting current component and the torque current component based on the speed estimation value .
Equipped with
The specific frequency component is a frequency component determined by the load, and is a frequency component that is several times the natural angular frequency of the rotation speed of the motor expressed by the rotation angle per unit time.
Load drive device. The load drive device according to claim 1 or 2 , wherein the current controller controls the motor current to match the current command value. The load drive device according to any one of claims 1 to 3 , wherein the filter is a low-pass filter. The load drive device according to any one of claims 1 to 3 , wherein the filter is a notch filter. A current detector for detecting the input current of the inverter is provided.
The load drive device according to any one of claims 1 to 5 , wherein the motor current is calculated based on a value detected by the current detector. The load drive device according to any one of claims 1 to 5 , further comprising a current detector for detecting the motor current. A refrigeration cycle device comprising the load drive device according to any one of claims 1 to 7. An air conditioner provided with the refrigeration cycle apparatus according to claim 8.

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