WO2022085048A1

WO2022085048A1 - Motor drive device, electric blower, electric vacuum cleaner, and hand dryer

Info

Publication number: WO2022085048A1
Application number: PCT/JP2020/039273
Authority: WO
Inventors: 裕次 ▲高▼山; 和徳畠山
Original assignee: 三菱電機株式会社
Priority date: 2020-10-19
Filing date: 2020-10-19
Publication date: 2022-04-28
Also published as: JP7462788B2; JPWO2022085048A1

Abstract

A motor drive device (2) is provided with an inverter (11) disposed between a battery (10) and a single-phase motor (12) and drives the single-phase motor (12) in a position sensorless manner. When the single-phase motor (12) is started, the inverter (11) applies a first voltage to the single-phase motor (12) in a first time period immediately after the single-phase motor (12) is started and applies a second voltage having the reverse polarity to the polarity of the first voltage in a second time period after the application of the first voltage. A suspension time period during which application of the first voltage is suspended is provided between the first time period and the second time period.

Description

Motor drive, electric blower, vacuum cleaner and hand dryer

The present disclosure relates to a motor drive device that drives a single-phase motor without a position sensor, an electric blower equipped with a single-phase motor driven by the motor drive device, an electric vacuum cleaner, and a hand dryer.

Conventionally, when starting a multi-phase brushless motor without a position sensor, there is a method of applying a high frequency voltage so that the motor rotates following the rotating magnetic field generated by the inverter. Further, in Patent Document 1 below, in the method of starting a three-phase sensorless brushless motor, the initial position of the rotor is set by one energization, and the rotation speed of the rotor is increased based on the information of the set initial position. A method of detecting the position of the rotor after the rotation speed has increased is disclosed.

Japanese Unexamined Patent Publication No. 1-308192

As mentioned above, various starting methods have been proposed for multi-phase motors. On the other hand, in the case of a single-phase motor, a rotating magnetic field cannot be generated by the inverter. Therefore, depending on the stop position of the rotor magnetic pole, it may not be possible to start in the correct direction. If an appropriate voltage corresponding to the stop position of the rotor magnetic pole is not applied at the time of starting, a steep current is generated, which may damage the single-phase motor. Further, if a steep current flows at the time of starting, the overcurrent cutoff function may work and the single-phase motor may be stopped. Therefore, when starting a single-phase motor without a position sensor, safe and reliable starting is required.

The present disclosure has been made in view of the above, and an object of the present invention is to obtain a motor drive device capable of safely and reliably starting a single-phase motor when the single-phase motor is started without a position sensor. ..

In order to solve the above-mentioned problems and achieve the object, the motor drive device according to the present disclosure is a motor drive device that drives a single-phase motor without a position sensor. The motor drive includes an inverter located between the DC power supply and the single-phase motor. When the single-phase motor is started, the inverter applies the first voltage to the single-phase motor in the first period immediately after the start, and reverses the polarity of the first voltage in the second period after the application of the first voltage. The single-phase motor is started by applying the second voltage. Between the first period and the second period, there is a stop period during which the application of the first voltage is stopped.

According to the motor drive device according to the present disclosure, when the single-phase motor is started without a position sensor, the effect is that the single-phase motor can be started safely and surely.

A block diagram showing a configuration of a motor drive system including a motor drive device according to the first embodiment. Sectional drawing which provides the explanation of the structure of the single-phase motor in Embodiment 1. The figure which shows the change of the rotor position when the single-phase motor shown in FIG. 2 is excited. The figure which shows the torque characteristic of the single-phase motor shown in FIG. Circuit diagram of the inverter shown in FIG. A circuit diagram showing a modified example of the inverter shown in FIG. A block diagram showing a functional part that generates a pulse width modulation (PWM) signal among the functional parts of the control unit shown in FIG. 1. A block diagram showing an example of the carrier comparison unit shown in FIG. The figure which shows the waveform example of the main part when it operated using the carrier comparison part shown in FIG. A block diagram showing another example of the carrier comparison unit shown in FIG. The figure which shows the waveform example of the main part when it operated using the carrier comparison part shown in FIG. A block diagram showing a functional configuration for calculating the advance phase input to the carrier comparison unit shown in FIG. 7. The figure which shows an example of the calculation method of the advance angle phase in Embodiment 1. The figure used to explain the relationship between the voltage command shown in FIG. 7 and the advance phase. The figure used for the operation explanation of the main part in Embodiment 1. FIG. 2 is a diagram used to explain the relationship between the rotor position when the single-phase motor shown in FIG. 2 is stopped and the start control in the first embodiment. The figure used for the operation explanation of the main part in the motor drive device which concerns on Embodiment 2. Configuration diagram of the vacuum cleaner according to the third embodiment Configuration diagram of the hand dryer according to the third embodiment

The motor drive device, electric blower, vacuum cleaner and hand dryer according to the embodiment of the present disclosure will be described in detail with reference to the accompanying drawings below.

Embodiment 1.
FIG. 1 is a block diagram showing a configuration of a motor drive system 1 including a motor drive device 2 according to the first embodiment. The motor drive system 1 shown in FIG. 1 includes a single-phase motor 12, a motor drive device 2, and a battery 10. The motor drive device 2 is a drive device that supplies AC power to the single-phase motor 12 to drive the single-phase motor 12. The battery 10 is a DC power source that supplies DC power to the motor drive device 2.

The motor drive device 2 includes an inverter 11, an analog-digital converter 30, a control unit 25, and a drive signal generation unit 32. The inverter 11 and the single-phase motor 12 are connected by two connecting

lines

18a and 18b.

The motor drive system 1 includes

voltage detectors

20 and 21 and

current detectors

22 and 24. The motor drive system 1 is a so-called position sensorless control drive system that does not use a position sensor signal for detecting the rotation position of the rotor 12a.

The voltage detector 20 is a detector that detects the DC voltage _Vdc output from the battery 10 to the motor drive device 2. The DC voltage V _dc is the output voltage of the battery 10 and is the voltage applied to the inverter 11.

The voltage detector 21 is a detector that detects the AC voltage _Vac generated between the

connection lines

18a and 18b. The AC voltage V _ac is a voltage obtained by superimposing the motor applied voltage applied by the inverter 11 to the single-phase motor 12 and the motor-induced voltage induced by the single-phase motor 12. When the inverter 11 is stopped and the single-phase motor 12 is rotating, the motor induced voltage is observed. In this paper, the state in which the inverter 11 has stopped operating and the inverter 11 is not outputting a voltage is referred to as "gate off". Further, the voltage output by the inverter 11 is appropriately referred to as an "inverter output voltage".

The current detector 22 is a detector that detects the motor current _Im . The motor current _Im is an alternating current flowing in and out between the inverter 11 and the single-phase motor 12. The motor current Im is equal to the alternating current flowing through the windings (not shown in FIG. 1) wound around the stator _12b of the single-phase motor 12. Examples of the current detector 22 include a current transformer (CT) or a current detector that detects a current using a shunt resistor.

The current detector 24 is a detector that detects the power supply current I _dc . The power supply current I _dc is a direct current flowing between the battery 10 and the inverter 11. The current detector 24 is generally configured to use a shunt resistor as shown in the figure. The detected value of the power supply current I _dc flowing through the current detector 24 is converted into a voltage value and input to the analog-digital converter 30. In this paper, the detection value of the current detector 24 is appropriately referred to as "shunt voltage". Further, the shunt voltage, which is the detected value of the power supply current I _dc , has a correlation with the motor current _Im . That is, if the motor current _Im increases, the shunt voltage also increases, and if the motor current _Im decreases, the shunt voltage also decreases. Therefore, in this paper, the shunt voltage may be described as "a physical quantity correlated with the motor current _Im ". Further, in this paper, the current detector 24 may be referred to as a "first detector", and the current detector 22 may be referred to as a "second detector".

The single-phase motor 12 is used as a rotary electric machine for rotating an electric blower (not shown). Electric blowers are mounted on devices such as vacuum cleaners and hand dryers.

The inverter 11 is a power converter that converts the DC voltage _Vdc applied from the battery 10 into an AC voltage. The inverter 11 supplies AC power to the single-phase motor 12 by applying the converted AC voltage to the single-phase motor 12.

The analog-to-digital converter 30 is a signal converter that converts analog data into digital data. The analog-digital converter 30 converts the detected value of the DC voltage V _dc detected by the voltage detector 20 and the detected value of the AC voltage V _ac detected by the voltage detector 21 into digital data, and causes the control unit 25. Output. Further, the analog-digital converter 30 converts the detected value of the motor current _{Im detected by the current detector 22 and the detected value of the power supply current I dc} _detected by the current detector 24 into digital data, and the control unit 25. Output to.

The control unit 25 is referred to as PWM signals Q1, Q2, Q3, Q4 (hereinafter, appropriately referred to as "Q1 to Q4") based on the digital output value 30a converted by the analog digital converter 30 and the voltage amplitude command V *. ) Is generated. The voltage amplitude command V * will be described later.

The drive signal generation unit 32 has drive signals S1, S2, S3, S4 for driving the switching element in the inverter 11 based on the PWM signals Q1 to Q4 output from the control unit 25 (hereinafter, appropriately "S1 to". S4 ") is generated.

The control unit 25 has a processor 31, a carrier generation unit 33, and a memory 34. The processor 31 generates PWM signals Q1 to Q4 for performing PWM control. The processor 31 is a processing unit that performs various operations related to PWM control and advance angle control. As the processor 31, a CPU (Central Processing Unit), a microprocessor, a microcomputer, a microcomputer, a DSP (Digital Signal Processor), or a system LSI (Large Scale Integration) can be exemplified.

The program read by the processor 31 is stored in the memory 34. The memory 34 is also used as a work area when the processor 31 performs arithmetic processing. The memory 34 is generally a non-volatile or volatile semiconductor memory such as a RAM (Random Access Memory), a flash memory, an EPROM (Erasable Project ROM), or an EEPROM (registered trademark) (Electrically EPROM). Details of the configuration of the carrier generation unit 33 will be described later.

FIG. 2 is a cross-sectional view for explaining the structure of the single-phase motor 12 in the first embodiment. FIG. 2 shows the cross-sectional shapes of the rotor 12a and the stator 12b of the single-phase permanent magnet brushless motor as an example of the single-phase motor 12 used in the embodiment.

The rotor 12a is fitted to the shaft 12c and is configured to be rotatable in the direction of the arrow shown in the figure, that is, counterclockwise. Four permanent magnets are arranged in the circumferential direction on the rotor 12a. These four permanent magnets are arranged so that the magnetizing directions are alternately reversed in the circumferential direction to form a magnetic pole in the rotor 12a. In the first embodiment, the case where the number of magnetic poles of the rotor 12a is 4 poles is illustrated, but the number of magnetic poles of the rotor 12a may be other than 4 poles.

A stator 12b is arranged around the rotor 12a. The stator 12b is configured by connecting four divided cores 12d in an annular shape.

The split core 12d has an asymmetrically shaped tooth 12e. A winding 12f is wound around the teeth 12e. The teeth 12e has a first tip portion 12e1 and a second tip portion 12e2 protruding toward the rotor 12a. The side ahead of the rotation direction is the first tip portion 12e1, and the side behind the rotation direction is the second tip portion 12e2. Here, the distance between the first tip portion 12e1 and the rotor 12a is referred to as a "first gap" and is represented by G1. Further, the distance between the second tip portion 12e2 and the rotor 12a is called a "second gap" and is represented by G2. There is a relationship of G1 <G2 between the first gap G1 and the second gap G2.

The single-phase motor 12 may be a motor having a structure in which a permanent magnet is arranged on the surface of the rotor 12a (Surface Permanent Magnet: SPM), or a magnet-embedded type (Interior) in which the permanent magnet is embedded inside the rotor 12a. It may be a motor having a Permanent Magnet (IPM) structure. When the single-phase motor 12 is a motor having an SPM structure, there is an effect that the torque pulsation due to the reluctance torque can be reduced. Further, when the single-phase motor 12 is a motor having an IPM structure, there is an effect that the structure for holding the permanent magnet becomes easy.

FIG. 3 is a diagram showing changes in the rotor position when the single-phase motor 12 shown in FIG. 2 is excited. FIG. 4 is a diagram showing torque characteristics of the single-phase motor 12 shown in FIG. The stop position of the rotor 12a is shown in the upper part of FIG. At the stop position of the rotor 12a, the magnetic pole center line representing the center of the magnetic pole and the tooth center line representing the structural center of the stator 12b are deviated so that the magnetic pole center line precedes the rotation direction. This occurs because the single-phase motor 12 has a structure having an asymmetrically shaped teeth 12e. With this structure, the torque characteristics as shown in FIG. 4 appear.

In FIG. 4, the curve K1 shown by the solid line represents the motor torque, and the curve K2 shown by the broken line represents the cogging torque. The motor torque is the torque generated in the rotor 12a by the current flowing through the winding of the stator 12b. The cogging torque is the torque generated in the rotor 12a by the magnetic force of the permanent magnet when no current is flowing in the winding of the stator 12b. Take the counterclockwise direction to the positive torque. Further, the horizontal axis of FIG. 4 represents the machine angle, and the stop position of the rotor 12a whose magnetic pole center line coincides with the teeth center line is the machine angle 0 °. As shown in FIG. 4, the cogging torque is positive when the mechanical angle is 0 °. Therefore, the rotor 12a rotates counterclockwise and stops at the position of the mechanical angle θ1 where the cogging torque becomes zero. The position of the mechanical angle θ1 is the stop position shown in the upper part of FIG.

In the case of the single-phase motor 12 shown in FIG. 2, there are two stop positions for the rotor 12a. One of the stop positions is the stop position shown in the upper part of FIG. 3 described above, and the other is the stop position shown in the lower part of FIG. When a DC voltage is applied to the winding 12f, it rotates counterclockwise, passes through the state of excitation shown in the middle part of FIG. 3, and stops in the state shown in the lower part of FIG. In the case of the example of FIG. 3, since the magnetic force generated in the teeth 12e by applying the DC voltage is the same pole as the magnetic poles of the opposing rotors 12a, torque is applied in the rotation direction and the rotor 12a rotates. Then, after a certain period of time elapses, the magnetic force generated in the teeth 12e and the magnetic poles of the rotors 12a facing each other are stably stopped at the position of the lower portion of FIG.

FIG. 5 is a circuit diagram of the inverter 11 shown in FIG. The inverter 11 has a plurality of switching

elements

51, 52, 53, 54 (hereinafter, appropriately referred to as “51 to 54”) to be bridge-connected.

The switching

elements

51 and 52 constitute the first leg, the leg 5A. The leg 5A is a series circuit in which a switching element 51, which is a first switching element, and a switching element 52, which is a second switching element, are connected in series.

The switching

elements

53 and 54 constitute the second leg, the leg 5B. The leg 5B is a series circuit in which a switching element 53, which is a third switching element, and a switching element 54, which is a fourth switching element, are connected in series.

The

legs

5A and 5B are connected between the DC bus 16a on the high potential side and the DC bus 16b on the low potential side so as to be in parallel with each other. As a result, the

legs

5A and 5B are connected in parallel to both ends of the battery 10.

The switching

elements

51 and 53 are located on the high potential side, and the

switching elements

52 and 54 are located on the low potential side. Generally, in an inverter circuit, the high potential side is referred to as an "upper arm" and the low potential side is referred to as a "lower arm". Therefore, the switching element 51 of the leg 5A may be referred to as a "first switching element of the upper arm", and the switching element 52 of the leg 5A may be referred to as a "second switching element of the lower arm". Similarly, the switching element 53 of the leg 5B may be referred to as a "third switching element of the upper arm", and the switching element 54 of the leg 5B may be referred to as a "fourth switching element of the lower arm".

The connection end 6A between the switching element 51 and the switching element 52 and the connection end 6B between the switching element 53 and the switching element 54 form an AC end in the bridge circuit. A single-phase motor 12 is connected between the connection end 6A and the connection end 6B.

For each of the switching elements 51 to 54, a MOSFET (Metal-Oxide-Semiconductor Field-Effective Transistor), which is a metal oxide film semiconductor field effect transistor, is used. MOSFET is an example of FET (Field-Effective Transistor).

The switching element 51 is formed with a body diode 51a connected in parallel between the drain and the source of the switching element 51. The switching element 52 is formed with a body diode 52a connected in parallel between the drain and the source of the switching element 52. The switching element 53 is formed with a body diode 53a connected in parallel between the drain and the source of the switching element 53. The switching element 54 is formed with a body diode 54a connected in parallel between the drain and the source of the switching element 54. Each of the plurality of

body diodes

51a, 52a, 53a, 54a is a parasitic diode formed inside the MOSFET and is used as a freewheeling diode. A separate freewheeling diode may be connected. Further, instead of the MOSFET, an insulated gate bipolar transistor (IGBT) may be used.

The switching elements 51 to 54 are not limited to MOSFETs formed of silicon-based materials, and may be MOSFETs formed of wide bandgap (Wide Band Gap: WBG) semiconductors such as silicon carbide, gallium nitride, gallium oxide, or diamond.

Generally, WBG semiconductors have higher withstand voltage and heat resistance than silicon semiconductors. Therefore, by using a WBG semiconductor for at least one of the plurality of switching elements 51 to 54, the withstand voltage resistance and the allowable current density of the switching element are increased, and the semiconductor module incorporating the switching element can be miniaturized. In addition, WBG semiconductors have high heat resistance. Therefore, it is possible to reduce the size of the heat radiating portion for radiating the heat generated in the semiconductor module. In addition, it is possible to simplify the heat dissipation structure that dissipates heat generated by the semiconductor module.

Further, FIG. 6 is a circuit diagram showing a modified example of the inverter 11 shown in FIG. The inverter 11A shown in FIG. 6 has

shunt resistors

55a and 55b added to the configuration of the inverter 11 shown in FIG. The shunt resistor 55a is a detector for detecting the current flowing through the leg 5A, and the shunt resistor 55b is a detector for detecting the current flowing through the leg 5B. As shown in FIG. 6, the shunt resistor 55a is connected between the terminal on the low potential side of the switching element 52 and the DC bus 16b, and the shunt resistor 55b is connected to the terminal on the low potential side of the switching element 54 and the DC bus. It is connected to 16b. When the inverter 11A provided with the

shunt resistors

55a and 55b is used, the current detector 22 shown in FIG. 1 can be omitted. In this configuration, the detected values of the

shunt resistors

55a and 55b are sent to the processor 31 via the analog-digital converter 30. The processor 31 implements activation control, which will be described later, based on the detected values of the

shunt resistors

55a and 55b.

The shunt resistor 55a is not limited to that of FIG. 6 as long as it can detect the current flowing through the leg 5A. The shunt resistor 55a is located between the DC bus 16a and the terminal on the high potential side of the switching element 51, between the terminal on the low potential side of the switching element 51 and the connection end 6A, or between the connection end 6A and the high potential of the switching element 52. It may be arranged between the terminal on the side. Similarly, the shunt resistor 55b is between the DC bus 16a and the terminal on the high potential side of the switching element 53, between the terminal on the low potential side of the switching element 53 and the connection end 6B, or between the connection end 6B and the switching element 54. It may be arranged between the terminal on the high potential side of the. Further, instead of the

shunt resistors

55a and 55b, the on-resistance of the MOFFET may be used to detect the current with the voltage generated across the on-resistance.

FIG. 7 is a block diagram showing a functional part that generates a PWM signal among the functional parts of the control unit 25 shown in FIG.

In FIG. 7, the carrier comparison unit 38 is input with the advance angle controlled advance phase θ _v and the reference phase θ _e used when generating the voltage command V _m described later. The reference phase θ _e is a phase obtained by converting the rotor mechanical angle θ _m , which is the angle of the rotor 12a from the reference position, into an electric angle. As described above, the motor drive device 2 according to the first embodiment has a so-called position sensorless control configuration that does not use the position sensor signal from the position sensor. Therefore, the rotor mechanical angle θ _m and the reference phase θ _e are estimated by calculation. Further, the "advance angle phase" referred to here is a phase representing the "advance angle" which is the "advance angle" of the voltage command _Vm . Further, the "advance angle" referred to here is a phase difference between the motor applied voltage applied to the winding 12f of the stator 12b and the motor induced voltage induced in the winding 12f of the stator 12b. The "advance angle" takes a positive value when the voltage applied to the motor is ahead of the voltage induced by the motor.

Further, in the carrier comparison unit 38, in addition to the advance phase θ _v and the reference phase θ _e , the carrier generated by the carrier generation unit 33, the DC voltage V _dc , and the voltage which is the amplitude value of the voltage command V _m . Amplitude command V * is input. The carrier comparison unit 38 generates PWM signals Q1 to Q4 based on the carrier, the advance phase θ _v , the reference phase θ _e , the DC voltage V _dc , and the voltage amplitude command V *.

FIG. 8 is a block diagram showing an example of the carrier comparison unit 38 shown in FIG. 7. FIG. 8 shows the detailed configuration of the carrier comparison unit 38A and the carrier generation unit 33.

In FIG. 8, the carrier frequency f _C [Hz], which is the frequency of the carrier, is set in the carrier generation unit 33. At the tip of the arrow of the carrier frequency f _C , as an example of the carrier waveform, a triangular wave carrier moving up and down between “0” and “1” is shown. The PWM control of the inverter 11 includes synchronous PWM control and asynchronous PWM control. In the case of synchronous PWM control, it is necessary to synchronize the carrier with the advance phase θ _v . On the other hand, in the case of asynchronous PWM control, it is not necessary to synchronize the carrier with the advance phase θ _v .

As shown in FIG. 8, the carrier comparison unit 38A has an absolute value calculation unit 38a, a division unit 38b, a multiplication unit 38c, a multiplication unit 38d, a multiplication unit 38f, an addition unit 38e, a comparison unit 38g, a comparison unit 38h, and an output inversion unit. It has 38i and an output inversion unit 38j.

The absolute value calculation unit 38a calculates the absolute value | V * | of the voltage amplitude command V *. In the division unit 38b, the absolute value | V * | is divided by the DC voltage V _dc detected by the voltage detector 20. In the configuration of FIG. 8, the output of the division unit 38b is the modulation factor. The battery voltage, which is the output voltage of the battery 10, fluctuates as the current continues to flow. On the other hand, by dividing the absolute value | V * | by the DC voltage _Vdc , the value of the modulation factor can be adjusted so that the motor applied voltage does not decrease due to the decrease in the battery voltage.

The multiplication unit 38c calculates a sine value of “θ _e + θ _v ”, which is the reference phase θ _e plus the advance phase θ _v . The calculated sine value of "θ _e + θ _v " is multiplied by the modulation factor which is the output of the division unit 38b. In the multiplication unit 38d, "1/2" is multiplied by the voltage command _Vm , which is the output of the multiplication unit 38c. In the addition unit 38e, "1/2" is added to the output of the multiplication unit 38d. In the multiplication unit 38f, "-1" is multiplied by the output of the addition unit 38e. The output of the addition unit 38e is input to the comparison unit _38g as a positive voltage command Vm1 for driving the two switching

elements

51 and 53 of the upper arm among the plurality of switching elements 51 to 54, and is input to the comparison unit 38g of the multiplication unit 38f. The output is input to the comparison unit _38h as a negative voltage command Vm2 for driving the two switching

elements

52 and 54 of the lower arm.

In the comparison unit 38 g, the positive voltage command V _m1 and the amplitude of the carrier are compared. The output of the output inversion unit 38i in which the output of the comparison unit 38g is inverted becomes the PWM signal Q1 to the switching element 51, and the output of the comparison unit 38g becomes the PWM signal Q2 to the switching element 52. Similarly, in the comparison unit 38h, the negative voltage command _Vm2 and the amplitude of the carrier are compared. The output of the output inversion unit 38j, which is the inverted output of the comparison unit 38h, is the PWM signal Q3 to the switching element 53, and the output of the comparison unit 38h is the PWM signal Q4 to the switching element 54. The output inverting unit 38i does not turn on the switching element 51 and the switching element 52 at the same time, and the output inverting unit 38j does not turn on the switching element 53 and the switching element 54 at the same time.

FIG. 9 is a diagram showing an example of waveforms of a main part when operated using the carrier comparison unit 38A shown in FIG. In FIG. 9, the waveform of the positive voltage command V _m1 output from the addition unit 38e, the waveform of the negative voltage command V _m2 output from the multiplication unit 38f, the waveforms of the PWM signals Q1 to Q4, and the inverter output are shown. The voltage waveform is shown.

The PWM signal Q1 becomes “Low” when the positive voltage command V _m1 is larger than the carrier, and becomes “High” when the positive voltage command V _m1 is smaller than the carrier. The PWM signal Q2 is an inverted signal of the PWM signal Q1. The PWM signal Q3 becomes “Low” when the negative voltage command V _m2 is larger than the carrier, and becomes “High” when the negative voltage command V _m2 is smaller than the carrier. The PWM signal Q4 is an inverted signal of the PWM signal Q3. As described above, the circuit shown in FIG. 8 is configured with "Low Active", but even if each signal is configured with "High Active" having opposite values. good.

As shown in FIG. 9, the waveform of the inverter output voltage shows a voltage pulse due to the difference voltage between the PWM signal Q1 and the PWM signal Q4 and a voltage pulse due to the difference voltage between the PWM signal Q3 and the PWM signal Q2. These voltage pulses are applied to the single-phase motor 12 as the motor applied voltage.

Bipolar modulation and unipolar modulation are known as modulation methods used when generating PWM signals Q1 to Q4. Bipolar modulation is a modulation method that outputs a voltage pulse that changes with a positive or negative potential every cycle of the voltage command _Vm . Unipolar modulation is a modulation method that outputs a voltage pulse that changes at three potentials in each cycle of the voltage command _Vm , that is, a voltage pulse that changes between a positive potential, a negative potential, and a zero potential. The waveform shown in FIG. 9 is due to unipolar modulation. In the motor drive device 2 according to the first embodiment, any modulation method may be used. In applications where it is necessary to control the motor current waveform to a more sinusoidal wave, it is preferable to adopt unipolar modulation having a lower harmonic content than bipolar modulation.

Further, the waveform shown in FIG. 9 shows four switchings of the switching

elements

51 and 52 constituting the leg 5A and the

switching elements

53 and 54 constituting the leg 5B during the half-cycle T / 2 period of the voltage command V _m . It is obtained by a method of switching the element. This method is called "both-sided PWM" because the switching operation is performed by both the positive side voltage command V _m1 and the negative side voltage command V _m2 . On the other hand, in one half cycle of one cycle T of the voltage command V _m , the switching operation of the switching

elements

51 and 52 is suspended, and in the other half cycle of the one cycle T of the voltage command V _m , the switching operation is suspended. There is also a method of suspending the switching operation of the switching

elements

53 and 54. This method is called "one-sided PWM". Hereinafter, "one-sided PWM" will be described. In the following description, the operation mode operated by double-sided PWM is referred to as "double-sided PWM mode", and the operation mode operated by one-sided PWM is referred to as "one-sided PWM mode". Further, the PWM signal by "two-sided PWM" may be called "two-sided PWM signal", and the PWM signal by "one-sided PWM" may be called "one-sided PWM signal".

FIG. 10 is a block diagram showing another example of the carrier comparison unit 38 shown in FIG. 7. FIG. 10 shows an example of a one-sided PWM signal generation circuit, and specifically, a detailed configuration of a carrier comparison unit 38B and a carrier generation unit 33 is shown. The configuration of the carrier generation unit 33 shown in FIG. 10 is the same as or equivalent to that shown in FIG. Further, in the configuration of the carrier comparison unit 38B shown in FIG. 10, the same or equivalent components as the carrier comparison unit 38A shown in FIG. 8 are designated by the same reference numerals.

As shown in FIG. 10, the carrier comparison unit 38B has an absolute value calculation unit 38a, a division unit 38b, a multiplication unit 38c, a multiplication unit 38k, an addition unit 38m, an addition unit 38n, a comparison unit 38g, a comparison unit 38h, and an output inversion. It has a unit 38i and an output inversion unit 38j.

The absolute value calculation unit 38a calculates the absolute value | V * | of the voltage amplitude command V *. In the division unit 38b, the absolute value | V * | is divided by the DC voltage V _dc detected by the voltage detector 20. Even in the configuration of FIG. 10, the output of the division unit 38b is the modulation factor.

The multiplication unit 38c calculates a sine value of “θ _e + θ _v ”, which is the reference phase θ _e plus the advance phase θ _v . The calculated sine value of "θ _e + θ _v " is multiplied by the modulation factor which is the output of the division unit 38b. In the multiplication unit 38k, "-1" is multiplied by the voltage command _Vm , which is the output of the multiplication unit 38c. In the addition unit 38m, “1” is added to the voltage command _Vm which is the output of the multiplication unit 38c. In the addition unit 38n, "1" is added to the output of the multiplication unit 38k, that is, the inverted output of the voltage command _Vm . The output of the addition unit 38m is input to the comparison unit 38g as a first voltage command _Vm3 for driving the two switching

elements

51 and 53 of the upper arm among the plurality of switching elements 51 to 54. The output of the addition unit 38n is input to the comparison unit 38h as a second voltage command _Vm4 for driving the two switching

elements

52 and 54 of the lower arm.

In the comparison unit 38 g, the first voltage command V _m3 and the amplitude of the carrier are compared. The output of the output inversion unit 38i in which the output of the comparison unit 38g is inverted becomes the PWM signal Q1 to the switching element 51, and the output of the comparison unit 38g becomes the PWM signal Q2 to the switching element 52. Similarly, in the comparison unit 38h, the second voltage command _Vm4 and the amplitude of the carrier are compared. The output of the output inversion unit 38j, which is the inverted output of the comparison unit 38h, is the PWM signal Q3 to the switching element 53, and the output of the comparison unit 38h is the PWM signal Q4 to the switching element 54. The output inverting unit 38i does not turn on the switching element 51 and the switching element 52 at the same time, and the output inverting unit 38j does not turn on the switching element 53 and the switching element 54 at the same time.

FIG. 11 is a diagram showing an example of waveforms of a main part when operated using the carrier comparison unit 38B shown in FIG. In FIG. 11, the waveform of the first voltage command V _m3 output from the adder 38 m, the waveform of the second voltage command V _m4 output from the adder 38n, the waveforms of the PWM signals Q1 to Q4, and the inverter output are shown. The voltage waveform is shown. In FIG. 11, for convenience, the waveform portion of the first voltage command V _m3 whose amplitude value is larger than the peak value of the carrier and the second voltage command V _m4 whose amplitude value is larger than the peak value of the carrier. The corrugated portion is represented by a flat straight line.

The PWM signal Q1 becomes “Low” when the first voltage command V _m3 is larger than the carrier, and becomes “High” when the first voltage command V _m3 is smaller than the carrier. The PWM signal Q2 is an inverted signal of the PWM signal Q1. The PWM signal Q3 becomes “Low” when the second voltage command V _m4 is larger than the carrier, and becomes “High” when the second voltage command V _m4 is smaller than the carrier. The PWM signal Q4 is an inverted signal of the PWM signal Q3. As described above, the circuit shown in FIG. 10 is configured with "Low Active", but even if each signal is configured with "High Active" having opposite values. good.

As shown in FIG. 11, the waveform of the inverter output voltage shows a voltage pulse due to the difference voltage between the PWM signal Q1 and the PWM signal Q4 and a voltage pulse due to the difference voltage between the PWM signal Q3 and the PWM signal Q2. These voltage pulses are applied to the single-phase motor 12 as the motor applied voltage.

In the waveform shown in FIG. 11, in one half cycle of one cycle T of the voltage command V _m , the switching operation of the switching

elements

51 and 52 is suspended, and the other of the one cycle T of the voltage command V _m is stopped. In the half cycle, the switching operation of the switching

elements

53 and 54 is suspended.

Further, in the waveform shown in FIG. 11, the switching element 52 is controlled to be always on in one half cycle of one cycle T of the voltage command V _m , and the one cycle T of the voltage command V _m is controlled. In the other half cycle, the switching element 54 is controlled to be always on. Note that FIG. 11 is an example, in which the switching element 51 is controlled to be always on in one half cycle, and the switching element 53 is controlled to be always on in the other half cycle. Is also possible. That is, the waveform shown in FIG. 11 is characterized in that at least one of the switching elements 51 to 54 is controlled to be in the ON state in the half cycle of the voltage command _Vm .

Further, in FIG. 11, the waveform of the inverter output voltage is unipolar modulation that changes at three potentials in each cycle of the voltage command _Vm . As described above, bipolar modulation may be used instead of unipolar modulation, but in applications where it is necessary to control the motor current waveform to a more sinusoidal wave, it is preferable to adopt unipolar modulation.

Next, the advance angle control in the first embodiment will be described with reference to the drawings of FIGS. 12 to 14. FIG. 12 is a block diagram showing a functional configuration for calculating the advance angle phase θ _v input to the carrier comparison unit 38 shown in FIG. 7. FIG. 13 is a diagram showing an example of a method for calculating the advance angle phase θ _v in the first embodiment. FIG. 14 is a diagram used to explain the relationship between the voltage command V _m shown in FIG. 7 and the advance phase θ _v .

As shown in FIG. 12, the function of calculating the advance angle phase θ _v can be realized by the rotation speed calculation unit 42 and the advance angle phase calculation unit 44. The rotation speed calculation unit 42 calculates the rotation speed ω of the single-phase _motor 12 based on the detection value of the motor current Im detected by the current detector 22. Further, the rotation speed calculation unit 42 calculates the rotor mechanical angle θ _m based on the detected value of the motor current _Im , and also calculates the reference phase θ _e obtained by converting the rotor mechanical angle θ _m into an electric angle.

Here, in the uppermost part of FIG. 14, the position of the rotor 12a is represented by a signal level. In the waveform of the uppermost stage, the edge portion where the signal falls from “H” to “L” is set as the reference position of the rotor 12a, and this reference position is set to “0 °” of the rotor mechanical angle θ _m . ing. Further, a reference phase θ _e , which is a phase obtained by converting the rotor mechanical angle θ _m into an electric angle, is shown at the lower part of the numerical sequence representing the rotor mechanical angle θ _m . The advance phase phase calculation unit 44 calculates the advance angle phase θ _v based on the rotation speed ω and the reference phase θ _e calculated by the rotation speed calculation unit 42.

The horizontal axis of FIG. 13 shows the rotation speed N, and the vertical axis of FIG. 13 shows the advance phase θ _v . As shown in FIG. 13, the advance phase θ _v can be determined by using a function in which the advance phase θ _v increases with respect to the increase in the rotation speed N. In the example of FIG. 13, the advance phase θ _v is determined by a linear function of the first order, but the phase is not limited to the linear function of the first order. A function other than the linear linear function of the first order may be used as long as the advance phase θ _v is the same or becomes larger according to the increase of the rotation speed N.

In the middle part of FIG. 14, two voltage command _Vm waveform examples are shown as “Example 1” and “Example 2”. Further, the lowermost portion of FIG. 14 shows a state in which the rotor mechanical angles θ _m when the rotor 12a is rotated clockwise are 0 °, 45 °, 90 °, 135 °, and 180 °. Four magnets are shown on the rotor 12a of the single-phase motor 12, and four teeth 12e are shown on the outer circumference of the rotor 12a. When the rotor 12a rotates clockwise, the rotor mechanical angle θ _m is estimated based on the detected value of the motor current _Im , and the reference phase θ _e converted into an electric angle based on the estimated rotor mechanical angle θ _m . Is calculated.

In the middle part of FIG. 14, the voltage command V _m shown as “Example 1” is a voltage command when the advance phase θ _v = 0. When the advance phase θ _v = 0, the voltage command V _m having the same phase as the reference phase θ _e is output. The amplitude of the voltage command V _m at this time is determined based on the voltage amplitude command V * described above.

Further, in the middle part of FIG. 14, the voltage command _Vm shown as “Example 2” is a voltage command when the advance phase θ _v = π / 4. When the advance phase θ _v = π / 4, the voltage command V _m advanced by π / 4, which is a component of the advance phase θ _v , is output from the reference phase θ _e .

Next, the start control of the single-phase motor 12, which is the main point of the drive control of the motor drive device 2 according to the first embodiment, will be described with reference to FIGS. 15 and 16. FIG. 15 is a diagram used for explaining the operation of the main part in the first embodiment. FIG. 16 is a diagram used to explain the relationship between the rotor position when the single-phase motor shown in FIG. 2 is stopped and the start control in the first embodiment. The above-mentioned FIGS. 2, 3 and 16 below are examples in which the single-phase motor 12 having the asymmetrically shaped teeth 12e is the drive target, but the single-phase motor to be driven is FIG. 2, FIG. It is not limited to the structure of 3 and FIG. That is, the method of the first embodiment is not limited to the case where the teeth 12e has an asymmetrical shape, and can be applied even when the teeth 12e has a symmetrical shape.

In FIG. 15, the waveform of the motor applied voltage or the motor induced voltage is shown in the upper part, and the waveform of the motor current is shown in the middle part. Further, the waveform of the shunt voltage is shown in the lower part.

In the first period, as shown in the upper part of FIG. 15, a voltage having a negative polarity is applied. In this paper, the voltage of this polarity is called "first voltage". The first period starts immediately after startup. Note that FIG. 15 illustrates a case where the first voltage is a voltage of one pulse, but the present invention is not limited to this. The first voltage may be the voltage of a plurality of PWM-controlled pulse trains.

In the case of a single-phase motor driven without a position sensor, the stop position of the rotor 12a is not uniquely determined regardless of whether the teeth have an asymmetrical shape. Therefore, when a first voltage of a certain polarity is applied, there are two cases: when the polarity is positive energization (pattern 1 in FIG. 16) and when the polarity is reverse energization (patterns 2 and 3 in FIG. 16). be. Positive energization means that the polarity of the first voltage rotates the rotor 12a in the intended rotation direction, and reverse energization means that the polarity of the first voltage rotates the rotor 12a in an unintended rotation direction. It means that there is. The example of FIG. 15 shows the case of pattern 2.

First, when the stop position of the rotor 12a is pattern 1, the polarity of the first voltage is the polarity of the intended positive energization. Therefore, the rate of increase of the motor current and the shunt voltage is smaller than that shown in FIG. Further, in this case, the polarity of the motor applied voltage is reversed before the shunt voltage reaches the first threshold value. Therefore, the single-phase motor 12 rotates without any problem.

The timing for switching the polarity of the motor applied voltage can be determined based on the command value of the rotation speed given to the single-phase motor 12. Further, during the rotation of the single-phase motor 12, the rotation speed can be calculated based on the motor-induced voltage detected by the voltage detector 21, and the switching timing can be determined based on the calculated rotation speed.

On the other hand, when the stop position of the rotor 12a is pattern 2, the polarity of the first voltage is the polarity of unintended reverse energization. Therefore, the waveforms of the motor current and the shunt voltage linearly decrease or increase as shown in FIG. Therefore, when the shunt voltage reaches the first threshold value, the inverter 11 gates off. As a result, the application of the first voltage to the single-phase motor 12 is stopped.

Immediately after the inverter 11 gates off, the current flowing through the single-phase motor 12 tends to continue flowing due to the influence of the motor inductance component. At this time, a regenerative current flows from the single-phase motor 12 toward the battery 10. In this paper, the period during which this regenerative current flows is referred to as the "regenerative period". In FIG. 15, each waveform during the regeneration period is surrounded by a vertically long ellipse.

As shown in FIG. 15, during the regeneration period, a voltage having a polarity opposite to the voltage applied to the motor is generated by the regeneration current, so that the motor-induced voltage induced in the single-phase motor 12 cannot be detected. Therefore, an "inverter output stop period" longer than the regeneration period is provided. The inverter output stop period is a period during which the application of the first voltage is stopped, and is started immediately after the shunt voltage reaches the first threshold value. The motor induced voltage can be detected by the voltage detector 21.

Instead of the method of directly detecting the motor-induced voltage by the voltage detector 21, the motor-induced voltage may be calculated based on the detection value of the voltage detector 20 or the detection value of the current detector 24. When the detection value of the voltage detector 20 is used, a control means for reducing the output voltage of the battery 10 to zero or a mechanism for disconnecting the electrical connection between the battery 10 and the inverter 11 is required.

When the inverter output stop period ends, the process shifts to the second period shown in FIG. In the second period, the inverter 11 applies a voltage having a polarity opposite to the first voltage applied in the first period to the single-phase motor 12. In this paper, this voltage of opposite polarity is called "second voltage". Note that FIG. 15 illustrates a case where the second voltage is a voltage of one pulse, but the present invention is not limited to this. The second voltage may be the voltage of a plurality of PWM-controlled pulse trains.

The polarity of the second voltage is the voltage of the polarity that rotates the single-phase motor 12 in the first direction, which is the intended rotation direction. The second voltage gives the single-phase motor 12 a rotational torque that rotates in the first direction. After that, the inverter 11 inverts the polarity of the voltage applied to the single-phase motor 12 each time the shunt voltage reaches the first threshold value.

Note that, in FIG. 15, in the second period, a waveform in which the value of the shunt voltage changes from an increase to a decrease and has a first peak smaller than the first threshold value is observed. This feature is a feature that appears in the single-phase motor 12 having the asymmetrically shaped teeth 12e as shown in FIG. In the first embodiment, the activation control using this feature is not particularly performed. The activation control using this feature will be described in the second embodiment.

Next, in the start control in the first embodiment, the significance of providing the inverter output stop period as shown in FIG. 15 will be described.

As described above, in the start control of the first embodiment, the first voltage is applied to the single-phase motor 12 during the first period at the time of starting the single-phase motor 12. Then, when the shunt voltage reaches the first threshold value, the application of the first voltage is stopped. By stopping the application of the first voltage, the stop position of the rotor magnetic pole can be grasped. By providing the inverter stop period in this way, it is possible to uniquely determine the stop position of the rotor magnetic pole.

In particular, when the single-phase motor 12 has a structure having an asymmetrically shaped teeth 12e, the rotor 12a is stably stopped at a position shifted in a specific direction due to the cogging torque. Therefore, the control method of the first embodiment can be suitably used for this type of motor.

If the first threshold value is too large, as shown in pattern 3 of FIG. 16, there is a possibility that the rotor position after energization will stop in a state of being shifted in the direction of reverse energization from pattern 2. If a voltage whose polarity is reversed is applied in this state, the single-phase motor 12 may rotate in the direction of reverse energization, that is, in the second direction opposite to the intended direction. Therefore, the first threshold value needs to be appropriately set so that the single-phase motor 12 does not rotate in an unintended second direction.

Further, when the first threshold value is too small, the motor applied voltage is switched in a state where the rotational torque generated by the winding 12f of the stator 12b is small. In this case, if the load torque is large, it may not be possible to give a sufficient rotational torque to the load. Therefore, the first threshold value needs to be appropriately set so that the single-phase motor 12 can give a sufficient rotational torque to the load.

As described above, according to the motor drive device according to the first embodiment, the inverter applies the first voltage to the single-phase motor during the first period at the time of starting the single-phase motor, and the first voltage is increased. In the second period after application, a second voltage in which the polarity of the first voltage is reversed is applied. Between the first period and the second period, a stop period for stopping the application of the first voltage is provided. With such control, when the single-phase motor is started without a position sensor, the single-phase motor can be started safely and surely. Further, by providing a stop section, it is possible to control the rotor magnetic pole to be located near the stable stop point. This enables stable driving even in a single-phase motor having a large cogging torque.

Further, according to the motor drive device according to the first embodiment, the stop period is set so as to start after the physical quantity correlated with the motor current reaches the first threshold value. The physical quantity that correlates with the motor current can be detected by the first current detector. When this control is realized by hardware, it can be implemented on a small circuit scale. This makes it possible to reduce the size and weight of the control board. Moreover, when this control is realized by software, it can be implemented even with an inexpensive microcomputer because a small amount of source code needs to be added. Therefore, the effect of suppressing the cost increase can be obtained.

Embodiment 2.
Next, start control in the motor drive device according to the second embodiment will be described. The configuration of the motor drive device according to the second embodiment is the same as the configuration of the motor drive device 2 according to the first embodiment. In the second embodiment, the parts different from the first embodiment will be mainly described, and the description of the overlapping contents will be omitted as appropriate.

FIG. 17 is a diagram used for explaining the operation of a main part in the motor drive device 2 according to the second embodiment. In FIG. 17, the waveform of the motor applied voltage or the motor induced voltage is shown in the upper part, and the waveform of the motor current is shown in the middle part. Further, the waveform of the shunt voltage is shown in the lower part.

In FIG. 17, the definitions of the first period, the second period, and the inverter output stop section are the same as those in the first embodiment. Further, the definitions of the first voltage, the second voltage, and the first threshold value are the same as those in the first embodiment.

When the stop position of the rotor 12a is pattern 1, the single-phase motor 12 can be smoothly driven by applying the first voltage, which is the polarity of positive energization, as in the first embodiment.

Further, when the stop position of the rotor 12a is the pattern 2, the inverter 11 applies the first voltage to the single-phase motor 12, and stops the application of the first voltage when the shunt voltage reaches the first threshold value. Then, the stop of applying the first voltage continues during the inverter output stop period. When the inverter output stop period ends, the process shifts to the second period. When the second period begins, the inverter 11 applies a second voltage to the single-phase motor 12. The operation up to this point is the same as that of the first embodiment.

In the second embodiment, the level of the first top mentioned above, that is, the level of the first top where the value of the shunt voltage changes from increasing to decreasing is detected, and the detection level is set as the second threshold value. The position and level at which the first top is generated are determined by the rotational torque generated in the rotor 12a and the motor-induced voltage generated by the rotation of the rotor 12a.

The inverter 11 reverses the polarity of the second voltage when the shunt voltage reaches the second threshold value after the setting of the second threshold value. After that, the inverter 11 inverts the polarity of the voltage applied to the single-phase motor 12 each time the shunt voltage reaches the second threshold value. As a result, the rotary torque can be continuously applied to the rotor 12a, so that the single-phase motor 12 can be smoothly driven.

At the first top, the value of di / dt, which is the rate of change in current, is small. Therefore, there is an advantage that the sampling interval for current detection is not reduced and the decrease in the threshold setting accuracy can be suppressed even with a small number of samplings. Since the appearance of the first top depends on the characteristics of the single-phase motor 12, the characteristics of the load connected to the single-phase motor 12, and the like, the second threshold value is set every second period, that is, when the motor is applied. It is preferable that the voltage is reset and reset each time the polarity is reversed.

As described above, according to the motor drive device according to the second embodiment, the inverter applies the first voltage to the single-phase motor during the first period at the time of starting the single-phase motor, and the first voltage is increased. In the second period after application, a second voltage in which the polarity of the first voltage is reversed is applied. Between the first period and the second period, a stop period for stopping the application of the first voltage is provided. Then, after the application of the second voltage, a second threshold value smaller than the first threshold value is set. The inverter reverses the polarity of the second voltage when the shunt voltage reaches the second threshold after the setting of the second threshold. After that, each time the shunt voltage reaches the second threshold value, the polarity of the voltage applied to the single-phase motor is reversed. By such control, in addition to the effect of the first embodiment, the effect that stable start can be realized even when the fluctuation of the power supply voltage is large or the motor constant of the single-phase motor is varied can be obtained. ..

Next, as a matter common to the first and second embodiments, the effect of driving the single-phase motor 12 without a position sensor will be described.

First, when an application example is a vacuum cleaner and a magnetic pole position sensor is used as a position sensor, the distance between the permanent magnet provided in the rotor and the substrate provided with the magnetic pole position sensor becomes close. In this case, the substrate is arranged at a position that obstructs the flow of the wind generated by the blades, which increases the pressure loss of the air passage. The increase in pressure loss becomes a factor that deteriorates the suction power of the vacuum cleaner and lowers the suction power.

On the other hand, in the case of no position sensor, since the position sensor is not provided, the degree of freedom in board placement is increased, so that the board can be placed parallel to the air passage. As a result, since the substrate does not block the air passage, the pressure loss of the air passage can be suppressed and the suction force can be improved. As a result, it becomes possible to improve the suction work rate of the vacuum cleaner.

Further, in the case where the application example is an electric blower, if the gas sucked by the electric blower contains a large amount of water, the amount of water that directly collides with the substrate increases. In this case, when a voltage is applied to the substrate, ionized metal moves between the electrodes to cause a short circuit, which may cause ion migration. Further, there is a concern about a short circuit caused by dust or dust accumulating on the substrate. As a countermeasure, a method of applying a moisture-proof agent to the substrate or a method of isolating the substrate from the air passage is adopted, but both of them lead to an increase in manufacturing cost.

On the other hand, in the case of no position sensor, since the position sensor is not provided, the degree of freedom in board placement is increased, so that the board can be placed while avoiding the air passage. As a result, the amount of water that directly collides with the substrate is reduced, so that the occurrence of ion migration can be suppressed and the amount of the moisture-proofing agent can be reduced. Further, since the degree of freedom in arranging the substrate is increased, the quality of the substrate can be improved by arranging the substrate outside the housing.

Further, when the position sensor is a magnetic pole position sensor, the accuracy of the mounting work for correctly detecting the magnetic pole position is required, and it is necessary to carry out the position adjusting work according to the mounting position. For this reason, it becomes difficult to control the manufacturing, and the manufacturing cost including the installation work increases.

On the other hand, in the case of no position sensor, the installation process for installing the position sensor and the adjustment process after installation are not required, so that the manufacturing cost can be significantly reduced. Moreover, since the influence of the secular variation of the position sensor does not occur, the quality of the product can be improved.

Furthermore, since the position sensorless type does not require a position sensor, the inverter and the single-phase motor can be configured separately. This makes it possible to reduce the restrictions when applying the product. For example, when the application example is a product used in a water place or the like, the inverter can be isolated from the position of the water place or the like and arranged.

Also, in the case of no position sensor, the configuration is equipped with a current detector. The current detector can detect a motor abnormality such as a shaft lock or a phase loss by detecting the motor current. This makes it possible to safely stop without a position sensor.

In order to detect a motor abnormality, for example, a third threshold value larger than the first threshold value is set. Then, when the shunt voltage reaches the third threshold value, it is determined that the motor is abnormal. Further, when it is determined that the motor is abnormal, the output of the inverter is cut off. By doing so, it is possible to detect a motor abnormality and safely stop the operation of the product.

Embodiment 3.
In the third embodiment, an application example of the motor drive device 2 described in the first and second embodiments will be described. The motor drive device 2 described above can be used, for example, in a vacuum cleaner. In the case of a product such as an electric vacuum cleaner that is used immediately after the power is turned on, the effect of shortening the start-up time of the motor drive device 2 according to the first and second embodiments is increased.

FIG. 18 is a configuration diagram of the vacuum cleaner 61 according to the third embodiment. The vacuum cleaner 61 shown in FIG. 18 is a so-called stick-type vacuum cleaner. In FIG. 18, the vacuum cleaner 61 includes a battery 10 shown in FIG. 1, a motor driving device 2 shown in FIG. 1, an electric blower 64 driven by a single-phase motor 12 shown in FIG. 1, and dust collector. A chamber 65, a sensor 68, a suction port 63, an extension pipe 62, and an operation unit 66 are provided.

The user who uses the vacuum cleaner 61 has an operation unit 66 and operates the vacuum cleaner 61. The motor drive device 2 of the vacuum cleaner 61 drives the electric blower 64 using the battery 10 as a power source. By driving the electric blower 64, dust is sucked from the suction port 63. The sucked dust is collected in the dust collecting chamber 65 via the extension pipe 62.

Although the stick-type vacuum cleaner is illustrated in FIG. 18, it is not limited to the stick-type vacuum cleaner. The technique of the present disclosure can be applied to any product as long as it is an electric device equipped with an electric blower.

Further, FIG. 18 shows a configuration in which the battery 10 is used as a power source, but the present invention is not limited to this. Instead of the battery 10, an AC power supply supplied from an outlet may be used.

Further, the above-mentioned motor drive device can be used for, for example, a hand dryer. In the case of a hand dryer, the shorter the time from inserting the hand to driving the electric blower, the better the user's usability. Therefore, the effect of shortening the start-up time of the motor drive device 2 according to the first and second embodiments is greatly exhibited.

FIG. 19 is a block diagram of the hand dryer 90 according to the third embodiment. In FIG. 19, the hand dryer 90 includes a motor drive device 2 shown in FIG. 1, a casing 91, a hand detection sensor 92, a water receiving unit 93, a drain container 94, a cover 96, a sensor 97, and an intake air. It includes a port 98 and an electric blower 95 driven by the single-phase motor 12 shown in FIG. Here, the sensor 97 is either a gyro sensor or a motion sensor. In the hand dryer 90, when a hand is inserted into the hand insertion portion 99 at the upper part of the water receiving portion 93, water is blown off by the blown air by the electric blower 95, and the blown water is collected by the water receiving portion 93. After that, it is stored in the drain container 94.

Since the above-mentioned electric vacuum cleaner 61 and the hand dryer 90 are both position sensorless products equipped with the motor drive device 2 according to the first and second embodiments, the following effects can be obtained.

First, in the case of a position sensorless configuration, since it can be started without a position sensor, it is possible to reduce costs such as material cost and processing cost of the position sensor. Further, since there is no position sensor, it is possible to eliminate the performance effect due to the misalignment of the position sensor. As a result, stable performance can be ensured.

Also, since the position sensor is a sensitive sensor, high-precision mounting accuracy is required for the installation position of the position sensor. In addition, after mounting, it is necessary to make adjustments according to the mounting position of the position sensor. On the other hand, in the case of the position sensorless configuration, the position sensor itself becomes unnecessary, and the adjustment step of the position sensor can be eliminated. As a result, the manufacturing cost can be significantly reduced. In addition, the quality of the product can be improved because the position sensor is not affected by the secular variation.

Also, in the case of a position sensorless configuration, since a position sensor is not required, the inverter and the single-phase motor can be configured separately. This makes it possible to relax restrictions on the product. For example, in the case of a product used in a water place with a large amount of water, the mounting position of the inverter in the product can be arranged at a place far from the water place. As a result, the possibility of failure of the inverter can be reduced, and the reliability of the device can be improved.

Further, in the case of the position sensorless configuration, the motor abnormality such as the shaft lock and the open phase can be detected by detecting the motor current or the inverter current by the current detector arranged instead of the position sensor. Therefore, the product can be safely stopped without the position sensor.

As described above, a configuration example in which the motor drive device 2 according to the first and second embodiments is applied to a vacuum cleaner and a hand dryer has been described, but the present invention is not limited to these examples. The motor drive device 2 can be widely applied to an electric device on which a motor is mounted. Examples of electrical equipment equipped with motors are incinerators, crushers, dryers, dust collectors, printing machines, cleaning machines, confectionery machines, tea making machines, woodworking machines, plastic extruders, cardboard machines, packaging machines, hot air generators. , OA equipment, and electric blowers. The electric blower is a blowing means for transporting an object, sucking dust, or for general blowing and exhausting.

The configuration shown in the above embodiments is an example, and can be combined with another known technique, or can be combined with each other, and deviates from the gist. It is also possible to omit or change a part of the configuration to the extent that it does not.

1 motor drive system, 2 motor drive device, 5A, 5B leg, 6A, 6B connection end, 10 battery, 11, 11A inverter, 12 single-phase motor, 12a rotor, 12b stator, 12c shaft, 12d split core, 12e teeth, 12e1 1st tip, 12e2 2nd tip, 12f winding, 16a, 16b DC bus, 18a, 18b connection line, 20,21 voltage detector, 22,24 current detector, 25 control unit, 30 analog digital conversion Instrument, 30a digital output value, 31 processor, 32 drive signal generation unit, 33 carrier generation unit, 34 memory, 38, 38A, 38B carrier comparison unit, 38a absolute value calculation unit, 38b division unit, 38c, 38d, 38f, 38k Multiplying unit, 38e, 38m, 38n addition unit, 38g, 38h comparison unit, 38i, 38j output inversion unit, 42 rotation speed calculation unit, 44 advance position calculation unit, 51, 52, 53, 54 switching element, 51a, 52a , 53a, 54a body diode, 55a, 55b shunt resistance, 61 electric vacuum cleaner, 62 extension tube, 63 suction port, 64,95 electric blower, 65 dust collection chamber, 66 operation unit, 68,97 sensor, 90 hand dryer , 91 casing, 92 hand detection sensor, 93 water receiving part, 94 drain container, 96 cover, 98 intake port, 99 hand insertion part.

Claims

A motor drive device that drives a single-phase motor without a position sensor.
Equipped with an inverter placed between the DC power supply and the single-phase motor,
When the single-phase motor is started, the inverter applies a first voltage to the single-phase motor in the first period immediately after the start, and in the second period after the application of the first voltage, the first voltage is applied. Apply a second voltage with the polarity of the voltage reversed,
A motor drive device in which a stop period for stopping the application of the first voltage exists between the first period and the second period.
A first detector for detecting a physical quantity correlated with the current flowing through the single-phase motor is provided.
The motor drive device according to claim 1, wherein the stop period is started after the physical quantity reaches the first threshold value.
The motor drive device according to claim 2, wherein the inverter applies the second voltage to the single-phase motor after the end of the stop period.
After applying the second voltage,
The motor drive device according to claim 3, wherein the inverter reverses the polarity of the voltage applied to the single-phase motor each time the physical quantity reaches the first threshold value.
After the application of the second voltage, a second threshold value smaller than the first threshold value is set, and the second threshold value is set.
The motor drive device according to claim 3, wherein the inverter reverses the polarity of the second voltage when the physical quantity reaches the second threshold value after the setting of the second threshold value.
The motor drive device according to claim 5, wherein the second threshold value is set to a value detected at the top of a portion where the detection waveform of the first detector turns from an increase to a decrease.
Claim 5 or 6 that after the application of the second voltage, the polarity of the voltage applied to the single-phase motor is reversed each time the physical quantity reaches the second threshold value after the setting of the second threshold value. The motor drive device according to.
The motor drive device according to claim 7, wherein the second threshold value is reset and reset each time the polarity of the voltage applied to the single-phase motor is inverted.
The motor drive device according to any one of claims 2 to 8, wherein the inverter stops operating when the physical quantity reaches a third threshold value larger than the first threshold value.
The inverter has a plurality of switching elements that are bridge-connected and has a plurality of switching elements.
The motor drive device according to any one of claims 1 to 9, wherein at least one of the plurality of switching elements is formed of a wide bandgap semiconductor.
The motor drive device according to claim 10, wherein the wide bandgap semiconductor is silicon carbide, gallium nitride, gallium oxide, or diamond.
An electric blower provided with the motor drive device according to any one of claims 1 to 11.
A vacuum cleaner equipped with the electric blower according to claim 12.
A hand dryer equipped with the electric blower according to claim 12.