JPH11330935A - Driving circuit for semiconductor device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To obtain the driving circuit which can obtain desired switching characteristics, for example, make constant the variation rate of the current of a main electrode or the voltage between main electrodes in the switching operation of the semiconductor device. SOLUTION: A switching signal generating means 4 generates an ON/OFF signal and a current variation rate setting means 19 commands a variation rate of the collector current Ic of an IGBT element 1. A 1st function generating means 20 generates and commands the reverse function of a transfer function from a current supplied to a gate 1a to switching characteristics of the IGBT element 1 set by the current variation rate setting means 19 to a gate current generating means 18.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an IGBT (Insulated Gate) which is a switching voltage drive type semiconductor device.
Bipolar Transistor (insulated gate transistor) element and MOSFET (Metal Oxicide Silicon Feild Effe)
The present invention relates to a gate drive circuit of a semiconductor device such as a ct transistor (field effect transistor) device.

[0002]

2. Description of the Related Art FIG. 28 is a circuit diagram showing a conventional driving circuit for a semiconductor device described in Japanese Patent Application Laid-Open No. 9-65644. In FIG. IGBT element, which is a voltage-driven self-extinguishing semiconductor element for controlling the conduction state of the IGBT element, 1a is a gate which is a first control electrode, 1b is a collector which is a first main electrode, 1c is a second control electrode , An emitter which is also an electrode also serving as a second main electrode, 2 is a DC voltage source for turning on which supplies a voltage for turning on the IGBT element 1, and 3 is an IGB
An off DC voltage source for supplying a voltage for turning off the T element 1, a switching signal generating means 4 for generating a signal for turning on / off the IGBT element 1, and a switching DC generating means 5 based on an output of the switching signal generating means 4. The gate 1a of the IGBT element 1 is switched by switching between the voltage source 2 and the off DC voltage source 3.
And 6 is a gate resistance switching means for switching the gate resistance. The gate resistance switching means 6 includes a resistor 9, a resistor 10, and a switching means 11. Reference numeral 7 denotes current detection means for detecting a current flowing through the main electrode of the IGBT element 1, 8 denotes control means for controlling the gate resistance switching means 6 based on the output of the current detection means 7, 12 denotes a DC voltage source, 13 denotes a load, 14
Is a freewheeling diode, and 15 is a wiring inductance.

First, the basic operation of the conventional example will be described with reference to FIGS. In FIG. 28, when the switching signal generating means 4 outputs an ON signal, the ON / OFF switching means 5 is switched to the DC voltage source 2 for ON, and the gate resistance switching means 6 is provided between the gate 1a and the emitter 1c of the IGBT element 1. The ON voltage is applied, and the IGBT element 1
Turns on. When the IGBT element 1 is turned on, the voltage Vce between the collector 1b and the emitter 1c of the IGBT element 1 becomes almost zero. At this time, the voltage Vdc of the DC voltage source 12 is applied to the load 13, and a current flows through a path of the DC voltage source 12, the wiring inductance 15, the load 13, the IGBT element 1, and the DC voltage source 12.

When the switching signal generating means 4 outputs an off signal, the on / off switching means 5 is switched to the off DC voltage source 3 side, and the gate 1a- of the IGBT element 1 is turned off.
An off voltage is applied between the emitters 1c via the gate resistance switching means 6, and the IGBT element 1 is turned off. IGB
When the T element 1 is turned off, the collector 1b of the IGBT element 1
Is almost zero. Until then IGBT
The current flowing through the element 1 is diverted to the freewheeling diode 14, and the freewheeling diode 14-load 13-freewheeling diode 1
The current flows through the path 4. At this time, the freewheeling diode 14
Is substantially zero, the voltage Vdc of the DC voltage source 12 is applied to the IGBT element 1. FIG.
29 shows a waveform at the time of turn-off when the IGBT element 1 transitions from the on-state to the off-state. FIG. 29A shows a case where the resistance value of the gate resistor is small, FIG. The case of switching is shown. The turn-off operation of the IGBT element 1 starts with an increase in the collector-emitter voltage Vce. During this period, the collector current Ic is substantially constant, and a first period in which the collector current changes slowly is formed. When the collector-emitter voltage Vce increases and reaches the voltage Vdc of the DC voltage source 12, the freewheeling diode 14 is turned on, and the current flowing through the load 13 starts flowing through the freewheeling diode 14. Since the current can be regarded as constant by the action of the inductance of the load 13, the freewheeling diode 14
The collector current Ic is reduced by the current Id flowing through the collector current Id. Therefore, as shown in FIG. 29, after the collector-emitter voltage Vce reaches the voltage Vdc of the DC voltage source 12, the collector current Ic rapidly decreases, and a second period in which the change of the collector current is sharp is formed. Is done.

In the second period, the collector current Ic
Is equal to the current flowing through the wiring inductance 15,
When the collector current Ic decreases rapidly, the current flowing through the wiring inductance 15 also decreases. For this reason, the wiring inductance 15 generates a surge voltage represented by (wiring inductance × current reduction rate). Since this surge voltage has the same polarity as the voltage of the DC voltage source 12, the collector-emitter voltage Vce has a waveform in which the surge voltage generated by the wiring inductance 15 on the DC voltage source 12 is superimposed. When the collector current Ic reaches zero, all the current of the load 13 flows to the freewheeling diode 14, and the turn-off operation is completed.

Next, the operation of the control means 8 will be described. At the time of the turn-off operation in which the IGBT element 1 changes from on to off as described above, the control means 8 controls the collector current I of the IGBT element 1 detected by the current detection means 7.
A differential value of c is obtained, and it is determined whether the value is smaller than a reference value. When the differential value of the collector current is smaller than the reference value, that is, in the first period in which the change of the collector current is slow, the control means 8 short-circuits the switching means 11 and uses the resistor 9 as a gate resistor. When the differential value of the collector current is larger than the reference value, that is, in the second period in which the change in the collector current is steep, the control means 8 opens the switching means 11 and uses the resistors 9 and 10 as gate resistors. . Since the resistors 9 and 10 are connected in series, the combined resistance value is larger than the resistance value of the resistor 9. By this operation, in the first period in which the change in the collector current is slow, the IGB
The T element 1 is driven, and the change in the collector current is sharp.
During the period of IGBT, the gate resistance is large and the IGBT
The element 1 is driven.

Next, the effect of switching the gate resistance in the conventional example will be described with reference to FIG. FIG.
The current Ic at the collector of the IGBT element 1 when the GBT element 1 is turned off, the collector-emitter voltage Vce of the IGBT element 1, and the switching loss P of the IGBT element 1 are illustrated. When the IGBT element 1 is driven when the switching means 11 is short-circuited, the IGBT element 1 is driven when the switching means 11 is open, and when the IGBT element 1 is driven when the switching means 11 is opened, (c). The case where the gate resistance is switched between the first period in which the change of the current Ic is slow and the second period in which the change of the current Ic is steep is shown. FIG.
According to the comparison between (a) and (b), as the gate resistance is increased, the rate of decrease of the collector current Ic at the time of turn-off becomes gentler, and the collector current generated by applying the rate of decrease of the collector current Ic to the wiring inductance is reduced. Although the surge voltage between the emitters is reduced, the rate of increase of the voltage Vce between the collector and the emitter also becomes slower at the same time. Therefore, the time integrated value of the switching loss P expressed by the multiplication of the collector current Ic and the voltage Vce between the collector and the emitter Is found to increase. As described above, the suppression of the surge voltage can be realized by simply increasing the gate resistance, but has a disadvantage that the switching loss increases.

According to the conventional semiconductor device drive circuit shown in FIG. 28, during the first period in which the change in the collector current Ic is slow, the IGBT device 1 is operated with the gate resistance shown in FIG. During the second period in which the collector current Ic is sharply changed, the IGBT element 1 is driven with the gate resistance shown in FIG. 29B being large. As a result, as shown in FIG. 29C, a voltage and a current waveform equivalent to those in FIG. 29A are obtained in the first period, and equivalent to those in FIG. 29B in the second period. Voltage and current waveforms are obtained. Therefore, the IGBT element 1 is driven at a high speed in the first period not related to the surge voltage to reduce the time integral value of the switching loss P, and the IGBT element 1 is driven in the second period related to the surge voltage. Surge voltage can be suppressed by driving at a low speed.

[0009]

The conventional driving circuit for a semiconductor device is constructed as described above. Although the gate resistance is switched between the first period and the second period, the first period is not changed. During each of the middle and second periods, the IGBT element is driven by a constant DC voltage source and a constant gate resistance. Since the relationship between the gate-emitter voltage and the collector current and the collector-emitter voltage of the IGBT element is non-linear, when the IGBT element is driven by a constant DC voltage source and a constant gate resistance, as shown in FIG. The rate of change of the collector-emitter voltage during the period 1 and the rate of change of the collector current during the second period change with time, and are locally high near the switching point between the first and second periods. A rate of change occurs. For this reason, when the maximum value of the current change rate is limited so that the surge voltage is equal to or less than a desired value, the average value of the current change rate is reduced more than necessary, and the switching loss can be sufficiently reduced. There was no problem. The rate of change of the collector-emitter voltage and the rate of change of the collector current are the
Since it also changes depending on the value of the voltage between the emitters, if the voltage / current operating point deviates from the selected voltage / current operating point due to fluctuations in the load or power supply voltage, the switching loss and surge voltage can be sufficiently reduced. There is a problem that the effect is small in applications that do not function and have a wide operating range of voltage / current.

SUMMARY OF THE INVENTION The present invention has been made to solve the above-mentioned problems, and it is intended to reduce the rate of change of the current of the main electrode or the rate of change of the voltage between the main electrodes during the switching operation.
It is an object of the present invention to provide a driving circuit for a semiconductor element that can obtain desired switching characteristics such as keeping the voltage substantially constant regardless of a change in load or power supply voltage.

[0011]

According to a first aspect of the present invention, there is provided a drive circuit for a semiconductor device, wherein the control electrode of a voltage-driven self-extinguishing semiconductor device controls a conduction state between main electrodes by a voltage applied between control electrodes. In the drive circuit connected between,
Switching signal generating means for generating a signal for turning on / off the voltage-driven self-extinguishing semiconductor element, and switching for generating a command relating to switching characteristics of the voltage-driven self-extinguishing semiconductor element by an output of the switching signal generating means Characteristic setting means, function generating means for generating a command for a current to be supplied to the control electrode by an output of the switching characteristic setting means, and current generating means for generating a current to be supplied to the control electrode by an output of the function generating means The function generating means generates an inverse function of a transfer function from a current supplied to the control electrode to a switching characteristic of the voltage-driven self-extinguishing semiconductor element set by the switching characteristic setting means. It was made.

According to a second aspect of the present invention, there is provided a driving circuit for a semiconductor element,
2. The current change rate setting means according to claim 1, wherein the switching characteristic setting means generates a command of a change rate of a current flowing through a main electrode of the voltage-driven self-extinguishing semiconductor element.
Voltage change rate setting means for generating a command for the rate of change of the voltage between the main electrodes, wherein the function generation means outputs the current change rate setting means to the control electrode of the voltage-driven self-extinguishing semiconductor device. A first function generating means for generating a command for a current to be supplied; and a second function generating means for generating a command for a current to be supplied to the control electrode by the voltage change rate setting means. Switching means for switching the second function generating means to send any of its outputs to the current generating means; and Current change point detection means for detecting a change point from a first period in which the change in the flowing current is slow to a second period in which the change in the subsequent current is steep, and an output of the current change point detection means Previous Switching means, in the first period, the to second function generating means, in the second period of time is obtained by the switched to the first function generating means.

According to a third aspect of the present invention, there is provided a driving circuit for a semiconductor device,
2. The current change rate setting means according to claim 1, wherein the switching characteristic setting means generates a command of a change rate of a current flowing through a main electrode of the voltage-driven self-extinguishing semiconductor element.
Voltage change rate setting means for generating a command for the rate of change of the voltage between the main electrodes, wherein the function generation means outputs the current change rate setting means to the control electrode of the voltage-driven self-extinguishing semiconductor device. A first function generating means for generating a command for a current to be supplied; and a second function generating means for generating a command for a current to be supplied to the control electrode by the voltage change rate setting means. Switching means for switching the second function generating means and sending any of the outputs to the current generating means; and switching the second function generating means to the main electrode during a transition from a non-conductive state to a conductive state between the main electrodes by voltage control of the control electrode. Current change point detection means for detecting a change point from a second period in which the flowing current is steep to a first period in which the subsequent change in current is slow, and the output of the current change point detection means Previous Switching means, in the second period of time said to first function generating means, in the first period is obtained by the switched to the second function generating means.

According to a fourth aspect of the present invention, there is provided a driving circuit for a semiconductor element,
The switching characteristic setting means according to any one of claims 1 to 3, wherein the switching characteristic setting means includes a current change rate setting means for generating a command of a change rate of a current flowing through a main electrode of the voltage-driven self-extinguishing semiconductor device. Function generating means, as a command of a current supplied to a control electrode of the voltage-driven self-extinguishing semiconductor element, a reciprocal of a derivative of a transfer function from a voltage between the control electrodes to a current of the main electrode, The apparatus has first function generation means for generating a product of the capacitance between the control electrodes and the output of the current change rate setting means. Claim 5
The driving circuit for a semiconductor device according to any one of claims 1 to 3, wherein the switching characteristic setting means issues a command for a rate of change of a voltage between main electrodes of the voltage-driven self-extinguishing semiconductor device. Having a voltage change rate setting means to generate, the function generating means, the capacitance between the control electrode and the main electrode as a command of the current supplied to the control electrode of the voltage-driven self-extinguishing semiconductor device, And a second function generating means for generating a product with the output of the voltage change rate setting means.

According to a sixth aspect of the present invention, there is provided a driving circuit for a semiconductor device,
4. The voltage-driven self-extinguishing device according to claim 1, further comprising a capacitor connected between control electrodes of the voltage-driven self-extinguishing semiconductor device, and wherein the switching characteristic setting means includes the voltage-driven self-extinguishing device. A current change rate setting unit for generating a command for a change rate of a current flowing through the main electrode of the semiconductor element, wherein the function generation unit determines a command for a current supplied to the control electrode from a voltage between the control electrodes as the main electrode. A first function generation for generating a product of a reciprocal of a differential of a transfer function to a current of the current, a sum of a capacitance between the control electrodes and a capacitance of the capacitor, and an output of the current change rate setting means. Means. A drive circuit for a semiconductor device according to claim 7, wherein the capacitor is connected between the control electrode and the main electrode of the voltage-driven self-extinguishing semiconductor device according to any one of claims 1 to 3. And the switching characteristic setting means includes voltage change rate setting means for generating a command for a change rate of the voltage between the main electrodes, and the function generating means controls the control as a command for a current supplied to the control electrode. And a second function generating means for generating a product of the sum of the capacitance between the electrode and the main electrode and the capacitance of the capacitor and the output of the voltage change rate setting means.

The driving circuit for a semiconductor device according to claim 8 is
4. The voltage-driven self-extinguishing semiconductor device according to claim 2, further comprising a capacitor connected between control electrodes of the voltage-driven self-extinguishing semiconductor device, and wherein the switching characteristic setting means includes:
The voltage-driven self-extinguishing semiconductor device has current change rate setting means for generating a command for a change rate of a current flowing through a main electrode, and the function generating means outputs the control electrode as a command for a current supplied to the control electrode. The product of the reciprocal of the differential of the transfer function from the voltage between the current to the main electrode, the sum of the capacitance between the control electrodes and the capacitance of the capacitor, and the output of the current change rate setting means. The first function generating means is provided, and the capacitance of the capacitor is set so that the outputs of the first and second function generating means at the time of the switching operation of the switching means are equal to each other. According to a ninth aspect of the present invention, a driving circuit for a semiconductor device according to the second or third aspect further comprises a capacitor connected between a control electrode and a main electrode of the voltage-driven self-extinguishing semiconductor device. Switching characteristic setting means,
A voltage change rate setting unit that generates a command for a change rate of a voltage between the main electrodes, wherein the function generation unit generates a command for a current to be supplied to the control electrode; A second function generating means for generating a product of a sum of a capacitance and an electrostatic capacity of the capacitor and an output of the voltage change rate setting means; The capacitance of the capacitor is set so that the outputs of the function generating means are equal to each other.

According to a tenth aspect of the present invention, in the driving circuit for driving a semiconductor device according to any one of the fourth, sixth and eighth aspects, the first function generating means is a voltage-driven self-extinguishing semiconductor. Instead of the reciprocal of the derivative of the transfer function from the voltage between the control electrodes of the element to the current of the main electrode, the voltage between the control electrodes is approximated by a linear expression and used.
The driving circuit for a semiconductor device according to claim 11 is a semiconductor device driving circuit according to claim 4,
According to any one of claims 6 and 8,
The first function generating means uses a constant which is a representative value instead of a reciprocal of a differential of a transfer function from a voltage between the control electrodes of the voltage-driven self-extinguishing semiconductor device to a current of the main electrode. It is like that. According to a twelfth aspect of the present invention, in the driving circuit for a semiconductor element according to any one of the fifth, seventh, and ninth aspects, the second function generating means includes:
Instead of the capacitance between the control electrode and the main electrode of the voltage-driven self-extinguishing semiconductor device, the voltage between the control electrode and the main electrode is approximated by a linear expression and used. is there. According to a thirteenth aspect of the present invention, in the driving circuit for a semiconductor device according to any one of the fifth, seventh, and ninth aspects, the second function generating means includes a voltage-driven self-extinguishing semiconductor device. Instead of the capacitance between the control electrode and the main electrode, the capacitance is approximated by a constant as a representative value.

[0018]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Prior to a specific description of an embodiment of the present invention, a basic operation will be described. FIG.
Is a voltage-driven self-extinguishing semiconductor device that controls the conduction state between main electrodes by a voltage applied between control electrodes.
FIG. 2 is a circuit diagram showing an example of a drive circuit (switching circuit) of a GBT (insulated gate transistor) element 1. 1a
1c are electrodes of the IGBT element 1, and 1a is a first electrode.
1b is a collector which is a first main electrode, and 1c is an emitter which is an electrode serving as both a second control electrode and a second main electrode. Reference numeral 2 denotes an on DC voltage source for supplying a voltage for turning on the IGBT element 1;
An off DC voltage source for supplying a voltage for turning off the GBT element 1, a switching signal generating means 4 for generating a signal for turning on / off the IGBT element 1, and a switching DC generating means 5 based on an output of the switching signal generating means 4. On / off switching means for switching between the voltage source 2 and the off DC voltage source 3 to apply a voltage to the gate 1a of the IGBT element 1, 9 is a gate resistor, 12 is a DC voltage source, 13 is a load, 14 is a freewheeling diode, 15 is This is the wiring inductance. Although the off DC voltage source 3 is shown as a negative voltage in the figure, it is assumed here that the voltage is zero. Also, the polarity of the two DC voltage sources may be reversed depending on the configuration of the PN polarity of the material of the semiconductor element.

First, a turn-off operation for changing the IGBT element from a conductive state (on state) to a non-conductive state (off state) will be described. FIG. 26 is a waveform diagram of the voltage and current of the IGBT element 1 at the time of turn-off in the drive circuit of FIG. In FIG. 25, when the switching signal generating means 4 outputs an ON signal, the ON / OFF switching means 5 is switched to the DC voltage source 2 for ON, and is turned ON via the gate resistor 9 between the gate 1a and the emitter 1c of the IGBT element 1. A voltage is applied, and the IGBT element 1 turns on.
When the IGBT element 1 is turned on, the voltage V
dc is applied to the load 13, and a current flows through a path of the DC voltage source 12-the wiring inductance 15-the load 13-the IGBT element 1-the DC voltage source 12.

The voltage and current of the IGBT element 1 at this time are represented by a point at time 0 in FIG. IG at time 0
Since the BT element 1 is on, the voltage Vce between the collector 1b and the emitter 1c is almost zero, and the collector current I
c is equal to the current IL of the load 13. The output Vs of the on / off switching means 5 is equal to the voltage of the DC voltage source 2 for ON. Since a capacitor is provided between the gate and the emitter of the IGBT element 1, if the ON time is sufficiently long, the gate and the emitter are turned off.
The emitter-to-emitter voltage Vge is charged up to the output Vs of the on / off switching means 5, that is, the voltage of the on-state DC voltage source 2,
The gate current Ig becomes zero.

Next, when the switching signal generating means 4 outputs an off signal at time t1, the on / off switching means 5 is switched to the off DC voltage source 3 side and the IG
An off-voltage is applied to the gate 1a of the BT element 1 via the gate resistor 9. Here, since the off DC voltage source 3 is set to zero volt, the output Vs of the on / off switching means 5 becomes zero. And the gate-emitter voltage V
ge goes to zero (gate resistance 9 × IGBT element 1)
(The capacitor between the gate and the emitter). However, normally, the on-state DC voltage source 2 is set sufficiently high with respect to the on-state voltage Von required to flow the load current IL. Until the voltage Von required to flow the IGBT element 1
Is not actually turned off.

At time t2, the gate-emitter voltage Vge becomes equal to the voltage V necessary for flowing the load current IL.
When it reaches on, the IGBT element 1 starts a turn-off operation, and the collector-emitter voltage Vce increases. At this time, the voltage applied to the load 13 is a voltage obtained by subtracting the collector-emitter voltage Vce from the voltage Vdc of the DC voltage source 12, but the current IL flowing due to the action of the inductance of the load 13 does not change suddenly. Therefore, the collector current Ic does not change suddenly, and the collector current Ic forms a first period in which the current changes slowly. In the first period, a feedback action to keep the gate-emitter voltage Vge at the voltage Von required to flow the load current IL acts so as not to change the collector current Ic, and the gate-emitter voltage Vge is maintained. Vge is a substantially constant voltage. This feedback effect is caused by the increase in the collector-emitter voltage Vce, which causes the collector-emitter voltage of the IGBT element 1 to rise.
Displacement current flows from the collector to the gate via the gate-to-gate capacitor, and this displacement current is realized by causing a voltage drop at the gate resistor 9.

At time t3, when collector-emitter voltage Vce reaches voltage Vdc of DC voltage source 12, free-wheeling diode 14 turns on, and part of the current flowing through load 13 starts commutating to free-wheeling diode 14. . The collector current I corresponding to the current Id commutated to the freewheeling diode 14
Since c decreases, the collector current Ic forms a second period in which the current changes sharply. In the second period, the feedback action for keeping the gate-emitter voltage Vge constant is lost, and the gate-emitter voltage Vge starts decreasing toward zero again.
Further, since the collector current Ic is equal to the current flowing through the wiring inductance 15, the current flowing through the wiring inductance 15 during the second period also decreases. For this reason,
The wiring inductance 15 generates a voltage represented by (wiring inductance × current reduction rate). Since this voltage has the same polarity as the voltage of the DC voltage source 12, the collector-emitter voltage Vce has a waveform in which a surge voltage generated by the wiring inductance 15 is superimposed on the DC voltage source 12.

At time t4, when the gate-emitter voltage Vge reaches the gate threshold voltage Vth of the IGBT element 1, the collector current Ic becomes zero, and the IGBT element 1 completes the turn-off operation. At time t4, the actual turn-off operation is completed, but since the gate-emitter voltage Vge has not yet reached zero, it continues to decrease toward zero. At time t5, the gate-emitter voltage Vge reaches zero, and all operations are completed. In the turn-off operation of the IGBT element 1, the collector current Ic changes slowly during the first period (t2
To t3), and a second period (from t3 to t4) in which the change of the collector current Ic subsequently generated is steep. Note that the first period and the second period do not mean the temporal order in which they occur, and in the case of turn-on described later, the order in which they occur is reversed.

Next, a turn-on operation for transitioning the IGBT element from the off state to the on state will be described. FIG.
FIG. 7 is a waveform diagram of voltage and current when the IGBT element 1 in the switching circuit of FIG. 25 is turned on. FIG.
When the switching signal generating means 4 outputs an off signal, the on / off switching means 5 is switched to the off DC voltage source 3 side, and the off voltage is reduced between the gate 1a and the emitter 1c of the IGBT element 1 via the gate resistor 9. Is applied, and the IGBT element 1 is turned off. When the IGBT element 1 is turned off, the collector current Ic flowing through the collector 1b of the IGBT element 1 becomes almost zero. Until then IGBT element 1
The current that has flowed through the diode is commutated to the freewheeling diode 14, and the current flows through the freewheeling diode 14, the load 13, and the freewheeling diode 14. At this time, the voltage between the terminals of the freewheeling diode 14 becomes almost zero, and thus the voltage Vd of the DC voltage source 12
c is applied to the IGBT element 1.

The voltage and current of the IGBT element 1 at this time are represented by a point at time 0 in FIG. IG at time 0
Since the BT element 1 is off, the collector current Ic is almost zero, and the collector-emitter voltage Vce is equal to the voltage Vdc of the DC voltage source 12. The output Vs of the on / off switching means 5 is equal to the voltage of the off DC voltage source 3. Here, since the off DC voltage source 3 is set to zero volt, the output Vs of the on / off switching means 5 becomes zero. Since the gate-emitter of the IGBT element 1 is a capacitor, if the off-time is sufficiently long, the gate-emitter voltage Vge becomes the output Vs of the on / off switching means 5, that is, zero, and the gate current Ig also becomes zero.

Next, when the switching signal generating means 4 outputs an ON signal at time t6, the ON / OFF switching means 5 is switched to the DC voltage source 2 for ON, and the IGB
An on-voltage is applied to the gate 1 a of the T element 1 via the gate resistor 9. Then, the gate-emitter voltage Vge
Is directed toward the voltage of the DC voltage source 2 for ON (gate resistance 9
Of the IGBT element 1 (the capacitance of the capacitor between the gate and the emitter of the IGBT element 1). But,
Since the collector current does not flow until the gate-emitter voltage Vge reaches the gate threshold voltage Vth of the IGBT element 1, the actual turn-on operation of the IGBT element 1 is not performed.

At time t7, when gate-emitter voltage Vge reaches gate threshold voltage Vth, IGBT element 1 starts turn-on operation, and collector current Ic increases, and collector current Ic changes sharply. Two periods are formed. During the second period, the current IL flowing due to the inductance of the load 13 does not change suddenly. Therefore, the current Id of the freewheeling diode 14 is a value obtained by subtracting the collector current Ic from the substantially constant load current IL. Since the freewheel diode 14 is on while the freewheel diode current Id is positive, the voltage of the DC voltage source 12 is applied to the IGBT element 1 and the wiring inductance 15. This second
In the period, the current flowing through the wiring inductance 15 is equal to the collector current Ic, so that the current flowing through the wiring inductance 15 increases with the collector current Ic.
Therefore, the wiring inductance 15 generates a voltage represented by (wiring inductance × current increase rate). Since this voltage has the opposite polarity to the voltage of the DC voltage source 12, the collector-emitter voltage Vce becomes the voltage Vce of the DC voltage source 12.
The waveform is obtained by subtracting the voltage generated by the wiring inductance 15 from dc. The second period during the turn-on operation is essentially the same as the above-described second period during the turn-off operation.

At time t8, the freewheeling diode 14 remains on for a while after the current Id of the freewheeling diode 14 reaches zero. This operation is known as a diode recovery operation. During this recovery operation, a negative current flows through the freewheel diode 14, so that the collector current Ic is changed to the current IL of the load 13 by the freewheel diode 1.
4 is the sum of the recovery currents.

When the recovery operation of the freewheeling diode 14 is completed at time t9, the freewheeling diode 14 is turned off, and the direct current voltage source 12-wiring inductance 15-load 1
Current flows through the path of the 3-IGBT element 1-DC voltage source 12. Since the current IL does not change suddenly due to the inductance of the load 13, the collector current Ic in the same current path does not change suddenly, and the collector current Ic forms a first period in which the current changes slowly. In the first period, a feedback action to keep the gate-emitter voltage Vge at the voltage Von required to flow the load current IL acts so as not to change the collector current Ic, and the gate-emitter voltage Vge is maintained. Vge is a substantially constant voltage. This feedback effect is realized by a displacement current flowing from the gate to the collector via the collector-gate capacitor of the IGBT element 1 due to a decrease in the collector-emitter voltage Vce, and this displacement current causes a voltage drop at the gate resistor 9. Is done. The first period during the turn-on operation is essentially the same as the above-described first period during the turn-off operation.

At time t10, when the collector-emitter voltage Vce reaches zero volts, the collector-emitter voltage Vce thereafter becomes zero.
Since the emitter-to-emitter voltage Vce is constant at zero volts and no displacement current is generated, the above-mentioned feedback effect is lost and the gate-emitter voltage Vge is restored to the on-state DC voltage source 2.
Start increasing towards the voltage of. At time t10, the actual turn-on operation has been completed, but since the gate-emitter voltage Vge has not yet reached the voltage of the DC voltage source 2 for ON, the gate-emitter voltage Vg continues.
e keeps increasing. At time t11, the gate-emitter voltage Vge reaches the voltage of the DC voltage source 2 for ON,
Complete all operations. In the turn-on operation of the IGBT element 1 described above, the collector current Ic changes continuously between the second period (from t7 to t8) where the change is steep and the recovery operation of the freewheeling diode. A slow first period (from t9 to t10) is formed.

Next, an embodiment of the present invention will be described with reference to FIG. FIG. 1 is a block diagram showing one basic configuration of a driving circuit of a semiconductor device according to an embodiment of the present invention. In FIG.
Switching signal generating means for generating a signal for turning off, 1
6 is an IGBT based on the output of the switching signal generating means 4.
It is a switching characteristic setting means for generating a command relating to the switching characteristic of the element 1. Reference numeral 17 denotes a gate 1a of the IGBT element 1 according to the output of the switching characteristic setting means 16.
Function generating means for generating a command for the current to be supplied to
An inverse function of a transfer function from the gate current of the IGBT element 1 to the switching characteristic set by the switching characteristic setting means 16 is generated. Reference numeral 18 denotes a gate current generating means as a current generating means for supplying a current corresponding to the output of the function generating means 17 to the gate 1a of the IGBT element 1. The above switching signal generating means 4 and switching characteristic setting means 1
6. Function generating means 17 and gate current generating means 18
Thus, the driving circuit of the IGBT element 1 is configured. Other details are the same as those in FIG.

The operation will be described with reference to FIGS. Here, a case where the rate of change of the collector current at the time of turn-off is set as a command regarding the switching characteristic generated by the switching characteristic setting means 16 will be described. I
It is during the second period in FIG. 26 that the collector current changes when the GBT element 1 is turned off. In this second period, the gate-emitter voltage Vge decreases, and the collector current Ic also decreases. In the second period, the collector current Ic and the gate-emitter voltage Vge have a functional relationship, and the expression (1) is established. In the equation (1), f is a transfer function from the gate-emitter voltage Vge to the collector current Ic. Ic = f (Vge) (1)

Since the gate-emitter of the IGBT element 1 is a capacitor, if the capacitance between the gate and the emitter is Cge, the following equation (2) is obtained between the gate current Ig and the gate-emitter voltage Vge. Holds. To be precise, the gate current Ig is equal to the gate-emitter capacitance Cg.
e, but also flows through the gate-collector capacitance Cgc in the second period, since the collector-emitter voltage Vce, that is, the gate-collector voltage Vgc is high in the second period, the gate-collector static voltage having strong voltage dependence is high. The capacitance Cgc is sufficiently small with respect to the gate-emitter capacitance Cge and can be ignored. Ig = Cge · (dVge / dt) (2) From the equations (1) and (2), the gate current I of the IGBT element 1 is obtained.
The transfer function from g to the collector current change rate dIc / dt is given by equation (3). Further, when the inverse function of equation (3) is obtained, equation (4) is obtained. In the equation (4), when dIc / dt is a constant value A, the equation (5) is obtained.

[0035]

(Equation 1)

As is clear from the definition of the inverse function, IGB
From the gate current Ig of the T element 1 to the collector current change rate dI
When the gate current Ig obtained from the inverse function (5) of the transfer function (3) to c / dt flows through the IGBT element 1 having the transfer function of the expression (3), the collector current change rate dIc /
dt is constant (dIc / dt = A). Therefore, FIG.
In the output of the switching signal generating means 4,
The switching characteristic setting means 16 generates a command of a constant value A, and the function generating means 17 generates a gate current command from an inverse function of a transfer function from the gate current Ig to the collector current change rate dIc / dt according to the equation (5). The IGBT element 1 having the transfer function of the equation (3)
If the current according to the output of the function generating means 17 is supplied to the circuit, the rate of change of the collector current can be kept constant.

Although only the operation at the time of turn-off has been described above, the turn-on only reverses the polarity of the rate of change of the voltage and current of each part of the IGBT element 1.
Obviously, a similar effect can be obtained. In the above description, the collector current change rate of the IGBT element 1 is kept constant. If necessary, the output of the switching characteristic setting means 16 is changed so that the collector current change rate follows a desired function. It is possible. Furthermore, although only the collector current change rate has been described above as an example, it is clear from the definition of the inverse function that similar effects can be obtained for other switching characteristics such as the collector-emitter voltage. A specific configuration in the embodiment will be described below.

Embodiment 1 FIG. 2 is a block diagram showing a drive circuit of the semiconductor device according to the first embodiment of the present invention, and shows a configuration in which the rate of change of the collector current during the switching operation is kept substantially constant. In FIG. 2, reference numeral 19 denotes current change rate setting means as switching characteristic setting means, 20 denotes first function generation means, and 18 denotes IGB.
The gate current generating means serves as a current generating means for supplying a current corresponding to the output of the first function generating means 20 to the gate 1a of the T element 1. Others are the same as those in FIG. FIG. 3 is a block diagram showing the first function generator 20 and the internal functional configuration of the IGBT element 1 shown in FIG. In FIG. 3, the first function generating means 20 comprises 20a and 20b, 20a is the reciprocal of the differential of the transfer function from the gate-emitter voltage of the IGBT element 1 to the collector current, and 20b is the gate of the IGBT element 1. This is the capacitance between the emitters. 1d, 1e,
1f and 1g are means for representing the characteristics of the IGBT element 1,
1d and 1e are means representing the equation (2), and are IGBTs respectively.
It is the reciprocal of the gate-emitter capacitance of the element 1 and the integrating means. 1f is a transfer function representing the equation (1), and 1g is a differentiating means.

Next, the operation will be described with reference to FIGS. FIG. 26 shows the IGBT element 1 as described above.
FIG. 3 is a waveform diagram showing a voltage and a current at the time of turn-off of FIG. FIG.
In the output of the switching signal generating means 4,
When the current change rate setting means 19 generates the command dIc / dt ^* , the output Ig ^* of the first function generating means 20 becomes IG
The reciprocal 20a of the differential of the transfer function f from the gate-emitter voltage to the collector current of the BT element 1 and the IGBT element 1
Of the gate-emitter capacitance 20b and the output of the current change rate setting means 19, and is expressed by equation (6). This output Ig ^* is a gate current command, and is the same as the equation (4).

[0040]

(Equation 2)

The gate current generating means 18 is an IGBT element 1
In this case, the current according to the output of the first function generating means 20 flows, so that the gate current Ig of the IGBT element 1 is expressed by the following equation (7). Ig = Ig ^* (7) In the second period of FIG. 26 in which the collector current Ic changes when the IGBT element 1 is turned off, the gate-emitter voltage Vge and the gate current Ig are expressed by the following equation (2). IGBT element 1 has a gate-emitter voltage V
Ge is given by equation (8) from the reciprocal 1d of the gate-emitter capacitance and the integrating means 1e.

[0042]

(Equation 3)

Since the collector current Ic and the gate-emitter voltage Vge are in the relationship of the equation (1), the IGB
The collector current Ic of the T element 1 is represented by a transfer function 1f from the gate-emitter voltage of the IGBT element 1 to the collector current.
Is given by equation (9). And the collector current I
The derivative of c with the differentiating means 1g is the collector current change rate dIc / dt, which can be modified as shown in equation (10).

[0044]

(Equation 4)

Therefore, in the configuration of FIG. 3, if the output dIc / dt ^* of the current change rate setting means 19 is kept constant,
The collector current change rate of the GBT element 1 can be kept constant.

FIG. 4 is a block diagram of FIG. 3 in which the transfer function from the gate-emitter voltage to the collector current is given by
It is the figure rewritten about the case represented by Formula 1).
In FIG. 4, reference numeral 20c denotes a reciprocal of a derivative of a transfer function from the gate-emitter voltage of the IGBT element 1 to the collector current, 20d denotes a multiplication means for obtaining a product of the output of the 20c and the output of the current change rate setting means 19, 20e is a detector for obtaining a gate-emitter voltage Vge used in 20c. Others are the same as FIG.

Next, the operation will be described in detail with reference to FIG. In the second period of FIG. 26 in which the collector current changes when the IGBT element 1 is turned off, the collector current Ic and the gate-emitter voltage Vge are substantially the following (1)
There is a functional relationship of the expression 1). Here, Kp is the transconductance, and Vth is the gate threshold voltage. Ic = Kp · (Vge−Vth) ² (11) Similarly to the equation (2), the gate-emitter voltage Vge
And the gate current Ig have a functional relationship of equation (12). Ig = Cge · (dVge / dt) (12) From equations (11) and (12), the transfer function from the gate electrode Ig of the IGBT element 1 to the collector current change rate dIc / dt is expressed by equation (13). Become. Then, when the inverse function of equation (13) is obtained, equation (14) is obtained.

[0048]

(Equation 5)

The equation (14) is similar to the equation (4) in that the IGBT
It is the product of the reciprocal 20c of the differential of the transfer function from the gate-emitter voltage of the element 1 to the collector current, the gate-emitter capacitance 20b of the IGBT element 1, and the output of the current change rate setting means 19. From equation (4), the first of FIG.
(14) in the same way as the configuration of the function generating means 20 is obtained.
The configuration of the first function generator 20 in FIG. 4 is obtained from the equation.

Although only the operation at the time of turn-off has been described above, it is apparent that the same effect can be obtained with the turn-on since the polarity of the rate of change of the voltage and current of each part of the IGBT element is only reversed. is there. This method
The simplest configuration is provided when there is no restriction on the switching characteristics of the IGBT element other than the collector current change rate.
Although the equations (1) to (14) are applied to the period in which the collector current changes sharply as in the second period in FIG. 26, the equations (1) to (14) are used in other periods. It is possible to drive the IGBT element with the gate current obtained by this.

Embodiment 2 FIG. 5 is a block diagram showing a driving circuit of a semiconductor device according to the second embodiment of the present invention, and shows a configuration in which the rate of change of the collector-emitter voltage during the switching operation is kept substantially constant.
In FIG. 5, reference numeral 21 denotes voltage change rate setting means as switching characteristic setting means, 22 denotes second function generation means,
Reference numeral 8 denotes a gate current generating means as a current generating means for supplying a current corresponding to the output of the second function generating means 22 to the gate 1a of the IGBT element 1. Others are the same as those in FIG. FIG. 6 is a block diagram including the function configuration inside the second function generating means 22 and the IGBT element 1 of FIG. In FIG.
The second function generating means 22 is a capacitance between the gate and the collector of the IGBT element 1. 1h, 1i, 1j, 1k, 1
m and 1n are means for indicating the characteristics of the IGBT element 1;
h is the gate-collector capacitance (g represents a function);
1i is reciprocal means, 1j is multiplication means, 1k is integration means, 1
m is an adding means, and 1n is a differentiating means.

Next, the operation will be described with reference to FIGS. FIG. 26 is a waveform diagram showing voltage and current when the IGBT element 1 is turned off. It is during the first period in FIG. 26 that the voltage between the collector and the emitter greatly changes when the IGBT element 1 is turned off. In the first period, a feedback action to keep the gate-emitter voltage Vge at the voltage Von required for flowing the load current IL acts so as not to change the collector current 1c, and the gate-emitter voltage Vge is maintained. Vge is a substantially constant voltage. This feedback effect is realized by a displacement current flowing from the collector to the gate via the collector-gate capacitor of the IGBT element 1 due to an increase in the collector-emitter voltage Vce, and this displacement current causes a voltage drop in the gate drive circuit. Is done. Since dVge / dt is almost zero and the current between the gate and the emitter is negligible, this displacement current is nothing less than the gate current and the following (1)
5) Formula holds. Ig = Cgc · dVgc / dt (15)

The following equation (16) holds for the voltage between the three terminals of the gate, collector and emitter of the IGBT element 1. Vce = Vgc + Vge (16) From the equations (15) and (16), the rate of change of the collector-emitter voltage dVce / from the gate current Ig of the IGBT element 1 is obtained.
The transfer function to dt and its inverse function are (17)
Equation (18) is obtained. dVce / dt = Ig / Cgc + dVge / dt (17) Ig = Cgc · (dVce / dt−dVge / dt) (18)

As described above, since the gate-emitter voltage Vge is substantially constant during the first period, the rate of change of the gate-emitter voltage dVge / dt.
Is a sufficiently small value and can be ignored. Therefore, (19)
Equation (20) is obtained. dVce / dt = Ig / Cgc (19) Ig = Cgc · dVce / dt (20) The means for generating the equation (20) is the second function generating means 22. In FIG. 6, the output of the switching signal generating means 4 causes the voltage change rate setting means 21 to issue a command dVc.
When e / dt ^* is generated, the output Ig ^* of the second function generating means 22 is given by the following equation (21). This output Ig ^* is a gate current command. Ig ^* = Cgc · dVce / dt ^* (21)

The gate current generating means 18 is an IGBT element 1
In this case, the current according to the output of the second function generating means 22 flows, so that the gate current Ig of the IGBT element 1 is given by the following equation (22). Ig = Ig ^* (22) As described above, in the first period of FIG. 26 in which the collector-emitter voltage Vce greatly changes when the IGBT element 1 is turned off, the gate-collector voltage Vgc and the gate Since the relationship with the current Ig is expressed by equation (15), IG
The gate-collector voltage Vce of the BT element 1 is represented by a gate-collector capacitance 1h, a reciprocal unit 1i, a multiplication unit 1
j, given by equation (23) from the integrating means 1k.

[0056]

(Equation 6)

Further, since the gate-collector voltage Vgc and the collector-emitter voltage Vce have the relationship of the formula (16), the collector-emitter voltage Vce is given by the adding means 1m by the formula (24). The collector-emitter voltage Vce is differentiated by the differentiating means 1n to obtain the collector-emitter voltage change rate dVce / dt. As described above, the gate-emitter voltage Vge during the first period is equal to Since the voltage is substantially constant, the voltage can be modified as shown in equation (25).

[0058]

(Equation 7)

Therefore, in the configuration of FIG. 6, if the output dVce / dt ^* of the voltage change rate setting means 21 is kept constant,
The rate of change of the collector-emitter voltage of the IGBT element 1 can be kept constant.

FIG. 7 is a diagram in which the block of FIG. 6 is rewritten for the case where the voltage characteristic of the gate-collector capacitance is expressed by the following equation (26). In FIG. 7, 2
2a is the gate-collector capacitance of the IGBT element 1,
22b is a multiplying means for obtaining the product of the output of 22a and the output of the voltage change rate setting means 21, and 22c is a detector for obtaining the gate-collector voltage Vgc used in 22a. Others are the same as FIG. Next, the operation will be described with reference to FIG. In the first period of FIG. 26 in which the voltage between the collector and the emitter greatly changes when the IGBT element 1 is turned off, the capacitance Cgc between the gate and the collector substantially has the voltage dependency of the equation (26). Here, Cjo is a zero bias capacitance, Vj is a junction potential difference, M
Is the junction gradient coefficient. The potential dependence of equation (26) is expressed as (1
Even when applied to the expression (5), the relationship of the expressions (15) to (25) holds.

[0061]

(Equation 8)

Although only the operation at the time of turning off has been described above, it is apparent that the same effect can be obtained with respect to turn-on since the polarity of the rate of change of the voltage and current of each part of the IGBT element is only reversed. is there. This method
The simplest configuration is provided when there is no restriction on the switching characteristics of the IGBT element other than the collector-emitter voltage change rate. Equations (15) to (26) are applied to the period in which the collector current is almost constant and the voltage between the collector and the emitter greatly changes as in the first period in FIG. ) To (26), it is possible to drive the IGBT element with the gate current obtained.

Embodiment 3 In this embodiment, during the turn-off switching operation, the change rate of the collector-emitter voltage is kept substantially constant during the first period in which the collector-emitter voltage largely changes, and the change of the collector current Ic is sharp. In the second period, the rate of change of the collector current is kept substantially constant. FIG. 8 is a block diagram showing a driving circuit of a semiconductor device according to a third embodiment of the present invention. In FIG. 8, reference numeral 23 denotes a switch between the output of the first function generator 20 and the output of the second function generator 22. The switching means 24 for supplying the IGBT element 1 from the on state to the off state during the transition of the IGBT element 1 from the on state to the off state from the first period in which the change in the current flowing through the IGBT element 1 is slow to A current change point detecting means 18 for detecting a change point of the current when transitioning to the steep second period is a gate current generating means 18 as a current generating means.
A current corresponding to the output of the first or second function generating means 20, 21 is supplied to the gate 1a of the IGBT element 1. Other features are the same as those in the first embodiment or the second embodiment, and thus description thereof is omitted.

Next, the operation will be described. The operation in the case where the switching means 23 is provided on the first function generating means 20 side is the same as that of the first embodiment, and the description is omitted. Further, the operation in the case of the second function generating means 22 is the same as that of the second embodiment, and the description is omitted. Here, the switching means 23 and the current change point detecting means 24 will be described. In FIG. 26 showing the turn-off waveform of the IGBT element 1, the collector-emitter voltage of the IGBT element 1 largely changes during the first period in FIG. 26, and the collector current does not change during this period. The collector current changes during the second period in FIG. 26. During this period, the collector-emitter voltage does not change significantly. As described above, the second period in which the first embodiment exhibits the effect and the first period in which the second embodiment exhibits the effect.
The period is divided at time t3 in FIG.

For this reason, the current change point detecting means 24
Time t3, which corresponds to the boundary between the period and the second period,
In a first period before time t3, the switching unit 23 is controlled to use the output of the second function generation unit 22, and in a second period after time t3, the output of the first function generation unit 20 is controlled. By controlling the switching means 23 so as to use the same effect, the same effects as in the second and first embodiments can be obtained in each period. Therefore, according to the configuration of FIG. 8, the first,
In the second period, the rate of change of the collector-emitter voltage or the rate of change of the collector current can be kept substantially constant.

FIGS. 9 to 16 show the current change point detecting means 24.
FIG. 7 is a block diagram showing first to seventh configuration examples of FIG. First,
A first configuration example of the current change point detecting means 24 will be described with reference to FIG. Reference numeral 24a denotes first voltage detection means for detecting the collector-emitter voltage Vce of the IGBT element 1, 24b denotes second voltage detection means for detecting the voltage Vdc of the DC voltage source 12, and 24c denotes comparison means. 26, at time t3, which is the boundary between the first period and the second period, the collector-emitter voltage Vce and the DC voltage source 1
Since the two voltages Vdc cross each other, the comparison means 24c uses the first voltage detection means 24a and the second voltage detection means 2
The time t3 can be detected by comparing the magnitude of the output of 4b. In applications where the voltage of the DC voltage source 12 is constant, the same effect can be obtained even if the second voltage detecting means 24b is replaced with another DC voltage source serving as a reference for comparison.

Next, a second configuration example of the current change point detecting means 24 will be described with reference to FIG. 24a is IG
Voltage detecting means for detecting the gate-emitter voltage Vge of the BT element 1, 24c is a comparing means, and 24d is a DC voltage source serving as a reference for comparison. From FIG. 26, at time t3, which is the boundary between the first period and the second period, the gate-emitter voltage Vge starts decreasing from the gate voltage Von required to flow the load current IL. By setting the voltage of the voltage source 24d to be slightly lower than Von, the comparison unit 24c can detect the time t3.

Next, a third configuration example of the current change point detecting means 24 will be described with reference to FIG. 24a is IG
Voltage detecting means for detecting the gate-emitter voltage Vge of the BT element 1; 24e, differentiating means for differentiating the output of the voltage detecting means 24a; 24c, comparing means;
This is a DC voltage source serving as a reference for comparing the voltage change rate of the emitter-to-emitter voltage Vge. According to FIG. 26, at time t3, which is the boundary between the first period and the second period, the gate-emitter voltage Vge starts to decrease from the gate voltage Von required to flow the load current IL. -Time t3 can be detected by differentiating the emitter-to-emitter voltage Vge to obtain the rate of change and comparing the difference with the reference value of the DC voltage source 24d.

Next, a fourth configuration example of the current change point detecting means 24 will be described with reference to FIG. 24f is IG
First current detection means for detecting the collector current Ic of the BT element 1, 24g is second current detection means for detecting the current IL of the load 13, and 24c is comparison means. From FIG. 26,
At time t3, which is the boundary between the first period and the second period, the collector current Ic starts to decrease from the current IL of the load 13, so that the comparing unit 24c causes the first current detecting unit 24f and the second voltage detecting The time t3 can be detected by comparing the output of the means 24g.

Next, a fifth configuration example of the current change point detecting means 24 will be described with reference to FIG. 24f is IG
Current detection means for detecting the collector current Ic of the BT element 1, 24c a comparison means, and 24h a DC voltage source serving as a reference for comparison. 26, at time t3, which is the boundary between the first period and the second period, the collector current Ic starts to decrease from the current IL of the load 13, so that the DC voltage source 24
By setting the voltage of h to be slightly lower than the load current IL, the comparing means 24c can detect the time t3.

Next, a sixth configuration example of the current change point detecting means 24 will be described with reference to FIG. 24f is IG
Current detecting means for detecting the collector current Ic of the BT element 1; 24e, a differentiating means for differentiating the output of the current detecting means 24f; 24c, a comparing means; 24h, a DC voltage serving as a reference for comparing the current change rate of the collector current Ic. Source. FIG.
6, time t3, which is the boundary between the first period and the second period,
Since the collector current Ic starts to decrease from the load current IL, the time t3 can be detected by differentiating the collector current Ic to obtain the rate of change, and comparing the difference with the reference value of the DC voltage source 24h. .

Next, a seventh configuration example of the current change point detecting means 24 will be described with reference to FIG. 24f is a current detecting means for detecting the current Id of the diode 14, 24c
Is a comparing means, and 24h is a DC voltage source serving as a reference for comparison. According to FIG. 26, at time t3, which is the boundary between the first period and the second period, the current Id of the diode 14 starts increasing from zero. Thus, the time t3 can be detected by the comparing means 24c.

Next, an eighth configuration example of the current change point detecting means 24 will be described with reference to FIG. 24f is a current detecting means for detecting the current Id of the diode 14, 24e
Is a differentiating means for differentiating the output of the current detecting means 24f;
c is a comparison means, and 24h is a DC voltage source serving as a reference for comparing the current change rate of the current Id of the diode 14. FIG.
6, time t3, which is the boundary between the first period and the second period,
Since the current Id of the diode 14 starts increasing from zero, the time t3 can be detected by differentiating the current Id of the diode 14 to obtain the rate of change, and comparing the change with the reference value of the DC voltage source 24h. it can.

Embodiment 4 In this embodiment, during the turn-on switching operation, the rate of change of the collector-emitter voltage is maintained substantially constant during the first period, and the rate of change of the collector current is maintained substantially constant during the second period. is there. The configuration in this embodiment is the same as that shown in FIG. In FIG. 8, reference numeral 24 denotes IG
When transitioning the BT element 1 from the off state to the on state,
This is a current change point detecting means for detecting a change point of the current when the current flowing through the GBT element 1 transitions from the second period in which the change of the current is steep to the first period in which the change of the subsequent current is slow. The specific configuration of the current change point detecting means 24 can be the same as or equivalent to that shown in FIGS. 9 to 16, and both of them are the DC voltage sources 24d,
The only difference is the voltage of 24h or the output polarity of the differentiating means 24e. The other parts are the same as those in the third embodiment, and the description is omitted.

Next, the operation will be described. The operation in the case where the switching means 23 is provided on the first function generating means 20 side is the same as that of the first embodiment, and the description is omitted. Further, the operation in the case of the second function generating means 22 is the same as that of the second embodiment, and the description is omitted. Here, the switching means 23 and the current change point detecting means 24 will be described. In FIG. 27 showing the turn-on waveform of the IGBT element 1, the collector-emitter voltage of the IGBT element 1 largely changes during the first period in FIG. 27, and the collector current does not change during this period. The collector current changes during the second period in FIG. 27, and during this period, the collector-emitter voltage does not change significantly. As described above, the second period in which the first embodiment exhibits the effect and the first period in which the second embodiment exhibits the effect.
The period is divided by the recovery operation period of the freewheeling diode 14 between the time t8 and the time t9 in FIG.

For this reason, the current change point detecting means 24
Free-wheel diode 14 at the boundary between the period and the second period
Of the recovery operation period (t8 to t9) at time t8
The switching means 23 is controlled so as to use the output of the first function generating means 20 in the previous second period, and the second function generating means 2 is controlled in the first period after time t9.
If the switching means 23 is controlled to use the second output, the same effect as in the first and second embodiments can be obtained in each period. Therefore, according to the configuration of FIG. 8, the rate of change of the collector current or the rate of change of the collector-emitter voltage can be kept substantially constant during the second and first periods during the turn-on of the IGBT element 1. In the third and fourth embodiments, the present invention is applied to turn-on and turn-off, respectively. However, these embodiments may be combined and applied to both turn-on and turn-off.

Embodiment 5 FIG. 17 is a block diagram of a drive circuit of a semiconductor device according to the fifth embodiment of the present invention, showing a configuration in a case where the rate of change of the collector current during the switching operation is kept substantially constant. In the figure, 2
Reference numeral 5 denotes a first capacitor connected between the gate and the emitter of the IGBT element 1, and the other components are the same as those in the first embodiment, and thus the description is omitted. Next, the operation will be described. FIG. 17 is the same as FIG. 2 except for the first capacitor 25, and the first capacitor 25 is connected in parallel with the gate-emitter capacitance Cge of the IGBT element 1. For this reason, the capacitance of the first capacitor 25 is set to C
If it is set to 1, this is equivalent to an increase in the gate-emitter capacitance of the IGBT element 1 from Cge to (Cge + C1). Therefore, the output Ig of the first function generating means 20
^* Is the gate-emitter voltage capacitance Cge of equation (6)
If the expression (27) is obtained by changing the above expression, the same effect as in the first embodiment can be obtained in the configuration of FIG.

[0078]

(Equation 9)

According to this embodiment, IGBT element 1
Can be adjusted by C1, the gate current Ig is adjusted to the collector current change rate command value dIc / d
The gate current value can be set independently of t ^* and the characteristics f and Cge of the IGBT element 1 due to the convenience of the drive circuit. In applications where switching is performed as in Embodiments 3 and 4, the gate current I before and after switching is used.
By setting the capacitance of the first capacitor 25 so that g becomes equal, it is possible to prevent a sudden change in the gate current Ig at the time of switching and realize smooth switching.

Embodiment 6 FIG. FIG. 18 is a block diagram of a drive circuit of a semiconductor device according to the sixth embodiment of the present invention, showing a configuration in which the rate of change of the collector-emitter voltage during the switching operation is kept substantially constant. In the figure, reference numeral 26 denotes a second capacitor connected between the collector and the gate of the IGBT element 1, and the other components are the same as those of the second embodiment, so that the description is omitted. Next, the operation will be described. FIG. 18 is the same as FIG. 5 except for the second capacitor 26, and the second capacitor 26 is an IGB
It is connected in parallel with the collector-gate capacitance Cgc of the T element 1. Therefore, assuming that the capacitance of the second capacitor 26 is C2, this is equivalent to an increase in the capacitance between the collector and the gate of the IGBT element 1 from Cgc to (Cgc + C2). Therefore, the output Ig ^* of the second function generating means 22 is changed to the collector-gate capacitance C of equation (21).
If the expression (28) is obtained by changing gc, the same effect as in the second embodiment can be obtained in the configuration of FIG. Ig ^* = (Cgc + C2) · dVce / dt ^* (28)

According to this embodiment, IGBT element 1
Can be adjusted by C2, so that the gate current Ig is controlled by the command value d of the rate of change of the collector-emitter voltage.
The gate current value can be set independently of Vce / dt ^* and the characteristic Cgc of the IGBT element 1 due to the convenience of the drive circuit. Further, in applications where switching is performed as in Embodiments 3 and 4, the capacitance of the second capacitor 26 is set so that the gate current Ig before and after switching is equalized, so that the gate at the time of switching is set. A sudden change in the current Ig can be prevented and smooth switching can be realized. further,
Since the second capacitor 26 is a capacitor having a constant capacitance C2, the influence of the voltage dependence of the collector-gate capacitance Cgc as shown in the equation (26) is suppressed, and the gate current Ig is reduced. It has the effect of reducing the change.

Embodiment 7 FIG. FIG. 19 is a circuit diagram showing a drive circuit of a semiconductor element according to the seventh embodiment of the present invention, showing a configuration in which the rate of change of the collector current during the switching operation is kept substantially constant. In FIG. 19, the gate current generating means 18 comprises transistors 18a and 18b constituting a current mirror circuit, the current change rate setting means 19 comprises a DC voltage source 19a and a transistor 19b, and a first function generating means 20 Is composed of a resistor 20f and transistors 20g and 20h forming a current mirror circuit. Other configurations are the same as those in the first embodiment, and thus description thereof is omitted. In FIG. 19, only a turn-off circuit portion is shown for simplification.

Next, the operation will be described. When the switching signal generating means 4 outputs an off signal, the transistor 19b turns off. The current mirror circuit including the transistors 20g and 20h is connected to the voltage Vc of the DC voltage source 19a.
c, and the transistor 20g is connected to a resistor 20f
, The collector current Ig ^* of the transistor 20h is given by Equation (29) if the base-emitter voltage drop of the transistor is ignored. Here, K1 is the current gain of the current mirror circuit, R1 is the resistance value of the resistor 20f, Vg
e is a gate-emitter voltage of the IGBT element 1. Ig ^* = K1 · (Vcc−Vge) / R1 (29)

The collector current Ig ^* of the transistor 20h in the above equation is an output of the first function generating means 20, and is a gate current command. Since the transistor 19b is off, all the collector current Ig ^* of the transistor 20h flows to the transistor 18a. Since the transistors 18a and 18b form a current mirror circuit,
A current obtained by multiplying the collector current Ig ^* of the transistor 20h by the gain of the current mirror circuit composed of the transistors 18a and 18b flows through the gate of the BT element 1.
Comparing equation (29) with equation (6) of the first embodiment,
The gate-emitter capacitance Cge of the GBT element 1 and the collector current change rate dIc / dt ^* are taken as constants, and the IGBT
It can be seen that the linear expression of the gate-emitter voltage is used as the reciprocal of the derivative of the transfer function from the gate-emitter voltage of the device 1 to the collector current. (A) of FIG.
Is a graph showing the relationship between the gate-emitter voltage of the IGBT element 1 and the inverse of the derivative of the transfer function from the gate-emitter voltage to the collector current. From the equation (14) shown in the first embodiment, the collector current change rate dIc / d
When t ^* is a constant A, the gate current command Ig ^* is given by (3)
0).

[0085]

(Equation 10)

The reciprocal of the derivative of the transfer function from the gate-emitter voltage of the IGBT element 1 to the collector current is (3
This is the part of 1 / (2 · Kp · (Vge−Vth)) on the right side of the equation (0), which is a hyperbolic function. (A) of FIG.
As can be seen from FIG.
Good approximation is possible by the linear expression of the voltage between the emitters. Therefore, by using the linear expression of the gate-emitter voltage as the reciprocal of the derivative of the transfer function from the gate-emitter voltage to the collector current, the same effect as in the first embodiment can be realized with a simple circuit.

Embodiment 8 FIG. FIG. 21 is a circuit diagram showing a drive circuit for a semiconductor element according to an eighth embodiment of the present invention, showing a configuration in which the rate of change of the collector current during the switching operation is kept substantially constant. In FIG. 21, the gate current generating means 18 comprises transistors 18a and 18b constituting a current mirror circuit, and the current change rate setting means 19 comprises a DC voltage source 19a and a transistor 19b. 20j is a resistor as the first function generating means. Other configurations are the same as those in the first embodiment, and a description thereof will be omitted. FIG. 21 shows only a turn-off circuit portion for simplification.

Next, the operation will be described. When the switching signal generating means 4 outputs an off signal, the transistor 19b turns off. Since the resistor 20j is connected to the voltage Vcc of the DC voltage source 19a, the current Ig ^* flowing through the resistor 20j is given by Expression (31) if the base-emitter voltage drop of the transistor is ignored. Where R
1 is the resistance value of the resistor 20j. Ig ^* = Vcc / R1 (31) The current Ig ^* of the resistor 20j in the above equation is an output of the first function generating means and is a gate current command. Transistor 1
Since 9b is off, the current Ig ^{* of the} resistor 20j becomes
All flows to the transistor 18a. Transistor 18a
And 18b constitute a current mirror circuit.
A current obtained by multiplying the current Ig ^{* of the} resistor 20j by the gain of the current mirror circuit including the transistors 18a and 18b flows through the gate of the GBT element 1. Comparing the equation (31) with the equation (6) of the first embodiment, the gate-emitter capacitance Cge and the collector current change rate dIc / dt ^* of the IGBT element 1 are constants,
It can be seen that a constant is also used for the reciprocal of the derivative of the transfer function from the voltage between the emitters to the collector current.

FIG. 20B is a graph showing the relationship between the gate-emitter voltage of the IGBT element 1 and the inverse of the derivative of the transfer function from the gate-emitter voltage to the collector current. The reciprocal of the derivative of the transfer function from the gate-emitter voltage to the collector current of the IGBT element 1 is a hyperbolic function as described in the seventh embodiment. As can be seen from FIG. 20B, when the reciprocal of the derivative of the transfer function from the gate-emitter voltage to the collector current of the IGBT element 1 is a hyperbolic function, it cannot be said that the constant is a good approximation. However, when the IGBT element has a characteristic with small change, it is possible to obtain generally good results even with a constant. In such a case, the same effect as in the first embodiment can be realized with a simple circuit by using a constant, which is a representative value, as the reciprocal of the differential of the transfer function from the gate-emitter voltage to the collector current.

Embodiment 9 FIG. 22 is a circuit diagram showing a drive circuit of a semiconductor device according to a ninth embodiment of the present invention, showing a configuration in which the rate of change of the collector-emitter voltage during the switching operation is kept substantially constant. In FIG. 22, the gate current generating means 18 comprises transistors 18a and 18b constituting a current mirror circuit, the voltage change rate setting means 21 comprises a DC voltage source 21a and a transistor 21b, and the second function generating means 22 Is composed of transistors 22f and 22g forming a current mirror circuit with the resistors 22d and 22e. The other parts are the same as those in the second embodiment, and the description is omitted. FIG.
Here, for simplicity, only the turn-off circuit is shown.

Next, the operation will be described. When the switching signal generating means 4 outputs an off signal, the transistor 21b turns off. The resistor 22d is connected to the voltage Vcc of the DC voltage source 21a, and the transistor 22g is connected to the collector of the IGBT element 1 via the resistor 22e. -If the voltage drop between emitters is ignored, (3
It is given by equation 2). Here, K2 is the current gain of the current mirror circuit, R2 is the resistance value of the resistor 22d, R3 is the resistance value of the resistor 22e, and Vce is the collector-emitter voltage of the IGBT element 1. Ig ^* = Vcc / R2-K2.Vce / R3 (32)

In the above equation, Ig ^* is the second function generator 22.
And a gate current command. Transistor 2
Since 1b is off, the gate current command Ig ^* of Expression (32) all flows through the transistor 18a. Since the transistors 18a and 18b constitute a current mirror circuit, the gate of the IGBT element 1 has a gate current command I
A current flows by multiplying g ^* by the gain of the current mirror circuit including the transistors 18a and 18b. (32)
When the equation is compared with the equation (21) of the second embodiment, the IGBT
Collector-emitter voltage change rate dVce / d of device 1
t ^* is a constant, and the collector-gate voltage Cgc of the collector-emitter voltage of the IGBT element 1 is 1
It can be seen that the following equation is used.

FIG. 23A is a graph showing the voltage dependence of the collector-gate capacitance Cgc of the IGBT element 1. From equation (26) of the second embodiment, the gate current command Ig ^* when the collector-emitter voltage change rate dVce / dt ^* is a constant B is given by equation (33).

[0094]

[Equation 11]

The capacitance Cgc between the collector and gate of the IGBT element 1 is given by Cjo · (1 + Vgc) on the right side of the equation (33).
/ Vj) ^-M , which is a hyperbolic function. As can be seen from FIG. 23A, a good approximation can be made by a linear expression of the collector-emitter voltage, for example, as in Expression (32). Therefore, by using a linear expression of the collector-emitter voltage as the collector-gate capacitance Cgc, the same effect as in the second embodiment can be realized with a simple circuit. The circuit of FIG. 22 can realize the same function as that of the seventh embodiment shown in FIG. 19 by changing the connection point of the resistor 22e from the collector of the IGBT element 1 to the gate.

Embodiment 10 FIG. FIG. 24 is a circuit diagram showing a drive circuit for a semiconductor device according to the tenth embodiment of the present invention, showing a configuration in which the rate of change of the collector-emitter voltage during the switching operation is kept substantially constant. In FIG. 24, gate current generating means 18 includes transistors 18a and 18b forming a current mirror circuit.
The voltage change rate setting means 21 includes a DC voltage source 21
a and the transistor 21b. 22h is a resistor as a second function generating means. The other parts are the same as those in the second embodiment, and the description is omitted. FIG. 24 shows only a turn-off circuit portion for simplification.

Next, the operation will be described. When the switching signal generating means 4 outputs an off signal, the transistor 19b turns off. Since the resistor 22h is connected to the voltage Vcc of the DC voltage source 19a, the current Ig ^* flowing through the resistor 22h is given by Expression (34) if the base-emitter voltage drop of the transistor is ignored. Where R
2 is the resistance value of the resistor 22h. Ig ^* = Vcc / R2 (34) The current Ig ^* of the resistor 22h in the above equation is an output of the second function generating means and is a gate current command. Transistor 1
Since 9b is off, the current Ig ^{* of the} resistor 22h becomes
All flows to the transistor 18a. Transistor 18a
And 18b constitute a current mirror circuit.
A current obtained by multiplying the current Ig ^{* of the} resistor 22h by the gain of the current mirror circuit including the transistors 18a and 18b flows through the gate of the GBT element 1. Comparing Equation (34) with Equation (21) of the second embodiment, the collector-emitter voltage change rate dVce / dt ^* of the IGBT element 1 is a constant, and the collector-gate capacitance Cgc of the IGBT element 1 is Also uses constants.

FIG. 23 (b) is a graph showing the voltage dependence of the collector-gate capacitance Cgc of the IGBT element 1. The collector-gate capacitance Cgc of the IGBT element 1 has a hyperbolic function as described in the ninth embodiment. As can be seen from FIG.
When the collector-gate capacitance Cgc of the GBT element 1 is a hyperbolic function, it cannot be said that a constant is a good approximation. However, when the IGBT element has a characteristic with a small change,
Generally good results can be obtained even with constants. Further, when a capacitor is connected in parallel to the collector-gate capacitance Cgc as shown in the sixth embodiment, the effect of the change in the collector-gate capacitance Cgc is reduced, so that the constant is almost constant. Good results can be obtained. Therefore, by using a constant that is a representative value as the capacitance Cgc between the collector and the gate, the same effect as in the second embodiment can be realized with a simple circuit.

In the seventh to tenth embodiments, the case where the current change rate of the collector is controlled or the case where the voltage change rate between the collector and the emitter is controlled has been described, but both may be combined. Although the circuit for turning off is shown, it can be controlled approximately similarly with a simple circuit for turning on, and both can be used together.

[0100]

According to the first aspect of the present invention, the inverse function of the transfer function from the current supplied to the control electrode to the switching characteristic of the voltage-driven self-extinguishing semiconductor device set by the switching characteristic setting means is represented by a function Since the generating means is generated, the non-linearity of the semiconductor element is corrected irrespective of the fluctuation of the load or the power supply voltage, and the desired switching characteristic set by the switching characteristic setting means, for example, a constant change rate of the main electrode current, Alternatively, characteristics such as a constant change rate of the voltage between the main electrodes are obtained. Therefore, it is possible to suppress the rate of change of voltage and current unnecessarily high or low during the switching operation, which is effective for reducing switching loss, reducing surge voltage at turn-off, and reducing recovery current of the freewheeling diode at turn-on. There is. Further, it is effective in reducing noise and leakage current caused by switching.

According to the second and third aspects of the present invention,
The voltage change rate setting means and the second function generation means are used in the first period in which the current change of the main electrode is slow, and the current change rate setting means and the first function generation are used in the second period in which the current change is steep. Means, the desired switching characteristics,
For example, characteristics such as keeping the voltage change rate between the main electrodes constant in the first period and keeping the current change rate of the main electrode constant in the second period are obtained.

According to the fourth aspect of the present invention, the first function generating means includes: a reciprocal of a differential of a transfer function from a voltage between the control electrodes to a current of the main electrode; Since the product of the output and the output of the current change rate setting means is generated, the same effects as those of the first to third aspects are obtained. According to the invention according to claim 5, the second function generating means generates the product of the capacitance between the control electrode and the main electrode and the output of the voltage change rate setting means. Claims 1-3
The same effect as that of the above is obtained.

According to the invention of claim 6, there is provided a capacitor connected between the control electrodes, and the first function generating means calculates the sum of the capacitance between the control electrodes and the capacitance of the capacitor. Since the function is used to generate the function, it is possible to set an appropriate gate current value according to the convenience of the drive circuit. According to the invention according to claim 7, there is provided a capacitor connected between the control electrode and the main electrode, and the second function generating means determines the capacitance between the control electrode and the main electrode and the capacitance of the capacitor. Since the function is generated using the sum of the capacitances, it is possible to set an appropriate gate current value according to the convenience of the drive circuit.

According to the eighth aspect of the present invention, there is provided a capacitor connected between the control electrodes, and when the first and second function generating means are switched, the output of the capacitor is set so that both outputs become equal to each other. Since the capacitance is set, smooth switching can be performed. According to the ninth aspect of the present invention, there is provided a capacitor connected between the control electrode and the main electrode, wherein the first and second function generating means are switched so that both outputs become equal to each other when switching. Since the capacitance is set, smooth switching can be performed.

According to the tenth aspect of the present invention, the first function generation means replaces the reciprocal of the differential of the transfer function from the voltage between the control electrodes to the current of the main electrode, and replaces the first order of the voltage between the control electrodes. Since it is used by approximating the equation, the current change rate of the main electrode can be controlled with a simple circuit. Claim 1
According to the invention according to the first aspect, the first function generating means approximates and uses the representative value of the constant instead of the reciprocal of the differential of the transfer function from the voltage between the control electrodes to the current of the main electrode. Thus, the current change rate of the main electrode can be controlled with a simple circuit. According to the twelfth aspect of the present invention, the second function generating means approximates the voltage between the control electrode and the main electrode by a linear expression instead of the capacitance between the control electrode and the main electrode. Since it is used, the rate of voltage change between the main electrodes can be controlled with a simple circuit. According to the thirteenth aspect of the present invention, the second function generating means uses the constant between the control electrode and the main electrode by approximating it with a constant which is a representative value. The circuit can control the voltage change rate between the main electrodes.

[Brief description of the drawings]

FIG. 1 is a block diagram showing a basic configuration of a drive circuit of a semiconductor device according to an embodiment of the present invention.

FIG. 2 is a block diagram of a driving circuit of the semiconductor element according to the first embodiment of the present invention.

FIG. 3 is a block diagram showing a functional configuration inside a semiconductor element and a driving circuit thereof according to the first embodiment of the present invention;

FIG. 4 is a block diagram showing a functional configuration inside a semiconductor element and a driving circuit thereof according to the first embodiment of the present invention;

FIG. 5 is a block diagram of a drive circuit for a semiconductor element according to a second embodiment of the present invention.

FIG. 6 is a block diagram showing a functional configuration inside a semiconductor element and a driving circuit thereof according to a second embodiment of the present invention;

FIG. 7 is a block diagram showing a functional configuration inside a semiconductor element and a driving circuit thereof according to a second embodiment of the present invention;

FIG. 8 is a block diagram of a drive circuit of a semiconductor device according to the third and fourth embodiments of the present invention.

FIG. 9 is a block diagram illustrating a configuration example of a current change point detection unit according to Embodiment 3 of the present invention.

FIG. 10 is a block diagram illustrating a configuration example of a current change point detection unit according to a third embodiment of the present invention.

FIG. 11 is a block diagram illustrating a configuration example of a current change point detection unit according to Embodiment 3 of the present invention.

FIG. 12 is a block diagram illustrating a configuration example of a current change point detection unit according to Embodiment 3 of the present invention.

FIG. 13 is a block diagram illustrating a configuration example of a current change point detection unit according to Embodiment 3 of the present invention.

FIG. 14 is a block diagram illustrating a configuration example of a current change point detection unit according to Embodiment 3 of the present invention.

FIG. 15 is a block diagram illustrating a configuration example of a current change point detection unit according to Embodiment 3 of the present invention.

FIG. 16 is a block diagram illustrating a configuration example of a current change point detection unit according to Embodiment 3 of the present invention.

FIG. 17 is a block diagram of a drive circuit of a semiconductor device according to a fifth embodiment of the present invention.

FIG. 18 is a block diagram of a drive circuit of a semiconductor device according to a sixth embodiment of the present invention.

FIG. 19 is a circuit diagram of a drive circuit of a semiconductor device according to a seventh embodiment of the present invention.

FIG. 20 is a graph showing the reciprocal of the derivative of the transfer function from the gate-emitter voltage to the collector voltage in the seventh and eighth embodiments of the present invention.

FIG. 21 is a circuit diagram of a drive circuit of a semiconductor element according to an eighth embodiment of the present invention.

FIG. 22 is a circuit diagram of a drive circuit of a semiconductor device according to a ninth embodiment of the present invention.

FIG. 23 is a graph showing the voltage dependence of the collector-gate capacitance in the ninth and tenth embodiments of the present invention.

FIG. 24 is a circuit diagram of a drive circuit for a semiconductor element according to a tenth embodiment of the present invention.

FIG. 25 is a circuit diagram for explaining switching characteristics of the IGBT element.

FIG. 26 is a waveform chart showing a turn-off operation of the IGBT element.

FIG. 27 is a waveform chart showing a turn-on operation of the IGBT element.

FIG. 28 is a circuit diagram of a conventional drive circuit for a semiconductor device.

FIG. 29 is a waveform chart showing the operation of a conventional semiconductor element drive circuit.

[Explanation of symbols]

1 IGBT element, 1a gate, 1b collector, 1
c emitter, 4 switching signal generating means, 16
Switching characteristic setting means, 17 function generating means, 18
Gate current generation means, 19 current change rate setting means, 2
0 first function generating means, 21 voltage change rate setting means,
22 second function generating means, 23 switching means, 24
Current change point detecting means, 25 first capacitor, 26
Second capacitor.

Claims (13)

[Claims] 1. A drive circuit connected between said control electrodes of a voltage-driven self-extinguishing semiconductor element for controlling conduction between main electrodes by a voltage applied between control electrodes, wherein said voltage-driven self-extinguishing is performed. Switching signal generating means for generating a signal for turning on / off the semiconductor element, switching characteristic setting means for generating a command relating to the switching characteristic of the voltage-driven self-extinguishing semiconductor element by an output of the switching signal generating means, A function generating means for generating a command for a current to be supplied to the control electrode by an output of the characteristic setting means; and a current generating means for generating a current to be supplied to the control electrode by an output of the function generating means. The generating means is configured to generate the voltage-driven self-voltage set by the switching characteristic setting means from a current supplied to the control electrode. A drive circuit for a semiconductor device, wherein an inverse function of a transfer function to a switching characteristic of an arc-extinguishing semiconductor device is generated. A switching characteristic setting means for generating a command for a change rate of a current flowing through a main electrode of the voltage-driven self-extinguishing semiconductor device; A voltage change rate setting means for generating a command, wherein the function generating means generates a command for a current supplied to a control electrode of the voltage-driven self-extinguishing semiconductor device by an output of the current change rate setting means. And a second function generating means for generating a command for the current supplied to the control electrode by the voltage change rate setting means, and switching between the first and second function generating means. Switching means for sending any output to a current generating means; and And a current change point detecting means for detecting a change point from the period of 1 to a second period in which the change of the current that follows is steep. 2. The driving circuit for a semiconductor device according to claim 1, wherein the switching to the second function generating means is performed during the period, and the switching to the first function generating means is performed during the second period. 3. The switching characteristic setting means includes: a current change rate setting means for generating a command for a change rate of a current flowing through a main electrode of the voltage-driven self-extinguishing semiconductor element; A voltage change rate setting means for generating a command, wherein the function generating means generates a command for a current supplied to a control electrode of the voltage-driven self-extinguishing semiconductor device by an output of the current change rate setting means. And a second function generating means for generating a command for the current supplied to the control electrode by the voltage change rate setting means, and switching between the first and second function generating means. A switching unit for sending any output to a current generation unit; and a switching unit for changing a current flowing through the main electrode during a transition from a non-conductive state to a conductive state between the main electrodes by voltage control of the control electrode. Current change point detection means for detecting a change point from the second period to the first period in which the current change is slow, and the switching means uses the output of the current change point detection means to switch the second current to the second period. 2. The driving circuit for a semiconductor device according to claim 1, wherein switching to the first function generating means is performed during the period, and switching to the second function generating means is performed during the first period. 4. The switching characteristic setting means has a current change rate setting means for generating a command of a change rate of a current flowing through a main electrode of the voltage-driven self-extinguishing semiconductor element, and the function generating means has the voltage driving type. A reciprocal of a derivative of a transfer function from a voltage between the control electrodes to a current of the main electrode as a command of a current supplied to a control electrode of the self-extinguishing semiconductor device, a capacitance between the control electrodes, 4. The driving circuit for a semiconductor device according to claim 1, further comprising a first function generating means for generating a product with an output of the change rate setting means. 5. The switching characteristic setting means includes a voltage change rate setting means for generating a command of a change rate of a voltage between main electrodes of the voltage-driven self-extinguishing semiconductor element, and the function generating means includes: Generation of a second function for generating a product of the capacitance between the control electrode and the main electrode and the output of the voltage change rate setting means as a command for the current supplied to the control electrode of the self-extinguishing semiconductor device. 4. The driving circuit for a semiconductor device according to claim 1, further comprising means. 6. A capacitor connected between control electrodes of a voltage-driven self-extinguishing semiconductor device, wherein the switching characteristic setting means changes a rate of change of a current flowing through a main electrode of the voltage-driven self-extinguishing semiconductor device. The current generation rate setting means for generating a command of the, the function generating means, as a command of the current supplied to the control electrode, the reciprocal of the derivative of the transfer function from the voltage between the control electrodes to the current of the main electrode, And a first function generating means for generating a product of a sum of a capacitance between the control electrodes and a capacitance of the capacitor and an output of the current change rate setting means. A driving circuit for a semiconductor device according to any one of claims 1 to 3. 7. A voltage-driven self-extinguishing semiconductor device having a capacitor connected between a control electrode and a main electrode of the semiconductor device, wherein the switching characteristic setting means generates a command for a rate of change in voltage between the main electrodes. Having a voltage change rate setting means to perform, the function generating means, as a command of the current supplied to the control electrode, the sum of the capacitance between the control electrode and the main electrode and the capacitance of the capacitor, 4. The driving circuit for a semiconductor device according to claim 1, further comprising a second function generating means for generating a product with an output of the voltage change rate setting means. 8. A voltage-driven self-extinguishing semiconductor device having a capacitor connected between control electrodes thereof, wherein the switching characteristic setting means changes a rate of change of a current flowing through a main electrode of the voltage-driven self-extinguishing semiconductor device. The current generation rate setting means for generating a command of the, the function generating means, as a command of the current supplied to the control electrode, the reciprocal of the derivative of the transfer function from the voltage between the control electrodes to the current of the main electrode, A first function generating means for generating a product of a sum of a capacitance between the control electrodes and a capacitance of the capacitor and an output of the current change rate setting means, and a switching operation of a switching means. 4. The driving circuit for a semiconductor device according to claim 2, wherein the capacitance of the capacitor is set so that the outputs of the first and second function generating means at the time are equal to each other. . 9. A capacitor having a capacitor connected between a control electrode and a main electrode of a voltage-driven self-extinguishing semiconductor device, and a switching characteristic setting means for issuing a command for a rate of change of a voltage between the main electrodes. Having a voltage change rate setting means to perform, the function generating means, as a command of the current supplied to the control electrode, the sum of the capacitance between the control electrode and the main electrode and the capacitance of the capacitor, A second function generating means for generating a product of the output of the voltage change rate setting means and the first and second functions at the time of the switching operation of the switching means;
4. The driving circuit for a semiconductor device according to claim 2, wherein the capacitance of the capacitor is set so that the outputs of the function generating means are equal to each other. 10. The driving circuit for a semiconductor device according to claim 4, wherein the first function generating means is provided between the control electrodes of the voltage-driven self-extinguishing semiconductor device. A drive circuit for a semiconductor element, wherein a voltage between the control electrodes is approximated by a linear expression instead of a reciprocal of a differential of a transfer function from a voltage to a current of the main electrode. 11. The driving circuit for a semiconductor device according to claim 4, wherein the first function generating means is provided between the control electrodes of the voltage-driven self-extinguishing semiconductor device. A driving circuit for a semiconductor element, wherein a driving function is approximated by a constant, which is a representative value, instead of a reciprocal of a differential of a transfer function from a voltage to a current of a main electrode. 12. The driving circuit for a semiconductor device according to claim 5, wherein
Is used in place of the capacitance between the control electrode and the main electrode of the voltage-driven self-extinguishing semiconductor device by approximating the voltage between the control electrode and the main electrode by a linear expression. A driving circuit for a semiconductor element, wherein 13. The driving circuit for a semiconductor device according to claim 5, wherein
Wherein the function generating means is used by approximating with a constant which is a representative value instead of the capacitance between the control electrode and the main electrode of the voltage-driven self-extinguishing semiconductor device. Driver circuit for semiconductor devices.