JP3650264B2

JP3650264B2 - Semiconductor device drive circuit

Info

Publication number: JP3650264B2
Application number: JP12761998A
Authority: JP
Inventors: 政弘木全
Original assignee: Mitsubishi Electric Corp
Current assignee: Mitsubishi Electric Corp
Priority date: 1998-05-11
Filing date: 1998-05-11
Publication date: 2005-05-18
Anticipated expiration: 2018-05-11
Also published as: JPH11330935A

Description

【０００１】
【発明の属する技術分野】
この発明は、スイッチング用電圧駆動形半導体素子であるＩＧＢＴ（Insulated Gate Bipolar Transistor、絶縁ゲート形トランジスタ）素子やＭＯＳＦＥＴ（Metal Oxicide Silicon Feild Effect Transistor、電界効果トランジスタ）素子などの半導体素子のゲート駆動回路に関するものである。
【０００２】
【従来の技術】
図２８は例えば特開平９−６５６４４号公報に述べられた従来の半導体素子の駆動回路を示す回路図であり、図において、１は制御電極間に印加する電圧によって主電極間の導通状態を制御する電圧駆動形自己消弧半導体素子であるＩＧＢＴ素子であり、１ａは第１の制御電極であるゲート、１ｂは第１の主電極であるコレクタ、１ｃは第２の制御電極と第２の主電極を兼ねた電極であるエミッタ、２はＩＧＢＴ素子１をオンさせるための電圧を供給するオン用直流電圧源、３はＩＧＢＴ素子１をオフさせるための電圧を供給するオフ用直流電圧源、４はＩＧＢＴ素子１をオン／オフする信号を発生するスイッチング信号発生手段、５はスイッチング信号発生手段４の出力によりオン用直流電圧源２とオフ用直流電圧源３を切り換えてＩＧＢＴ素子１のゲート１ａに電圧を印加するオンオフ切り換え手段、６はゲート抵抗を切り換えるゲート抵抗切り換え手段であり、ゲート抵抗切り換え手段６は、抵抗９と、抵抗１０と、切り換え手段１１から構成されている。７はＩＧＢＴ素子１の主電極に流れる電流を検出する電流検出手段、８は電流検出手段７の出力によりゲート抵抗切り換え手段６を制御する制御手段、１２は直流電圧源、１３は負荷、１４は環流ダイオード、１５は配線インダクタンスである。
【０００３】
まず図２８、図２９を用いて従来例の基本動作について説明する。図２８でスイッチング信号発生手段４がオン信号を出力すると、オンオフ切り換え手段５はオン用直流電圧源２側に切り換えられ、ＩＧＢＴ素子１のゲート１ａ−エミッタ１ｃ間にゲート抵抗切り換え手段６を介してオン電圧が印加され、ＩＧＢＴ素子１はオンする。ＩＧＢＴ素子１がオンすると、ＩＧＢＴ素子１のコレクタ１ｂ−エミッタ１ｃ間電圧Ｖｃｅはほぼゼロとなる。この時、直流電圧源１２の電圧Ｖｄｃは負荷１３に印加され、直流電圧源１２−配線インダクタンス１５−負荷１３−ＩＧＢＴ素子１−直流電圧源１２の経路で電流が流れる。
【０００４】
スイッチング信号発生手段４がオフ信号を出力すると、オンオフ切り換え手段５はオフ用直流電圧源３側に切り換えられ、ＩＧＢＴ素子１のゲート１ａ−エミッタ１ｃ間にゲート抵抗切り換え手段６を介してオフ電圧が印加され、ＩＧＢＴ素子１はオフする。ＩＧＢＴ素子１がオフすると、ＩＧＢＴ素子１のコレクタ１ｂを流れる電流Ｉｃはほぼゼロとなる。それまでＩＧＢＴ素子１に流れていた電流は環流ダイオード１４に転流し、環流ダイオード１４−負荷１３−環流ダイオード１４の経路で電流が流れる。この時、環流ダイオード１４の端子間電圧はほぼゼロとなるため、直流電圧源１２の電圧ＶｄｃがＩＧＢＴ素子１に印加される。
図２９は、ＩＧＢＴ素子１がオン状態からオフ状態に遷移するターンオフ時の波形を示したものであり、（ａ）はゲート抵抗の抵抗値が小、（ｂ）は大、（ｃ）は図２９のように切り換える場合を示す。ＩＧＢＴ素子１のターンオフ動作は、まずコレクタ−エミッタ間電圧Ｖｃｅが増加することから始まる。この期間中コレクタ電流Ｉｃはほぼ一定であり、コレクタ電流の変化が緩慢な第１の期間が形成される。コレクタ−エミッタ間電圧Ｖｃｅが増加して直流電圧源１２の電圧Ｖｄｃに達すると、環流ダイオード１４がオンして負荷１３を流れる電流が環流ダイオード１４に流れ始める。負荷１３の持つインダクタンスの作用で電流は一定と見なせるため、環流ダイオード１４に流れる電流Ｉｄの分だけコレクタ電流Ｉｃは減少する。従って図２９に示したように、コレクタ−エミッタ間電圧Ｖｃｅが直流電圧源１２の電圧Ｖｄｃに達した後はコレクタ電流Ｉｃは急速に減少し、コレクタ電流の変化が急峻な第２の期間が形成される。
【０００５】
この第２の期間においてコレクタ電流Ｉｃは配線インダクタンス１５に流れる電流と等しいので、コレクタ電流Ｉｃが急速に減少すると配線インダクタンス１５に流れる電流も減少する。このため、配線インダクタンス１５は（配線インダクタンス×電流減少率）で表されるサージ電圧を発生する。このサージ電圧は直流電圧源１２の電圧と同極性であるため、コレクタ−エミッタ間電圧Ｖｃｅには、直流電圧源１２に配線インダクタンス１５が発生するサージ電圧が重畳された波形となる。コレクタ電流Ｉｃがゼロに達すると、負荷１３の電流は全て環流ダイオード１４に流れるようになり、ターンオフ動作が完了する。
【０００６】
次に、制御手段８の動作について説明する。以上に示したＩＧＢＴ素子１がオンからオフに遷移するターンオフ動作の際に、制御手段８は電流検出手段７によって検出されたＩＧＢＴ素子１のコレクタ電流Ｉｃの微分値を求め、その値が基準値より小さいか否かを判別する。コレクタ電流の微分値が基準値より小さい場合、すなわちコレクタ電流の変化が緩慢な第１の期間においては、制御手段８は切り換え手段１１を短絡状態にし、抵抗９をゲート抵抗として使用する。コレクタ電流の微分値が基準値より大きい場合、すなわちコレクタ電流の変化が急峻な第２の期間においては、制御手段８は切り換え手段１１を開放状態にし、抵抗９と抵抗１０をゲート抵抗として使用する。抵抗９と抵抗１０は直列接続されるため、合成抵抗値は抵抗９の抵抗値よりも大きくなる。この動作により、コレクタ電流の変化が緩慢な第１の期間においてはゲート抵抗が小さい状態でＩＧＢＴ素子１が駆動され、コレクタ電流の変化が急峻な第２の期間においては、ゲート抵抗が大きい状態でＩＧＢＴ素子１が駆動される。
【０００７】
次に、図２９を用いて従来例におけるゲート抵抗切り換えの効果について説明する。図２９は、ＩＧＢＴ素子１のターンオフ時におけるＩＧＢＴ素子１のコレクタで電流Ｉｃと、ＩＧＢＴ素子１のコレクタ−エミッタ間電圧Ｖｃｅと、ＩＧＢＴ素子１のスイッチング損失Ｐとを図示しており、（ａ）はゲート抵抗が小さい場合すなわち切り換え手段１１を短絡状態でＩＧＢＴ素子１を駆動した場合を、（ｂ）はゲート抵抗が大きい場合すなわち切り換え手段１１を開放状態でＩＧＢＴ素子１を駆動した場合を、（ｃ）はコレクタ電流Ｉｃの変化が緩慢な第１の期間と急峻な第２の期間とでゲート抵抗を切り換えた場合をそれぞれ示している。図２９（ａ）、（ｂ）の比較より、ゲート抵抗を大きくする程ターンオフ時のコレクタ電流Ｉｃの減少率が緩やかになり、配線インダクタンスに前記コレクタ電流Ｉｃの減少率が印加されることによって生じるコレクタ−エミッタ間のサージ電圧は低減されるが、同時にコレクタ−エミッタ間電圧Ｖｃｅの増加率も緩やかになるため、コレクタ電流Ｉｃとコレクタ−エミッタ間電圧Ｖｃｅの乗算で表されるスイッチング損失Ｐの時間積算値は増加することがわかる。このように、サージ電圧の抑制は単純にゲート抵抗を大きくすることでも実現できるが、スイッチング損失が増加するという欠点がある。
【０００８】
図２８に示した従来の半導体素子の駆動回路によれば、コレクタ電流Ｉｃの変化が緩慢な第１の期間においては、図２９（ａ）のゲート抵抗が小さい状態でＩＧＢＴ素子１が駆動され、コレクタ電流Ｉｃの変化が急峻な第２の期間においては、図２９（ｂ）のゲート抵抗が大きい状態でＩＧＢＴ素子１が駆動される。この結果、図２９（ｃ）に示したように、第１の期間においては図２９（ａ）と等価な電圧、電流波形が得られ、第２の期間においては図２９（ｂ）と等価な電圧、電流波形が得られる。従って、サージ電圧に関係しない第１の期間においてはＩＧＢＴ素子１を高速で駆動してスイッチング損失Ｐの時間積分値を低減し、かつ、サージ電圧に関係する第２の期間においてはＩＧＢＴ素子１を低速で駆動してサージ電圧を抑制することができる。
【０００９】
【発明が解決しようとする課題】
従来の半導体素子の駆動回路は以上のように構成されていて、第１の期間と第２の期間とでゲート抵抗を切り換えてはいるが、第１の期間中および第２の期間中それぞれにおいては一定の直流電圧源と一定のゲート抵抗でＩＧＢＴ素子を駆動している。ＩＧＢＴ素子のゲート−エミッタ間電圧とコレクタ電流およびコレクタ−エミッタ間電圧との関係は非線形であるため、一定の直流電圧源と一定のゲート抵抗で駆動した場合には図２９に示したように第１の期間中のコレクタ−エミッタ間電圧の変化率および第２の期間中のコレクタ電流の変化率が時間と共に変化し、第１の期間と第２の期間の切換点近傍にて局所的に高い変化率が発生する。このため、サージ電圧が所望の値以下となるように電流変化率の最大値を制限した場合、電流変化率の平均値は必要以上に低減されることになり、スイッチング損失の低減が充分に果たせないという問題点があった。また、コレクタ−エミッタ間電圧の変化率およびコレクタ電流の変化率は、コレクタ電流およびコレクタ−エミッタ間電圧の値によっても変化するため、負荷や電源電圧の変動により電圧／電流動作点がゲート抵抗値を選定した電圧／電流動作点からずれるとスイッチング損失の低減およびサージ電圧の抑制が充分に機能せず、電圧／電流の動作範囲が広い用途においては効果が少ないという問題点があった。
【００１０】
この発明は上記のような問題点を解消するためになされたもので、スイッチング動作中における主電極の電流の変化率あるいは主電極間電圧の変化率を、負荷や電源電圧の変動によらず常に概ね一定に保つなど所望のスイッチング特性が得られる半導体素子の駆動回路を得ることを目的としている。
【００１１】
【課題を解決するための手段】
請求項１に係る半導体素子の駆動回路は、ゲート−エミッタ間に印加する電圧によってコレクターエミッタ間の導通状態を制御する電圧駆動形自己消弧半導体素子の前記ゲート−エミッタ間に接続される駆動回路において、前記電圧駆動形自己消弧半導体素子をオン／オフする信号を発生するスイッチング信号発生手段と、このスイッチング信号発生手段の出力により一定値の前記電圧駆動形自己消弧半導体素子のコレクタ電流変化率の指令を発生する電流変化率設定手段と、この電流変化率設定手段の出力により前記電圧駆動形自己消弧半導体素子のゲート電流の指令を発生する第１の関数発生手段と、この第１の関数発生手段の出力により前記ゲート電流を発生するゲート電流発生手段とを有し、前記第１の関数発生手段は、前記ゲート電流から前記電圧駆動形自己消弧半導体素子のコレクタ電流変化率への伝達関数の逆関数に、前記電流変化率設定手段の出力を代入して得られるゲート電流の指令を発生するようにしたものである。
【００１２】
請求項２に係る半導体素子の駆動回路は、ゲート−エミッタ間に印加する電圧によってコレクタ−エミッタ間の導通状態を制御する電圧駆動形自己消弧半導体素子の前記ゲート−エミッタ間に接続される駆動回路において、前記電圧駆動形自己消弧半導体素子をオン／オフする信号を発生するスイッチング信号発生手段と、このスイッチング信号発生手段の出力により一定値の前記電圧駆動形自己消弧半導体素子のコレクタ−エミッタ間電圧変化率の指令を発生する電圧変化率設定手段と、この電圧変化率設定手段の出力により前記電圧駆動形自己消弧半導体素子のゲート電流の指令を発生する第２の関数発生手段と、この第２の関数発生手段の出力により前記ゲート電流を発生するゲート電流発生手段とを有し、前記第２の関数発生手段は、前記ゲート電流から前記電圧駆動形自己消弧半導体素子のコレクタ−エミッタ間電圧変化率への伝達関数の逆関数に、前記電圧変化率設定手段の出力を代入して得られるゲート電流の指令を発生するようにしたものである。
【００１３】
請求項３に係る半導体素子の駆動回路は、ゲート−エミッタ間に印加する電圧によってコレクタ−エミッタ間の導通状態を制御する電圧駆動形自己消弧半導体素子の前記ゲート−エミッタ間に接続される駆動回路において、前記電圧駆動形自己消弧半導体素子をオン／オフする信号を発生するスイッチング信号発生手段と、前記スイッチング信号発生手段の出力により一定値の前記電圧駆動形自己消弧半導体素子のコレクタ電流変化率の指令を発生する電流変化率設定手段と、この電流変化率設定手段の出力により前記電圧駆動形自己消弧半導体素子のゲート電流の指令を発生する第１の関数発生手段と、前記スイッチング信号発生手段の出力により一定値の前記電圧駆動形自己消弧半導体素子のコレクタ−エミッタ間電圧変化率の指令を発生する電圧変化率設定手段と、この電圧変化率設定手段の出力により前記電圧駆動形自己消弧半導体素子のゲート電流の指令を発生する第２の関数発生手段と、前記第１、第２の関数発生手段を切り換えてそのいずれかの出力を出力する切り換え手段と、この切り換え手段の出力により前記ゲート電流を発生するゲート電流発生手段と、前記ゲート−エミッタ間の電圧制御による前記コレクタ−エミッタ間の導通状態から非導通状態への遷移時に前記コレクタ電流の変化が緩慢な第１の期間からそれに続くコレクタ電流の変化が急峻な第２の期間への変化点を検出する電流変化点検出手段とを有し、前記第１の関数発生手段は、前記ゲート電流から前記電圧駆動形自己消弧半導体素子のコレクタ電流変化率への伝達関数の逆関数に、前記電流変化率設定手段の出力を代入して得られるゲート電流の指令を発生し、前記第２の関数発生手段は、前記ゲート電流から前記電圧駆動形自己消弧半導体素子のコレクタ−エミッタ間電圧変化率への伝達関数の逆関数に、前記電圧変化率設定手段の出力を代入して得られるゲート電流の指令を発生するもので、前記切り換え手段が、前記第１の期間においては前記第２の関数発生手段へ、前記第２の期間においては前記第１の関数発生手段へ切り換えるようにしたものである。
【００１４】
請求項４に係る半導体素子の駆動回路は、ゲート−エミッタ間に印加する電圧によってコレクタ−エミッタ間の導通状態を制御する電圧駆動形自己消弧半導体素子の前記ゲート−エミッタ間に接続される駆動回路において、前記電圧駆動形自己消弧半導体素子をオン／オフする信号を発生するスイッチング信号発生手段と、前記スイッチング信号発生手段の出力により一定値の前記電圧駆動形自己消弧半導体素子のコレクタ電流変化率の指令を発生する電流変化率設定手段と、この電流変化率設定手段の出力により前記電圧駆動形自己消弧半導体素子のゲート電流の指令を発生する第１の関数発生手段と、前記スイッチング信号発生手段の出力により一定値の前記電圧駆動形自己消弧半導体素子のコレクタ−エミッタ間電圧変化率の指令を発生する電圧変化率設定手段と、この電圧変化率設定手段の出力により前記電圧駆動形自己消弧半導体素子のゲート電流の指令を発生する第２の関数発生手段と、前記第１、第２の関数発生手段を切り換えてそのいずれかの出力を出力する切り換え手段と、この切り換え手段の出力により前記ゲート電流を発生するゲート電流発生手段と、前記ゲート−エミッタ間の電圧制御による前記コレクタ−エミッタ間の非導通状態から導通状態への遷移時に前記コレクタ電流の変化が急峻な第２の期間からそれに続くコレクタ電流の変化が変化が緩慢な第１の期間への変化点を検出する電流変化点検出手段とを有し、前記第１の関数発生手段は、前記ゲート電流から前記電圧駆動形自己消弧半導体素子のコレクタ電流変化率への伝達関数の逆関数に、前記電流変化率設定手段の出力を代入して得られるゲート電流の指令を発生し、前記第２の関数発生手段は、前記ゲート電流から前記電圧駆動形自己消弧半導体素子のコレクタ−エミッタ間電圧変化率への伝達関数の逆関数に、前記電圧変化率設定手段の出力を代入して得られるゲート電流の指令を発生するもので、前記切り換え手段が、前記第２の期間においては前記第１の関数発生手段へ、前記第１の期間においては前記第２の関数発生手段へ切り換えるようにしたものである。
【００１５】
請求項５に係る半導体素子の駆動回路は、請求項１、請求項３、請求項４のいずれかにおいて、前記第１の関数発生手段で使用する前記ゲート電流から前記電圧駆動形自己消弧半導体素子のコレクタ電流変化率への伝達関数の逆関数が、前記電圧駆動形自己消弧半導体素子のゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数と、ゲート−エミッタ間の静電容量と、コレクタ電流変化率との積である。
【００１６】
請求項６に係る半導体素子の駆動回路は、請求項２、請求項３、請求項４のいずれかにおいて、前記第２の関数発生手段で使用する前記ゲート電流から前記電圧駆動形自己消弧半導体素子のコレクタ−エミッタ間電圧変化率への伝達関数の逆関数が、前記電圧駆動形自己消弧半導体素子のゲート−コレクタ間の静電容量と、コレクタ−エミッタ間電圧変化率との積である。
【００１７】
請求項７に係る半導体素子の駆動回路は、請求項１、請求項３、請求項４のいずれかにおいて、前記電圧駆動形自己消弧半導体素子のゲート−エミッタ間に接続されたコンデンサを有すると共に、前記第１の関数発生手段で使用する前記ゲート電流から前記電圧駆動形自己消弧半導体素子のコレクタ電流変化率への伝達関数の逆関数が、前記電圧駆動形自己消弧半導体素子のゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数と、ゲート−エミッタ間の静電容量および前記コンデンサの静電容量の和と、コレクタ電流変化率との積である。
【００１８】
請求項８に係る半導体素子の駆動回路は、請求項２、請求項３、請求項４のいずれかにおいて、前記電圧駆動形自己消弧半導体素子のゲート−コレクタ間に接続されたコンデンサを有すると共に、前記第２の関数発生手段で使用する前記ゲート電流から前記電圧駆動形自己消弧半導体素子のコレクタ−エミッタ間電圧変化率への伝達関数の逆関数が、前記電圧駆動形自己消弧半導体素子のゲート−コレクタ間の静電容量および前記コンデンサの静電容量の和と、コレクタ−エミッタ間電圧変化率との積である。
【００１９】
請求項９に係る半導体素子の駆動回路は、請求項３または請求項４において、前記電圧駆動形自己消弧半導体素子のゲート−エミッタ間に接続されたコンデンサを有すると共に、前記第１の関数発生手段で使用する前記ゲート電流から前記電圧駆動形自己消弧半導体素子のコレクタ電流変化率への伝達関数の逆関数を、前記電圧駆動形自己消弧半導体素子のゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数と、ゲート−エミッタ間の静電容量および前記コンデンサの静電容量の和と、コレクタ電流変化率との積とし、かつ、前記切り換え手段の切り換え動作時における第１、第２の関数発生手段の出力が互いに等しくなるように前記コンデンサの静電容量を設定したものである。
【００２０】
請求項１０に係る半導体素子の駆動回路は、請求項３または請求項４において、前記電圧駆動形自己消弧半導体素子のゲート−コレクタ間に接続されたコンデンサを有すると共に、前記第２の関数発生手段で使用する前記ゲート電流から前記電圧駆動形自己消弧半導体素子のコレクタ−エミッタ間電圧変化率への伝達関数の逆関数を、前記電圧駆動形自己消弧半導体素子のゲート−コレクタ間の静電容量および前記コンデンサの静電容量の和と、コレクタ−エミッタ間電圧変化率との積とし、かつ、前記切り換え手段の切り換え動作時における第１、第２の関数発生手段の出力が互いに等しくなるように前記コンデンサの静電容量を設定したものである。
【００２１】
請求項１１に係る半導体素子の駆動回路は、請求項５、請求項７、請求項９のいずれかにおいて、前記第１の関数発生手段で使用する前記ゲート電流から前記電圧駆動形自己消弧半導体素子のコレクタ電流変化率への伝達関数の逆関数において、前記電圧駆動形自己消弧半導体素子のゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数に代えて、前記ゲート−エミッタ間電圧の一次の近似式を使用するようにしたものである。
【００２２】
請求項１２に係る半導体素子の駆動回路は、請求項５、請求項７、請求項９のいずれかにおいて、前記第１の関数発生手段で使用する前記ゲート電流から前記電圧駆動形自己消弧半導体素子のコレクタ電流変化率への伝達関数の逆関数において、前記電圧駆動形自己消弧半導体素子のゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数に代えて、代表値である定数で近似して使用するようにしたものである。
【００２３】
請求項１３に係る半導体素子の駆動回路は、請求項６、請求項８、請求項１０のいずれかにおいて、前記第２の関数発生手段で使用する前記ゲート電流から前記電圧駆動形自己消弧半導体素子のコレクタ−エミッタ間電圧変化率への伝達関数の逆関数において、前記電圧駆動形自己消弧半導体素子のゲート−コレクタ間の静電容量に代えて、前記ゲート−コレクタ間電圧の一次の近似式を使用するようにしたものである。
【００２４】
請求項１４に係る半導体素子の駆動回路は、請求項６、請求項８、請求項１０のいずれかにおいて、前記第２の関数発生手段で使用する前記ゲート電流から前記電圧駆動形自己消弧半導体素子のコレクタ−エミッタ間電圧変化率への伝達関数の逆関数において、前記電圧駆動形自己消弧半導体素子のゲート−コレクタ間の静電容量に代えて、代表値である定数で近似して使用するようにしたものである。
【００２５】
【発明の実施の形態】
この発明の実施の形態の具体的説明に先行して、基本的動作について説明する。図２５は、制御電極間に印加する電圧によって主電極間の導通状態を制御する電圧駆動形自己消弧半導体素子であるＩＧＢＴ（絶縁ゲート形トランジスタ）素子１の駆動回路（スイッチング回路）の一例を示す回路図である。１ａ〜１ｃはＩＧＢＴ素子１の各電極であって、１ａは第１の制御電極であるゲート、１ｂは第１の主電極であるコレクタ、１ｃは第２の制御電極と第２の主電極を兼ねた電極であるエミッタである。２はＩＧＢＴ素子１をオンさせるための電圧を供給するオン用直流電圧源、３はＩＧＢＴ素子１をオフさせるための電圧を供給するオフ用直流電圧源、４はＩＧＢＴ素子１をオン／オフする信号を発生するスイッチング信号発生手段、５はスイッチング信号発生手段４の出力によりオン用直流電圧源２とオフ用直流電圧源３を切り換えてＩＧＢＴ素子１のゲート１ａに電圧を印加するオンオフ切り換え手段、９はゲート抵抗、１２は直流電圧源、１３は負荷、１４は環流ダイオード、１５は配線インダクタンスである。なお、図ではオフ用直流電圧源３を負電圧として図示しているが、ここではゼロであるものとする。また、半導体素子の材料のＰＮ極性の構成などによって両直流電圧源の正負極性が逆になることもある。
【００２６】
まず、ＩＧＢＴ素子を導通状態（オン状態）から非導通状態（オフ状態）へ遷移させるターンオフ動作について説明する。
図２６は、図２５の駆動回路におけるターンオフ時のＩＧＢＴ素子１の電圧、電流の波形図である。図２５において、スイッチング信号発生手段４がオン信号を出力すると、オンオフ切り換え手段５はオン用直流電圧源２側に切り換えられ、ＩＧＢＴ素子１のゲート１ａ−エミッタ１ｃ間にゲート抵抗９を介してオン電圧が印加され、ＩＧＢＴ素子１はオンする。ＩＧＢＴ素子１がオンすると、直流電圧源１２の電圧Ｖｄｃが負荷１３に印加され、直流電圧源１２−配線インダクタンス１５−負荷１３−ＩＧＢＴ素子１−直流電圧源１２の経路で電流が流れる。
【００２７】
この時のＩＧＢＴ素子１の電圧、電流を図２６では時刻０の点で表している。時刻０においてＩＧＢＴ素子１はオンしているため、コレクタ１ｂ−エミッタ１ｃ間電圧Ｖｃｅはほぼゼロであり、コレクタ電流Ｉｃは負荷１３の電流ＩＬと等しい。オンオフ切り換え手段５の出力Ｖｓは、オン用直流電圧源２の電圧と等しい。そして、ＩＧＢＴ素子１のゲート−エミッタ間はコンデンサであるため、オン時間が十分長ければゲート−エミッタ間電圧Ｖｇｅはオンオフ切り換え手段５の出力Ｖｓすなわちオン用直流電圧源２の電圧まで充電され、ゲート電流Ｉｇはゼロとなる。
【００２８】
次いで、時刻ｔ１においてスイッチング信号発生手段４がオフ信号を出力すると、オンオフ切り換え手段５はオフ用直流電圧源３側に切り換えられ、ＩＧＢＴ素子１のゲート１ａにゲート抵抗９を介してオフ電圧が印加される。ここでは、オフ用直流電圧源３をゼロボルトとしているため、オンオフ切り換え手段５の出力Ｖｓはゼロとなる。そして、ゲート−エミッタ間電圧Ｖｇｅはゼロに向かって（ゲート抵抗９×ＩＧＢＴ素子１のゲート−エミッタ間コンデンサ）の時定数で減少を開始する。しかし、通常はオン用直流電圧源２は負荷電流ＩＬを流すために必要とされるオン電圧Ｖｏｎに対して十分に高く設定されているため、時刻ｔ２においてゲート−エミッタ間電圧Ｖｇｅが負荷電流ＩＬを流すのに必要とされる電圧Ｖｏｎに達するまでは、ＩＧＢＴ素子１の実際のターンオフ動作は行われない。
【００２９】
時刻ｔ２において、ゲート−エミッタ間電圧Ｖｇｅが負荷電流ＩＬを流すのに必要とされる電圧Ｖｏｎに達すると、ＩＧＢＴ素子１はターンオフ動作を開始し、コレクタ−エミッタ間電圧Ｖｃｅが増加する。この時、負荷１３に印加される電圧は直流電圧源１２の電圧Ｖｄｃからコレクタ−エミッタ間電圧Ｖｃｅを減算した電圧となるが、負荷１３の持つインダクタンスの作用で流れる電流ＩＬは急変しない。このため、コレクタ電流Ｉｃも急変せず、コレクタ電流Ｉｃは電流の変化が緩慢な第１の期間を形成する。この第１の期間においては、コレクタ電流Ｉｃを変化させないようにゲート−エミッタ間電圧Ｖｇｅを負荷電流ＩＬを流すのに必要とされる電圧Ｖｏｎに保とうとするフィードバック作用が働き、ゲート−エミッタ間電圧Ｖｇｅはほぼ一定の電圧となる。このフィードバック作用は、コレクタ−エミッタ間電圧Ｖｃｅの増加によりＩＧＢＴ素子１のコレクタ−ゲート間コンデンサを介してコレクタからゲートに変位電流が流れ、この変位電流がゲート抵抗９で電圧降下を生じることにより実現される。
【００３０】
時刻ｔ３において、コレクタ−エミッタ間電圧Ｖｃｅが直流電圧源１２の電圧Ｖｄｃに達すると環流ダイオード１４がオンして、負荷１３を流れる電流の一部が環流ダイオード１４に転流を開始する。環流ダイオード１４に転流した電流Ｉｄの分だけコレクタ電流Ｉｃは減少するため、コレクタ電流Ｉｃは電流の変化が急峻な第２の期間を形成する。この第２の期間においては、上記のゲート−エミッタ間電圧Ｖｇｅを一定に保とうとするフィードバック作用は失われ、ゲート−エミッタ間電圧Ｖｇｅは再びゼロに向かって減少を開始する。また、コレクタ電流Ｉｃは配線インダクタンス１５に流れる電流と等しいので、この第２の期間において配線インダクタンス１５に流れる電流も減少する。このため、配線インダクタンス１５は（配線インダクタンス×電流減少率）で表される電圧を発生する。この電圧は直流電圧源１２の電圧と同極性であるため、コレクタ−エミッタ間電圧Ｖｃｅには、直流電圧源１２に配線インダクタンス１５が発生するサージ電圧が重畳された波形となる。
【００３１】
時刻ｔ４において、ゲート−エミッタ間電圧ＶｇｅがＩＧＢＴ素子１のゲート閾電圧Ｖｔｈに達するとコレクタ電流Ｉｃはゼロとなり、ＩＧＢＴ素子１はターンオフ動作を完了する。時刻ｔ４において実際のターンオフ動作は完了しているが、ゲート−エミッタ間電圧Ｖｇｅはまだゼロに達していないため、引き続きゼロに向かって減少を続ける。時刻ｔ５においてゲート−エミッタ間電圧Ｖｇｅがゼロに達して、全ての動作を完了する。以上のＩＧＢＴ素子１のターンオフ動作において、コレクタ電流Ｉｃは変化が緩慢な第１の期間（ｔ２からｔ３）と、それに引き続いて発生するコレクタ電流Ｉｃの変化が急峻な第２の期間（ｔ３からｔ４）を形成する。なお、第１の期間および第２の期間は、これらが生じる時間的順序を意味するものではなく、後述のターンオンの場合は発生順序が逆になる。
【００３２】
次に、ＩＧＢＴ素子をオフ状態からオン状態に遷移させるターンオン動作について説明する。図２７は、図２５のスイッチング回路におけるＩＧＢＴ素子１のターンオン時の電圧、電流の波形図である。図２５において、スイッチング信号発生手段４がオフ信号を出力すると、オンオフ切り換え手段５はオフ用直流電圧源３側に切り換えられ、ＩＧＢＴ素子１のゲート１ａ−エミッタ１ｃ間にゲート抵抗９を介してオフ電圧が印加され、ＩＧＢＴ素子１はオフする。ＩＧＢＴ素子１がオフすると、ＩＧＢＴ素子１のコレクタ１ｂを流れるコレクタ電流Ｉｃはほぼゼロとなる。それまでＩＧＢＴ素子１に流れていた電流は環流ダイオード１４に転流し、環流ダイオード１４−負荷１３−環流ダイオード１４の経路で電流が流れる。この時、環流ダイオード１４の端子間電圧はほぼゼロとなるため、直流電圧源１２の電圧ＶｄｃがＩＧＢＴ素子１に印加される。
【００３３】
この時のＩＧＢＴ素子１の電圧、電流を図２７では時刻０の点で表している。時刻０においてＩＧＢＴ素子１はオフしているため、コレクタ電流Ｉｃはほぼゼロであり、コレクタ−エミッタ間電圧Ｖｃｅは直流電圧源１２の電圧Ｖｄｃと等しい。オンオフ切り換え手段５の出力Ｖｓは、オフ用直流電圧源３の電圧と等しい。ここでは、オフ用直流電圧源３をゼロボルトとしているため、オンオフ切り換え手段５の出力Ｖｓはゼロとなる。そして、ＩＧＢＴ素子１のゲート−エミッタ間はコンデンサであるため、オフ時間が十分長ければゲート−エミッタ間電圧Ｖｇｅはオンオフ切り換え手段５の出力Ｖｓすなわちゼロとなり、ゲート電流Ｉｇもゼロとなる。
【００３４】
次いで、時刻ｔ６においてスイッチング信号発生手段４がオン信号を出力すると、オンオフ切り換え手段５はオン用直流電圧源２に切り換えられ、ＩＧＢＴ素子１のゲート１ａにゲート抵抗９を介してオン電圧が印加される。そして、ゲート−エミッタ間電圧Ｖｇｅはオン用直流電圧源２の電圧に向かって（ゲート抵抗９の抵抗値×ＩＧＢＴ素子１のゲート−エミッタ間コンデンサの静電容量）の時定数で増加を開始する。しかし、ゲート−エミッタ間電圧ＶｇｅがＩＧＢＴ素子１のゲート閾電圧Ｖｔｈに達するまではコレクタ電流は流れないため、ＩＧＢＴ素子１の実際のターンオン動作は行われない。
【００３５】
時刻ｔ７において、ゲート−エミッタ間電圧Ｖｇｅがゲート閾電圧Ｖｔｈに達すると、ＩＧＢＴ素子１はターンオン動作を開始してコレクタ電流Ｉｃが増加し、コレクタ電流Ｉｃは電流の変化が急峻な第２の期間を形成する。この第２の期間中、負荷１３の持つインダクタンスのため流れる電流ＩＬは急変しない。このため、環流ダイオード１４の電流Ｉｄはほぼ一定の負荷電流ＩＬからコレクタ電流Ｉｃを減算した値となる。環流ダイオード電流Ｉｄが正の期間は環流ダイオード１４がオンしているため、直流電圧源１２の電圧はＩＧＢＴ素子１と配線インダクタンス１５に印加される。この第２の期間において、配線インダクタンス１５に流れる電流はコレクタ電流Ｉｃと等しいので、配線インダクタンス１５に流れる電流もコレクタ電流Ｉｃと共に増加する。このため、配線インダクタンス１５は（配線インダクタンス×電流増加率）で表される電圧を発生する。この電圧は直流電圧源１２の電圧と逆極性であるため、コレクタ−エミッタ間電圧Ｖｃｅは、直流電圧源１２の電圧Ｖｄｃから配線インダクタンス１５が発生する電圧を減算した波形となる。このターンオン動作中の第２の期間は本質的に前述のターンオフ動作中の第２の期間と同様のものである。
【００３６】
時刻ｔ８において、環流ダイオード１４の電流Ｉｄがゼロに達した後もしばらくの間は環流ダイオード１４はオンを維持する。この動作は、ダイオードのリカバリ動作として知られている。このリカバリ動作中は、環流ダイオード１４に負の電流が流れるため、コレクタ電流Ｉｃは負荷１３の電流ＩＬに環流ダイオード１４のリカバリ電流を加算した電流となる。
【００３７】
時刻ｔ９において環流ダイオード１４のリカバリ動作が完了すると環流ダイオード１４はオフとなり、直流電圧源１２−配線インダクタンス１５−負荷１３−ＩＧＢＴ素子１−直流電圧源１２の経路で電流が流れるようになる。負荷１３の持つインダクタンスのためその電流ＩＬは急変しないので、同一電流経路にあるコレクタ電流Ｉｃも急変せず、コレクタ電流Ｉｃは電流の変化が緩慢な第１の期間を形成する。この第１の期間においては、コレクタ電流Ｉｃを変化させないようにゲート−エミッタ間電圧Ｖｇｅを負荷電流ＩＬを流すのに必要とされる電圧Ｖｏｎに保とうとするフィードバック作用が働き、ゲート−エミッタ間電圧Ｖｇｅはほぼ一定の電圧となる。このフィードバック作用は、コレクタ−エミッタ間電圧Ｖｃｅの減少によりＩＧＢＴ素子１のコレクタ−ゲート間コンデンサを介してゲートからコレクタに変位電流が流れ、この変位電流がゲート抵抗９で電圧降下を生じることにより実現される。このターンオン動作中の第１の期間は本質的に前述のターンオフ動作中の第１の期間と同様のものである。
【００３８】
時刻ｔ１０において、コレクタ−エミッタ間電圧Ｖｃｅがゼロボルトに達すると、以後コレクタ−エミッタ間電圧Ｖｃｅがゼロボルト一定となり変位電流を生じないため、上記のフィードバック作用は失われゲート−エミッタ間電圧Ｖｇｅは再びオン用直流電圧源２の電圧に向かって増加を開始する。時刻ｔ１０において実際のターンオン動作は完了しているが、ゲート−エミッタ間電圧Ｖｇｅがまだオン用直流電圧源２の電圧に達していないため、引き続きゲート−エミッタ間電圧Ｖｇｅは増加を続ける。時刻ｔ１１においてゲート−エミッタ間電圧Ｖｇｅがオン用直流電圧源２の電圧に達して、全ての動作を完了する。以上のＩＧＢＴ素子１のターンオン動作において、コレクタ電流Ｉｃは変化が急峻な第２の期間（ｔ７からｔ８）と、環流ダイオードのリカバリ動作を間に挟んで、引き続いて発生するコレクタ電流Ｉｃの変化が緩慢な第１の期間（ｔ９からｔ１０）を形成する。
【００３９】
次に、この発明の実施の形態を図１について説明する。図１は、この発明の実施の形態における半導体素子の駆動回路の一つの基本的な構成を示すブロック図であり、図において、４はＩＧＢＴ素子１をオン／オフする信号を発生するスイッチング信号発生手段、１６はスイッチング信号発生手段４の出力によりＩＧＢＴ素子１のスイッチング特性に関する指令を発生するスイッチング特性設定手段である。１７はスイッチング特性設定手段１６の出力によりＩＧＢＴ素子１のゲート１ａへ供給する電流の指令を発生する関数発生手段であり、ＩＧＢＴ素子１のゲート電流からスイッチング特性設定手段１６で設定したスイッチング特性への伝達関数の逆関数を発生する。１８はＩＧＢＴ素子１のゲート１ａに関数発生手段１７の出力に応じた電流を供給する電流発生手段としてのゲート電流発生手段である。以上のスイッチング信号発生手段４、スイッチング特性設定手段１６、関数発生手段１７およびゲート電流発生手段１８で、ＩＧＢＴ素子１の駆動回路を構成している。その他については図２５と同様であるので説明を省略する。
【００４０】
図１、図２６により動作について説明する。ここではスイッチング特性設定手段１６が発生するスイッチング特性に関する指令として、ターンオフ時のコレクタ電流の変化率を設定する場合を取り上げる。ＩＧＢＴ素子１のターンオフの際にコレクタ電流が変化するのは図２６の第２の期間である。この第２の期間において、ゲート−エミッタ間電圧Ｖｇｅが減少し、コレクタ電流Ｉｃも減少する。この第２の期間において、コレクタ電流Ｉｃとゲート−エミッタ間電圧Ｖｇｅとは関数関係にあり（１）式が成り立つ。（１）式でｆはゲート−エミッタ間電圧Ｖｇｅからコレクタ電流Ｉｃへの伝達関数である。
Ｉｃ＝ｆ（Ｖｇｅ）・・・（１）
【００４１】
また、ＩＧＢＴ素子１のゲート−エミッタ間はコンデンサであるため、ゲート−エミッタ間静電容量をＣｇｅとすると、ゲート電流Ｉｇとゲート−エミッタ間電圧Ｖｇｅの間には（２）式が成り立つ。正確には、ゲート電流Ｉｇはゲート−エミッタ間静電容量Ｃｇｅだけではなくゲート−コレクタ間静電容量Ｃｇｃにも流れるが、第２の期間においてはコレクタ−エミッタ間電圧Ｖｃｅすなわちゲート−コレクタ間電圧Ｖｇｃが高いため、電圧依存性が強いゲート−コレクタ間静電容量Ｃｇｃはゲート−エミッタ間静電容量Ｃｇｅに対して充分に小さな値となり無視できる。
Ｉｇ＝Ｃｇｅ・（ｄＶｇｅ／ｄｔ）・・・（２）
（１）、（２）式より、ＩＧＢＴ素子１のゲート電流Ｉｇからコレクタ電流変化率ｄＩｃ／ｄｔへの伝達関数は（３）式となる。さらに、（３）式の逆関数を求めると（４）式となる。（４）式において、ｄＩｃ／ｄｔを一定値Ａとした場合、（５）式となる。
【００４２】
【数１】

【００４３】
逆関数の定義から明らかなように、ＩＧＢＴ素子１のゲート電流Ｉｇからコレクタ電流変化率ｄＩｃ／ｄｔへの伝達関数（３）式の逆関数（５）式から得られたゲート電流Ｉｇを、（３）式の伝達関数を有するＩＧＢＴ素子１に流すと、コレクタ電流変化率ｄＩｃ／ｄｔは一定（ｄＩｃ／ｄｔ＝Ａ）となる。従って、図１において、スイッチング信号発生手段４の出力により、スイッチング特性設定手段１６が一定値Ａの指令を発生し、関数発生手段１７が（５）式に従って、ゲート電流Ｉｇからコレクタ電流変化率ｄＩｃ／ｄｔへの伝達関数の逆関数からゲート電流指令を発生し、ゲート電流発生手段１８が（３）式の伝達関数を有するＩＧＢＴ素子１に関数発生手段１７の出力通りの電流を流せば、コレクタ電流変化率を一定に保つことができる。
【００４４】
以上はターンオフの際の動作のみについて述べたが、ターンオンに関してもＩＧＢＴ素子１各部の電圧、電流の変化率の極性が逆になるだけであるため、同様の効果が得られることは明らかである。また、以上はＩＧＢＴ素子１のコレクタ電流変化率を一定に保つことについて述べたが、必要とあればスイッチング特性設定手段１６の出力を変化させることで、コレクタ電流変化率を所望の関数に従わせることが可能である。さらに、以上は例としてコレクタ電流変化率のみについて述べたが、逆関数の定義から、コレクタ−エミッタ間電圧など、その他のスイッチング特性についても同様の効果が得られることは明らかである。
実施の形態における具体的構成について、以下に説明する。
【００４５】
実施の形態１．
図２は、この発明の実施の形態１における半導体素子の駆動回路を示すブロック図であり、スイッチング動作中におけるコレクタ電流の変化率を概ね一定に保つ場合の構成を示す。図２において、１９はスイッチング特性設定手段としての電流変化率設定手段、２０は第１の関数発生手段、１８はＩＧＢＴ素子１のゲート１ａへ第１の関数発生手段２０の出力に応じた電流を供給する電流発生手段としてのゲート電流発生手段である。その他については図１の場合と同様であるため説明は省略する。図３は、図２の第１の関数発生手段２０とＩＧＢＴ素子１の内部の機能構成を含めて示したブロック図である。図３において、第１の関数発生手段２０は２０ａと２０ｂから構成され、２０ａはＩＧＢＴ素子１のゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数、２０ｂはＩＧＢＴ素子１のゲート−エミッタ間静電容量である。１ｄ、１ｅ、１ｆ、１ｇはＩＧＢＴ素子１の特性を表す手段であり、１ｄ、１ｅは（２）式を表す手段で、それぞれＩＧＢＴ素子１のゲート−エミッタ間静電容量の逆数と積分手段である。１ｆは（１）式を表す伝達関数、１ｇは微分手段である。
【００４６】
次に、図３および図２６により動作について説明する。図２６は前述したように、ＩＧＢＴ素子１のターンオフ時の電圧、電流を示す波形図である。図３において、スイッチング信号発生手段４の出力により、電流変化率設定手段１９が指令ｄＩｃ／ｄｔ^＊を発生した場合、第１の関数発生手段２０の出力Ｉｇ^＊は、ＩＧＢＴ素子１のゲート−エミッタ間電圧からコレクタ電流への伝達関数ｆの微分の逆数２０ａと、ＩＧＢＴ素子１のゲート−エミッタ間静電容量２０ｂと、電流変化率設定手段１９の出力の積となり、（６）式で表される。この出力Ｉｇ^＊は、ゲート電流指令であり、（４）式と同じとなる。
【００４７】
【数２】

【００４８】
ゲート電流発生手段１８はＩＧＢＴ素子１に第１の関数発生手段２０の出力通りの電流を流すため、ＩＧＢＴ素子１のゲート電流Ｉｇは次の（７）式となる。
Ｉｇ＝Ｉｇ^＊・・・（７）
ＩＧＢＴ素子１のターンオフの際にコレクタ電流Ｉｃが変化する図２６の第２の期間において、ゲート−エミッタ間電圧Ｖｇｅとゲート電流Ｉｇとは（２）式の関係にあるため、ＩＧＢＴ素子１のゲート−エミッタ間電圧Ｖｇｅは、ゲート−エミッタ間静電容量の逆数１ｄ、積分手段１ｅより（８）式で与えられる。
【００４９】
【数３】

【００５０】
また、コレクタ電流Ｉｃとゲート−エミッタ間電圧Ｖｇｅとは（１）式の関係にあるため、ＩＧＢＴ素子１のコレクタ電流Ｉｃは、ＩＧＢＴ素子１のゲート−エミッタ間電圧からコレクタ電流への伝達関数１ｆにより（９）式で与えられる。そして、コレクタ電流Ｉｃを微分手段１ｇで微分したのが、コレクタ電流変化率ｄＩｃ／ｄｔであり、（１０）式のように変形できる。
【００５１】
【数４】

【００５２】
従って、図３の構成において、電流変化率設定手段１９の出力ｄＩｃ／ｄｔ^＊を一定に保てば、ＩＧＢＴ素子１のコレクタ電流変化率を一定に保つことができる。
【００５３】
図４は図３のブロック図を、ゲート−エミッタ間電圧からコレクタ電流への伝達関数が下記の（１１）式で表される場合について、書き換えた図である。図４において、２０ｃはＩＧＢＴ素子１のゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数、２０ｄは２０ｃの出力と電流変化率設定手段１９の出力の積を得るための乗算手段、２０ｅは２０ｃで使用するゲート−エミッタ間電圧Ｖｇｅを得るための検出器である。その他については図３と同じである。
【００５４】
次に、図４により動作について詳しく説明する。ＩＧＢＴ素子１のターンオフの際にコレクタ電流が変化する図２６の第２の期間において、コレクタ電流Ｉｃとゲート−エミッタ間電圧Ｖｇｅとは概ね次の（１１）式の関数関係にある。ここで、Ｋｐはトランスコンダクタンス、Ｖｔｈはゲート閾電圧である。
Ｉｃ＝Ｋｐ・（Ｖｇｅ−Ｖｔｈ）^２・・・（１１）
また（２）式と同様に、ゲート−エミッタ間電圧Ｖｇｅとゲート電流Ｉｇとは（１２）式の関数関係にある。
Ｉｇ＝Ｃｇｅ・（ｄＶｇｅ／ｄｔ）・・・（１２）
（１１）、（１２）式より、ＩＧＢＴ素子１のゲート電極Ｉｇからコレクタ電流変化率ｄＩｃ／ｄｔへの伝達関数は（１３）式となる。そして（１３）式の逆関数を求めると、（１４）式となる。
【００５５】
【数５】

【００５６】
（１４）式は（４）式と同様に、ＩＧＢＴ素子１のゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数２０ｃと、ＩＧＢＴ素子１のゲート−エミッタ間静電容量２０ｂと、電流変化率設定手段１９の出力の積となっており、（４）式から図３の第１の関数発生手段２０の構成を得たのと同様に、（１４）式から図４の第１の関数発生手段２０の構成が得られる。
【００５７】
以上、ターンオフの際の動作のみについて述べたが、ターンオンに関してもＩＧＢＴ素子各部の電圧、電流の変化率の極性が逆になるだけであるため、同様の効果が得られることは明らかである。この方法は、コレクタ電流変化率以外のＩＧＢＴ素子のスイッチング特性に制約が無い場合には最も簡便な構成を提供する。なお、（１）〜（１４）式は、図２６における第２の期間のようにコレクタ電流が急峻に変化する期間に適用したが、他の期間においても（１）〜（１４）式を用いて得られるゲート電流でＩＧＢＴ素子を駆動することは可能である。
【００５８】
実施の形態２．
図５は、この発明の実施の形態２における半導体素子の駆動回路を示すブロック図であり、スイッチング動作中におけるコレクタ−エミッタ間電圧の変化率を概ね一定に保つ場合の構成を示す。図５において、２１はスイッチング特性設定手段としての電圧変化率設定手段、２２は第２の関数発生手段、１８はＩＧＢＴ素子１のゲート１ａへ第２の関数発生手段２２の出力に応じた電流を供給する電流発生手段としてのゲート電流発生手段である。その他については図１の場合と同様であるため説明は省略する。図６は、図５の第２の関数発生手段２２とＩＧＢＴ素子１の内部の機能構成を含めて示したブロック図である。図６において、第２の関数発生手段２２はＩＧＢＴ素子１のゲート−コレクタ間静電容量である。１ｈ、１ｉ、１ｊ、１ｋ、１ｍ、１ｎはＩＧＢＴ素子１の特性を表す手段であり、１ｈはゲート−コレクタ間静電容量（ｇは関数を表す）、１ｉは逆数手段、１ｊは乗算手段、１ｋは積分手段、１ｍは加算手段、１ｎは微分手段である。
【００５９】
次に、図６および図２６により動作について説明する。図２６はＩＧＢＴ素子１のターンオフ時の電圧、電流を示す波形図である。ＩＧＢＴ素子１のターンオフの際にコレクタ−エミッタ間電圧が大きく変化するのは図２６の第１の期間である。この第１の期間においては、コレクタ電流１ｃを変化させないようにゲート−エミッタ間電圧Ｖｇｅを負荷電流ＩＬを流すのに必要とされる電圧Ｖｏｎに保とうとするフィードバック作用が働き、ゲート−エミッタ間電圧Ｖｇｅはほぼ一定の電圧となる。このフィードバック作用は、コレクタ−エミッタ間電圧Ｖｃｅの増加によりＩＧＢＴ素子１のコレクタ−ゲート間コンデンサを介してコレクタからゲートに変位電流が流れ、この変位電流がゲート駆動回路で電圧降下を生じることにより実現される。ｄＶｇｅ／ｄｔはほぼゼロでありゲート−エミッタ間の電流は無視できるので、この変位電流はゲート電流に他ならず、次の（１５）式が成り立つ。
Ｉｇ＝Ｃｇｃ・ｄＶｇｃ／ｄｔ・・・（１５）
【００６０】
また、ＩＧＢＴ素子１のゲート、コレクタ、エミッタの３端子間電圧には次の（１６）式が成り立つ。
Ｖｃｅ＝Ｖｇｃ＋Ｖｇｅ・・・（１６）
（１５）、（１６）式より、ＩＧＢＴ素子１のゲート電流Ｉｇからコレクタ−エミッタ間電圧変化率ｄＶｃｅ／ｄｔへの伝達関数およびその逆関数はそれぞれ（１７）式、（１８）式となる。
ｄＶｃｅ／ｄｔ＝Ｉｇ／Ｃｇｃ＋ｄＶｇｅ／ｄｔ・・・（１７）
Ｉｇ＝Ｃｇｃ・（ｄＶｃｅ／ｄｔ−ｄＶｇｅ／ｄｔ）・・・（１８）
【００６１】
上述のように、この第１の期間においてはゲート−エミッタ間電圧Ｖｇｅはほぼ一定の電圧となるため、ゲート−エミッタ間電圧の変化率ｄＶｇｅ／ｄｔは充分に小さな値となり無視できる。よって、（１９）式、（２０）式が得られる。
ｄＶｃｅ／ｄｔ＝Ｉｇ／Ｃｇｃ・・・（１９）
Ｉｇ＝Ｃｇｃ・ｄＶｃｅ／ｄｔ・・・（２０）
この（２０）式を発生する手段が、第２の関数発生手段２２である。図６において、スイッチング信号発生手段４の出力により、電圧変化率設定手段２１が指令ｄＶｃｅ／ｄｔ^＊を発生した場合、第２の関数発生手段２２の出力Ｉｇ^＊は（２１）式となる。この出力Ｉｇ^＊は、ゲート電流指令である。
Ｉｇ^＊＝Ｃｇｃ・ｄＶｃｅ／ｄｔ^＊・・・（２１）
【００６２】
ゲート電流発生手段１８はＩＧＢＴ素子１に第２の関数発生手段２２の出力通りの電流を流すため、ＩＧＢＴ素子１のゲート電流Ｉｇは（２２）式となる。
Ｉｇ＝Ｉｇ^＊・・・（２２）
上述のように、ＩＧＢＴ素子１のターンオフの際にコレクタ−エミッタ間電圧Ｖｃｅが大きく変化する図２６の第１の期間において、ゲート−コレクタ間電圧Ｖｇｃとゲート電流Ｉｇとは（１５）式の関係にあるため、ＩＧＢＴ素子１のゲート−コレクタ間電圧Ｖｃｅは、ゲート−コレクタ間静電容量１ｈ、逆数手段１ｉ、乗算手段１ｊ、積分手段１ｋより（２３）式で与えられる。
【００６３】
【数６】

【００６４】
また、ゲート−コレクタ間電圧Ｖｇｃとコレクタ−エミッタ間電圧Ｖｃｅとは（１６）式の関係にあるため、コレクタ−エミッタ間電圧Ｖｃｅは加算手段１ｍにより（２４）式で与えられる。そして、コレクタ−エミッタ間電圧Ｖｃｅを微分手段１ｎで微分したのが、コレクタ−エミッタ間電圧変化率ｄＶｃｅ／ｄｔであり、上述のように、この第１の期間においてはゲート−エミッタ間電圧Ｖｇｅはほぼ一定の電圧となるため（２５）式の様に変形できる。
【００６５】
【数７】

【００６６】
従って、図６の構成において、電圧変化率設定手段２１の出力ｄＶｃｅ／ｄｔ^＊を一定に保てば、ＩＧＢＴ素子１のコレクタ−エミッタ間電圧変化率を一定に保つことができる。
【００６７】
図７は図６のブロックを、ゲート−コレクタ間静電容量の電圧特性が下記の（２６）式で表される場合について、書き換えた図である。図７において、２２ａはＩＧＢＴ素子１のゲート−コレクタ間静電容量、２２ｂは２２ａの出力と電圧変化率設定手段２１の出力の積を得るための乗算手段、２２ｃは２２ａで使用するゲート−コレクタ間電圧Ｖｇｃを得るための検出器である。その他については図６と同じである。次に、図７により動作について説明する。ＩＧＢＴ素子１のターンオフの際にコレクタ−エミッタ間電圧が大きく変化する図２６の第１の期間において、ゲート−コレクタ間静電容量Ｃｇｃは概ね（２６）式の電圧依存性を持つ。ここで、Ｃｊｏはゼロバイアス容量、Ｖｊは接合電位差、Ｍは接合傾斜係数である。（２６）式の電位依存性を（１５）式に適用した場合にも（１５）〜（２５）式の関係は成り立つ。
【００６８】
【数８】

【００６９】
以上、ターンオフの際の動作のみについて述べたが、ターンオンに関してもＩＧＢＴ素子各部の電圧、電流の変化率の極性が逆になるだけであるため、同様の効果が得られることは明らかである。この方法は、コレクタ−エミッタ間電圧変化率以外のＩＧＢＴ素子のスイッチング特性に制約が無い場合には最も簡便な構成を提供する。なお、（１５）〜（２６）式は、図２６における第１の期間のようにコレクタ電流がほぼ一定でコレクタ−エミッタ間電圧が大きく変化する期間に適用したが、他の期間においても（１５）〜（２６）式を用いて得られるゲート電流でＩＧＢＴ素子を駆動することは可能である。
【００７０】
実施の形態３．
この実施の形態は、ターンオフのスイッチング動作中において、コレクタ−エミッタ間電圧が大きく変化する第１の期間ではコレクタ−エミッタ間電圧の変化率を概ね一定に保つとともに、コレクタ電流Ｉｃの変化が急峻な第２の期間ではコレクタ電流の変化率を概ね一定に保つものである。
図８はこの発明の実施の形態３における半導体素子の駆動回路を示すブロック図であり、図において、２３は第１の関数発生手段２０の出力と第２の関数発生手段２２の出力とを切り換えてゲート電流発生手段１８に供給する切り換え手段、２４はＩＧＢＴ素子１をオン状態からオフ状態に遷移させる際にＩＧＢＴ素子１に流れる電流の変化が緩慢な第１の期間からそれに続く電流の変化が急峻な第２の期間に遷移する時の電流の変化点を検出する電流変化点検出手段、１８は電流発生手段としてのゲート電流発生手段であり、切り換え手段２３により切り換えられて、第１または第２の関数発生手段２０、２１の出力に応じた電流を、ＩＧＢＴ素子１のゲート１ａへ供給する。その他については実施の形態１または実施の形態２と同様であるため説明は省略する。
【００７１】
次に、動作について説明する。切り換え手段２３を、第１の関数発生手段２０の側にした場合の動作については実施の形態１と同じであるため説明を省略する。また、第２の関数発生手段２２の側にした場合の動作については実施の形態２と同じであるため説明を省略する。ここでは、切り換え手段２３と電流変化点検出手段２４について説明する。ＩＧＢＴ素子１のターンオフ波形を示した図２６において、ＩＧＢＴ素子１のコレクタ−エミッタ間電圧が大きく変化するのは図２６の第１の期間であり、この期間にはコレクタ電流は変化しない。コレクタ電流が変化するのは図２６の第２の期間であり、この期間にはコレクタ−エミッタ間電圧が大きくは変化しない。このように、実施の形態１が効果を発揮する第２の期間と、実施の形態２が効果を発揮する第１の期間とは、図２６の時刻ｔ３で切り分けられる。
【００７２】
このため、電流変化点検出手段２４が第１の期間と第２の期間の境目にあたる時刻ｔ３を検出し、時刻ｔ３以前の第１の期間においては第２の関数発生手段２２の出力を使用するように切り換え手段２３を制御し、時刻ｔ３以後の第２の期間においては第１の関数発生手段２０の出力を使用するように切り換え手段２３を制御すれば、それぞれの期間において実施の形態２および実施の形態１と同じ効果が得られる。よって、図８の構成によれば、ＩＧＢＴ素子１のターンオフ中の第１、第２の期間においてそれぞれコレクタ−エミッタ間電圧の変化率あるいはコレクタ電流の変化率を概ね一定に保つことができる。
【００７３】
図９〜図１６は、電流変化点検出手段２４の第１〜第７の構成例を示すブロック図である。まず、図９を用いて、電流変化点検出手段２４の第１の構成例について説明する。２４ａはＩＧＢＴ素子１のコレクタ−エミッタ間電圧Ｖｃｅを検出する第１の電圧検出手段、２４ｂは直流電圧源１２の電圧Ｖｄｃを検出する第２の電圧検出手段、２４ｃは比較手段である。図２６より、第１の期間と第２の期間の境目である時刻ｔ３において、コレクタ−エミッタ間電圧Ｖｃｅと直流電圧源１２の電圧Ｖｄｃは交差するため、比較手段２４ｃにより、第１の電圧検出手段２４ａと第２の電圧検出手段２４ｂの出力を大小比較することで時刻ｔ３を検出することができる。直流電圧源１２の電圧が一定の用途においては、第２の電圧検出手段２４ｂを比較の基準となる別の直流電圧源に置き換えても同じ効果が得られる。
【００７４】
次に、図１０を用いて、電流変化点検出手段２４の第２の構成例について説明する。２４ａはＩＧＢＴ素子１のゲート−エミッタ間電圧Ｖｇｅを検出する電圧検出手段、２４ｃは比較手段、２４ｄは比較の基準となる直流電圧源である。図２６より、第１の期間と第２の期間の境目である時刻ｔ３において、ゲート−エミッタ間電圧Ｖｇｅは、負荷電流ＩＬを流すのに必要とされるゲート電圧Ｖｏｎから低下を始めるため、直流電圧源２４ｄの電圧をＶｏｎより若干低く設定しておくことで、比較手段２４ｃにより時刻ｔ３を検出することができる。
【００７５】
次に、図１１を用いて、電流変化点検出手段２４の第３の構成例について説明する。２４ａはＩＧＢＴ素子１のゲート−エミッタ間電圧Ｖｇｅを検出する電圧検出手段、２４ｅは電圧検出手段２４ａの出力を微分する微分手段、２４ｃは比較手段、２４ｄはゲート−エミッタ間電圧Ｖｇｅの電圧変化率の比較の基準となる直流電圧源である。図２６より、第１の期間と第２の期間の境目である時刻ｔ３において、ゲート−エミッタ間電圧Ｖｇｅは、負荷電流ＩＬを流すのに必要とされるゲート電圧Ｖｏｎから低下を始めるため、ゲート−エミッタ間電圧Ｖｇｅを微分してその変化率を求め、直流電圧源２４ｄによる基準値と大小比較することで時刻ｔ３を検出することができる。
【００７６】
次に、図１２を用いて、電流変化点検出手段２４の第４の構成例について説明する。２４ｆはＩＧＢＴ素子１のコレクタ電流Ｉｃを検出する第１の電流検出手段、２４ｇは負荷１３の電流ＩＬを検出する第２の電流検出手段、２４cは比較手段である。図２６より、第１の期間と第２の期間の境目である時刻ｔ３において、コレクタ電流Ｉｃは負荷１３の電流ＩＬから低下を始めるため、比較手段２４ｃにより、第１の電流検出手段２４ｆと第２の電圧検出手段２４ｇの出力を比較することで時刻ｔ３を検出することができる。
【００７７】
次に、図１３を用いて、電流変化点検出手段２４の第５の構成例について説明する。２４ｆはＩＧＢＴ素子１のコレクタ電流Ｉｃを検出する電流検出手段、２４ｃは比較手段、２４ｈは比較の基準となる直流電圧源である。図２６より、第１の期間と第２の期間の境目である時刻ｔ３において、コレクタ電流Ｉｃは負荷１３の電流ＩＬから低下を始めるため、直流電圧源２４ｈの電圧を負荷電流ＩＬより若干低く設定しておくことで、比較手段２４ｃにより時刻ｔ３を検出することができる。
【００７８】
次に、図１４を用いて、電流変化点検出手段２４の第６の構成例について説明する。２４ｆはＩＧＢＴ素子１のコレクタ電流Ｉｃを検出する電流検出手段、２４ｅは電流検出手段２４ｆの出力を微分する微分手段、２４ｃは比較手段、２４ｈはコレクタ電流Ｉｃの電流変化率の比較の基準となる直流電圧源である。図２６より、第１の期間と第２の期間の境目である時刻ｔ３において、コレクタ電流Ｉｃは、負荷電流ＩＬから低下を始めるため、コレクタ電流Ｉｃを微分してその変化率を求め、直流電圧源２４ｈによる基準値と大小比較することで時刻ｔ３を検出することができる。
【００７９】
次に、図１５を用いて、電流変化点検出手段２４の第７の構成例について説明する。２４ｆはダイオード１４の電流Ｉｄを検出する電流検出手段、２４ｃは比較手段、２４ｈは比較の基準となる直流電圧源である。図２６より、第１の期間と第２の期間の境目である時刻ｔ３において、ダイオード１４の電流Ｉｄはゼロから増加を始めるため、直流電圧源２４ｈの電圧をゼロより若干高く設定しておくことで、比較手段２４ｃにより時刻ｔ３を検出することができる。
【００８０】
次に、図１６を用いて、電流変化点検出手段２４の第８の構成例について説明する。２４ｆはダイオード１４の電流Ｉｄを検出する電流検出手段、２４ｅは電流検出手段２４ｆの出力を微分する微分手段、２４ｃは比較手段、２４ｈはダイイオード１４の電流Ｉｄの電流変化率の比較の基準となる直流電圧源である。図２６より、第１の期間と第２の期間の境目である時刻ｔ３において、ダイオード１４の電流Ｉｄはゼロから増加を始めるため、ダイオード１４の電流Ｉｄを微分してその変化率を求め、直流電圧源２４ｈによる基準値と大小比較することで時刻ｔ３を検出することができる。
【００８１】
実施の形態４．
この実施の形態は、ターンオンのスイッチング動作中において、第１の期間ではコレクタ−エミッタ間電圧の変化率を概ね一定に保つとともに、第２の期間ではコレクタ電流の変化率を概ね一定に保つものである。
この実施の形態における構成は、図８に示したのと同様である。図８において、２４はＩＧＢＴ素子１をオフ状態からオン状態に遷移させる際にＩＧＢＴ素子１に流れる電流の変化が急峻な第２の期間からそれに続く電流の変化が緩慢な第１の期間に遷移する時の電流の変化点を検出する電流変化点検出手段である。電流変化点検出手段２４の具体的な構成については、図９〜図１６と同じもしくは相当する構成とすることができ、両者は比較の基準となる直流電圧源２４ｄ、２４ｈの電圧、もしくは微分手段２４ｅの出力極性が異なるのみである。その他については実施の形態３と同様であるので説明を省略する。
【００８２】
次に、動作について説明する。切り換え手段２３を、第１の関数発生手段２０の側にした場合の動作については実施の形態１と同じであるため説明を省略する。また、第２の関数発生手段２２の側にした場合の動作については実施の形態２と同じであるため説明を省略する。ここでは、切り換え手段２３と電流変化点検出手段２４について説明する。ＩＧＢＴ素子１のターンオン波形を示した図２７において、ＩＧＢＴ素子１のコレクタ−エミッタ間電圧が大きく変化するのは図２７の第１の期間であり、この期間にはコレクタ電流は変化しない。コレクタ電流が変化するのは図２７の第２の期間であり、この期間にはコレクタ−エミッタ間電圧が大きくは変化しない。このように、実施の形態１が効果を発揮する第２の期間と、実施の形態２が効果を発揮する第１の期間とは、図２７の時刻ｔ８から時刻ｔ９の間の環流ダイオード１４のリカバリ動作期間で切り分けられる。
【００８３】
このため、電流変化点検出手段２４が第１の期間と第２の期間の境目にあたる環流ダイオード１４のリカバリ動作期間（ｔ８〜ｔ９）を検出し、時刻ｔ８以前の第２の期間においては第１の関数発生手段２０の出力を使用するように切り換え手段２３を制御し、時刻ｔ９以後の第１の期間においては第２の関数発生手段２２の出力を使用するように切り換え手段２３を制御すれば、それぞれの期間において実施の形態１および実施の形態２と同じ効果が得られる。よって、図８の構成によれば、ＩＧＢＴ素子１のターンオン中の第２、第１の期間においてそれぞれコレクタ電流の変化率あるいはコレクタ−エミッタ間電圧の変化率を概ね一定に保つことができる。
なお、実施の形態３および４ではそれぞれターンオン、ターンオフに適用したが、これらを組み合わせてターンオン、ターンオフの両方に適用することもできる。
【００８４】
実施の形態５．
図１７は、この発明の実施の形態５における半導体素子の駆動回路のブロック図であり、スイッチング動作中におけるコレクタ電流の変化率を概ね一定に保つ場合の構成を示す。図において、２５はＩＧＢＴ素子１のゲート−エミッタ間に接続された第１のコンデンサであり、その他については実施の形態１と同様であるので説明を省略する。
次に、動作について説明する。図１７は、第１のコンデンサ２５以外は図２と同じであり、第１のコンデンサ２５はＩＧＢＴ素子１のゲート−エミッタ間静電容量Ｃｇｅと並列に接続される。このため、第１のコンデンサ２５の静電容量をＣ１とすると、ＩＧＢＴ素子１のゲート−エミッタ間静電容量がＣｇｅから（Ｃｇｅ＋Ｃ１）に増加したことと等価である。従って、第１の関数発生手段２０の出力Ｉｇ^＊を（６）式のゲート−エミッタ間電圧静電容量Ｃｇｅを変更して得られる（２７）式とすれば、図１７の構成においても実施の形態１と同様の効果が得られる。
【００８５】
【数９】

【００８６】
この実施の形態によれば、ＩＧＢＴ素子１のゲート電流ＩｇをＣ１により調整できるため、ゲート電流Ｉｇをコレクタ電流変化率の指令値ｄＩｃ／ｄｔ^＊、およびＩＧＢＴ素子１の特性ｆ、Ｃｇｅと独立に、駆動回路の都合でゲート電流値を設定することが可能となる。また、実施の形態３、実施の形態４のように切り換えを行う用途では、切り換え前後のゲート電流Ｉｇが等しくなるように第１のコンデンサ２５の静電容量を設定することで、切り換え時のゲート電流Ｉｇの急変を防止し滑らかな切り換えを実現できる。
【００８７】
実施の形態６．
図１８は、この発明の実施の形態６における半導体素子の駆動回路のブロック図であり、スイッチング動作中におけるコレクタ−エミッタ間電圧の変化率を概ね一定に保つ場合の構成を示す。図において、２６はＩＧＢＴ素子１のコレクタ−ゲート間に接続された第２のコンデンサ、その他については実施の形態２と同様であるので説明を省略する。
次に、動作について説明する。図１８は、第２のコンデンサ２６以外は図５と同じであり、第２のコンデンサ２６はＩＧＢＴ素子１のコレクタ−ゲート間静電容量Ｃｇｃと並列に接続される。このため、第２のコンデンサ２６の静電容量をＣ２とすると、ＩＧＢＴ素子１のコレクタ−ゲート間静電容量がＣｇｃから（Ｃｇｃ＋Ｃ２）に増加したことと等価である。従って、第２の関数発生手段２２の出力Ｉｇ^＊を（２１）式のコレクタ−ゲート間静電容量Ｃｇｃを変更して得られる（２８）式とすれば、図１８の構成においても実施の形態２と同じ効果が得られる。
Ｉｇ^＊＝（Ｃｇｃ＋Ｃ２）・ｄＶｃｅ／ｄｔ^＊・・・（２８）
【００８８】
この実施の形態によれば、ＩＧＢＴ素子１のゲート電流ＩｇをＣ２により調整できるため、ゲート電流Ｉｇをコレクタ−エミッタ間電圧変化率の指令値ｄＶｃｅ／ｄｔ^＊、およびＩＧＢＴ素子１の特性Ｃｇｃと独立に、駆動回路の都合でゲート電流値を設定することが可能となる。また、実施の形態３、実施の形態４のように切り換えを行う用途では、切り換え前後のゲート電流Ｉｇが等しくなるように第２のコンデンサ２６の静電容量を設定することで、切り換え時のゲート電流Ｉｇの急変を防止し滑らかな切り換えを実現できる。さらに、第２のコンデンサ２６は一定の静電容量Ｃ２を持つコンデンサであるため、（２６）式で示したようなコレクタ−ゲート間静電容量Ｃｇｃの電圧依存性の影響を抑制し、ゲート電流Ｉｇの変化を軽減する効果がある。
【００８９】
実施の形態７．
図１９は、この発明の実施の形態７における半導体素子の駆動回路を示す回路図であり、スイッチング動作中におけるコレクタ電流の変化率を概ね一定に保つ場合の構成を示す。図１９において、ゲート電流発生手段１８は、カレントミラー回路を構成するトランジスタ１８ａと１８ｂから成り、電流変化率設定手段１９は、直流電圧源１９ａとトランジスタ１９ｂから成り、そして第１の関数発生手段２０は、抵抗２０ｆとカレントミラー回路を構成するトランジスタ２０ｇ、２０ｈから成る。その他は実施の形態１と同様であるので説明を省略する。図１９では、簡単化のため、ターンオフ用の回路部分のみを示した。
【００９０】
次に、動作について説明する。スイッチング信号発生手段４がオフ信号を出力すると、トランジスタ１９ｂがオフする。トランジスタ２０ｇ、２０ｈから成るカレントミラー回路は直流電圧源１９ａの電圧Ｖｃｃに接続されており、トランジスタ２０ｇは抵抗２０ｆを介してＩＧＢＴ素子１のゲートに接続されているため、トランジスタ２０ｈのコレクタ電流Ｉｇ^＊は、トランジスタのベース−エミッタ間電圧降下を無視すれば（２９）式で与えられる。ここで、Ｋ１はカレントミラー回路の電流ゲイン、Ｒ１は抵抗２０ｆの抵抗値、ＶｇｅはＩＧＢＴ素子１のゲート−エミッタ間電圧である。
Ｉｇ^＊＝Ｋ１・（Ｖｃｃ−Ｖｇｅ）／Ｒ１・・・（２９）
【００９１】
上式のトランジスタ２０ｈのコレクタ電流Ｉｇ^＊は、第１の関数発生手段２０の出力であり、ゲート電流指令である。トランジスタ１９ｂがオフしているため、トランジスタ２０ｈのコレクタ電流Ｉｇ^＊は、全てトランジスタ１８ａに流れる。トランジスタ１８ａと１８ｂはカレントミラー回路を構成しているため、ＩＧＢＴ素子１のゲートにはトランジスタ２０ｈのコレクタ電流Ｉｇ^＊にトランジスタ１８ａと１８ｂとから成るカレントミラー回路のゲインを乗算した電流が流れる。（２９）式を実施の形態１の（６）式と比較すると、ＩＧＢＴ素子１のゲート−エミッタ間静電容量Ｃｇｅとコレクタ電流変化率ｄＩｃ／ｄｔ^＊を定数とし、ＩＧＢＴ素子１のゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数として、ゲート−エミッタ間電圧の１次式を使用していることがわかる。
図２０の（ａ）は、ＩＧＢＴ素子１のゲート−エミッタ間電圧と、ゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数との関係を示すグラフである。実施の形態１で示した（１４）式から、コレクタ電流変化率ｄＩｃ／ｄｔ^＊を定数Ａとした時のゲート電流指令Ｉｇ^＊は次の（３０）式となる。
【００９２】
【数１０】

【００９３】
ＩＧＢＴ素子１のゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数は、（３０）式右辺の１／（２・Ｋｐ・（Ｖｇｅ−Ｖｔｈ））の部分であり、双曲線関数となっている。図２０の（ａ）からわかるように、例えば（２９）式のようなゲート−エミッタ間電圧の１次式により良好な近似が可能である。従って、ゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数としてゲート−エミッタ間電圧の１次式を用いることで、実施の形態１と同様の効果を簡単な回路で実現できる。
【００９４】
実施の形態８．
図２１は、この発明の実施の形態８における半導体素子の駆動回路を示す回路図であり、スイッチング動作中におけるコレクタ電流の変化率を概ね一定に保つ場合の構成を示す。図２１において、ゲート電流発生手段１８は、カレントミラー回路を構成するトランジスタ１８ａと１８ｂから成り、電流変化率設定手段１９は、直流電圧源１９ａとトランジスタ１９ｂから成る。２０ｊは第１の関数発生手段としての抵抗である。その他は実施の形態１と同じであるので説明を省略する。図２１では、簡単化のため、ターンオフ用の回路部分のみを示した。
【００９５】
次に、動作について説明する。スイッチング信号発生手段４がオフ信号を出力すると、トランジスタ１９ｂがオフする。抵抗２０ｊは直流電圧源１９ａの電圧Ｖｃｃに接続されているため、抵抗２０ｊを流れる電流Ｉｇ^＊は、トランジスタのベース−エミッタ間電圧降下を無視すれば（３１）式で与えられる。ここで、Ｒ１は抵抗２０ｊの抵抗値である。
Ｉｇ^＊＝Ｖｃｃ／Ｒ１・・・（３１）
上式の抵抗２０ｊの電流Ｉｇ^＊は、第１の関数発生手段の出力であり、ゲート電流指令である。トランジスタ１９ｂがオフしているため、抵抗２０ｊの電流Ｉｇ^＊は、全てトランジスタ１８ａに流れる。トランジスタ１８ａと１８ｂはカレントミラー回路を構成しているため、ＩＧＢＴ素子１のゲートには抵抗２０ｊの電流Ｉｇ^＊にトランジスタ１８ａと１８ｂとから成るカレントミラー回路のゲインを乗算した電流が流れる。（３１）式を実施の形態１の（６）式と比較すると、ＩＧＢＴ素子１のゲート−エミッタ間静電容量Ｃｇｅとコレクタ電流変化率ｄＩｃ／ｄｔ^＊を定数とし、ＩＧＢＴ素子１のゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数にも、定数を使用していることがわかる。
【００９６】
図２０の（ｂ）は、ＩＧＢＴ素子１のゲート−エミッタ間電圧と、ゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数との関係を示すグラフである。ＩＧＢＴ素子１のゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数は、実施の形態７で説明したように、双曲線関数となっている。図２０の（ｂ）からわかるように、ＩＧＢＴ素子１のゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数が双曲線関数の場合、定数では良好な近似とはいえない。しかし、ＩＧＢＴ素子が変化の小さい特性を有する場合、定数でも概ね良好な結果を得ることが可能である。このような場合、ゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数として、代表値である定数を用いることで、実施の形態１と同様の効果を簡単な回路で実現できる。
【００９７】
実施の形態９．
図２２は、この発明の実施の形態９における半導体素子の駆動回路を示す回路図であり、スイッチング動作中におけるコレクタ−エミッタ間電圧の変化率を概ね一定に保つ場合の構成を示す。図２２において、ゲート電流発生手段１８は、カレントミラー回路を構成するトランジスタ１８ａと１８ｂから成り、電圧変化率設定手段２１は、直流電圧源２１ａとトランジスタ２１ｂから成り、そして第２の関数発生手段２２は、抵抗２２ｄ、２２ｅとカレントミラー回路を構成するトランジスタ２２ｆ、２２ｇから成る。その他は実施の形態２と同様であるので説明を省略する。図２２では、簡単化のため、ターンオフ用の回路部分のみを示した。
【００９８】
次に、動作について説明する。スイッチング信号発生手段４がオフ信号を出力すると、トランジスタ２１ｂがオフする。抵抗２２ｄは直流電圧源２１ａの電圧Ｖｃｃに接続されており、トランジスタ２２ｇは抵抗２２ｅを介してＩＧＢＴ素子１のコレクタに接続されているため、第２の関数発生手段２２の出力は、トランジスタのベース−エミッタ間電圧降下を無視すれば（３２）式で与えられる。ここで、Ｋ２はカレントミラー回路の電流ゲイン、Ｒ２は抵抗２２ｄの抵抗値、Ｒ３は抵抗２２ｅの抵抗値、ＶｃｅはＩＧＢＴ素子１のコレクタ−エミッタ間電圧である。
Ｉｇ^＊＝Ｖｃｃ／Ｒ２−Ｋ２・Ｖｃｅ／Ｒ３・・・（３２）
【００９９】
上式のＩｇ^＊は、第２の関数発生手段２２の出力であり、ゲート電流指令である。トランジスタ２１ｂがオフしているため、（３２）式のゲート電流指令Ｉｇ^＊は、全てトランジスタ１８ａに流れる。トランジスタ１８ａと１８ｂはカレントミラー回路を構成しているため、ＩＧＢＴ素子１のゲートにはゲート電流指令Ｉｇ^＊にトランジスタ１８ａと１８ｂとから成るカレントミラー回路のゲインを乗算した電流が流れる。（３２）式を実施の形態２の（２１）式と比較すると、ＩＧＢＴ素子１のコレクタ−エミッタ間電圧変化率ｄＶｃｅ／ｄｔ^＊を定数とし、ＩＧＢＴ素子１のコレクタ−ゲート間静電容量Ｃｇｃとして、コレクタ−エミッタ間電圧の１次式を使用していることがわかる。
【０１００】
図２３の（ａ）は、ＩＧＢＴ素子１のコレクタ−ゲート間静電容量Ｃｇｃの電圧依存性を示したグラフである。実施の形態２の（２６）式から、コレクタ−エミッタ間電圧変化率ｄＶｃｅ／ｄｔ^＊を定数Ｂとした時のゲート電流指令Ｉｇ^＊は（３３）式となる。
【０１０１】
【数１１】

【０１０２】
ＩＧＢＴ素子１のコレクタ−ゲート間静電容量Ｃｇｃは、（３３）式右辺のＣｊｏ・（１＋Ｖｇｃ／Ｖｊ）^−Ｍの部分であり、双曲線関数となっている。図２３の（ａ）からわかるように、例えば（３２）式のようなコレクタ−エミッタ間電圧の１次式により良好な近似が可能である。従って、コレクタ−ゲート間静電容量Ｃｇｃとしてコレクタ−エミッタ間電圧の１次式を用いることで、実施の形態２と同様の効果を簡単な回路で実現できる。また、図２２の回路は、抵抗２２ｅの接続点をＩＧＢＴ素子１のコレクタからゲートに変更することで、図１９で示した実施の形態７と同様の機能を実現できる。
【０１０３】
実施の形態１０．
図２４は、この発明の実施の形態１０における半導体素子の駆動回路を示す回路図であり、スイッチング動作中におけるコレクタ−エミッタ間電圧の変化率を概ね一定に保つ場合の構成を示す。図２４において、ゲート電流発生手段１８は、カレントミラー回路を構成するトランジスタ１８ａと１８ｂから成り、電圧変化率設定手段２１は、直流電圧源２１ａとトランジスタ２１ｂから成る。２２ｈは第２の関数発生手段としての抵抗である。その他は実施の形態２と同様であるので説明を省略する。図２４では、簡単化のため、ターンオフ用の回路部分のみを示した。
【０１０４】
次に、動作について説明する。スイッチング信号発生手段４がオフ信号を出力すると、トランジスタ１９ｂがオフする。抵抗２２ｈは直流電圧源１９ａの電圧Ｖｃｃに接続されているため、抵抗２２ｈを流れる電流Ｉｇ^＊は、トランジスタのベース−エミッタ間電圧降下を無視すれば（３４）式で与えられる。ここで、Ｒ２は抵抗２２ｈの抵抗値である。
Ｉｇ^＊＝Ｖｃｃ／Ｒ２・・・（３４）
上式の抵抗２２ｈの電流Ｉｇ^＊は、第２の関数発生手段の出力であり、ゲート電流指令である。トランジスタ１９ｂがオフしているため、抵抗２２ｈの電流Ｉｇ^＊は、全てトランジスタ１８ａに流れる。トランジスタ１８ａと１８ｂはカレントミラー回路を構成しているため、ＩＧＢＴ素子１のゲートには抵抗２２ｈの電流Ｉｇ^＊にトランジスタ１８ａと１８ｂとから成るカレントミラー回路のゲインを乗算した電流が流れる。（３４）式を実施の形態２の（２１）式と比較すると、ＩＧＢＴ素子１のコレクタ−エミッタ間電圧変化率ｄＶｃｅ／ｄｔ^＊を定数とし、ＩＧＢＴ素子１のコレクタ−ゲート間静電容量Ｃｇｃにも、定数を使用していることがわかる。
【０１０５】
図２３の（ｂ）は、ＩＧＢＴ素子１のコレクタ−ゲート間静電容量Ｃｇｃの電圧依存性を示したグラフである。ＩＧＢＴ素子１のコレクタ−ゲート間静電容量Ｃｇｃは、実施の形態９で説明したように双曲線関数となっている。図２３の（ｂ）からわかるように、ＩＧＢＴ素子１のコレクタ−ゲート間静電容量Ｃｇｃが双曲線関数の場合、定数では良好な近似とはいえない。しかし、ＩＧＢＴ素子が変化の小さい特性を有する場合、定数でも概ね良好な結果を得ることが可能である。また、実施の形態６に示したようにコレクタ−ゲート間静電容量Ｃｇｃにコンデンサを並列接続する場合には、コレクタ−ゲート間静電容量Ｃｇｃの変化の影響が軽減されるため、定数でも概ね良好な結果を得ることが可能である。従って、コレクタ−ゲート間静電容量Ｃｇｃとして、代表値である定数を用いることで、実施の形態２と同様の効果を簡単な回路で実現できる。
【０１０６】
なお、実施の形態７〜１０ではコレクタの電流変化率を制御する場合、またはコレクタ−エミッタ間電圧変化率を制御する場合について示したが、両方を組み合わせてもよい。また、ターンオフ用の回路で示したが、ターンオン用でも同様に簡単な回路で近似的に制御することができ、さらに両方を併せて用いることもできる。
【０１０７】
【発明の効果】
請求項１に係る発明によれば、ゲート電流から電圧駆動形自己消弧半導体素子のコレクタ電流変化率への伝達関数の逆関数に電流変化率設定手段の出力を代入して得られるゲート電流の指令を、第１の関数発生手段が発生するようにしたので、負荷や電源電圧の変動にかかわらず半導体素子の非線形性を補正して、設定した所望のスイッチング特性である、コレクタ電流の変化率一定の特性が得られる。また、請求項２に係る発明によれば、ゲート電流から電圧駆動形自己消弧半導体素子のコレクタ−エミッタ間電圧変化率への伝達関数の逆関数に電圧変化率設定手段の出力を代入して得られるゲート電流の指令を、第２の関数発生手段が発生するようにしたので、負荷や電源電圧の変動にかかわらず半導体素子の非線形性を補正して、設定した所望のスイッチング特性である、コレクタ−エミッタ間電圧の変化率一定の特性が得られる。このため、スイッチング動作中における不要に高いもしくは低い電圧および電流の変化率を抑制することができ、スイッチング損失の低減、ターンオフ時のサージ電圧の低減、ターンオン時の環流ダイオードのリカバリ電流の低減に効果がある。さらに、スイッチングに起因するノイズや漏洩電流の低減にも効果がある。
【０１０８】
請求項３、請求項４に係る発明によれば、コレクタ電流の変化が緩慢な第１の期間で電圧変化率設定手段と第２の関数発生手段を利用し、コレクタ電流の変化が急峻な第２の期間で電流変化率設定手段と第１の関数発生手段を利用するようにしたので、所望のスイッチング特性、例えば第１の期間でコレクタ−エミッタ間電圧の変化率を一定に保つとともに、第２の期間でコレクタ電流の変化率を一定に保つなどの特性が得られる。
【０１０９】
請求項５に係る発明によれば、第１の関数発生手段で使用するゲート電流からコレクタ電流変化率への伝達関数の逆関数が、ゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数と、ゲート−エミッタ間の静電容量と、コレクタ電流変化率との積であるので、請求項１、３、４のものと同様の効果が得られる。
請求項６に係る発明によれば、第２の関数発生手段で使用するゲート電流からコレクタ−エミッタ間電圧変化率への伝達関数の逆関数が、ゲート−コレクタ間の静電容量と、コレクタ−エミッタ間電圧変化率との積であるので、請求項２、３、４のものと同様の効果が得られる。
【０１１０】
請求項７に係る発明によれば、ゲート−エミッタ間に接続されるコンデンサを有し、第１の関数発生手段で使用するゲート電流からコレクタ電流変化率への伝達関数の逆関数が、ゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数と、ゲート−エミッタ間の静電容量およびコンデンサの静電容量の和と、コレクタ電流変化率との積であるので、駆動回路の都合に合わせた適切なゲート電流値を設定することができる。
請求項８に係る発明によれば、ゲート−コレクタ間に接続されるコンデンサを有し、第２の関数発生手段で使用するゲート電流からコレクタ−エミッタ間電圧変化率への伝達関数の逆関数が、ゲート−コレクタ間の静電容量およびコンデンサの静電容量の和と、コレクタ−エミッタ間電圧変化率との積であるので、駆動回路の都合に合わせた適切なゲート電流値を設定することができる。
【０１１１】
請求項９に係る発明によれば、ゲート−エミッタ間に接続されるコンデンサを有し、第１、第２の関数発生手段の切り換え時に、両方の出力が互いに等しくなるように前記コンデンサの静電容量を設定したので、滑らかな切り換えをすることができる。
請求項１０に係る発明によれば、ゲート−コレクタ間に接続されるコンデンサを有し、第１、第２の関数発生手段の切り換え時に、両方の出力が互いに等しくなるように前記コンデンサの静電容量を設定したので、滑らかな切り換えをすることができる。
【０１１２】
請求項１１に係る発明によれば、第１の関数発生手段で使用するゲート電流からコレクタ電流変化率への伝達関数の逆関数において、ゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数に代えて、ゲート−エミッタ間電圧の一次式で近似して使用するようにしたので、簡単な回路でコレクタ電流変化率を制御することができる。
請求項１２に係る発明によれば、第１の関数発生手段で使用するゲート電流からコレクタ電流変化率への伝達関数の逆関数において、ゲート−エミッタ間電圧からコレクタ電流への伝達関数の微分の逆数に代えて、代表値である定数で近似して使用するようにしたので簡単な回路でコレクタ電流変化率を制御することができる。
請求項１３に係る発明によれば、第２の関数発生手段で使用するゲート電流からコレクタ−エミッタ間電圧変化率への伝達関数の逆関数において、ゲート−コレクタ間の静電容量に代えて、ゲート−コレクタ間電圧の一次式で近似して使用するようにしたので簡単な回路でコレクタ−エミッタ間の電圧変化率を制御することができる。
請求項１４に係る発明によれば、第２の関数発生手段で使用するゲート電流からコレクタ−エミッタ間電圧変化率への伝達関数の逆関数において、ゲート−コレクタ間の静電容量に代えて、代表値である定数で近似して使用するようにしたので簡単な回路でコレクタ−エミッタ間の電圧変化率を制御することができる。
【図面の簡単な説明】
【図１】この発明の実施の形態における半導体素子の駆動回路の基本的構成を示すブロック図である。
【図２】この発明の実施の形態１における半導体素子の駆動回路のブロック図である。
【図３】この発明の実施の形態１における半導体素子とその駆動回路の内部の機能構成を示すブロック図である。
【図４】この発明の実施の形態１における半導体素子とその駆動回路の内部の機能構成を示すブロック図である。
【図５】この発明の実施の形態２における半導体素子の駆動回路のブロック図である。
【図６】この発明の実施の形態２における半導体素子とその駆動回路の内部の機能構成を示すブロック図である。
【図７】この発明の実施の形態２における半導体素子とその駆動回路の内部の機能構成を示すブロック図である。
【図８】この発明の実施の形態３および４における半導体素子の駆動回路のブロック図である。
【図９】この発明の実施の形態３における電流変化点検出手段の構成例を示すブロック図である。
【図１０】この発明の実施の形態３おける電流変化点検出手段の構成例を示すブロック図である。
【図１１】この発明の実施の形態３における電流変化点検出手段の構成例を示すブロック図である。
【図１２】この発明の実施の形態３における電流変化点検出手段の構成例を示すブロック図である。
【図１３】この発明の実施の形態３における電流変化点検出手段の構成例を示すブロック図である。
【図１４】この発明の実施の形態３における電流変化点検出手段の構成例を示すブロック図である。
【図１５】この発明の実施の形態３における電流変化点検出手段の構成例を示すブロック図である。
【図１６】この発明の実施の形態３における電流変化点検出手段の構成例を示すブロック図である。
【図１７】この発明の実施の形態５における半導体素子の駆動回路のブロック図である。
【図１８】この発明の実施の形態６における半導体素子の駆動回路のブロック図である。
【図１９】この発明の実施の形態７における半導体素子の駆動回路の回路図である。
【図２０】この発明の実施の形態７および８におけるゲート−エミッタ間電圧からコレクタ電圧への伝達関数の微分の逆数を示すグラフである。
【図２１】この発明の実施の形態８における半導体素子の駆動回路の回路図である。
【図２２】この発明の実施の形態９における半導体素子の駆動回路の回路図である。
【図２３】この発明の実施の形態９および１０におけるコレクタ−ゲート間静電容量の電圧依存性を示すグラフである。
【図２４】この発明の実施の形態１０における半導体素子の駆動回路の回路図である。
【図２５】ＩＧＢＴ素子のスイッチング特性を説明するための回路図である。
【図２６】ＩＧＢＴ素子のターンオフ動作を示す波形図である。
【図２７】ＩＧＢＴ素子のターンオン動作を示す波形図である。
【図２８】従来の半導体素子の駆動回路の回路図である。
【図２９】従来の半導体素子の駆動回路の動作を示す波形図である。
【符号の説明】
１ＩＧＢＴ素子、１ａゲート、１ｂコレクタ、１ｃエミッタ、
４スイッチング信号発生手段、１６スイッチング特性設定手段、
１７関数発生手段、１８ゲート電流発生手段、１９電流変化率設定手段、
２０第１の関数発生手段、２１電圧変化率設定手段、
２２第２の関数発生手段、２３切り換え手段、２４電流変化点検出手段、
２５第１のコンデンサ、２６第２のコンデンサ。[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a gate drive circuit for a semiconductor element such as an IGBT (Insulated Gate Bipolar Transistor) element or a MOSFET (Metal Oxicide Silicon Field Effect Transistor) element, which is a voltage-driven semiconductor element for switching. Is.
[0002]
[Prior art]
FIG. 28 is a circuit diagram showing a conventional semiconductor element driving circuit described in, for example, Japanese Patent Laid-Open No. 9-65644. In FIG. 28, reference numeral 1 designates a conduction state between main electrodes by a voltage applied between control electrodes. 1a is a gate as a first control electrode, 1b is a collector as a first main electrode, 1c is a second control electrode and a second main electrode. An emitter that is also an electrode serving as an electrode, 2 is a DC voltage source for turning on to supply a voltage for turning on the IGBT element 1, 3 is a DC voltage source for turning off that supplies a voltage for turning off the IGBT element 1, 4 Is a switching signal generating means for generating a signal for turning on / off the

IGBT element

1, and 5 is an IGBT that switches between the DC voltage source for turning on 2 and the DC voltage source for turning off by the output of the switching signal generating means 4. An on / off switching means for applying a voltage to the gate 1a of the element 1, 6 is a gate resistance switching means for switching the gate resistance, and the gate resistance switching means 6 comprises a resistor 9, a resistor 10, and a switching means 11. . 7 is a current detecting means for detecting the current flowing through the main electrode of the

IGBT element

1, 8 is a control means for controlling the gate resistance switching means 6 by the output of the current detecting means 7, 12 is a DC voltage source, 13 is a load, 14 is A free-wheeling diode 15 is a wiring inductance.
[0003]
First, the basic operation of the conventional example will be described with reference to FIGS. When the switching signal generating means 4 outputs an ON signal in FIG. 28, the ON / OFF switching means 5 is switched to the ON DC voltage source 2 side, and the gate resistance switching means 6 is connected between the gate 1a and the emitter 1c of the IGBT element 1. The on-voltage is applied, and the IGBT element 1 is turned on. When the IGBT element 1 is turned on, the voltage Vce between the collector 1b and the emitter 1c of the IGBT element 1 becomes substantially zero. At this time, the voltage Vdc of the DC voltage source 12 is applied to the load 13, and a current flows through a path of the DC voltage source 12 -wiring inductance 15 -load 13 -IGBT element 1 -DC voltage source 12.
[0004]
When the switching signal generating means 4 outputs an off signal, the on / off switching means 5 is switched to the off DC voltage source 3 side, and the off voltage is switched between the gate 1a and the emitter 1c of the IGBT element 1 via the gate resistance switching means 6. When applied, the IGBT element 1 is turned off. When the IGBT element 1 is turned off, the current Ic flowing through the collector 1b of the IGBT element 1 becomes substantially zero. The current that has been flowing through the IGBT element 1 until then is commutated to the freewheeling diode 14, and the current flows through the path of the freewheeling diode 14, the load 13, and the freewheeling diode 14. At this time, since the voltage between the terminals of the freewheeling diode 14 becomes substantially zero, the voltage Vdc of the DC voltage source 12 is applied to the IGBT element 1.
FIGS. 29A and 29B show waveforms at the turn-off time when the IGBT element 1 transitions from the on state to the off state. FIG. 29A shows a small resistance value of the gate resistance, FIG. 29B shows a large value, and FIG. The case of switching as shown in FIG. The turn-off operation of the IGBT element 1 starts when the collector-emitter voltage Vce first increases. During this period, the collector current Ic is substantially constant, and a first period in which the collector current changes slowly is formed. When the collector-emitter voltage Vce increases and reaches the voltage Vdc of the DC voltage source 12, the freewheeling diode 14 is turned on and the current flowing through the load 13 starts to flow through the freewheeling diode 14. Since the current can be regarded as constant due to the inductance of the load 13, the collector current Ic decreases by the amount of the current Id flowing through the freewheeling diode 14. Therefore, as shown in FIG. 29, after the collector-emitter voltage Vce reaches the voltage Vdc of the DC voltage source 12, the collector current Ic decreases rapidly, and a second period in which the collector current changes sharply is formed. Is done.
[0005]
Since the collector current Ic is equal to the current flowing through the wiring inductance 15 in the second period, the current flowing through the wiring inductance 15 decreases when the collector current Ic decreases rapidly. For this reason, the wiring inductance 15 generates a surge voltage represented by (wiring inductance × current reduction rate). Since the surge voltage has the same polarity as the voltage of the DC voltage source 12, the collector-emitter voltage Vce has a waveform in which the surge voltage generated by the wiring inductance 15 is superimposed on the DC voltage source 12. When the collector current Ic reaches zero, all the current of the load 13 flows to the freewheeling diode 14, and the turn-off operation is completed.
[0006]
Next, the operation of the control means 8 will be described. In the above-described turn-off operation in which the IGBT element 1 transitions from on to off, the control means 8 obtains a differential value of the collector current Ic of the IGBT element 1 detected by the current detection means 7, and the value is the reference value. It is determined whether or not it is smaller. When the derivative value of the collector current is smaller than the reference value, that is, in the first period in which the change in the collector current is slow, the control means 8 short-circuits the switching means 11 and uses the resistor 9 as the gate resistance. When the differential value of the collector current is larger than the reference value, that is, in the second period in which the change in the collector current is steep, the control means 8 opens the switching means 11 and uses the resistors 9 and 10 as gate resistors. . Since the resistor 9 and the resistor 10 are connected in series, the combined resistance value is larger than the resistance value of the resistor 9. By this operation, the IGBT element 1 is driven in a state where the gate resistance is small in the first period in which the change in the collector current is slow, and in the second period in which the change in the collector current is steep, the state in which the gate resistance is large. The IGBT element 1 is driven.
[0007]
Next, the effect of switching the gate resistance in the conventional example will be described with reference to FIG. FIG. 29 illustrates the current Ic at the collector of the IGBT element 1, the collector-emitter voltage Vce of the IGBT element 1, and the switching loss P of the IGBT element 1 when the IGBT element 1 is turned off. (B) shows a case where the gate resistance is small, that is, the IGBT element 1 is driven with the switching means 11 short-circuited, and (b) shows a case where the gate resistance is high, that is, the IGBT element 1 is driven with the switching means 11 opened. c) shows a case where the gate resistance is switched between the first period in which the change in the collector current Ic is slow and the second period in which the change is steep. 29A and 29B, as the gate resistance is increased, the decrease rate of the collector current Ic at the turn-off time becomes gradual, and the decrease rate of the collector current Ic is applied to the wiring inductance. Although the collector-emitter surge voltage is reduced, at the same time, the rate of increase of the collector-emitter voltage Vce also becomes moderate, so that the switching loss P time represented by the multiplication of the collector current Ic and the collector-emitter voltage Vce is obtained. It can be seen that the integrated value increases. As described above, suppression of the surge voltage can be realized by simply increasing the gate resistance, but there is a disadvantage that the switching loss increases.
[0008]
According to the conventional semiconductor element driving circuit shown in FIG. 28, the IGBT element 1 is driven in a state where the gate resistance of FIG. 29A is small in the first period in which the change in the collector current Ic is slow. In the second period in which the change in the collector current Ic is steep, the IGBT element 1 is driven with a large gate resistance as shown in FIG. As a result, as shown in FIG. 29C, a voltage and current waveform equivalent to those in FIG. 29A are obtained in the first period, and equivalent to those in FIG. 29B in the second period. Voltage and current waveforms can be obtained. Therefore, the IGBT element 1 is driven at a high speed in the first period not related to the surge voltage to reduce the time integral value of the switching loss P, and the IGBT element 1 is set in the second period related to the surge voltage. The surge voltage can be suppressed by driving at a low speed.
[0009]
[Problems to be solved by the invention]
The conventional semiconductor element driving circuit is configured as described above, and the gate resistance is switched between the first period and the second period, but in each of the first period and the second period. Drives an IGBT element with a constant DC voltage source and a constant gate resistance. Since the relationship between the gate-emitter voltage, the collector current, and the collector-emitter voltage of the IGBT element is non-linear, when it is driven by a constant DC voltage source and a constant gate resistance, as shown in FIG. The rate of change of the collector-emitter voltage during the period 1 and the rate of change of the collector current during the second period change with time, and are locally high near the switching point between the first period and the second period. A rate of change occurs. For this reason, if the maximum value of the current change rate is limited so that the surge voltage is less than or equal to the desired value, the average value of the current change rate is reduced more than necessary, and the switching loss can be sufficiently reduced. There was no problem. In addition, the rate of change of collector-emitter voltage and the rate of change of collector current also change depending on the values of collector current and collector-emitter voltage. When the voltage / current operating point is deviated from the selected voltage / current operating point, switching loss reduction and surge voltage suppression do not sufficiently function, and there is a problem that the effect is small in applications where the voltage / current operating range is wide.
[0010]
The present invention has been made to solve the above-described problems, and the rate of change of the current of the main electrode or the rate of change of the voltage between the main electrodes during the switching operation is always set regardless of the fluctuation of the load or the power supply voltage. An object of the present invention is to obtain a drive circuit for a semiconductor element that can obtain desired switching characteristics such as being kept substantially constant.
[0011]
[Means for Solving the Problems]
2. A driving circuit for a semiconductor device according to claim 1, wherein the driving circuit is connected between the gate and the emitter of the voltage-driven self-extinguishing semiconductor device for controlling a conduction state between the collector and the emitter by a voltage applied between the gate and the emitter. A switching signal generating means for generating a signal for turning on / off the voltage-driven self-extinguishing semiconductor element, and a collector current change of the voltage-driven self-extinguishing semiconductor element having a constant value by an output of the switching signal generating means Current change rate setting means for generating a rate command, first function generating means for generating a gate current command for the voltage-driven self-extinguishing semiconductor element by the output of the current change rate setting means, Gate current generating means for generating the gate current according to the output of the function generating means, wherein the first function generating means comprises the gate Inverse function of the transfer function of the collector current change rate of the voltage drive self-turn-off semiconductor element from the flow Command of the gate current obtained by substituting the output of the current change rate setting means Is generated.
[0012]
According to a second aspect of the present invention, there is provided a drive circuit connected between the gate and the emitter of a voltage-driven self-extinguishing semiconductor device that controls a conduction state between the collector and the emitter by a voltage applied between the gate and the emitter. In the circuit, a switching signal generating means for generating a signal for turning on / off the voltage driven self-extinguishing semiconductor element, and a collector of the voltage driven self-extinguishing semiconductor element having a constant value by the output of the switching signal generating means. Voltage change rate setting means for generating a command for the voltage change rate between the emitters, and second function generating means for generating a command for the gate current of the voltage-driven self-extinguishing semiconductor element by the output of the voltage change rate setting means; And a gate current generating means for generating the gate current by the output of the second function generating means, and the second function generating means Inverse function of the transfer function of the emitter voltage change rate - collector of said voltage-driven self-turn-off semiconductor element from the gate current To the gate current command obtained by substituting the output of the voltage change rate setting means Is generated.
[0013]
According to a third aspect of the present invention, there is provided a drive circuit for a semiconductor device, wherein the drive circuit is connected between the gate and emitter of a voltage-driven self-extinguishing semiconductor device that controls the conduction state between the collector and emitter by a voltage applied between the gate and emitter. In the circuit, switching signal generating means for generating a signal for turning on / off the voltage-driven self-extinguishing semiconductor element, and a collector current of the voltage-driven self-extinguishing semiconductor element having a constant value by an output of the switching signal generating means Current change rate setting means for generating a change rate command, first function generating means for generating a gate current command for the voltage-driven self-extinguishing semiconductor element by an output of the current change rate setting means, and the switching A command for the rate of change in voltage between the collector and the emitter of the voltage-driven self-extinguishing semiconductor element of a constant value is issued by the output of the signal generating means. Voltage change rate setting means, second function generating means for generating a gate current command of the voltage-driven self-extinguishing semiconductor element based on an output of the voltage change rate setting means, and the first and second functions Switching means for switching the generating means to output one of the outputs, gate current generating means for generating the gate current by the output of the switching means, and between the collector and emitter by voltage control between the gate and emitter Current change point detecting means for detecting a change point from the first period in which the change in the collector current is slow at the transition from the conductive state to the non-conductive state to the second period in which the subsequent change in the collector current is steep. And the first function generating means includes an inverse function of a transfer function from the gate current to a rate of change in collector current of the voltage-driven self-extinguishing semiconductor element. Command of the gate current obtained by substituting the output of the current change rate setting means And the second function generating means is an inverse function of a transfer function from the gate current to the collector-emitter voltage change rate of the voltage-driven self-extinguishing semiconductor element. To the gate current command obtained by substituting the output of the voltage change rate setting means The switching means switches to the second function generating means in the first period and to the first function generating means in the second period.
[0014]
According to a fourth aspect of the present invention, there is provided a drive circuit for a semiconductor device, wherein the drive circuit is connected between the gate and emitter of a voltage-driven self-extinguishing semiconductor device that controls the conduction state between the collector and emitter by a voltage applied between the gate and emitter. In the circuit, switching signal generating means for generating a signal for turning on / off the voltage-driven self-extinguishing semiconductor element, and a collector current of the voltage-driven self-extinguishing semiconductor element having a constant value by an output of the switching signal generating means Current change rate setting means for generating a change rate command, first function generating means for generating a gate current command for the voltage-driven self-extinguishing semiconductor element by an output of the current change rate setting means, and the switching A command for the rate of change in voltage between the collector and the emitter of the voltage-driven self-extinguishing semiconductor element of a constant value is issued by the output of the signal generating means. Voltage change rate setting means, second function generating means for generating a gate current command of the voltage-driven self-extinguishing semiconductor element based on an output of the voltage change rate setting means, and the first and second functions Switching means for switching the generating means to output one of the outputs, gate current generating means for generating the gate current by the output of the switching means, and between the collector and emitter by voltage control between the gate and emitter Current change point detecting means for detecting a change point from the second period in which the change in the collector current is steep at the transition from the non-conduction state to the conduction state to the first period in which the subsequent change in the collector current is slow. And the first function generation means includes an inverse function of a transfer function from the gate current to a collector current change rate of the voltage-driven self-extinguishing semiconductor element. Command of the gate current obtained by substituting the output of the current change rate setting means And the second function generating means is an inverse function of a transfer function from the gate current to the collector-emitter voltage change rate of the voltage-driven self-extinguishing semiconductor element. To the gate current command obtained by substituting the output of the voltage change rate setting means The switching means switches to the first function generating means during the second period and to the second function generating means during the first period.
[0015]
According to a fifth aspect of the present invention, there is provided the driving circuit for a semiconductor device according to any one of the first, third, and fourth aspects, wherein the first function generating unit is used. Inverse function of the transfer function from the gate current used in the above to the collector current change rate of the voltage-driven self-extinguishing semiconductor device Is the reciprocal of the derivative of the transfer function from the gate-emitter voltage to the collector current of the voltage-driven self-extinguishing semiconductor element, the gate-emitter capacitance, Is the product of collector current change rate .
[0016]
According to a sixth aspect of the present invention, there is provided the semiconductor element drive circuit according to any one of the second, third, and fourth aspects, wherein the second function generating means is provided. Inverse function of the transfer function from the gate current used in the above to the voltage change rate between the collector and the emitter of the voltage-driven self-extinguishing semiconductor device A capacitance between the gate and the collector of the voltage-driven self-extinguishing semiconductor element, This is the product of the collector-emitter voltage change rate. .
[0017]
According to a seventh aspect of the present invention, there is provided a drive circuit for a semiconductor element according to any one of the first, third, and fourth aspects, including a capacitor connected between a gate and an emitter of the voltage-driven self-extinguishing semiconductor element. , The first function generating means Inverse function of the transfer function from the gate current used in the above to the collector current change rate of the voltage-driven self-extinguishing semiconductor device Is the inverse of the derivative of the transfer function from the gate-emitter voltage to the collector current of the voltage-driven self-extinguishing semiconductor element, the sum of the gate-emitter capacitance and the capacitance of the capacitor, Is the product of collector current change rate .
[0018]
According to an eighth aspect of the present invention, there is provided a semiconductor element driving circuit according to any one of the second, third, and fourth aspects, further comprising a capacitor connected between a gate and a collector of the voltage-driven self-extinguishing semiconductor element. , The second function generating means Inverse function of the transfer function from the gate current used in the above to the voltage change rate between the collector and the emitter of the voltage-driven self-extinguishing semiconductor device A sum of the gate-collector capacitance of the voltage-driven self-extinguishing semiconductor element and the capacitance of the capacitor, This is the product of the collector-emitter voltage change rate. .
[0019]
According to a ninth aspect of the present invention, there is provided a drive circuit for a semiconductor device according to the third or fourth aspect, further comprising a capacitor connected between a gate and an emitter of the voltage-driven self-extinguishing semiconductor device, and the first function generation. means The inverse function of the transfer function from the gate current used in the above to the collector current change rate of the voltage-driven self-extinguishing semiconductor device is Reciprocal of the derivative of the transfer function from the gate-emitter voltage to the collector current of the voltage-driven self-extinguishing semiconductor element, the sum of the gate-emitter capacitance and the capacitance of the capacitor, Product of collector current change rate In addition, the capacitance of the capacitor is set so that the outputs of the first and second function generating means during the switching operation of the switching means are equal to each other.
[0020]
According to a tenth aspect of the present invention, there is provided a semiconductor device driving circuit according to the third or fourth aspect, further comprising a capacitor connected between a gate and a collector of the voltage-driven self-extinguishing semiconductor device, and the second function generation. means The inverse function of the transfer function from the gate current used in the above to the voltage change rate between the collector and the emitter of the voltage-driven self-extinguishing semiconductor device is A sum of the gate-collector capacitance of the voltage-driven self-extinguishing semiconductor element and the capacitance of the capacitor; Product of collector-emitter voltage change rate In addition, the capacitance of the capacitor is set so that the outputs of the first and second function generating means during the switching operation of the switching means are equal to each other.
[0021]
A drive circuit for a semiconductor device according to an eleventh aspect is the one according to any one of the fifth, seventh, and ninth aspects, wherein the first function generating means. In the inverse function of the transfer function from the gate current to the collector current change rate of the voltage-driven self-extinguishing semiconductor device The first-order approximate expression of the gate-emitter voltage is used instead of the inverse of the derivative of the transfer function from the gate-emitter voltage to the collector current of the voltage-driven self-extinguishing semiconductor element. is there.
[0022]
A drive circuit for a semiconductor device according to a twelfth aspect of the present invention is the drive circuit for the semiconductor element according to any one of the fifth, seventh, and ninth aspects. In the inverse function of the transfer function from the gate current to the collector current change rate of the voltage-driven self-extinguishing semiconductor device Instead of the inverse of the derivative of the transfer function from the gate-emitter voltage to the collector current of the voltage-driven self-extinguishing semiconductor element, it is used by approximating it with a constant that is a representative value.
[0023]
According to a thirteenth aspect of the present invention, there is provided a drive circuit for a semiconductor device according to any one of the sixth, eighth, and tenth aspects, wherein the second function generating means. In the inverse function of the transfer function from the gate current to the collector-emitter voltage change rate of the voltage-driven self-extinguishing semiconductor device Instead of the gate-collector capacitance of the voltage-driven self-extinguishing semiconductor element, a first-order approximation of the gate-collector voltage is used.
[0024]
A drive circuit for a semiconductor device according to a fourteenth aspect is the one according to the sixth, eighth, or tenth aspect, wherein the second function generating means is provided. In the inverse function of the transfer function from the gate current to the collector-emitter voltage change rate of the voltage-driven self-extinguishing semiconductor device Instead of the capacitance between the gate and the collector of the voltage driven self-extinguishing semiconductor element, it is approximated by a constant which is a representative value.
[0025]
DETAILED DESCRIPTION OF THE INVENTION
Prior to specific description of the embodiment of the present invention, basic operation will be described. FIG. 25 shows an example of a drive circuit (switching circuit) of an IGBT (insulated gate transistor) element 1 which is a voltage-driven self-extinguishing semiconductor element that controls a conduction state between main electrodes by a voltage applied between control electrodes. FIG. 1a to 1c are electrodes of the IGBT element 1, 1a is a gate which is a first control electrode, 1b is a collector which is a first main electrode, 1c is a second control electrode and a second main electrode. It is an emitter which is also an electrode. 2 is a DC voltage source for turning on that supplies a voltage for turning on the IGBT element 1, 3 is a DC voltage source for turning off that supplies a voltage for turning off the IGBT element 1, and 4 is for turning on and off the IGBT element 1. A switching signal generating means 5 for generating a signal, an on / off switching means for applying a voltage to the gate 1a of the IGBT element 1 by switching the DC voltage source 2 for turning on and the DC voltage source 3 for turning off by the output of the switching signal generating means 4; 9 is a gate resistor, 12 is a DC voltage source, 13 is a load, 14 is a freewheeling diode, and 15 is a wiring inductance. In the figure, the off DC voltage source 3 is shown as a negative voltage, but here it is assumed to be zero. Moreover, the positive / negative polarity of both DC voltage sources may be reversed depending on the PN polarity configuration of the material of the semiconductor element.
[0026]
First, a turn-off operation for causing the IGBT element to transition from a conductive state (on state) to a non-conductive state (off state) will be described.
FIG. 26 is a waveform diagram of the voltage and current of the IGBT element 1 at the time of turn-off in the drive circuit of FIG. In FIG. 25, when the switching signal generating means 4 outputs an ON signal, the ON / OFF switching means 5 is switched to the ON DC voltage source 2 side, and is turned ON via the gate resistor 9 between the gate 1a and the emitter 1c of the IGBT element 1. A voltage is applied, and the IGBT element 1 is turned on. When the IGBT element 1 is turned on, the voltage Vdc of the DC voltage source 12 is applied to the load 13, and a current flows through a path of the DC voltage source 12 -wiring inductance 15 -load 13 -IGBT element 1 -DC voltage source 12.
[0027]
The voltage and current of the IGBT element 1 at this time are represented by a point at time 0 in FIG. Since IGBT element 1 is on at time 0, voltage Vce between collector 1b and emitter 1c is almost zero, and collector current Ic is equal to current IL of load 13. The output Vs of the on / off switching means 5 is equal to the voltage of the on-state DC voltage source 2. Since the gate-emitter of the IGBT element 1 is a capacitor, the gate-emitter voltage Vge is charged to the output Vs of the on / off switching means 5, that is, the voltage of the on-state DC voltage source 2 if the on-time is sufficiently long. The current Ig becomes zero.
[0028]
Next, when the switching signal generating means 4 outputs an off signal at time t1, the on / off switching means 5 is switched to the off DC voltage source 3 side, and an off voltage is applied to the gate 1a of the IGBT element 1 via the gate resistor 9. Is done. Here, since the DC voltage source for off 3 is set to zero volts, the output Vs of the on / off switching means 5 becomes zero. The gate-emitter voltage Vge starts to decrease with a time constant of zero (gate resistance 9 × the capacitor between the gate and emitter of the IGBT element 1). However, normally, the on-state DC voltage source 2 is set sufficiently high with respect to the on-voltage Von required for flowing the load current IL, so that the gate-emitter voltage Vge becomes the load current IL at time t2. The actual turn-off operation of the IGBT element 1 is not performed until the voltage Von required to flow the current is reached.
[0029]
At time t2, when gate-emitter voltage Vge reaches voltage Von required to flow load current IL, IGBT element 1 starts a turn-off operation, and collector-emitter voltage Vce increases. At this time, the voltage applied to the load 13 is a voltage obtained by subtracting the collector-emitter voltage Vce from the voltage Vdc of the DC voltage source 12, but the current IL flowing due to the inductance of the load 13 does not change suddenly. Therefore, the collector current Ic does not change suddenly, and the collector current Ic forms a first period in which the change in current is slow. In this first period, a feedback action is performed to keep the gate-emitter voltage Vge at the voltage Von required to flow the load current IL so as not to change the collector current Ic. Vge is a substantially constant voltage. This feedback action is realized by a displacement current flowing from the collector to the gate via the collector-gate capacitor of the IGBT element 1 due to an increase in the collector-emitter voltage Vce, and this displacement current causes a voltage drop at the gate resistor 9. Is done.
[0030]
When the collector-emitter voltage Vce reaches the voltage Vdc of the DC voltage source 12 at time t3, the freewheeling diode 14 is turned on, and a part of the current flowing through the load 13 starts to commutate to the freewheeling diode 14. Since the collector current Ic decreases by the amount of the current Id commutated to the freewheeling diode 14, the collector current Ic forms a second period in which the current change is steep. In the second period, the feedback action for keeping the gate-emitter voltage Vge constant is lost, and the gate-emitter voltage Vge starts decreasing toward zero again. Further, since the collector current Ic is equal to the current flowing through the wiring inductance 15, the current flowing through the wiring inductance 15 also decreases during this second period. For this reason, the wiring inductance 15 generates a voltage represented by (wiring inductance × current reduction rate). Since this voltage has the same polarity as the voltage of the DC voltage source 12, the collector-emitter voltage Vce has a waveform in which a surge voltage generated by the wiring inductance 15 is superimposed on the DC voltage source 12.
[0031]
When gate-emitter voltage Vge reaches gate threshold voltage Vth of IGBT element 1 at time t4, collector current Ic becomes zero, and IGBT element 1 completes the turn-off operation. Although the actual turn-off operation is completed at time t4, since the gate-emitter voltage Vge has not yet reached zero, it continues to decrease toward zero. At time t5, the gate-emitter voltage Vge reaches zero, completing all operations. In the above-described turn-off operation of the IGBT element 1, the collector current Ic changes slowly in the first period (t2 to t3) and the subsequent collector current Ic changes abruptly in the second period (t3 to t4). ). Note that the first period and the second period do not mean the temporal order in which these occur, but the order of occurrence is reversed in the case of turn-on described later.
[0032]
Next, a turn-on operation for causing the IGBT element to transition from the off state to the on state will be described. FIG. 27 is a waveform diagram of voltage and current when the IGBT element 1 is turned on in the switching circuit of FIG. In FIG. 25, when the switching signal generating means 4 outputs an off signal, the on / off switching means 5 is switched to the off DC voltage source 3 side and is turned off via the gate resistor 9 between the gate 1a and the emitter 1c of the IGBT element 1. A voltage is applied, and the IGBT element 1 is turned off. When the IGBT element 1 is turned off, the collector current Ic flowing through the collector 1b of the IGBT element 1 becomes substantially zero. The current that has been flowing through the IGBT element 1 until then is commutated to the freewheeling diode 14, and the current flows through the path of the freewheeling diode 14, the load 13, and the freewheeling diode 14. At this time, since the voltage between the terminals of the freewheeling diode 14 becomes substantially zero, the voltage Vdc of the DC voltage source 12 is applied to the IGBT element 1.
[0033]
The voltage and current of the IGBT element 1 at this time are represented by a point at time 0 in FIG. Since IGBT element 1 is off at time 0, collector current Ic is substantially zero, and collector-emitter voltage Vce is equal to voltage Vdc of DC voltage source 12. The output Vs of the on / off switching means 5 is equal to the voltage of the off DC voltage source 3. Here, since the DC voltage source for off 3 is set to zero volts, the output Vs of the on / off switching means 5 becomes zero. Since the gate-emitter of the IGBT element 1 is a capacitor, the gate-emitter voltage Vge becomes zero, that is, the output Vs of the on / off switching means 5, that is, zero if the off time is sufficiently long, and the gate current Ig also becomes zero.
[0034]
Next, when the switching signal generating means 4 outputs an ON signal at time t6, the ON / OFF switching means 5 is switched to the ON DC voltage source 2, and the ON voltage is applied to the gate 1a of the IGBT element 1 via the gate resistor 9. The Then, the gate-emitter voltage Vge starts increasing with a time constant of (the resistance value of the gate resistor 9 × the capacitance of the gate-emitter capacitor of the IGBT element 1) toward the voltage of the on-state DC voltage source 2. . However, since the collector current does not flow until the gate-emitter voltage Vge reaches the gate threshold voltage Vth of the IGBT element 1, the actual turn-on operation of the IGBT element 1 is not performed.
[0035]
At time t7, when the gate-emitter voltage Vge reaches the gate threshold voltage Vth, the IGBT element 1 starts a turn-on operation, the collector current Ic increases, and the collector current Ic is in the second period in which the current change is steep. Form. During this second period, the current IL flowing due to the inductance of the load 13 does not change suddenly. For this reason, the current Id of the freewheeling diode 14 is a value obtained by subtracting the collector current Ic from the substantially constant load current IL. Since the freewheeling diode 14 is on while the freewheeling diode current Id is positive, the voltage of the DC voltage source 12 is applied to the IGBT element 1 and the wiring inductance 15. In the second period, since the current flowing through the wiring inductance 15 is equal to the collector current Ic, the current flowing through the wiring inductance 15 also increases with the collector current Ic. For this reason, the wiring inductance 15 generates a voltage represented by (wiring inductance × current increase rate). Since this voltage is opposite in polarity to the voltage of the DC voltage source 12, the collector-emitter voltage Vce has a waveform obtained by subtracting the voltage generated by the wiring inductance 15 from the voltage Vdc of the DC voltage source 12. The second period during the turn-on operation is essentially the same as the second period during the aforementioned turn-off operation.
[0036]
At time t8, even after the current Id of the freewheeling diode 14 reaches zero, the freewheeling diode 14 remains on for a while. This operation is known as a diode recovery operation. During this recovery operation, since a negative current flows through the freewheeling diode 14, the collector current Ic is a current obtained by adding the recovery current of the freewheeling diode 14 to the current IL of the load 13.
[0037]
When the recovery operation of the freewheeling diode 14 is completed at time t <b> 9, the freewheeling diode 14 is turned off, and a current flows through the path of the DC voltage source 12 -wiring inductance 15 -load 13 -IGBT element 1 -DC voltage source 12. Since the current IL does not change suddenly due to the inductance of the load 13, the collector current Ic in the same current path also does not change suddenly, and the collector current Ic forms a first period in which the change in current is slow. In this first period, a feedback action is performed to keep the gate-emitter voltage Vge at the voltage Von required to flow the load current IL so as not to change the collector current Ic. Vge is a substantially constant voltage. This feedback action is realized by a displacement current flowing from the gate to the collector via the collector-gate capacitor of the IGBT element 1 due to a decrease in the collector-emitter voltage Vce, and this displacement current causes a voltage drop at the gate resistor 9. Is done. The first period during the turn-on operation is essentially the same as the first period during the turn-off operation.
[0038]
When the collector-emitter voltage Vce reaches zero volt at time t10, the collector-emitter voltage Vce thereafter becomes constant at zero volt and no displacement current is generated. Therefore, the feedback action is lost and the gate-emitter voltage Vge is turned on again. The increase starts toward the voltage of the direct current voltage source 2. Although the actual turn-on operation is completed at time t10, the gate-emitter voltage Vge continues to increase because the gate-emitter voltage Vge has not yet reached the voltage of the on-state DC voltage source 2. At time t11, the gate-emitter voltage Vge reaches the voltage of the on-state DC voltage source 2, and all operations are completed. In the above-described turn-on operation of the IGBT element 1, the collector current Ic is continuously changed between the second period (t7 to t8) in which the change is steep and the recovery operation of the freewheeling diode. A slow first period (t9 to t10) is formed.
[0039]
Next, an embodiment of the present invention will be described with reference to FIG. FIG. 1 is a block diagram showing one basic configuration of a semiconductor element driving circuit according to an embodiment of the present invention. In the figure, reference numeral 4 denotes a switching signal generation for generating a signal for turning on / off the IGBT element 1. Means 16 is a switching characteristic setting means for generating a command relating to the switching characteristic of the IGBT element 1 by the output of the switching signal generating means 4. Reference numeral 17 denotes a function generating means for generating a command for a current to be supplied to the gate 1a of the IGBT element 1 based on the output of the switching characteristic setting means 16. Generate the inverse of the transfer function. Reference numeral 18 denotes gate current generating means as current generating means for supplying a current corresponding to the output of the function generating means 17 to the gate 1a of the IGBT element 1. The switching signal generating means 4, the switching characteristic setting means 16, the function generating means 17 and the gate current generating means 18 constitute a drive circuit for the IGBT element 1. Others are the same as in FIG.
[0040]
The operation will be described with reference to FIGS. Here, the case where the change rate of the collector current at the time of turn-off is set as a command related to the switching characteristics generated by the switching characteristics setting means 16 is taken up. When the IGBT element 1 is turned off, the collector current changes during the second period of FIG. In this second period, the gate-emitter voltage Vge decreases and the collector current Ic also decreases. In this second period, the collector current Ic and the gate-emitter voltage Vge are in a functional relationship, and equation (1) is established. In the equation (1), f is a transfer function from the gate-emitter voltage Vge to the collector current Ic.
Ic = f (Vge) (1)
[0041]
Further, since the gate-emitter of the IGBT element 1 is a capacitor, Equation (2) is established between the gate current Ig and the gate-emitter voltage Vge, where Cge is the gate-emitter capacitance. To be exact, the gate current Ig flows not only in the gate-emitter capacitance Cge but also in the gate-collector capacitance Cgc, but in the second period, the collector-emitter voltage Vce, that is, the gate-collector voltage. Since Vgc is high, the gate-collector capacitance Cgc having strong voltage dependency is sufficiently small with respect to the gate-emitter capacitance Cge and can be ignored.
Ig = Cge · (dVge / dt) (2)
From the equations (1) and (2), the transfer function from the gate current Ig of the IGBT element 1 to the collector current change rate dIc / dt is represented by the equation (3). Further, when an inverse function of the equation (3) is obtained, the equation (4) is obtained. In the equation (4), when dIc / dt is a constant value A, the equation (5) is obtained.
[0042]
[Expression 1]

[0043]
As is clear from the definition of the inverse function, the transfer function from the gate current Ig of the IGBT element 1 to the collector current change rate dIc / dt is expressed as follows. 3) When flowing through the IGBT element 1 having the transfer function of the equation (3), the collector current change rate dIc / dt becomes constant (dIc / dt = A). Accordingly, in FIG. 1, the switching characteristic setting means 16 generates a command of a constant value A by the output of the switching signal generation means 4, and the function generation means 17 converts the gate current Ig to the collector current change rate dIc according to the equation (5). If the gate current command is generated from the inverse function of the transfer function to / dt, and the gate current generating means 18 causes the current as output from the function generating means 17 to flow to the IGBT element 1 having the transfer function of equation (3), the collector The current change rate can be kept constant.
[0044]
Although only the operation at the time of turn-off has been described above, it is obvious that the same effect can be obtained with respect to the turn-on because the polarity of the change rate of the voltage and current of each part of the IGBT element 1 is reversed. Further, the above description has been made about keeping the collector current change rate of the IGBT element 1 constant. If necessary, the collector current change rate can be made to follow a desired function by changing the output of the switching characteristic setting means 16. It is possible. Furthermore, although only the collector current change rate has been described above as an example, it is clear from the definition of the inverse function that the same effect can be obtained for other switching characteristics such as the collector-emitter voltage.
A specific configuration in the embodiment will be described below.
[0045]
Embodiment 1 FIG.
FIG. 2 is a block diagram showing a drive circuit for a semiconductor element according to the first embodiment of the present invention, and shows a configuration when the rate of change of the collector current during the switching operation is kept substantially constant. In FIG. 2, 19 is a current change rate setting means as a switching characteristic setting means, 20 is a first function generating means, and 18 is a current corresponding to the output of the first function generating means 20 to the gate 1a of the IGBT element 1. Gate current generating means as current generating means to be supplied. Others are the same as in the case of FIG. FIG. 3 is a block diagram including the internal functional configuration of the first function generating means 20 and the IGBT element 1 of FIG. In FIG. 3, the first function generating means 20 is composed of 20a and 20b, 20a is the reciprocal of the derivative of the transfer function from the gate-emitter voltage of the IGBT element 1 to the collector current, and 20b is the gate of the IGBT element 1. It is the capacitance between emitters. 1d, 1e, 1f, and 1g are means for expressing the characteristics of the

IGBT element

1, and 1d and 1e are means for expressing the expression (2). The reciprocal of the capacitance between the gate and the emitter of the IGBT element 1 and the integrating means, respectively. is there. 1f is a transfer function representing the equation (1), and 1g is a differentiating means.
[0046]
Next, the operation will be described with reference to FIGS. FIG. 26 is a waveform diagram showing the voltage and current when the IGBT element 1 is turned off, as described above. In FIG. 3, according to the output of the switching signal generating means 4, the current change rate setting means 19 is instructed by the command dIc / dt. ^* Is generated, the output Ig of the first function generating means 20 ^* Is the product of the inverse 20a of the derivative of the transfer function f from the gate-emitter voltage to the collector current of the IGBT element 1, the gate-emitter capacitance 20b of the IGBT element 1, and the output of the current change rate setting means 19. And expressed by equation (6). This output Ig ^* Is a gate current command, which is the same as equation (4).
[0047]
[Expression 2]

[0048]
Since the gate current generation means 18 causes the current as output from the first function generation means 20 to flow through the IGBT element 1, the gate current Ig of the IGBT element 1 is expressed by the following equation (7).
Ig = Ig ^* ... (7)
In the second period of FIG. 26 in which the collector current Ic changes when the IGBT element 1 is turned off, the gate-emitter voltage Vge and the gate current Ig are in the relationship of equation (2), so the gate of the IGBT element 1 The -emitter voltage Vge is given by equation (8) from the reciprocal 1d of the gate-emitter electrostatic capacitance and the integrating means 1e.
[0049]
[Equation 3]

[0050]
Further, since the collector current Ic and the gate-emitter voltage Vge are in the relationship of the expression (1), the collector current Ic of the IGBT element 1 is the transfer function 1f from the gate-emitter voltage of the IGBT element 1 to the collector current. Is given by equation (9). The collector current Ic is differentiated by the differentiating means 1g to obtain the collector current change rate dIc / dt, which can be transformed as shown in the equation (10).
[0051]
[Expression 4]

[0052]
Therefore, in the configuration of FIG. 3, the output dIc / dt of the current change rate setting means 19 ^* Is kept constant, the collector current change rate of the IGBT element 1 can be kept constant.
[0053]
FIG. 4 is a diagram obtained by rewriting the block diagram of FIG. 3 in the case where the transfer function from the gate-emitter voltage to the collector current is expressed by the following equation (11). In FIG. 4, 20c is the reciprocal of the derivative of the transfer function from the gate-emitter voltage of the IGBT element 1 to the collector current, 20d is a multiplying means for obtaining the product of the output of 20c and the output of the current change rate setting means 19, 20e is a detector for obtaining the gate-emitter voltage Vge used in 20c. Others are the same as in FIG.
[0054]
Next, the operation will be described in detail with reference to FIG. In the second period of FIG. 26 in which the collector current changes when the IGBT element 1 is turned off, the collector current Ic and the gate-emitter voltage Vge generally have a functional relationship of the following equation (11). Here, Kp is transconductance and Vth is a gate threshold voltage.
Ic = Kp · (Vge−Vth) ² (11)
Similarly to the equation (2), the gate-emitter voltage Vge and the gate current Ig have a functional relationship of the equation (12).
Ig = Cge · (dVge / dt) (12)
From the equations (11) and (12), the transfer function from the gate electrode Ig of the IGBT element 1 to the collector current change rate dIc / dt becomes the equation (13). Then, when the inverse function of equation (13) is obtained, equation (14) is obtained.
[0055]
[Equation 5]

[0056]
Equation (14), like Equation (4), is the inverse 20c of the derivative of the transfer function from the gate-emitter voltage of the IGBT element 1 to the collector current, the gate-emitter capacitance 20b of the IGBT element 1, This is the product of the output of the current change rate setting means 19, and from the expression (4), the first function generating means 20 of FIG. The function generator 20 is obtained.
[0057]
Although only the operation at the time of turn-off has been described above, it is apparent that the same effect can be obtained with respect to the turn-on because the polarity of the change rate of the voltage and current of each part of the IGBT element is only reversed. This method provides the simplest configuration when there are no restrictions on the switching characteristics of the IGBT elements other than the collector current change rate. The equations (1) to (14) are applied to a period in which the collector current changes sharply as in the second period in FIG. 26, but the equations (1) to (14) are also used in other periods. It is possible to drive the IGBT element with the gate current obtained.
[0058]
Embodiment 2. FIG.
FIG. 5 is a block diagram showing a drive circuit for a semiconductor element according to the second embodiment of the present invention, and shows a configuration when the rate of change of the collector-emitter voltage during the switching operation is kept substantially constant. In FIG. 5, 21 is a voltage change rate setting means as a switching characteristic setting means, 22 is a second function generating means, 18 is a current corresponding to the output of the second function generating means 22 to the gate 1a of the IGBT element 1. Gate current generating means as current generating means to be supplied. Others are the same as in the case of FIG. FIG. 6 is a block diagram including the internal functional configuration of the second function generating means 22 and the IGBT element 1 of FIG. In FIG. 6, the second function generating means 22 is a gate-collector capacitance of the IGBT element 1. 1h, 1i, 1j, 1k, 1m, 1n are means for representing the characteristics of the IGBT element 1, 1h is a gate-collector capacitance (g represents a function), 1i is a reciprocal means, 1j is a multiplication means, 1k is an integrating means, 1m is an adding means, and 1n is a differentiating means.
[0059]
Next, the operation will be described with reference to FIGS. FIG. 26 is a waveform diagram showing the voltage and current when the IGBT element 1 is turned off. It is during the first period of FIG. 26 that the collector-emitter voltage changes greatly when the IGBT element 1 is turned off. In this first period, a feedback action is performed to keep the gate-emitter voltage Vge at the voltage Von required to flow the load current IL so as not to change the collector current 1c. Vge is a substantially constant voltage. This feedback action is realized by a displacement current flowing from the collector to the gate via the collector-gate capacitor of the IGBT element 1 due to an increase in the collector-emitter voltage Vce, and this displacement current causes a voltage drop in the gate drive circuit. Is done. Since dVge / dt is almost zero and the current between the gate and the emitter can be ignored, this displacement current is nothing but the gate current, and the following equation (15) holds.
Ig = Cgc · dVgc / dt (15)
[0060]
Further, the following equation (16) holds for the voltage between the three terminals of the gate, collector, and emitter of the IGBT element 1.
Vce = Vgc + Vge (16)
From the equations (15) and (16), the transfer function from the gate current Ig of the IGBT element 1 to the collector-emitter voltage change rate dVce / dt and its inverse function are the equations (17) and (18), respectively.
dVce / dt = Ig / Cgc + dVge / dt (17)
Ig = Cgc · (dVce / dt−dVge / dt) (18)
[0061]
As described above, since the gate-emitter voltage Vge is substantially constant during the first period, the change rate dVge / dt of the gate-emitter voltage is sufficiently small and can be ignored. Therefore, Equation (19) and Equation (20) are obtained.
dVce / dt = Ig / Cgc (19)
Ig = Cgc · dVce / dt (20)
The means for generating the equation (20) is the second function generating means 22. In FIG. 6, the voltage change rate setting means 21 receives the command dVce / dt by the output of the switching signal generating means 4. ^* Is generated, the output Ig of the second function generating means 22 ^* Becomes the equation (21). This output Ig ^* Is a gate current command.
Ig ^* = Cgc · dVce / dt ^* ... (21)
[0062]
Since the gate current generating means 18 causes the current as output from the second function generating means 22 to flow through the IGBT element 1, the gate current Ig of the IGBT element 1 is expressed by equation (22).
Ig = Ig ^* (22)
As described above, in the first period of FIG. 26 in which the collector-emitter voltage Vce changes greatly when the IGBT element 1 is turned off, the relationship between the gate-collector voltage Vgc and the gate current Ig is expressed by the equation (15). Therefore, the gate-collector voltage Vce of the IGBT element 1 is given by the equation (23) from the gate-collector capacitance 1h, the reciprocal means 1i, the multiplying means 1j, and the integrating means 1k.
[0063]
[Formula 6]

[0064]
Further, since the gate-collector voltage Vgc and the collector-emitter voltage Vce are in the relationship of equation (16), the collector-emitter voltage Vce is given by equation (24) by the adding means 1m. The collector-emitter voltage Vce is differentiated by the differentiating means 1n to obtain the collector-emitter voltage change rate dVce / dt. As described above, during this first period, the gate-emitter voltage Vge is Since the voltage is almost constant, it can be transformed as shown in equation (25).
[0065]
[Expression 7]

[0066]
Therefore, in the configuration of FIG. 6, the output dVce / dt of the voltage change rate setting means 21 ^* Is kept constant, the collector-emitter voltage change rate of the IGBT element 1 can be kept constant.
[0067]
FIG. 7 is a diagram in which the block in FIG. 6 is rewritten when the voltage characteristic of the gate-collector capacitance is expressed by the following equation (26). In FIG. 7, 22a is the gate-collector capacitance of the

IGBT element

1, 22b is a multiplication means for obtaining the product of the output of 22a and the output of the voltage change rate setting means 21, and 22c is a gate-collector used in 22a. This is a detector for obtaining the inter-voltage Vgc. Others are the same as in FIG. Next, the operation will be described with reference to FIG. In the first period of FIG. 26 in which the collector-emitter voltage changes greatly when the IGBT element 1 is turned off, the gate-collector capacitance Cgc generally has a voltage dependency of equation (26). Here, Cjo is a zero bias capacity, Vj is a junction potential difference, and M is a junction slope coefficient. Even when the potential dependence of the equation (26) is applied to the equation (15), the relationship of the equations (15) to (25) is established.
[0068]
[Equation 8]

[0069]
Although only the operation at the time of turn-off has been described above, it is apparent that the same effect can be obtained with respect to the turn-on because the polarity of the change rate of the voltage and current of each part of the IGBT element is only reversed. This method provides the simplest configuration when there is no restriction on the switching characteristics of the IGBT element other than the collector-emitter voltage change rate. The equations (15) to (26) are applied to a period in which the collector current is substantially constant and the collector-emitter voltage changes greatly as in the first period in FIG. ) To (26) It is possible to drive the IGBT element with the gate current obtained using the equations.
[0070]
Embodiment 3 FIG.
In this embodiment, during the turn-off switching operation, the change rate of the collector-emitter voltage is kept substantially constant during the first period in which the collector-emitter voltage changes greatly, and the collector current Ic changes rapidly. In the second period, the change rate of the collector current is kept substantially constant.
FIG. 8 is a block diagram showing a semiconductor element drive circuit according to the third embodiment of the present invention. In FIG. 8, reference numeral 23 designates switching between the output of the first function generating means 20 and the output of the second function generating means 22. The switching means 24 for supplying the gate current generating means 18 with a current change from the first period when the change of the current flowing through the IGBT element 1 is slow when the IGBT element 1 is changed from the on state to the off state. A current change point detecting means 18 for detecting a current change point at the time of transition to the steep second period is a gate current generating means 18 as a current generating means. The current corresponding to the outputs of the two function generating means 20 and 21 is supplied to the gate 1 a of the IGBT element 1. Others are the same as those in the first embodiment or the second embodiment, and thus the description thereof is omitted.
[0071]
Next, the operation will be described. Since the operation when the switching unit 23 is on the first function generating unit 20 side is the same as that of the first embodiment, the description thereof is omitted. Further, since the operation when the second function generating means 22 is performed is the same as that of the second embodiment, the description thereof is omitted. Here, the switching means 23 and the current change point detection means 24 will be described. In FIG. 26 showing the turn-off waveform of the IGBT element 1, the collector-emitter voltage of the IGBT element 1 greatly changes during the first period of FIG. 26, and the collector current does not change during this period. The collector current changes during the second period of FIG. 26, and the collector-emitter voltage does not change significantly during this period. As described above, the second period in which the first embodiment exhibits the effect and the first period in which the second embodiment exhibits the effect are separated at time t3 in FIG.
[0072]
For this reason, the current change point detection means 24 detects the time t3 which is the boundary between the first period and the second period, and uses the output of the second function generation means 22 in the first period before the time t3. If the switching means 23 is controlled as described above and the switching means 23 is controlled to use the output of the first function generating means 20 in the second period after time t3, the second embodiment and The same effect as in the first embodiment can be obtained. Therefore, according to the configuration of FIG. 8, the change rate of the collector-emitter voltage or the change rate of the collector current can be kept substantially constant in the first and second periods during the turn-off of the IGBT element 1.
[0073]
9 to 16 are block diagrams showing first to seventh configuration examples of the current change point detecting means 24. First, a first configuration example of the current change point detection unit 24 will be described with reference to FIG. Reference numeral 24a denotes first voltage detection means for detecting the collector-emitter voltage Vce of the

IGBT element

1, 24b denotes second voltage detection means for detecting the voltage Vdc of the

DC voltage source

12, and 24c denotes comparison means. From FIG. 26, since the collector-emitter voltage Vce and the voltage Vdc of the DC voltage source 12 intersect at time t3 which is the boundary between the first period and the second period, the first voltage detection is performed by the comparison unit 24c. The time t3 can be detected by comparing the outputs of the means 24a and the second voltage detection means 24b. In applications where the voltage of the DC voltage source 12 is constant, the same effect can be obtained even if the second voltage detection means 24b is replaced with another DC voltage source as a reference for comparison.
[0074]
Next, a second configuration example of the current change point detection unit 24 will be described with reference to FIG. Reference numeral 24a denotes voltage detection means for detecting the gate-emitter voltage Vge of the

IGBT element

1, 24c denotes comparison means, and 24d denotes a DC voltage source serving as a reference for comparison. From FIG. 26, at time t3, which is the boundary between the first period and the second period, the gate-emitter voltage Vge starts to decrease from the gate voltage Von required to flow the load current IL. By setting the voltage of the voltage source 24d slightly lower than Von, the time t3 can be detected by the comparison unit 24c.
[0075]
Next, a third configuration example of the current change point detection unit 24 will be described with reference to FIG. 24a is a voltage detection means for detecting the gate-emitter voltage Vge of the

IGBT element

1, 24e is a differentiation means for differentiating the output of the voltage detection means 24a, 24c is a comparison means, and 24d is a voltage change rate of the gate-emitter voltage Vge. This is a DC voltage source that is a reference for comparison. As shown in FIG. 26, the gate-emitter voltage Vge starts to decrease from the gate voltage Von required to flow the load current IL at time t3 which is the boundary between the first period and the second period. The time t3 can be detected by differentiating the emitter-to-emitter voltage Vge to determine its rate of change and comparing it with the reference value from the DC voltage source 24d.
[0076]
Next, a fourth configuration example of the current change point detection unit 24 will be described with reference to FIG. Reference numeral 24f denotes first current detection means for detecting the collector current Ic of the IGBT element 1, 24g denotes second current detection means for detecting the current IL of the

load

13, and 24c denotes comparison means. 26, since the collector current Ic starts to decrease from the current IL of the load 13 at time t3 which is the boundary between the first period and the second period, the comparison unit 24c and the first current detection unit 24f The time t3 can be detected by comparing the outputs of the two voltage detection means 24g.
[0077]
Next, a fifth configuration example of the current change point detection unit 24 will be described with reference to FIG. Reference numeral 24f denotes current detection means for detecting the collector current Ic of the IGBT element 1, reference numeral 24c denotes comparison means, and reference numeral 24h denotes a DC voltage source serving as a reference for comparison. 26, at time t3, which is the boundary between the first period and the second period, the collector current Ic starts to decrease from the current IL of the load 13, so the voltage of the DC voltage source 24h is set slightly lower than the load current IL. By doing so, the time t3 can be detected by the comparison means 24c.
[0078]
Next, a sixth configuration example of the current change point detection unit 24 will be described with reference to FIG. 24f is a current detection means for detecting the collector current Ic of the

IGBT element

1, 24e is a differentiation means for differentiating the output of the current detection means 24f, 24c is a comparison means, and 24h is a reference for comparison of the current change rate of the collector current Ic. DC voltage source. 26, at time t3, which is the boundary between the first period and the second period, the collector current Ic starts to decrease from the load current IL. Therefore, the collector current Ic is differentiated to obtain the rate of change, and the DC voltage The time t3 can be detected by comparing the reference value with the source 24h.
[0079]
Next, a seventh configuration example of the current change point detection unit 24 will be described with reference to FIG. Reference numeral 24f denotes current detection means for detecting the current Id of the diode 14, reference numeral 24c denotes comparison means, and reference numeral 24h denotes a DC voltage source serving as a reference for comparison. From FIG. 26, at time t3, which is the boundary between the first period and the second period, the current Id of the diode 14 starts to increase from zero, so the voltage of the DC voltage source 24h is set slightly higher than zero. Thus, the time t3 can be detected by the comparison means 24c.
[0080]
Next, an eighth configuration example of the current change point detection unit 24 will be described with reference to FIG. 24f is a current detection means for detecting the current Id of the

diode

14, 24e is a differentiation means for differentiating the output of the current detection means 24f, 24c is a comparison means, and 24h is a reference for comparison of the current change rate of the current Id of the diode 14. DC voltage source. 26, at time t3 which is the boundary between the first period and the second period, the current Id of the diode 14 starts to increase from zero. Therefore, the current Id of the diode 14 is differentiated to obtain the rate of change, and the direct current The time t3 can be detected by comparing the reference value with the voltage source 24h.
[0081]
Embodiment 4 FIG.
In this embodiment, during the turn-on switching operation, the change rate of the collector-emitter voltage is kept substantially constant during the first period, and the change rate of the collector current is kept almost constant during the second period. is there.
The configuration in this embodiment is the same as that shown in FIG. In FIG. 8, reference numeral 24 denotes a transition from the second period in which the current flowing through the IGBT element 1 changes sharply to the first period in which the subsequent change in current slows down when the IGBT element 1 is changed from the OFF state to the ON state. Current changing point detecting means for detecting a current changing point when The specific configuration of the current change point detecting means 24 can be the same as or equivalent to that shown in FIGS. 9 to 16, and both are the voltages of the

DC voltage sources

24d and 24h serving as a reference for comparison, or differentiating means. Only the output polarity of 24e is different. Others are the same as those of the third embodiment, and thus description thereof is omitted.
[0082]
Next, the operation will be described. Since the operation when the switching unit 23 is on the first function generating unit 20 side is the same as that of the first embodiment, the description thereof is omitted. Further, since the operation when the second function generating means 22 is performed is the same as that of the second embodiment, the description thereof is omitted. Here, the switching means 23 and the current change point detection means 24 will be described. In FIG. 27 showing the turn-on waveform of the IGBT element 1, the collector-emitter voltage of the IGBT element 1 greatly changes during the first period of FIG. 27, and the collector current does not change during this period. The collector current changes during the second period of FIG. 27, and the collector-emitter voltage does not change significantly during this period. As described above, the second period in which the first embodiment exhibits the effect and the first period in which the second embodiment exhibits the effect are those of the freewheeling diode 14 between time t8 and time t9 in FIG. It is separated by the recovery operation period.
[0083]
For this reason, the current change point detecting means 24 detects the recovery operation period (t8 to t9) of the freewheeling diode 14 which is the boundary between the first period and the second period, and the first period is the first period before the time t8. The switching means 23 is controlled so as to use the output of the function generating means 20, and the switching means 23 is controlled so as to use the output of the second function generating means 22 in the first period after time t9. In each period, the same effect as in the first and second embodiments can be obtained. Therefore, according to the configuration of FIG. 8, the change rate of the collector current or the change rate of the collector-emitter voltage can be kept substantially constant in the second and first periods during the turn-on of the IGBT element 1, respectively.
Although the embodiments 3 and 4 are applied to turn-on and turn-off, respectively, they can be applied to both turn-on and turn-off in combination.
[0084]
Embodiment 5 FIG.
FIG. 17 is a block diagram of a semiconductor element drive circuit according to the fifth embodiment of the present invention, and shows a configuration in the case where the rate of change of the collector current during the switching operation is kept substantially constant. In the figure, reference numeral 25 denotes a first capacitor connected between the gate and the emitter of the IGBT element 1, and the other components are the same as those in the first embodiment, and the description thereof is omitted.
Next, the operation will be described. FIG. 17 is the same as FIG. 2 except for the first capacitor 25, and the first capacitor 25 is connected in parallel with the gate-emitter capacitance Cge of the IGBT element 1. For this reason, assuming that the capacitance of the first capacitor 25 is C1, this is equivalent to an increase in the gate-emitter capacitance of the IGBT element 1 from Cge to (Cge + C1). Therefore, the output Ig of the first function generating means 20 ^* If the equation (27) obtained by changing the gate-emitter voltage capacitance Cge in the equation (6) is obtained, the same effect as that of the first embodiment can be obtained even in the configuration of FIG.
[0085]
[Equation 9]

[0086]
According to this embodiment, since the gate current Ig of the IGBT element 1 can be adjusted by C1, the gate current Ig is set to the command value dIc / dt of the collector current change rate. ^* Independent of the characteristics f and Cge of the IGBT element 1, the gate current value can be set for the convenience of the drive circuit. Further, in applications where switching is performed as in the third and fourth embodiments, the capacitance at the time of switching is set by setting the capacitance of the first capacitor 25 so that the gate currents Ig before and after switching are equal. Smooth switching can be realized by preventing a sudden change in the current Ig.
[0087]
Embodiment 6 FIG.
FIG. 18 is a block diagram of a semiconductor element drive circuit according to the sixth embodiment of the present invention, and shows a configuration in which the rate of change of the collector-emitter voltage during the switching operation is kept substantially constant. In the figure, reference numeral 26 denotes a second capacitor connected between the collector and gate of the IGBT element 1, and the others are the same as those in the second embodiment, so that the description thereof is omitted.
Next, the operation will be described. FIG. 18 is the same as FIG. 5 except for the second capacitor 26, and the second capacitor 26 is connected in parallel to the collector-gate capacitance Cgc of the IGBT element 1. Therefore, assuming that the capacitance of the second capacitor 26 is C2, this is equivalent to the increase in the capacitance between the collector and gate of the IGBT element 1 from Cgc to (Cgc + C2). Therefore, the output Ig of the second function generating means 22 ^* 18 is the expression (28) obtained by changing the collector-gate capacitance Cgc of the expression (21), the same effect as that of the second embodiment can be obtained in the configuration of FIG.
Ig ^* = (Cgc + C2) · dVce / dt ^* ... (28)
[0088]
According to this embodiment, since the gate current Ig of the IGBT element 1 can be adjusted by C2, the gate current Ig is changed to the command value dVce / dt of the collector-emitter voltage change rate. ^* Independently of the characteristics Cgc of the IGBT element 1, the gate current value can be set for the convenience of the drive circuit. Further, in applications where switching is performed as in the third and fourth embodiments, the capacitance at the time of switching is set by setting the capacitance of the second capacitor 26 so that the gate currents Ig before and after switching are equal. Smooth switching can be realized by preventing a sudden change in the current Ig. Further, since the second capacitor 26 is a capacitor having a constant capacitance C2, the influence of the voltage dependency of the collector-gate capacitance Cgc as shown in the equation (26) is suppressed, and the gate current is reduced. There is an effect of reducing the change in Ig.
[0089]
Embodiment 7 FIG.
FIG. 19 is a circuit diagram showing a drive circuit for a semiconductor element according to the seventh embodiment of the present invention, and shows a configuration when the rate of change of the collector current during the switching operation is kept substantially constant. In FIG. 19, the gate current generating means 18 comprises transistors 18a and 18b constituting a current mirror circuit, the current change rate setting means 19 comprises a DC voltage source 19a and a transistor 19b, and first function generating means 20 Consists of a resistor 20f and

transistors

20g and 20h constituting a current mirror circuit. Others are the same as those in the first embodiment, and thus the description thereof is omitted. In FIG. 19, only the circuit part for turn-off is shown for simplification.
[0090]
Next, the operation will be described. When the switching signal generating means 4 outputs an off signal, the transistor 19b is turned off. Since the current mirror circuit composed of the

transistors

20g and 20h is connected to the voltage Vcc of the DC voltage source 19a, and the transistor 20g is connected to the gate of the IGBT element 1 through the resistor 20f, the collector current Ig of the transistor 20h. ^* Is given by equation (29) if the base-emitter voltage drop of the transistor is ignored. Here, K1 is the current gain of the current mirror circuit, R1 is the resistance value of the resistor 20f, and Vge is the gate-emitter voltage of the IGBT element 1.
Ig ^* = K1 · (Vcc−Vge) / R1 (29)
[0091]
Collector current Ig of transistor 20h in the above equation ^* Is an output of the first function generating means 20 and a gate current command. Since the transistor 19b is off, the collector current Ig of the transistor 20h ^* All flow through the transistor 18a. Since the transistors 18a and 18b form a current mirror circuit, the collector current Ig of the transistor 20h is connected to the gate of the IGBT element 1. ^* Current that is multiplied by the gain of the current mirror circuit composed of transistors 18a and 18b. Comparing equation (29) with equation (6) of the first embodiment, the gate-emitter capacitance Cge and the collector current change rate dIc / dt of the IGBT element 1 ^* Is a constant, and the primary expression of the gate-emitter voltage is used as the reciprocal of the derivative of the transfer function from the gate-emitter voltage of the IGBT element 1 to the collector current.
FIG. 20A is a graph showing the relationship between the gate-emitter voltage of the IGBT element 1 and the inverse of the derivative of the transfer function from the gate-emitter voltage to the collector current. From the equation (14) shown in the first embodiment, the collector current change rate dIc / dt ^* Gate current command Ig when A is a constant A ^* Is the following equation (30).
[0092]
[Expression 10]

[0093]
The reciprocal of the derivative of the transfer function from the gate-emitter voltage to the collector current of the IGBT element 1 is a 1 / (2 · Kp · (Vge−Vth)) portion on the right side of the equation (30), which is a hyperbolic function. ing. As can be seen from FIG. 20A, a good approximation can be made by a linear expression of the gate-emitter voltage, for example, the expression (29). Therefore, by using the linear expression of the gate-emitter voltage as the reciprocal of the derivative of the transfer function from the gate-emitter voltage to the collector current, the same effect as in the first embodiment can be realized with a simple circuit.
[0094]
Embodiment 8 FIG.
FIG. 21 is a circuit diagram showing a semiconductor element drive circuit according to the eighth embodiment of the present invention, and shows a structure in the case where the rate of change of the collector current during the switching operation is kept substantially constant. In FIG. 21, the gate current generating means 18 comprises transistors 18a and 18b constituting a current mirror circuit, and the current change rate setting means 19 comprises a DC voltage source 19a and a transistor 19b. Reference numeral 20j denotes a resistor as the first function generating means. Others are the same as those of the first embodiment, and thus the description thereof is omitted. In FIG. 21, only the circuit part for turn-off is shown for simplification.
[0095]
Next, the operation will be described. When the switching signal generating means 4 outputs an off signal, the transistor 19b is turned off. Since the resistor 20j is connected to the voltage Vcc of the DC voltage source 19a, the current Ig flowing through the resistor 20j ^* Is given by equation (31) if the base-emitter voltage drop of the transistor is ignored. Here, R1 is the resistance value of the resistor 20j.
Ig ^* = Vcc / R1 (31)
Current Ig of resistor 20j in the above equation ^* Is an output of the first function generating means and a gate current command. Since the transistor 19b is off, the current Ig of the resistor 20j ^* All flow through the transistor 18a. Since the transistors 18a and 18b constitute a current mirror circuit, the current Ig of the resistor 20j is connected to the gate of the IGBT element 1. ^* Current that is multiplied by the gain of the current mirror circuit composed of transistors 18a and 18b. When the equation (31) is compared with the equation (6) of the first embodiment, the gate-emitter capacitance Cge of the IGBT element 1 and the collector current change rate dIc / dt ^* Is a constant, and the constant is also used for the reciprocal of the derivative of the transfer function from the gate-emitter voltage of the IGBT element 1 to the collector current.
[0096]
FIG. 20B is a graph showing the relationship between the gate-emitter voltage of the IGBT element 1 and the inverse of the derivative of the transfer function from the gate-emitter voltage to the collector current. The reciprocal of the derivative of the transfer function from the gate-emitter voltage to the collector current of the IGBT element 1 is a hyperbolic function as described in the seventh embodiment. As can be seen from FIG. 20B, when the reciprocal of the derivative of the transfer function from the gate-emitter voltage of the IGBT element 1 to the collector current is a hyperbolic function, a constant cannot be said to be a good approximation. However, when the IGBT element has a characteristic with little change, it is possible to obtain a good result even with a constant. In such a case, by using a constant that is a representative value as the inverse of the derivative of the transfer function from the gate-emitter voltage to the collector current, the same effect as in the first embodiment can be realized with a simple circuit.
[0097]
Embodiment 9 FIG.
FIG. 22 is a circuit diagram showing a drive circuit for a semiconductor element according to the ninth embodiment of the present invention, and shows a configuration when the rate of change of the collector-emitter voltage during the switching operation is kept substantially constant. In FIG. 22, the gate current generating means 18 is composed of transistors 18a and 18b constituting a current mirror circuit, the voltage change rate setting means 21 is composed of a DC voltage source 21a and a transistor 21b, and second function generating means 22 Consists of resistors 22d and 22e and transistors 22f and 22g constituting a current mirror circuit. Others are the same as those of the second embodiment, and thus description thereof is omitted. In FIG. 22, only the circuit part for turn-off is shown for simplification.
[0098]
Next, the operation will be described. When the switching signal generating means 4 outputs an off signal, the transistor 21b is turned off. Since the resistor 22d is connected to the voltage Vcc of the DC voltage source 21a, and the transistor 22g is connected to the collector of the IGBT element 1 via the resistor 22e, the output of the second function generator 22 is the base of the transistor. -If the voltage drop between emitters is disregarded, it is given by equation (32). Here, K2 is the current gain of the current mirror circuit, R2 is the resistance value of the resistor 22d, R3 is the resistance value of the resistor 22e, and Vce is the collector-emitter voltage of the IGBT element 1.
Ig ^* = Vcc / R2-K2 / Vce / R3 (32)
[0099]
Ig of the above formula ^* Is an output of the second function generating means 22 and a gate current command. Since the transistor 21b is off, the gate current command Ig in the equation (32) ^* All flow through the transistor 18a. Since the transistors 18a and 18b constitute a current mirror circuit, the gate of the IGBT element 1 is connected to the gate current command Ig. ^* Current that is multiplied by the gain of the current mirror circuit composed of transistors 18a and 18b. Comparing the equation (32) with the equation (21) of the second embodiment, the collector-emitter voltage change rate dVce / dt of the IGBT element 1 ^* Is a constant, and the primary expression of the collector-emitter voltage is used as the collector-gate capacitance Cgc of the IGBT element 1.
[0100]
FIG. 23A is a graph showing the voltage dependence of the collector-gate capacitance Cgc of the IGBT element 1. From the equation (26) of the second embodiment, the collector-emitter voltage change rate dVce / dt ^* Is a constant B when the gate current command Ig ^* Becomes the equation (33).
[0101]
[Expression 11]

[0102]
The capacitance Cgc between the collector and gate of the IGBT element 1 is Cjo · (1 + Vgc / Vj) on the right side of the equation (33). ^-M This is a hyperbolic function. As can be seen from (a) of FIG. 23, for example, a good approximation can be made by a first-order expression of the collector-emitter voltage such as the expression (32). Therefore, by using a primary expression of the collector-emitter voltage as the collector-gate capacitance Cgc, the same effect as in the second embodiment can be realized with a simple circuit. 22 can change the connection point of the resistor 22e from the collector of the IGBT element 1 to the gate, thereby realizing the same function as that of the seventh embodiment shown in FIG.
[0103]
Embodiment 10 FIG.
FIG. 24 is a circuit diagram showing a semiconductor element drive circuit according to the tenth embodiment of the present invention, and shows a configuration in the case where the rate of change of the collector-emitter voltage during the switching operation is kept substantially constant. In FIG. 24, the gate current generating means 18 comprises transistors 18a and 18b constituting a current mirror circuit, and the voltage change rate setting means 21 comprises a DC voltage source 21a and a transistor 21b. 22h is a resistor as the second function generating means. Others are the same as those of the second embodiment, and thus description thereof is omitted. In FIG. 24, only the circuit part for turn-off is shown for simplification.
[0104]
Next, the operation will be described. When the switching signal generating means 4 outputs an off signal, the transistor 19b is turned off. Since the resistor 22h is connected to the voltage Vcc of the DC voltage source 19a, the current Ig flowing through the resistor 22h ^* Is given by equation (34) if the base-emitter voltage drop of the transistor is ignored. Here, R2 is the resistance value of the resistor 22h.
Ig ^* = Vcc / R2 (34)
Current Ig of resistor 22h in the above equation ^* Is an output of the second function generating means and a gate current command. Since the transistor 19b is off, the current Ig of the resistor 22h ^* All flow through the transistor 18a. Since the transistors 18a and 18b constitute a current mirror circuit, the current Ig of the resistor 22h is connected to the gate of the IGBT element 1. ^* Current that is multiplied by the gain of the current mirror circuit composed of transistors 18a and 18b. Comparing the equation (34) with the equation (21) of the second embodiment, the collector-emitter voltage change rate dVce / dt of the IGBT element 1 ^* It can be seen that the constant is also used for the collector-gate capacitance Cgc of the IGBT element 1.
[0105]
FIG. 23B is a graph showing the voltage dependence of the collector-gate capacitance Cgc of the IGBT element 1. The collector-gate capacitance Cgc of the IGBT element 1 is a hyperbolic function as described in the ninth embodiment. As can be seen from (b) of FIG. 23, when the collector-gate capacitance Cgc of the IGBT element 1 is a hyperbolic function, it cannot be said that the constant is a good approximation. However, when the IGBT element has a characteristic with little change, it is possible to obtain a good result even with a constant. Further, as shown in the sixth embodiment, when a capacitor is connected in parallel to the collector-gate capacitance Cgc, the influence of the change in the collector-gate capacitance Cgc is reduced. Good results can be obtained. Therefore, by using a constant which is a representative value as the collector-gate capacitance Cgc, the same effect as in the second embodiment can be realized with a simple circuit.
[0106]
In the seventh to tenth embodiments, the case of controlling the current change rate of the collector or the case of controlling the collector-emitter voltage change rate has been described, but both may be combined. In addition, although a circuit for turn-off is shown, it can be controlled approximately by a simple circuit for turn-on, and both can be used together.
[0107]
【The invention's effect】
According to the invention of claim 1, the inverse function of the transfer function from the gate current to the collector current change rate of the voltage-driven self-extinguishing semiconductor device The command of the gate current obtained by substituting the output of the current change rate setting means into Since the first function generating means is generated, the non-linearity of the semiconductor element is corrected regardless of variations in the load and the power supply voltage, and the desired switching characteristic, which is a set switching characteristic, is constant. Characteristics are obtained. According to the invention of claim 2, the inverse function of the transfer function from the gate current to the collector-emitter voltage change rate of the voltage-driven self-extinguishing semiconductor device The command of the gate current obtained by substituting the output of the voltage change rate setting means into Since the second function generating means is generated, the non-linearity of the semiconductor element is corrected regardless of fluctuations in the load and the power supply voltage, and the change in the collector-emitter voltage which is the desired switching characteristic set. A constant rate characteristic is obtained. For this reason, the rate of change of unnecessarily high or low voltage and current during switching operation can be suppressed, which is effective for reducing switching loss, reducing surge voltage at turn-off, and reducing recovery current of the freewheeling diode at turn-on. There is. Furthermore, it is effective in reducing noise and leakage current caused by switching.
[0108]
According to the third and fourth aspects of the present invention, the voltage change rate setting means and the second function generating means are used in the first period during which the collector current changes slowly, and the collector current changes rapidly. Since the current change rate setting means and the first function generating means are used in the period 2, the desired switching characteristics, for example, the change rate of the collector-emitter voltage is kept constant in the first period, and the first Characteristics such as keeping the rate of change of the collector current constant in the period 2 can be obtained.
[0109]
According to the invention of claim 5, the first function generating means Of transfer function from gate current to collector current change rate Is the reciprocal of the derivative of the transfer function from the gate-emitter voltage to the collector current, the gate-emitter capacitance, Is the product of collector current change rate Therefore, the same effects as those of the first, third, and fourth aspects can be obtained.
According to the invention of claim 6, the second function generating means Of transfer function from gate current to collector-emitter voltage change rate Is the gate-collector capacitance, This is the product of the collector-emitter voltage change rate. Therefore, the same effects as those of the second, third and fourth aspects can be obtained.
[0110]
According to the invention of claim 7, the first function generating means has a capacitor connected between the gate and the emitter. The inverse function of the transfer function from the gate current to the collector current change rate used in Fig. 4 is the inverse of the derivative of the transfer function from the gate-emitter voltage to the collector current, the gate-emitter capacitance, and the capacitance of the capacitor. It is the product of the sum of capacitance and the collector current change rate. Therefore, an appropriate gate current value can be set according to the convenience of the drive circuit.
According to the eighth aspect of the present invention, the second function generating means has a capacitor connected between the gate and the collector. The inverse function of the transfer function from the gate current to the collector-emitter voltage change rate used in the above is the product of the sum of the gate-collector capacitance and the capacitor capacitance and the collector-emitter voltage change rate. Is Therefore, an appropriate gate current value can be set according to the convenience of the drive circuit.
[0111]
According to the ninth aspect of the present invention, the capacitor is connected between the gate and the emitter, and the electrostatic capacity of the capacitor is set so that both outputs are equal to each other when the first and second function generating means are switched. Since the capacity is set, smooth switching can be performed.
According to the tenth aspect of the present invention, the capacitor is connected between the gate and the collector, and when the first and second function generating means are switched, the electrostatic capacity of the capacitor is set so that both outputs are equal to each other. Since the capacity is set, smooth switching can be performed.
[0112]
According to the invention of claim 11, the first function generating means In the inverse function of the transfer function from the gate current to the collector current change rate used in Instead of the inverse of the derivative of the transfer function from the gate-emitter voltage to the collector current, it is approximated by a linear expression of the gate-emitter voltage, so the collector current change rate is controlled with a simple circuit. be able to.
According to the invention of claim 12, the first function generating means In the inverse function of the transfer function from the gate current to the collector current change rate used in Instead of the inverse of the derivative of the transfer function from the gate-emitter voltage to the collector current, it is approximated by a constant which is a representative value, so that the collector current change rate can be controlled with a simple circuit.
According to the invention of claim 13, the second function generating means In the inverse function of the transfer function from the gate current to the collector-emitter voltage change rate used in Instead of the gate-collector capacitance, the gate-collector voltage is approximated by a linear expression so that the voltage change rate between the collector and the emitter can be controlled with a simple circuit.
According to the invention of claim 14, the second function generating means In the inverse function of the transfer function from the gate current to the collector-emitter voltage change rate used in Instead of the capacitance between the gate and the collector, it is approximated with a constant which is a representative value, so that the voltage change rate between the collector and the emitter can be controlled with a simple circuit.
[Brief description of the drawings]
FIG. 1 is a block diagram showing a basic configuration of a drive circuit for a semiconductor element according to an embodiment of the present invention.
FIG. 2 is a block diagram of a semiconductor element drive circuit according to the first embodiment of the present invention.
FIG. 3 is a block diagram showing an internal functional configuration of the semiconductor element and its drive circuit according to the first embodiment of the present invention.
FIG. 4 is a block diagram showing an internal functional configuration of a semiconductor element and its drive circuit according to Embodiment 1 of the present invention.
FIG. 5 is a block diagram of a semiconductor element drive circuit according to a second embodiment of the present invention.
FIG. 6 is a block diagram showing an internal functional configuration of a semiconductor element and its drive circuit according to Embodiment 2 of the present invention.
FIG. 7 is a block diagram showing an internal functional configuration of a semiconductor element and its drive circuit according to Embodiment 2 of the present invention.
FIG. 8 is a block diagram of a semiconductor element drive circuit in the third and fourth embodiments of the present invention.
FIG. 9 is a block diagram illustrating a configuration example of a current change point detection unit according to Embodiment 3 of the present invention.
FIG. 10 is a block diagram showing a configuration example of a current change point detection unit in Embodiment 3 of the present invention.
FIG. 11 is a block diagram illustrating a configuration example of a current change point detection unit according to Embodiment 3 of the present invention.
FIG. 12 is a block diagram showing a configuration example of a current change point detection unit according to Embodiment 3 of the present invention.
FIG. 13 is a block diagram illustrating a configuration example of a current change point detection unit according to Embodiment 3 of the present invention.
FIG. 14 is a block diagram illustrating a configuration example of a current change point detection unit according to Embodiment 3 of the present invention.
FIG. 15 is a block diagram illustrating a configuration example of a current change point detection unit according to Embodiment 3 of the present invention.
FIG. 16 is a block diagram showing a configuration example of a current change point detection unit according to Embodiment 3 of the present invention.
FIG. 17 is a block diagram of a semiconductor element drive circuit according to a fifth embodiment of the present invention.
FIG. 18 is a block diagram of a semiconductor element drive circuit according to a sixth embodiment of the present invention.
FIG. 19 is a circuit diagram of a semiconductor element drive circuit according to a seventh embodiment of the present invention.
FIG. 20 is a graph showing the reciprocal of the derivative of the transfer function from the gate-emitter voltage to the collector voltage in the seventh and eighth embodiments of the present invention.
FIG. 21 is a circuit diagram of a semiconductor element drive circuit according to an eighth embodiment of the present invention.
FIG. 22 is a circuit diagram of a semiconductor element drive circuit according to a ninth embodiment of the present invention.
FIG. 23 is a graph showing the voltage dependence of the collector-gate capacitance in the ninth and tenth embodiments of the present invention.
FIG. 24 is a circuit diagram of a semiconductor element drive circuit according to a tenth embodiment of the present invention.
FIG. 25 is a circuit diagram for illustrating switching characteristics of the IGBT element.
FIG. 26 is a waveform diagram showing a turn-off operation of the IGBT element.
FIG. 27 is a waveform diagram showing a turn-on operation of the IGBT element.
FIG. 28 is a circuit diagram of a conventional semiconductor element drive circuit.
FIG. 29 is a waveform diagram showing an operation of a conventional semiconductor element drive circuit.
[Explanation of symbols]
1 IGBT element, 1a gate, 1b collector, 1c emitter,
4 switching signal generating means, 16 switching characteristic setting means,
17 function generating means, 18 gate current generating means, 19 current change rate setting means,
20 first function generating means, 21 voltage change rate setting means,
22 second function generating means, 23 switching means, 24 current change point detecting means,
25 First capacitor, 26 Second capacitor.

Claims

In a drive circuit connected between the gate and emitter of a voltage-driven self-extinguishing semiconductor element that controls the conduction state between the collector and the emitter by a voltage applied between the gate and the emitter, the voltage-driven self-extinguishing semiconductor element is Switching signal generating means for generating an on / off signal, and current change rate setting means for generating a command for a collector current change rate of the voltage-driven self-extinguishing semiconductor element having a constant value by the output of the switching signal generating means; A first function generating means for generating a gate current command for the voltage-driven self-extinguishing semiconductor element by the output of the current change rate setting means; and the gate current is generated by the output of the first function generating means. And the first function generating means generates the voltage-driven self-extinguishing semiconductor element from the gate current. Of the inverse of the transfer function of the collector current change rate, the drive circuit of the semiconductor device is characterized in that so as to generate the command of the gate current obtained by substituting the output of the current change rate setting means.

In the drive circuit connected between the gate and the emitter of the voltage-driven self-extinguishing semiconductor element for controlling the conduction state between the collector and the emitter by the voltage applied between the gate and the emitter, the voltage-driven self-extinguishing semiconductor element Switching signal generating means for generating a signal for turning on / off the voltage, and a voltage change for generating a command for a rate of change in voltage between the collector and the emitter of the voltage-driven self-extinguishing semiconductor element having a constant value by the output of the switching signal generating means Rate setting means, second function generating means for generating a command of the gate current of the voltage-driven self-extinguishing semiconductor element by the output of the voltage change rate setting means, and output of the second function generating means Gate current generating means for generating a gate current, and the second function generating means is configured to detect the voltage-driven self-current from the gate current. The collector of the arc semiconductor device - semiconductor to an inverse function of a transfer function of the emitter voltage change rate, characterized by being adapted to generate a command gate current obtained by substituting the output of the voltage change rate setting means Device drive circuit.

A voltage-driven self-extinguishing semiconductor device connected between the gate and emitter of a voltage-driven self-extinguishing semiconductor device that controls a conduction state between a collector and an emitter by a voltage applied between a gate and an emitter. Switching signal generating means for generating an on / off signal, and current change rate setting means for generating a command of a collector current change rate of the voltage-driven self-extinguishing semiconductor element having a constant value by the output of the switching signal generating means First function generating means for generating a gate current command of the voltage-driven self-extinguishing semiconductor element based on the output of the current change rate setting means; and the voltage driving at a constant value based on the output of the switching signal generating means. Voltage change rate setting means for generating a command of the voltage change rate between the collector and the emitter of the self-extinguishing semiconductor element, and the voltage change The second function generating means for generating a command for the gate current of the voltage-driven self-extinguishing semiconductor element by the output of the rate setting means and the first and second function generating means are switched to output one of them. Switching means for outputting, gate current generating means for generating the gate current by the output of the switching means, and at the time of transition from the collector-emitter conduction state to the non-conduction state by the gate-emitter voltage control Current change point detecting means for detecting a change point from the first period in which the change in the collector current is slow to the second period in which the subsequent change in the collector current is steep, and the first function generating means includes: , the inverse function of the transfer function from the gate current to the collector current change rate of the voltage drive self-turn-off semiconductor device, obtained by substituting the output of the current change rate setting means Generates a command over preparative current, the second function generating means, from the gate current collector of said voltage-driven self-turn-off semiconductor devices - the inverse of the transfer function of the emitter voltage change rate, the voltage A command for a gate current obtained by substituting the output of the change rate setting means is generated, and the switching means sends the second function generating means in the first period, and in the second period. A circuit for driving a semiconductor device, wherein the switching is made to the first function generating means.

A voltage-driven self-extinguishing semiconductor device connected between the gate and emitter of a voltage-driven self-extinguishing semiconductor device that controls a conduction state between a collector and an emitter by a voltage applied between a gate and an emitter. Switching signal generating means for generating an on / off signal, and current change rate setting means for generating a command of a collector current change rate of the voltage-driven self-extinguishing semiconductor element having a constant value by the output of the switching signal generating means First function generating means for generating a gate current command of the voltage-driven self-extinguishing semiconductor element based on the output of the current change rate setting means; and the voltage driving at a constant value based on the output of the switching signal generating means. Voltage change rate setting means for generating a command of the voltage change rate between the collector and the emitter of the self-extinguishing semiconductor element, and the voltage change The second function generating means for generating a command for the gate current of the voltage-driven self-extinguishing semiconductor element by the output of the rate setting means and the first and second function generating means are switched to output one of them. Switching means for outputting, gate current generating means for generating the gate current by the output of the switching means, and at the time of transition from the non-conducting state between the collector and the emitter to the conducting state by voltage control between the gate and the emitter Current change point detecting means for detecting a change point from the second period in which the collector current changes sharply to the first period in which the subsequent change in collector current changes slowly, and generating the first function the resulting unit is the inverse function of the transfer function from the gate current to the collector current change rate of the voltage drive self-turn-off semiconductor devices, by substituting the output of the current change rate setting means Generates a command gate current, the second function generating means, from the gate current collector of said voltage-driven self-turn-off semiconductor devices - the inverse of the transfer function of the emitter voltage change rate, the voltage A command for a gate current obtained by substituting the output of the rate of change setting means is generated, and the switching means is directed to the first function generating means in the second period, and in the first period. A circuit for driving a semiconductor device, wherein the switching is made to the second function generating means.

The inverse function of the transfer function from the gate current used in the first function generating means to the collector current change rate of the voltage-driven self-extinguishing semiconductor element is the gate-emitter of the voltage-driven self-extinguishing semiconductor element. 5. The product of the reciprocal of the derivative of the transfer function from the inter-voltage to the collector current, the capacitance between the gate and the emitter, and the rate of change of the collector current. A drive circuit for a semiconductor device according to any one of the above.

The inverse function of the transfer function from the gate current used in the second function generating means to the collector-emitter voltage change rate of the voltage-driven self-extinguishing semiconductor element is the voltage-driven self-extinguishing semiconductor element. 5. The drive circuit for a semiconductor device according to claim 2, wherein the drive circuit is a product of a capacitance between a gate and a collector and a voltage change rate between the collector and the emitter .

A collector current change of the voltage-driven self-extinguishing semiconductor element from the gate current used in the first function generating means , having a capacitor connected between the gate and emitter of the voltage-driven self-extinguishing semiconductor element The inverse function of the transfer function to the ratio is the reciprocal of the derivative of the transfer function from the gate-emitter voltage to the collector current of the voltage-driven self-extinguishing semiconductor device, the capacitance between the gate-emitter and the capacitor. 5. The semiconductor element driving circuit according to claim 1, wherein the driving circuit is a product of a sum of capacitances and a collector current change rate .

A collector connected between the gate and collector of the voltage-driven self-extinguishing semiconductor element, and the collector-emitter of the voltage-driven self-extinguishing semiconductor element from the gate current used in the second function generating means The inverse function of the transfer function to the rate of change in voltage between voltages is the sum of the capacitance between the gate and collector of the voltage-driven self-extinguishing semiconductor element and the capacitance of the capacitor, and the rate of change in voltage between the collector and emitter. 5. The semiconductor element driving circuit according to claim 2, wherein the driving circuit is a product of:

A collector current change of the voltage-driven self-extinguishing semiconductor element from the gate current used in the first function generating means , having a capacitor connected between the gate and emitter of the voltage-driven self-extinguishing semiconductor element The inverse function of the transfer function to the rate is expressed as the inverse of the derivative of the transfer function from the gate-emitter voltage to the collector current of the voltage-driven self-extinguishing semiconductor element, the capacitance between the gate-emitter and the capacitor. The capacitance of the capacitor is set to be the product of the sum of the capacitance and the collector current change rate , and the outputs of the first and second function generating means are equal to each other during the switching operation of the switching means. 5. The semiconductor element drive circuit according to claim 3, wherein the drive circuit is set.

A collector connected between the gate and collector of the voltage-driven self-extinguishing semiconductor device, and the collector and emitter of the voltage-driven self-extinguishing semiconductor device from the gate current used in the second function generating means The inverse function of the transfer function to the voltage change rate between the voltage, the sum of the gate-collector capacitance of the voltage-driven self-extinguishing semiconductor element and the capacitance of the capacitor, the collector-emitter voltage change rate, The capacitance of the capacitor is set so that the outputs of the first and second function generating means are equal to each other during the switching operation of the switching means. Item 5. A drive circuit for a semiconductor element according to Item 4.

In the inverse function of the transfer function from the gate current used in the first function generating means to the collector current change rate of the voltage-driven self-extinguishing semiconductor element, the gate-emitter of the voltage-driven self-extinguishing semiconductor element 9. The first-order approximate expression of the gate-emitter voltage is used in place of the reciprocal of the derivative of the transfer function from the inter-voltage to the collector current. A drive circuit for a semiconductor device according to any one of the above.

In the inverse function of the transfer function from the gate current used in the first function generating means to the collector current change rate of the voltage-driven self-extinguishing semiconductor element, the gate-emitter of the voltage-driven self-extinguishing semiconductor element 10. The method according to claim 5, wherein the inverse of the derivative of the transfer function from the inter-voltage to the collector current is approximated by a constant that is a representative value. A drive circuit for a semiconductor device according to claim 1.

In the inverse function of the transfer function from the gate current used in the second function generating means to the collector-emitter voltage change rate of the voltage-driven self-extinguishing semiconductor element, the voltage-driven self-extinguishing semiconductor element 11. The first-order approximate expression of the gate-collector voltage is used in place of the gate-collector capacitance. The driving circuit of the semiconductor element.

In the inverse function of the transfer function from the gate current used in the second function generating means to the collector-emitter voltage change rate of the voltage-driven self-extinguishing semiconductor element, the voltage-driven self-extinguishing semiconductor element 11. The semiconductor element according to claim 6, wherein the semiconductor element is approximated by a constant which is a representative value instead of the capacitance between the gate and the collector. Drive circuit.