JPH09116867A - Frequency modulation circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a frequency modulation circuit which can set a deviated frequency to be non-regulated by adopting prescribed circuit constitution. SOLUTION: The modulation circuit consists of a current luminance signal generation block 1, a differential current amplifier circuit 2 and a current control type oscillation circuit 3. Voltage current conversion circuits Q8, Q6 and R2, offset current generation circuits Q9, Q11 and R3 and current addition circuits Q9, Q5 and Q8 are included in the current luminance signal generation block 1, The differential current amplifier circuit 2 is provided with a constant current source where a current value is adjusted to 2 × ICAR and it outputs control current proportional to the change of addition current outputted from the current addition circuit ICAR becomes the current value of control current in which the oscillation frequency of the current control type oscillation circuit 3 becomes the carrier frequency of the sink chip voltage of a voltage luminance signal, which is previosuly regulated. The oscillation frequency of the current control type oscillation circuit 3 is varied by outputted control current.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a frequency modulation circuit used in a video tape recorder or the like, and is particularly effective when applied to adjustment of the shift frequency in a frequency modulation circuit composed of a semiconductor integrated circuit. Related technology.

[0002]

2. Description of the Related Art In a video tape recorder (hereinafter referred to as "VTR"), a luminance signal is modulated by a frequency modulation circuit (FM modulation) because a tape head system cannot record up to direct current and a reproduction output includes level fluctuation. The circuit is frequency-modulated (FM-modulated) to a predetermined carrier frequency for recording.

In this case, the frequency modulation circuit is usually composed of a semiconductor integrated circuit (hereinafter referred to as IC) in order to cope with recent miniaturization and cost reduction.

FIG. 6 is a schematic diagram for explaining the relationship between a voltage luminance signal frequency-modulated by a frequency modulation circuit and its carrier frequency in an NTSC or VHS type VTR.

As shown in FIG. 6, in the NTSC and VHS type VTR, for example, the carrier frequency of the sync tip voltage (VST) is within 3.4 MHz ± 100 KHz, 100
The carrier frequency of the amplitude voltage (VWP) of the% luminance (white peak level) signal is within 4.4 MHz ± 100 KHz,
The shift frequency is specified within 1 MHz ± 100 KHz.

However, this frequency modulation circuit is
In the case of C, it is difficult to realize the above-mentioned carrier frequency without adjustment because of the absolute accuracy of the built-in resistor and capacitor, and it is necessary to adjust the carrier frequency outside the IC, for example, an adjusting component for adjusting the carrier frequency. The carrier frequency of the frequency modulator has been adjusted from the outside by providing a variable resistor, etc. and adjusting the adjusting parts.

[0007]

However, the method of adjusting the carrier frequency of the frequency modulator from the outside imposes a burden on the step of mounting the adjusting component, the adjusting step, etc., which causes a cost increase.

Therefore, instead of adjusting the carrier frequency of the frequency modulator from the outside, a method of adjusting the carrier frequency of the frequency modulator by trimming inside the IC can be considered, but in this method as well, the chip size of the IC is reduced. This also increases the number of manufacturing processes, which causes a cost increase.

Therefore, as a fundamental measure, there is a strong demand for no adjustment of the frequency modulator.

The present invention has been made to solve the above-mentioned problems of the prior art, and an object of the present invention is to provide a technique capable of making the shift frequency non-adjustable in an FM modulation circuit. .

The above and other objects and novel features of the present invention will become apparent from the description of the present specification and the accompanying drawings.

[0012]

SUMMARY OF THE INVENTION Among the inventions disclosed in the present application, the outline of a representative one will be briefly described.
It is as follows.

(1) A voltage-current conversion circuit for converting a voltage-brightness signal into a current-brightness signal in which the DC level of the sync chip is fixed to a predetermined potential and the amplitude level of the 100% brightness signal is controlled to be constant. An offset current generation circuit that generates a constant offset current, a current addition circuit that adds the offset current output from the offset current generation circuit to the current brightness signal output from the voltage current conversion circuit at a predetermined addition ratio, A differential current amplifier circuit having a constant current source whose current value is adjusted to (2 × ICAR) and outputting a control current proportional to a change in the added current output from the current addition circuit; A current control type oscillation circuit in which an oscillation frequency is varied by a control current output from a current amplification circuit, and (ICAR) is the current control type oscillation circuit. Wherein the oscillation frequency is the current value of the control current as the carrier frequency of the sync tip voltage of the predefined voltage luminance signal.

(2) In the means of (1) above, the voltage-current conversion circuit includes a pair of semiconductor elements, a pair of constant current sources cascade-connected to the pair of semiconductor elements, and the pair of constant current sources. A first differential amplifier circuit having a resistor connected between the pair of semiconductor elements and a pair of semiconductor elements, and the offset current generating circuit is connected to the pair of semiconductor elements in a cascade connection. A second differential amplifier circuit having a pair of constant current sources having the same current value as that of the constant current source of the differential amplifier circuit, and a resistor connected between the pair of constant current sources. However, a predetermined addition ratio of the current luminance signal and the offset current is obtained by the resistance ratio of the resistances of the first differential amplifier circuit and the second differential amplifier circuit.

According to the above-mentioned means (1), in the frequency modulation circuit including the current control type oscillation circuit in which the oscillation frequency is changed by the control current, in the voltage / current conversion circuit, the DC level of the sync tip is at a predetermined potential. Fixed to, and
The voltage luminance signal whose amplitude level of 100% luminance signal is controlled to be constant is converted into a current luminance signal, and the current addition circuit adds the offset current generated by the offset current generation circuit to the current luminance signal at a predetermined addition ratio. The current value is (2 × I
CAR) (ICAR is a differential current having a constant current source adjusted to an oscillation frequency of the current control type oscillation circuit, which is a current value of a control current which becomes a carrier frequency of a sync tip voltage of a voltage luminance signal defined in advance) The amplifier circuit outputs a control current proportional to the change in the added current output from the current addition circuit, and the oscillation frequency of the current control type oscillation circuit is changed by the control current output from the differential current amplification circuit. Therefore, the shift frequency can be made unadjusted.

According to the above-mentioned means (2), the voltage-current conversion circuit and the offset current generation circuit are further provided, a pair of semiconductor elements, and a pair of constant current sources subordinately connected to the pair of semiconductor elements. Since the constant current sources have the same constant current (bias current), the differential amplifier circuit has a resistor connected between the pair of constant current sources. Only by the resistance ratio of the resistors, it is possible to accurately obtain a predetermined addition ratio of the current luminance signal and the offset current.

As a result, it becomes possible to greatly contribute to downsizing and cost reduction of the video tape recorder.

[0018]

Embodiments of the present invention will be described below in detail with reference to the drawings.

In all the drawings for explaining the embodiments of the present invention, components having the same function are designated by the same reference numeral, and the repeated description thereof will be omitted.

FIG. 1 is a diagram showing a circuit configuration of a frequency modulation circuit according to an embodiment of the present invention.

In FIG. 1, 1 is a current / luminance signal generation block, 2 is a differential current amplifier circuit, and 3 is a current control type oscillation circuit.

The frequency modulation circuit according to the embodiment of the present invention is composed of a semiconductor integrated circuit (IC).

First, the current control type oscillation circuit 3 shown in FIG.
External adjustment of the carrier frequency of the conventional frequency modulator will be described with reference to FIG.

The current-controlled oscillator circuit 3 shown in FIG. 1 is an emitter-coupled astable multivibrator circuit in which the charging / discharging current of a capacitor (C) that determines the oscillation frequency is variable, and is an npn transistor. (Q
30) and the emitter of the npn transistor (Q31) are connected, and the collector of the npn transistor (Q30) is connected to the base of the npn transistor (Q31) via the emitter follower circuit. The collector is connected to the base of the npn transistor (Q30) via the emitter follower circuit.

The npn transistor (Q30) and n
An npn transistor (Q32) and an npn transistor (Q33) are connected in cascade to the emitters of the pn transistor (Q31).

In the embodiment of the present invention, this np
n transistor (Q32) and npn transistor (Q3
3) constitutes a current mirror circuit together with the npn transistor (Q15) of the differential current amplifier circuit 2,
n transistor (Q32) and npn transistor (Q3
The same current as the control current (IMOD), which is the current flowing through the npn transistor (Q15), flows through 3).

Therefore, the npn transistor (Q1
By varying the control current (IMOD) flowing through 5), the current flowing through the npn transistor (Q32) and the npn transistor (Q33) is varied.

The collector of the npn transistor (Q30) has an npn transistor (Q34) and a resistor (R).
30) is connected in a parallel circuit, and the npn transistor (Q31) has a collector connected to the npn transistor (Q31).
A parallel circuit of 35) and the resistor (R31) is cascade-connected.

Here, the potential of (-VMOD) is applied to the bases of the npn transistor (Q34) and the npn transistor (Q35) by the frequency modulation reference voltage source (VSM).

Further, the collectors of the npn transistor (Q34) and the npn transistor (Q35) are cascade-connected to the diode-type connected npn transistor (Q36).

In this current control type oscillation circuit 3, np
When the n-transistor (Q30) is turned on and the npn-transistor (Q31) is turned off, the capacitor (C) has a power source → npn transistor (Q36) → npn transistor (Q34) → npn transistor (Q30) → capacitor (C) → npn. It is charged in the path of the transistor (Q33), and the emitter potential of the npn transistor (Q31) drops.

Here, when the emitter potential of the npn transistor (Q31) drops to the potential of (VCC-VMOD-3VBE), the npn transistor (Q31) becomes O.
The N and npn transistors (Q30) are turned off.

As a result, the capacitor (C) is connected to the power source →
The npn transistor (Q36)-> npn transistor (Q35)-> npn transistor (Q31)-> capacitor (C)-> npn transistor (Q32) is charged along the path, and the emitter potential of the npn transistor (Q30) drops.

When the emitter potential of the npn transistor (Q30) drops to the potential of (VCC-VMOD-3VBE), the npn transistor (Q30) turns on and npn.
The transistor (Q31) is turned off, and this operation is repeated thereafter.

Here, VMOD is the reference voltage of the frequency modulation reference voltage source (VSM), and VBE is the base-emitter voltage of the bipolar transistor.

The current control type oscillating circuit 3 is a general oscillating circuit used for frequency modulation, has a good modulation linearity, and is suitable for an IC.

Here, the oscillation frequency (fMOD) of the current control type oscillation circuit 3 depends on each npn transistor (Q3).
2, the current flowing through Q33) (control current) is IMOD, which is given by the following equation (1).

[0038]

## EQU1 ## fMOD = IMOD / 4C.VMOD (1) From the equation (1), the oscillation frequency (fMOD) of the current control type oscillation circuit 3 becomes the control current (IMOD). It is understood that the oscillation frequency (fMOD) of the current control type oscillation circuit 3 can be changed by changing the control current (IMOD) in proportion.

Here, as described above, NTSC, VH
In the S type VTR, the carrier frequency of the sync tip voltage (VST) is within 3.4 MHz ± 100 KHz, 100%.
The carrier frequency of the amplitude voltage (VWP) of the luminance signal is 4.4M.
Within ± 100 KHz, deviation frequency is 1 MHz ± 1
It is regulated to be within 00 KHz.

However, since only the value of the capacitor (C) normally fluctuates by about 15%, it is difficult to realize the carrier frequency without adjustment, and the control current (IMOD) is adjusted to It was necessary to adjust the oscillation frequency (fMOD) of the current control type oscillation circuit 3.

Therefore, conventionally, two variable resistors are provided outside the IC for adjusting the carrier frequency of the sync tip voltage (VST) and for adjusting the deviation frequency, and each npn transistor (Q
This has been dealt with by adjusting the control current (IMOD) flowing through 32, Q33).

However, in recent years, there has been a demand for no adjustment of the frequency modulation circuit by a pure circuit method. Regarding the carrier frequency of the sync tip voltage (VST), the crystal oscillation mounted for chroma signal processing is required. Using a circuit, this crystal oscillation output and the frequency output during the sync tip period,
It was found that this can be realized by performing comparative feedback control after each frequency / voltage conversion.

However, with respect to the shift frequency, since the luminance signal does not always have 100% luminance signal,
By the feedback control as described above, the carrier frequency of the shift frequency cannot be adjusted, and the carrier frequency of the shift frequency needs to be adjusted by another method.

Therefore, in the embodiment of the present invention, in the frequency modulation circuit including the current control type oscillation circuit in which the oscillation frequency is changed by the control current, it is used that the modulation characteristic of the current control type oscillation circuit is linear. Then, a voltage-current conversion circuit that converts the voltage brightness signal into a current brightness signal, an offset current generation circuit that generates a constant offset current,
A current adding circuit for adding an offset current to a current brightness signal at a predetermined addition ratio, and a current value of (2 × ICAR) (ICA
R has a constant current source in which the oscillation frequency of the current control type oscillation circuit is adjusted to a current value of a control current which is a carrier frequency of a sync tip voltage of a voltage luminance signal, which is defined in advance. A differential current amplification circuit that outputs a control current proportional to the change in the addition current output from the addition circuit is provided, and the oscillation frequency of the current control type oscillation circuit is changed by the control current output from the differential current amplification circuit ( By controlling), the shift frequency is not adjusted.

The embodiments of the present invention will be described below step by step.

Current / luminance signal generation block 1 shown in FIG.
Has a pair of differential amplifier circuits having resistors connected between the emitters, and one differential amplifier circuit has a resistor (R
2) is connected to the npn transistor (Q3) and npn
The transistor (Q6), and the other differential amplifier circuit has an npn transistor (Q9) and an npn transistor (Q11) in which a resistor (R3) is connected between the emitters.

An npn transistor (Q4) and an npn transistor (Q7), which are constant current sources, are connected to the emitters of the npn transistor (Q3) and the npn transistor (Q6) in the differential amplifier circuit, respectively. The emitters of the npn transistor (Q9) and the npn transistor (Q11) in the other differential amplifier circuit each have an npn transistor (Q
10) and the nnpn transistor (Q12) are connected in cascade.

Here, each npn transistor (Q4, Q
7, Q10, Q12) are npn transistors (Q1)
And a current mirror circuit, and each npn transistor (Q4, Q7, Q10, Q12) has the same current (bias current) as the current flowing through the npn transistor (Q1).
Flows.

The collector of the npn transistor (Q3) has a pnp transistor (Q2), and the collector of the npn transistor (Q6) has a pnp transistor (Q).
5), the collector of the npn transistor (Q9) has pn
The p-transistors (Q8) are connected in cascade, and the pnp transistors (Q2, Q5, Q8) form a current mirror circuit.

A video signal is input to the input terminal of the current / luminance signal generation block 1. This video signal is automatically gain controlled (AGC), and the luminance signal and chrominance signal are separated (Y / C).
This is a video signal after processing such as separation), clamp, emphasis, white / dark clip, and the like.

Therefore, the video signal input to the input terminal is a luminance signal (VIN) in which the direct current (DC) level and the amplitude level are controlled to be constant, and in the embodiment of the present invention, for example, as shown in FIG. Sync tip voltage (VST)
Is 2 V and the amplitude voltage (VWP) of the 100% luminance signal is 0.
It is set to 5V.

Here, the input voltage luminance signal (VI
N) is applied to the base of the npn transistor (Q3), and the reference voltage (VREF) of the brightness signal reference voltage source (VSR) is applied to the base of the npn transistor (Q6).
Is applied.

The reference voltage (VREF) of the luminance signal reference voltage source (VSR) is applied to the base of the npn transistor (Q9), and the luminance signal reference voltage is applied to the base of the npn transistor (Q11). Voltage source (VS
The reference voltage (VREF) of R) is set to resistors (R4) and (R5)
The voltage divided by and is applied.

In the current / luminance signal generation block 1 shown in FIG. 1, each npn transistor (Q3, Q4, Q6, Q
7) and the resistor (R2) form a voltage-current conversion circuit that converts the voltage brightness signal into a current brightness signal.

Further, each npn transistor (Q9, Q1
0, Q11, Q12) and the resistors (R3) to (R5) form an offset current generation circuit that generates a constant offset (fixed current).

Further, each pnp transistor (Q2, Q
5, Q8) form a current addition circuit.

The differential current amplifier circuit 2 shown in FIG. 1 is a general circuit often used in a variable gain amplifier circuit. This differential current amplifier circuit 2 is a pnp transistor (Q14).
And a pnp transistor (Q16), the base of the pnp transistor (Q14) is connected to the collector of the npn transistor (Q6) of the current / luminance signal generation block 1, and the pnp transistor (Q16) is connected. The base is connected to the collector of the npn transistor (Q3) of the current / luminance signal generation block 1.

The base of the pnp transistor (Q14) is connected to the emitter of the pnp transistor (Q13) connected in a diode form, and the base of the pnp transistor (Q16) is connected to the diode form of p.
It is connected to the emitter of the np transistor (Q17).

The collector of the pnp transistor (Q13) connected in the diode form and the collector of the pnp transistor (Q17) connected in the diode form are
Through the resistor (R6), a reference voltage source (VS for luminance signal)
R).

In this pnp transistor (Q13),
npn transistor (Q6) and pnp transistor (Q
The current (I2) flowing out from the connection point with 5) flows in and p
The current (I1) flowing out from the connection point between the npn transistor (Q3) and the pnp transistor (Q2) flows into the np transistor (Q17).

Further, the current source (IS) is any means,
For example, the above-mentioned comparative feedback control, or
The control current (IMOD) flowing through the npn transistor (Q32) and the npn transistor (Q33) is adjusted by the adjustment of the carrier frequency adjustment parts provided outside the IC.
When R, the oscillation frequency of the current controlled oscillator circuit 3 (fM
OD) is a frequency modulation current source adjusted so that it becomes 3.4 MHz, which is the carrier frequency of the sync tip voltage (VST).

Further, the resistor (R6) is a resistor for ensuring the dynamic range of the collector-emitter voltage (VCE) of the npn transistor (Q3).

Next, the operation of the embodiment of the present invention will be described.

As described above, the voltage brightness signal (VIN) input to the input terminal of the current brightness signal generation block 1 is
It is applied to the base of the npn transistor (Q3).

Further, the reference voltage (VREF) of the luminance signal reference voltage source (VSR) is applied to the base of the npn transistor (Q6), where the reference voltage (VSR) of the luminance signal reference voltage source (VSR) is applied. VREF) is set to 2V, which is the same as the sync tip voltage (VST).

Therefore, the npn transistor (Q3)
The voltage luminance signal (VIN) input to the base of is converted into a differential current luminance signal with reference to 2V which is the reference voltage (VREF) of the luminance signal reference voltage source (VSR).

In this case, each npn transistor (Q
1, Q4, Q7, Q10, Q12), the current (bias current) flowing in the current mirror circuit is IB, and the amplitude voltage (VWP) of 100% luminance signal is 0.
If the current generated by the resistor (R2) at 5V is IDEV ', the collector current of the npn transistor (Q3) is IB + IDEV' and the collector current of the npn transistor (Q6) is IB-IDEV '.

Further, as described above, the reference voltage (VREF) of the reference signal source for luminance signal (VSR) is applied to the base of the npn transistor (Q9) of the differential amplifier circuit of the offset current generating circuit, and the npn transistor (npn transistor). (Q1
A voltage obtained by dividing the reference voltage (VREF) of the reference signal source for luminance signal (VSR) by resistors (R4) and (R5) is applied to the base of 1).

Here, an npn transistor (Q9)
The voltage between the npn transistor (Q11) and the base is
0.5V, which is the same as the amplitude voltage (VWP) of 100% luminance signal
It is said.

If the current generated by the resistor (R3) by this voltage is ICAR ', the collector current of the npn transistor (Q9) becomes IB + ICAR'.

This collector current is reflected by a current mirror circuit formed of a pnp transistor (Q2), a pnp transistor (Q5) and a pnp transistor (Q8).

In this case, the npn transistor (Q3)
Since the collector current is IB + IDEV 'and the collector current of the npn transistor (Q6) is IB-IDEV', from the connection point between the npn transistor (Q3) and the pnp transistor (Q2), the current ( I
1) flows out, and the npn transistor (Q6) and p
From the connection point with the np transistor (Q5), see (3) below.
The current (I2) shown in the formula flows out.

[0073]

## EQU00002 ## I1 = IB + ICAR '-(IB + IDEV') = ICAR'-IDEV '...................................... (2)

[0074]

## EQU00003 ## I2 = IB + ICAR '-(IB-IDEV') = ICAR '+ IDEV' (3) where the input voltage luminance signal (VIN) is When it is the sync tip voltage (VST), the current (IDE
Since V ′) is 0, the current (I1) and the current (I2) are equal to the current (ICAR ′) as shown in the following formula (4).
Becomes equal to

[0075]

## EQU00004 ## I1 = I2 = ICAR '(4) FIG. 3 shows the voltage input to the voltage-current conversion circuit 1 according to the embodiment of the present invention. Brightness signal (VIN) and current (I
2 is a graph showing the relationship with (1, I2).

The current (I1) and the current (I2) are respectively supplied to the diode type pnp transistor (Q17) and the diode type pnp transistor (Q13) of the differential current amplifier circuit 2. Inflow.

As a result, the pnp transistor (Q1
4) and the base of the pnp transistor (Q16) are supplied with a base potential, and the pnp transistor (Q1
According to the potential difference between the base potential of 4) and the base potential of the pnp transistor (Q16), the current (2ICAR) from the frequency modulation current source (IS) causes the pnp transistor (Q14) and the pnp transistor (Q16). ).

Here, the control current (I) which is the collector current of the pnp transistor (Q16) of the differential current amplifier circuit 2 is
MOD) assumes that the base potential of the pnp transistor (Q16) is Vb1, the base potential of the pnp transistor (Q14) is Vb2, and the grounded-emitter current amplification factor (hfe) of the pnp transistor is so large that it can be ignored. Are given by the following equations (5) and (6).

[0079]

[Formula 5] IMOD = 2ICAR / (1 + Xcot) …………………… (5)

[0080]

Xcot = exp {-q · (Vb2-Vb1) / kT} (6) Here, Vb2-Vb1 is given by the following equation (7).

[0081]

## EQU00007 ## Vb2-Vb1 = -kT.multidot. {Log (I1 / I2)} / q (7) When this equation (7) is substituted into equation (6), the following equation (8) is obtained.
Is obtained.

[0082]

Xcot = exp {log (I1 / I2)} Xcot = I1 / I2 ………………………………………… (8) Therefore, the control current (IMOD) is as follows. It is represented by (9).

[0083]

## EQU00009 ## IMOD = 2ICAR.multidot.I2 / (I1 + I2) = ICAR.multidot.I2 / ICAR '... (9) Therefore, the control current (IMOD) is proportional to the change of the current (I2). , Again, IDEV = ICAR · IDEV '
Assuming that / ICAR ', the above equation (9) is given by the following (10)
It is expressed like a formula.

[0084]

(10) IMOD = ICAR · (ICAR ′ + IDEV ′) / ICAR ′ = ICAR (1 + IDEV ′ / ICAR ′) = ICAR + IDEV ………………………… (10) Here, the voltage input When the luminance signal (VIN) is the sync tip voltage (VST), the current (IDE)
V ') is 0, so the current (IDEV) is also 0,
As shown in the following equation (11), the control current (IMOD)
Becomes equal to the current (ICAR).

[0085]

(11) IMOD = ICAR (11) FIG. 4 shows the current (I2) in the embodiment of the present invention.
Control current (IMOD) output from the differential current amplifier circuit 2
6 is a graph showing a relationship with the graph.

The oscillation frequency of the current control type oscillation circuit 3 is linearly changed according to the control current (IMOD),
As a result, the input voltage brightness signal (VIN) is linearly frequency-converted.

FIG. 5 is a graph showing the relationship between the control current (IMOD) and the oscillation frequency of the current control type oscillation circuit 3 in the embodiment of the present invention, showing the frequency when the control current (IMOD) is ICAR. fCAR, control current (IMO
The frequency when D) is ICAR + IDEV is fCAR +
This is a graph as fDEV.

As described above, in the embodiment of the present invention, first, in the current / luminance signal generation block 1, VST →
Convert from ICAR ', VWP to IDEV'.

In the differential current amplifier circuit 2, the following (1
In the current control type oscillation circuit 3, the relationship represented by the equation (2) is satisfied by the relationship represented by the following equation (13).

[0090]

## EQU12 ## IDEV = ICAR.IDEV '/ ICAR' ICAR '/ IDEV' = ICAR / IDEV ICAR ': IDEV' = ICAR: IDEV ............ (12)

[0091]

[Equation 13] ICAR: IDEV = fCAR: fDEV (13) Therefore, in the current / luminance signal generation block 1,
It suffices to perform voltage-current conversion at the ratio shown in (14) below.

[0092]

ICAR ′: IDEV ′ = fCAR: fDEV = 3.4 MHz: 1 MHz = 3.4: 1 (14) In this case, the pnp transistor (Q14) and the pnp transistor ( Current (ICAR) distributed from the frequency modulation current source (IS) to each pnp transistor (Q14, Q16) when the base potential is the same as that of Q16).
Is adjusted by some means so that the oscillation frequency (fMOD) of the current control type oscillation circuit 3 becomes the carrier frequency of the sync tip voltage (VST), 3.4 MHz, so that fDEV can be set to 1 MHz. it can.

Here, the current (ICAR ') is 100%
The current (IDEV ') is a current generated in the resistor (R3) due to the DC voltage of 0.5V, which is the same as the amplitude voltage (VWP) of the luminance signal, and the current (IDEV') is 100% of the amplitude voltage (VW of the luminance signal.
Since it is a current generated in the resistance (R2) by 0.5V which is P), if the resistance value of the resistance (R2) and the resistance value of the resistance (R3) are set to the resistance values satisfying the following expression (15). Good.

[0094]

[Equation 15] R2: R3 = 3.4: 1 …………………………………… (15) As the resistance value that satisfies the equation (15), for example, the resistance (R2) The resistance value may be 34 KΩ and the resistance value of the resistor (R3) may be 10 KΩ.

For the DC voltage of 0.5 V for generating the current (ICAR '), the resistance value of the resistor (R4) may be 10 KΩ and the resistance value of the resistor (R5) may be 30 KΩ.

According to the embodiment of the present invention, two sets of resistors are used, and the target current addition ratio can be obtained from the ratio (resistance ratio) of the resistance values of the two sets of resistors.

In the IC, since the resistance values of the resistors arranged adjacent to each other have high relative accuracy, the target resistance ratio can be accurately realized, and thus the target current addition ratio can be accurately realized.

FIG. 2 is a diagram showing another circuit configuration of the voltage-current conversion circuit according to the embodiment of the present invention.

The voltage-current conversion circuit 1'shown in FIG. 2 is different from the voltage-current conversion circuit 1 shown in FIG. 1 in the configuration of the current addition circuit, but the principle is the same.

In the voltage-current conversion circuit 1'shown in FIG. 2, the currents flowing through the npn transistor (Q3), the npn transistor (Q6) and the npn transistor (Q9) are respectively returned by the current mirror circuit to generate the current (I1). And a current (I2) is generated.

In the voltage-current conversion circuit 1'shown in FIG. 2, the collector of the npn transistor (Q3) of the differential amplifier circuit is cascode-connected with the pnp transistor (Q18) connected in the diode form, and the pnp transistor (Q18) is connected. Q18) forms a current mirror circuit with the pnp transistor (Q20).

Further, the collector of the npn transistor (Q6) of the differential amplifier circuit is connected to a pnp transistor (Q19) connected in a diode form in a cascade manner, and the pnp transistor (Q19) is connected.
The transistor (Q19) is a pnp transistor (Q2
3) and a current mirror circuit.

Further, the collector of the npn transistor (Q9) of the differential amplifier circuit has a pnp transistor (Q9).
8) is cascade-connected, and the pnp transistor (Q8) is
A current mirror circuit is formed with the pnp transistor (Q2) and the pnp transistor (Q5).

The npn transistor (Q21) and the npn transistor (Q22) form a current mirror circuit, and the npn transistor (Q21) is a pnp transistor (Q20) and an npn transistor (Q22).
Is cascade-connected to the pnp transistor (Q2).

Therefore, the npn transistor (Q3)
The current flowing through the current mirror circuit is composed of a pnp transistor (Q18) and a pnp transistor (Q20), and an npn transistor (Q21) and np transistor.
It is folded back by a current mirror circuit composed of an n-transistor (Q22) and also an npn-transistor (Q9).
The current flowing through the pnp transistor (Q8) and p
Since it is folded back by the current mirror circuit composed of the np transistor (Q2), the pnp transistor (Q2)
From the connection point between the npn transistor (Q22) and
The current (I1) which is the difference between the current flowing through the n-transistor (Q9) and the current flowing through the npn-transistor (Q3) is a diode-connected p of the differential current amplifier circuit 2 '.
It flows into the np transistor (Q17).

The npn transistor (Q24) and the npn transistor (Q25) form a current mirror circuit, and the npn transistor (Q24) is a pnp transistor (Q23) and an npn transistor (Q25).
Is cascade-connected to the pnp transistor (Q5).

Therefore, the npn transistor (Q6)
The current flowing through the current mirror circuit is composed of a pnp transistor (Q19) and a pnp transistor (Q23), and an npn transistor (Q24) and np transistor.
It is folded back by a current mirror circuit composed of an n-transistor (Q25) and also an npn transistor (Q9).
The current flowing through the pnp transistor (Q8) and p
Since it is folded back by the current mirror circuit composed of the np transistor (Q5), the pnp transistor (Q5)
From the connection point between the npn transistor (Q25) and
The current (I2) which is the difference between the current flowing in the n-transistor (Q9) and the current flowing in the npn-transistor (Q6) is connected to the differential current amplifier circuit 2 ′ in the diode type p.
It flows into the np transistor (Q13).

In the voltage-current conversion circuit 1'shown in FIG. 2, a pnp transistor (Q2), a pnp transistor (Q5), a pnp transistor (Q20) and a pnp transistor.
Since the collector potential of the transistor (Q23) is all the base-emitter voltage (VBE) of the bipolar transistor, the current fluctuation due to the Early effect is canceled out, and it becomes stable against changes in the power supply voltage and temperature.

Further, since the problem of the dynamic range of the npn transistor (Q3) is eliminated, it is suitable for lowering the voltage.

In the embodiment of the present invention, the current / luminance signal generation block (1, 1 ′), the differential current amplifier circuit (2, 2 ′) and the current control type oscillation circuit 3 are composed of bipolar transistors. However, the configuration is not limited to this, and the current / luminance signal generation block (1,
1 ′), the differential current amplifier circuits (2, 2 ′) and the current control type oscillation circuit 3 may be entirely or partially configured to have a field effect transistor (FET).

Further, the current / luminance signal generation block (1,
1 '), the differential current amplifier circuit (2, 2'), and the current control type oscillation circuit 3 may have a circuit configuration in which a bipolar transistor and a field effect transistor (FET) are combined.

As described above, according to the embodiment of the present invention, it is possible to realize the adjustment of the shift frequency of the frequency modulation circuit by a pure circuit method and a simple means. Becomes

This makes it possible to greatly contribute to the miniaturization and cost reduction of the VTR.

Although the present invention has been specifically described based on the embodiments of the present invention, the present invention is not limited to the embodiments of the present invention, and various modifications can be made without departing from the scope of the invention. Not to mention getting it.

[0115]

The effects obtained by typical ones of the inventions disclosed in the present application will be briefly described as follows.

(1) According to the present invention, in the frequency modulation circuit including the current control type oscillation circuit in which the oscillation frequency is changed by the control current, the voltage-current conversion circuit converts the voltage luminance signal into the current luminance signal. The current addition circuit adds the offset current generated by the offset current generation circuit to the current luminance signal at a predetermined addition ratio, and the current value becomes (2 × ICA
R) (ICAR is a differential current having a constant current source adjusted to an oscillation frequency of the current control type oscillation circuit, which is a current value of a control current which is a carrier frequency of a sync tip voltage of a voltage luminance signal defined in advance) The amplifier circuit outputs a control current proportional to the change in the added current output from the current addition circuit,
Since the oscillation frequency of the current control type oscillation circuit is made variable by the control current output from the differential current amplification circuit,
It is possible to make the shift frequency non-adjustable by a pure circuit method and a simple means.

(2) According to the present invention, the voltage-current conversion circuit and the offset current generation circuit are provided with a pair of semiconductor elements.
A differential amplifier circuit having a pair of constant current sources subordinately connected to the pair of semiconductor elements and a resistor connected between the pair of constant current sources, and a constant amplifier of the constant current source. Since the currents (bias currents) are the same, it is possible to accurately obtain the predetermined addition ratio of the current luminance signal and the offset current only by the resistance ratio of the resistors of each differential amplifier circuit.

(3) According to the present invention, it is possible to greatly contribute to downsizing and cost reduction of a video tape recorder.

[Brief description of the drawings]

FIG. 1 is a diagram showing a circuit configuration of a frequency modulation circuit according to an embodiment of the present invention.

FIG. 2 is a diagram showing another circuit configuration of the voltage-current conversion circuit according to the embodiment of the present invention.

FIG. 3 is a diagram illustrating a voltage luminance signal (VIN) input to the voltage-current conversion circuit 1 according to the embodiment of the present invention and a current (I).
2 is a graph showing the relationship with (1, I2).

FIG. 4 is a current (I2) in the embodiment of the present invention.
And the control current (IMO which is the output of the differential current amplifier circuit 2).
It is a graph which shows the relationship with D).

FIG. 5 is a graph showing the relationship between the control current (IMOD) of the current controlled oscillator circuit 3 and the oscillation frequency in the embodiment of the present invention.

FIG. 6 is a schematic diagram for explaining a relationship between a voltage luminance signal frequency-modulated by a frequency modulation circuit and a carrier frequency thereof in an NTSC / VHS type VTR.

[Explanation of symbols]

1, 1 '... Current luminance signal generation block, 2, 2' ... Differential current amplification circuit, 3 ... Current control type oscillation circuit, Q1-Q2
5, Q30 to Q36 ... Transistors, R1 to R6, R3
0, R31 ... Resistor, C ... Capacitor, VSR ... Luminance signal reference voltage source, VSM ... Frequency modulation reference voltage source, IS
… Current source for frequency modulation adjusted to the carrier frequency of the sync tip.

Claims (2)

[Claims] 1. A voltage-current conversion circuit for converting a voltage-brightness signal into a current-brightness signal, in which the direct current level of the sync tip is fixed to a predetermined potential and the amplitude level of the 100% brightness signal is controlled to be constant, and a constant value. An offset current generating circuit that generates an offset current, a current adding circuit that adds the offset current output from the offset current generating circuit to a current brightness signal output from the voltage-current conversion circuit at a predetermined addition ratio, and a current A differential current amplifier circuit having a constant current source whose value is adjusted to (2 × ICAR) and outputting a control current proportional to a change in the added current output from the current addition circuit; A current control type oscillation circuit in which an oscillation frequency is varied by a control current output from an amplification circuit, and (ICAR) is an oscillation of the current control type oscillation circuit. Frequency modulation circuit, wherein the wave number is a current value of the control current as a carrier frequency of the sync tip voltage of the predefined voltage luminance signal. 2. The voltage-current conversion circuit includes a pair of semiconductor elements, a pair of constant current sources subordinately connected to the pair of semiconductor elements, and a resistor connected between the pair of constant current sources. A first differential amplifier circuit having: and the offset current generation circuit is connected to a pair of semiconductor elements and the pair of semiconductor elements in a cascade connection, and a constant current source of the first differential amplifier circuit. And a second differential amplifier circuit having a pair of constant current sources having the same current value, and a resistor connected between the pair of constant current sources. By the resistance ratio of the resistance of the circuit and the second differential amplifier circuit,
The frequency modulation circuit according to claim 1, wherein a predetermined addition ratio of the current luminance signal and the offset current is obtained.