JPH08148970A - Phase shifter - Google Patents

Abstract

PURPOSE: To provide a phase shifter capable of changing only a phase without the attenuation of signal amplitude for which a phase shift amount per stage is large. CONSTITUTION: This phase shifter 100 is provided with an FET 10 for which input signals are inputted to a gate, a resistor 12 and an inductance 14 connected between the source and drain of the FET 10, a source resistor 16 and a drain resistor 18. By making the respective resistance values of the source resistor 16 and the drain resistor 18 approximately the same, signals whose amplitude is equal to the input signals provided with a phase lead angle which is the double of the phase difference of an input voltage and the both-terminal voltage of the inductor 14 are outputted from an output terminal.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an all-pass type phase shifter used for a phase shift oscillator or the like.

[0002] In this specification, when the phase of the input signal is used as a reference, the case where the phase of the output signal advances by up to 180 ° is "advanced type", and the phase of the output signal is 18 degrees.
The case where there is a delay within the range of 0 ° will be referred to as a “phase delay type” for explanation.

[0003]

2. Description of the Related Art Conventionally, there has been known a phase shift oscillator that oscillates by setting the phase of a signal that makes a round through a feedback loop to 0 ° or 360 ° and setting the loop gain to 1 or more. This phase-shift oscillator is composed of, for example, a portion that performs inverting amplification and a portion that rotates the phase shift. Has become.

FIG. 28 is a diagram showing an example of a configuration for rotating the phase shift used in the above-mentioned phase shift oscillator or the like. FIG. 3A shows a configuration in which CR high pass filters are connected in three stages. A 1-stage CR high-pass filter can theoretically shift the phase by up to 90 ° when the frequency is infinite, but it is not practical to operate at such a frequency, so in practice, 3 stages are connected to form an overall filter. As 180 °
The phase shift is performed (60 ° per stage).

Further, FIG. 1B shows a configuration in which CR low-pass filters are connected in three stages. Similar to the case of the high-pass filter, the one-stage CR low-pass filter can theoretically shift the phase by up to 90 ° when the frequency is infinite, but in reality, three stages are connected to make a total of 180 degrees.
° It is designed to perform a phase shift.

The phase shift amount is 90 ° or 18
If it is less than 0 °, C of 1 or 2 steps
It is also possible to shift the phase shift with an R high-pass filter or a low-pass filter.

[0007]

By the way, the above-mentioned C
R high-pass filter or low-pass filter,
Since the signal amplitude is attenuated at the same time as the phase shift, there is a problem when it is used for the purpose of shifting the phase. Therefore, for example, when it is used for a phase shift oscillator, the attenuation is compensated by setting a large amplification factor of the inverting amplifier.

Further, since the attenuation amount of the signal amplitude also changes in accordance with the phase shift amount, that is, the attenuation amount of the signal amplitude also changes in accordance with the frequency of the input / output signal, an arbitrary phase shift amount can be set. When it is used as a phaser, an additional circuit for stabilizing the signal amplitude is necessary, which is inconvenient.

Further, considering the amount of signal attenuation, the amount of phase shift by the filter per stage could not be set too large. Therefore, there has been a situation in which the number of stages of filters to be connected has to be changed for each required phase shift amount.

The present invention has been made in view of the above points, and its purpose is to change only the phase without attenuation of the signal amplitude and to have a large amount of phase shift per stage. To provide a container.

[0011]

In order to solve the above-mentioned problems, a phase shifter according to a first aspect of the present invention is such that a source and a drain are respectively connected to first and second resistors having substantially equal resistance values. A transistor having a gate to which an input signal is input, a third resistor and an inductor, and a series circuit connected between a source and a drain of the transistor, and forming the series circuit. It is characterized in that a signal appearing at a connection point between the third resistor and the inductor is taken out as an output signal.

According to another aspect of the phase shifter of the present invention, the first and second resistors having substantially the same resistance value are connected to the emitter and the collector, respectively, and a transistor to which an input signal is input to the base and a third resistor. And a series circuit connected between the source and drain of the transistor, which is composed of a resistor and an inductor, and outputs a signal appearing at a connection point between the third resistor and the inductor forming the series circuit. It is characterized in that it is taken out as an output signal.

A phase shifter according to a third aspect is the phase shifter according to the first or second aspect, further comprising a DC blocking capacitor connected in series to the series circuit.

According to a fourth aspect of the present invention, in the phase shifter according to any of the first to third aspects, the third resistor in the series circuit is connected to the source side or the emitter side of the transistor. It is characterized in that the phase of the input signal is advanced by a predetermined angle before being output.

A phase shifter according to claim 5 is the phase shifter according to any one of claims 1 to 3, wherein the inductor in the series circuit is connected to the source side or the emitter side of the transistor, and It is characterized in that the phase is delayed by a predetermined angle before being output.

A phase shifter according to a sixth aspect of the present invention is the phase shifter according to any one of the first to fifth aspects, wherein the resistance value of the third resistor is changed to change the entire phase shift amount. Characterize.

A phase shifter according to a seventh aspect is the phase shifter according to any one of the first to fifth aspects, wherein the inductance of the inductor is changed to change the entire phase shift amount.

The phase shifter according to claim 8 is the phase shifter according to any one of claims 1 to 5, wherein the resistance value of the third resistor and the inductance of the inductor are changed to obtain a total phase shift amount. It is characterized by changing.

According to a ninth aspect of the present invention, in the phase shifter according to the sixth or eighth aspect, the third resistance is formed by a channel of a field effect transistor, and a gate voltage is changed to change a channel resistance. , The total amount of phase shift is changed.

The phase shifter of claim 10 is the phase shifter of claim 7 or 8.
In the phase shifter of (1), the inductor is formed in a spiral shape in a substantially planar shape on a substrate, and on the substrate in a substantially concentric shape with the inductor conductor, and a predetermined DC bias is applied. A control conductor through which an electric current is passed, and an insulating magnetic body formed so as to cover the inductor conductor and the control conductor are provided, and a DC bias current flowing in the control conductor is changed to change the inductor conductor. It is characterized in that the total phase shift amount is changed by changing the inductance appearing at both ends.

The phase shifter of claim 11 is the phase shifter of claim 7 or 8.
In the phase shifter of (1), the inductor is formed in a spiral shape in a substantially planar shape on a substrate, and on the substrate in a substantially concentric shape with the inductor conductor, and a predetermined DC bias is applied. A control conductor through which a current is passed, and a conductive magnetic body formed so as to cover the inductor conductor and the control conductor with an insulating film interposed therebetween, and change the DC bias current flowing in the control conductor. The overall phase shift amount is changed by changing the inductance appearing at both ends of the inductor conductor.

According to a twelfth aspect of the present invention, in the phase shifter according to any one of the first to fifth aspects, the inductor is an amplifier having a predetermined gain and an inductor connected in parallel between the input and output of the amplifier. And an element, and the inductance seen from the input side of the amplifier is made different from the actual inductance of the inductor element.

According to a thirteenth aspect of the phase shifter of the twelfth aspect, by setting the gain of the amplifier to a negative value, the inductance seen from the input side of the amplifier is actually set by the inductor element. It is characterized in that it is smaller than the inductance that it has.

According to a fourteenth aspect of the present invention, in the phase shifter according to the twelfth aspect, the gain seen from the input side of the amplifier is actually set by setting the gain of the amplifier between 0 and 1. It is characterized in that it is made larger than the inductance of the element.

The phase shifter of claim 15 is the phase shifter of claim 12 or 13, wherein the amplifier has a non-inverting input terminal connected to a fixed potential and an output terminal connected to one end of the inductor element. A first operational amplifier, a fourth resistor inserted between the other end of the inductor element and the inverting input terminal of the first operational amplifier, an output terminal of the first operational amplifier and an inverting input. Fifth inserted between terminals
And a resistance of 1), and operates as an inverting amplifier having a negative gain.

According to a sixteenth aspect of the present invention, in the phase shifter of the fifteenth aspect, a buffer is further provided between the other end of the inductor element and the fourth resistor before the amplifier. It is characterized by increasing the input impedance.

According to a seventeenth aspect of the present invention, in the phase shifter according to the sixteenth aspect, the buffer has a non-inverting input terminal connected to the other end of the inductor element and an output terminal connected to the fourth terminal.
Of the second operational amplifier, and a sixth resistor connected between the output terminal and the inverting input terminal of the second operational amplifier.

According to an eighteenth aspect of the present invention, in the phase shifter according to the sixteenth aspect, the buffer is constituted by an emitter follower circuit or a source follower circuit.

The phase shifter of claim 19 is the phase shifter of claims 15-18.
In any one of the phase shifters described above, by forming at least one of the fourth, fifth and sixth resistors by a variable resistor, the inductance of the inductor can be varied to change the total phase shift amount. Characterize.

The phase shifter of claim 20 is the phase shifter of claim 12 or 14, wherein the amplifier is connected between two non-inverting amplifiers arranged in series and the two non-inverting amplifiers. And a first voltage dividing circuit, and the overall gain is set between 0 and 1 according to the voltage dividing ratio of the first voltage dividing circuit.

According to a twenty-first aspect of the present invention, in the phase shifter according to the twelfth or fourteenth aspects, the amplifier has an emitter follower circuit in which a second voltage dividing circuit is connected to the emitter side or a second side to the source side. It is composed of a source follower circuit to which a voltage dividing circuit is connected, and by taking out the output from the second voltage dividing circuit, the overall gain is changed from 0 to 1 in accordance with the voltage dividing ratio of the second voltage dividing circuit. It is characterized by setting in between.

A phase shifter according to a twenty-second aspect is the phase shifter according to the twentieth or twenty-first aspect, in which the voltage dividing circuit is formed to include a variable resistor, whereby the inductance of the inductor is varied and the entire phase shift is performed. It is characterized by changing the amount.

The phase shifter of claim 23 is the phase shifter of claim 19 or 22, wherein the resistance value of the third resistor included in the second series circuit is changed and the inductance of the inductor is changed. It is characterized in that the entire phase shift amount is changed by changing the amount.

A phase shifter according to a twenty-fourth aspect is the phase shifter according to the twenty-third aspect, in which the third resistance is formed by a channel of a field effect transistor, and a gate voltage is changed to change the channel resistance, whereby It is characterized in that the phase shift amount of is changed.

A phase shifter according to a twenty-fifth aspect is the phase shifter according to any one of the first through twenty-fourth aspects, wherein the constituent parts are integrally formed on a semiconductor substrate.

[0036]

According to the first aspect of the invention, when an input signal is input to the gate of the transistor, an AC signal having the same phase as the input signal is generated at the source, and an AC signal having the opposite phase to the input signal is generated at the drain. As a result, a voltage having twice the amplitude of the input signal is generated between the source and drain.

In particular, the phases of the voltages appearing at both ends of the third resistor and the inductor, which constitute the series circuit connected between the source and the drain, are deviated from each other by 90 °, and these respective voltages are vectorially expressed. The combined result is equal to the voltage having the double amplitude of the input signal described above. Therefore, the divided voltage by the series circuit is generated so as to be located on the circumference of the semicircle whose radius is the AC component of the input voltage. When the difference between the voltage divided by this series circuit and the input voltage is taken out as an output signal,
This output signal is obtained by connecting the center of the above-mentioned semicircle and the voltage dividing point by the second series circuit, and its amplitude becomes equal to that of the input signal (radius of the semicircle). Does not decay. Moreover, the output amplitude is always equal to the input signal even when the phase shift amount changes according to the frequency, and a stable output that does not change depending on the frequency between the input and output signals can be obtained.

Looking at the amount of phase shift between the input and output signals, the angle formed by the diameter connecting the line connecting the center of the semicircle and the voltage dividing point of the series circuit is the amount of phase shift as it is. A phase difference of about twice the phase angle shifted by the third resistor and the inductor occurs between the input and output signals, and a large amount of phase shift can be set.

The invention of claim 2 uses a bipolar transistor instead of the transistor (field effect transistor) used in claim 1, and a phase shifter can be constructed based on exactly the same operating principle. It is possible to realize a phase shifter having a large amount of phase shift without attenuation of signal amplitude. Particularly, when the bipolar transistor is used, the maximum operating frequency can be increased, and conversely, when the field effect transistor is used, the input impedance can be increased.

According to a third aspect of the present invention, a direct current blocking capacitor is further inserted in the series circuit between the source and drain or between the emitter and collector of the above-mentioned transistor, and the capacitor is connected between the source and drain or between the emitter and collector. Can be prevented from being inoperable due to DC short circuit.

According to the fourth aspect of the invention, the third resistor in the series circuit is connected to the source side or the emitter side of the transistor, and a voltage with a phase advanced from the input signal is generated across the inductor. As a result, it is possible to easily realize a phase-advancing type phase shifter having a phase shift amount twice this phase lead angle.

According to the invention of claim 5, the inductor in the series circuit is connected to the source side or the emitter side of the transistor, and a voltage delayed in phase with respect to the input signal is generated across the third resistor. As a result, it is possible to easily realize a delay type phase shifter having a phase shift amount twice this phase delay angle.

According to the sixth aspect of the invention, the phase shift amount can be easily changed by changing the resistance value of the third resistor which constitutes the series circuit. Similarly, in the invention of claim 7, the phase shift amount can be easily changed by changing the inductance of the inductor forming the series circuit.

According to the invention of claim 8, the phase shift amount can be easily changed by changing both the resistance value of the third resistor and the inductance of the inductor which form the series circuit.

In the ninth aspect of the invention, the third resistor whose resistance value can be changed is specifically formed by the channel of the field effect transistor, and the third resistor can be easily changed by changing the gate voltage. The resistance value of the resistor can be changed.

The invention according to claim 10 or 11 is
Disclosed is a specific example of an inductor with variable inductance, in which a magnetic body is formed so as to cover the inductor conductor and the control conductor, and the DC bias current flowing through the control conductor is variably controlled to control the magnetic property. The saturation magnetization characteristic of the inductor conductor having the body as a magnetic path changes, and the inductance of the inductor conductor can be changed.

Further, the invention of claim 12 discloses another specific example of the inductor, wherein the inductor described above is constituted by connecting an inductor element having a predetermined inductance and an amplifier or the like in parallel. Let the inductance of the inductor element be L0 and the transfer function of the amplifier be K.
Then, this configuration is converted based on Miller's theorem, and when viewed from the input side of the amplifier, the inductance of the converted configuration is L0 / (1-K). Therefore, the apparent inductance can be made different from the actual inductance L0 according to the transfer function K, that is, the gain of the amplifier.
In fact, it is possible to make a small inductance look large. In particular, when the entire phase shifter is integrated on the semiconductor substrate, only an inductor element having a small inductance can be formed on the semiconductor substrate. Therefore, if this small inductance can be converted into a large inductance, it is possible to integrate it. It is particularly convenient, and there is also an effect of cost reduction by reducing the mounting area.

The thirteenth or fourteenth aspect of the present invention specifically shows the gain of the amplifier when the inductance is changed. That is, since the apparent inductance is L0 / (1-K), L0 is set to a smaller value by setting the gain K of the amplifier to a negative value, and L0 is set to 0 by setting the gain of the amplifier between 0 and 1. Can be changed to a larger one.

According to the fifteenth aspect of the present invention, the amplifier according to the twelfth or thirteenth aspect of the invention is specifically configured by an inverting amplifier using an operational amplifier, and the gain of the amplifier can always be negative.

In the sixteenth aspect of the invention, the input impedance can be increased by inserting a buffer on the input side of the operational amplifier. In the seventeenth or eighteenth aspect of the invention, the buffer is formed by using the operational amplifier or the emitter follower. It can be easily configured by a circuit or a source follower circuit.

According to the nineteenth aspect of the present invention, the amplification factor of the inverting amplifier can be varied by configuring the resistance constituting the inverting amplifier together with the above-mentioned operational amplifier by a variable resistor, whereby the apparent inductance is obtained. Can be continuously changed to continuously change the phase shift amount of the entire phase shifter.

According to the invention of claim 20 or 21, the amplifier according to the invention of claim 12 or 14 is specifically configured by two non-inverting amplifiers or an emitter follower circuit, a source follower circuit and a voltage dividing circuit. By using a voltage divider circuit, the overall gain can be set between 0 and 1 and the apparent inductance can be changed to a value larger than L0.

According to the twenty-second aspect of the present invention, the overall gain can be changed by forming the resistor forming the above voltage dividing circuit by the variable resistor, whereby the apparent inductance can be continuously changed. The phase shift amount of the entire phase shifter can be continuously changed.

According to the twenty-third aspect of the present invention, the inductance of the inductor is changed and the resistance value of the third resistor included in the series circuit is also changed, and the element constants of these two elements are changed. As a result, the phase shift amount can be easily changed. According to a twenty-fourth aspect of the present invention, the third resistance whose resistance value can be changed is specifically formed by a channel of a field effect transistor,
The resistance value of the third resistor can be easily changed only by changing the gate voltage.

According to the twenty-fifth aspect of the present invention, the entire phase shifter is integrally formed on the semiconductor substrate, so that the circuit can be downsized and the manufacturing cost can be reduced by the integration.

[0056]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS A phaser according to an embodiment of the present invention will be specifically described below with reference to the drawings.

[First Embodiment] FIG. 1 is a diagram showing a circuit configuration of a phase advance type phase shifter 100 of the first embodiment. As shown in the figure, the phase shifter 100 of the first embodiment has a FET (field effect transistor) 1 whose gate is connected to the input terminal 20.
0, a resistor 12 and an inductor 14 connected in series between the source and drain of the FET 10, and the FET 10
Of the resistor 16 connected between the source of the
A resistor 18 connected between the drain of T10 and the positive power supply
It is comprised including.

Here, the resistance values of the two resistors 16 and 18 connected to the source and drain of the FET 10 described above are set to be substantially equal, and focusing on the AC component of the input voltage applied to the input terminal 20, A signal having substantially the same amplitude and the same phase is output from the source of the FET 10, and a signal having substantially the same amplitude and the phase inverted (shifted by 180 °) is output from the drain of the FET 10.

The capacitor 24 inserted between the inductor 14 and the drain of the FET 10 is
This is for blocking a direct current flowing by bypassing between the source and drain of, and its impedance is extremely small at the operating frequency, that is, set to a large capacitance so as not to affect the frequency characteristics.

The phase shifter 100 of the first embodiment has such a structure, and its operation will be described below.

When an input signal is input to the input terminal 20,
That is, when a predetermined AC voltage (input voltage Ei) is applied to the gate of the FET 10, an AC voltage having the same phase as the input voltage Ei appears at the source of the FET 10 and conversely the FET 1
An AC voltage having a phase opposite to the input voltage Ei appears at the drain of 0.

A series circuit composed of a resistor 12 and an inductor 14 is connected between the source and drain of the FET 10. Therefore, the source of the FET 10
A signal having a voltage component (AC component) obtained by dividing the potential difference between the drains by the resistor 12 and the inductor 14 is output from the output terminal 22.

FIG. 2 is a vector diagram showing the relationship between the input / output voltage of the phase shifter and the voltage appearing in the inductor or the like.

Since an AC voltage having the same phase and an opposite phase as the input voltage Ei appears at the source and drain of the FET 10, the potential difference (AC component) between the source and drain is 2Ei.
Becomes In addition, the voltage VL1 appearing across the inductor 14
And the voltage VR1 appearing at both ends of the resistor 12 are 90 ° out of phase with each other, and a vector combination (addition) of these results in a potential difference 2 between the source and drain of the FET 10.
It is equal to Ei.

Therefore, as shown in FIG. 2, twice the input voltage Ei is set as the hypotenuse, and the voltage VL1 across the inductor 14 is increased.
And a voltage VR1 across the resistor 12 form a right-angled triangle that constitutes two sides orthogonal to each other. Therefore, when the amplitude of the input signal is constant (Ei is constant) and only the frequency changes, the inductor 14 moves along the circumference of the semicircle shown in FIG.
The voltage VL1 between both ends of V and the voltage VR1 between both ends of the resistor 12 change.

By the way, if the variation of the voltage between the input and output is taken out as the output voltage Eo, this output voltage Eo
The o can be represented by a vector whose starting point is the center point of the semicircle shown in FIG. 2 and whose end point is one point on the circumference where the voltage VL1 and the voltage VR1 intersect. It is equal to the radius Ei. Therefore, in the phase shifter 100 of the present embodiment, the amplitude of the input signal and the amplitude of the output signal are equal, and it is understood that signal attenuation does not occur between the input and output signals. Moreover, even if the frequency of the input signal fluctuates, the end point of this vector only moves on the circumference, so a stable output in which the output amplitude does not fluctuate according to the frequency can be obtained.

As is clear from FIG. 2, the voltage VL1
And the voltage VR1 intersect at a right angle on the circumference, theoretically, the phase difference between the input voltage Ei and the voltage VL1 changes from 90 ° to 0 ° as the angular velocity ω changes from 0 to ∞. The phase shift amount φ of the entire phase shifter 100 is twice that, and changes from 180 ° to 0 ° depending on the frequency.

Next, the relationship between the input and output voltages described above will be verified quantitatively. FIG. 3 is an equivalent view of the phase shifter 100 of this embodiment.

For the source and drain of the FET 10,
Since two in-phase or opposite-phase voltages having the same amplitude as the input voltage Ei are generated, these two voltages are generated.
It can be considered by replacing the two voltage sources 24 and 26. At this time, if the resistance value of the resistor 12 is R and the inductance of the inductor 14 is L, the current I flowing in the closed loop of the equivalent circuit shown in FIG.

[Equation 1] Becomes Therefore, when the potential difference (difference) Eo between the two points shown in FIG.

[Equation 2] Becomes Substituting equation (1) into equation (2) above,

(Equation 3) Becomes Substituting s = jω in this equation (3) and transforming it,

[Equation 4] Becomes When the absolute value of the output voltage Eo is calculated from the equation (4),

(Equation 5) Becomes That is, the expression (5) is obtained by the phase shifter 10 of the present embodiment.
The value 0 indicates that the amplitude of the output signal is equal to the amplitude of the input signal and is constant no matter how the phase between the input and the output is rotated.

Further, when the phase shift amount φ of the output voltage Eo with respect to the input voltage Ei is calculated from the equation (4),

(Equation 6) Becomes From the equation (6), the phase shift amount φ is 90 ° at a frequency such that ω = R / L.

As described above, the phase shifter 100 of this embodiment is
Only the phase can be advanced without attenuating the input signal. Moreover, the amount of phase shift at this time is
It is possible to realize the phase shifter 100 that is set in a range of up to 180 ° and has a large amount of phase per stage.

Second Embodiment The phase shifter 100 of the first embodiment described above is for advancing the phase of the output signal with respect to the input signal in the range of 0 ° to 180 °. The phase shifter of the second embodiment is characterized in that the phase of the output signal is delayed in the range of 0 ° to 180 ° with respect to the input signal.

FIG. 4 shows a phase shifter 20 of the slow phase type according to the second embodiment.
It is a figure which shows the circuit structure of 0. As shown in FIG.
The phase shifter 200 of the embodiment includes a FET 30 having a gate connected to the input terminal 20, an inductor 32 and a resistor 34 connected in series between the source and drain of the FET 30.
And a resistor 36 connected between the source of the FET 30 and the negative power supply, and a resistor 38 connected between the drain of the FET 30 and the positive power supply.

Here, the resistance values of the two resistors 36 and 38 connected to the source and drain of the above-mentioned FET 30 are set to be substantially equal, and the inductor 32.
Like the phase shifter 100 of the first embodiment described above, a capacitor 24 for blocking a direct current is further inserted in the series circuit including the resistor 34 and the resistor 34.

The phase shifter 200 of the second embodiment has such a structure, and its operation will be described below.

When an input signal is input to the input terminal 20,
That is, when a predetermined AC voltage (input voltage Ei) is applied to the gate of the FET 30, an AC voltage having the same phase as the input voltage Ei appears at the source of the FET 30, and conversely the FET 3
An AC voltage having a phase opposite to the input voltage Ei appears at the drain of 0.

A series circuit composed of an inductor 32 and a resistor 34 is connected between the source and drain of the FET 30. Therefore, the source of the FET 30
A signal having a voltage (AC component) obtained by dividing the potential difference between the drains by the inductor 32 and the resistor 34 is output from the output terminal 22.

FIG. 5 is a vector diagram showing the relationship between the input / output voltage of the phase shifter of this embodiment and the voltage appearing in the inductor or the like.

The FET 1 shown in FIG. 1 in the first embodiment.
Similar to 0, since an AC voltage having the same phase and an opposite phase to the input voltage Ei appears at the source and drain of the FET 30, the potential difference (AC component) between the source and drain is 2Ei.
Becomes Further, the voltage VR2 appearing across the resistor 34 and the voltage VL2 appearing across the inductor 32 are 90 ° out of phase with each other, and a vector combination (addition) of these results between the source and drain of the FET 30. Potential difference 2
It is equal to Ei.

Therefore, as shown in FIG. 5, double the input voltage Ei is used as a hypotenuse to form a right-angled triangle that forms two sides where the voltage VR2 across the resistor 34 and the voltage VL2 across the inductor 32 are orthogonal to each other. become. Therefore, when the amplitude of the input signal is constant (Ei is constant) and only the frequency is changed, the voltage VR2 across the resistor 34 and the voltage VL2 across the inductor 32 are distributed along the circumference of the semicircle shown in FIG. Changes.

By the way, if the variation of the voltage between the input and output is taken out as the output voltage Eo, this output voltage Eo
The o can be represented by a vector whose starting point is the center point of the semicircle shown in FIG. 5 and whose end point is one point on the circumference where the voltage VR2 and the voltage VL2 intersect. It is equal to the radius Ei. Therefore, in the phase shifter 200 of the present embodiment, the amplitude of the input signal and the amplitude of the output signal are equal, and it can be seen that no signal attenuation occurs between the input and output signals. Moreover, even if the frequency of the input signal fluctuates, the end point of this vector only moves on the circumference, so a stable output in which the output amplitude does not fluctuate according to the frequency can be obtained.

As is clear from FIG. 5, the voltage VR2
And the voltage VL2 intersect at a right angle on the circumference, theoretically, the phase difference between the input voltage Ei and the voltage VR2 changes from 0 ° to 90 ° as the angular velocity ω changes from 0 to ∞. The phase shift amount φ ′ of the entire phase shifter 200 is twice that and changes from 0 ° to 180 ° depending on the frequency.

Next, the relationship between the input and output voltages described above will be verified quantitatively. FIG. 6 is an equivalent view of the phase shifter 200 of this embodiment.

For the source and drain of the FET 30,
Since in-phase and anti-phase voltages having the same amplitude as the input voltage Ei are generated, the two voltage sources 24 and 26 can be used instead as in the case shown in FIG.
At this time, the current I flowing in the closed loop of the equivalent circuit shown in FIG.
Assuming that the resistance value of R is R, the above equation (1) can be applied as it is.

Therefore, when the potential difference (difference) Eo between the two points shown in FIG. 6 is obtained,

(Equation 7) Becomes Substituting equation (1) into equation (7) above,

(Equation 8) Becomes Substituting s = jω in the equation (8) and transforming it,

[Equation 9] Becomes The expression (9) differs from the expression (4) shown in the first embodiment only in the sign. Therefore, as the absolute value of the output voltage Eo, the equation (5) can be applied as it is, and the phase shifter 200 of the second embodiment can output the amplitude of the output signal no matter how the phase between the input and the output rotates. It can be seen that is constant and equal to the amplitude of the input signal.

When the phase shift amount φ'of the output voltage Eo with respect to the input voltage Ei is calculated,

[Equation 10] Becomes Therefore, for example, the phase shift amount φ ′ at the frequency where ω = R / L is 90 °.

In this way, the phase shifter 200 of this embodiment is
Only the phase can be delayed without attenuating the input signal. Moreover, the amount of phase shift at this time is
It is possible to realize the phase shifter 200 that is set in the range of 0 ° to a maximum of 180 ° and has a large amount of phase per stage.

[Other Embodiments] Next, other embodiments in which the phase shifters of the above-described embodiments are modified will be described.

In the phase shifters 100 and 200 of the respective embodiments shown in FIG. 1 or 4, a constant amount of phase shift determined by the element constants of the resistor 12 and the inductor 14 or the inductor 32 and the resistor 34 is set. , The amount of phase shift is fixedly determined for a certain frequency. Therefore, if the element constant of the resistor or the inductor described above can be changed, the amount of phase shift can be arbitrarily changed.

FIG. 7 is a diagram showing the construction of a phase-advancing type phase shifter in which the amount of phase shift can be continuously changed by using a variable resistor. The phase shifter 100 of the first embodiment is shown in FIG. It corresponds.

The phase shifter 100a shown in FIG.
The resistance 12 of the phase shifter 100 shown in FIG. 2 is replaced by a variable resistance 12a. By continuously changing the resistance value of the variable resistance 12a, the phase shift amount of the entire phase shifter 100a is also continuous. Changes to. That is, when the resistance value of the variable resistor 12a increases, the variable resistor 12a and the inductor 1
Since the potential of the connection point 4 moves counterclockwise on the circumference shown in FIG. 2, the phase shift amount φ of the entire phase shifter 100a continuously increases accordingly. On the contrary, variable resistance 1
When the resistance value of 2a decreases, the potential of the connection point between the variable resistor 12a and the inductor 14 moves clockwise on the circumference shown in FIG. 2, and accordingly, the phase shift amount φ of the entire phase shifter 100a. Also becomes smaller continuously.

Further, the phase shifter 100b shown in FIG.
More specifically has a configuration in which the variable resistor 12a is replaced with a MOS type or junction type FET 12b.
By using the channel of the FET 12b as a resistor and variably controlling the gate voltage to change the channel resistance, the phase shift amount of the entire phase shifter 100b can be continuously changed.

The phase shifter 100b shown in FIG. 7B is used.
In the above, the variable resistance is constituted by the p-channel or n-channel FET 12b.
The variable resistor may be configured by a transmission gate in which T and an n-channel FET are connected in parallel.

Similarly, FIG. 8 is a diagram showing the structure of a delay type phase shifter in which the amount of phase shift can be continuously changed by using a variable resistor. The phase shifter of the second embodiment is shown in FIG. It corresponds to 200.

The phase shifter 200a shown in FIG.
The resistance 34 of the phase shifter 200 shown in FIG. 2 is replaced with a variable resistance 34a. By continuously changing the resistance value of the variable resistance 34a, the phase shift amount of the entire phase shifter 200a is also continuously changed. Changes to. That is, when the resistance value of the variable resistor 34 a increases, the inductor 32 and the variable resistor 34 a
Since the potential of the connection point of a moves counterclockwise on the circumference shown in FIG. 5, the phase shift amount φ ′ of the entire phase shifter 200a continuously decreases accordingly. On the contrary, when the resistance value of the variable resistor 34a decreases, the potential at the connection point between the inductor 32 and the variable resistor 34a moves clockwise on the circumference shown in FIG. The phase shift amount φ ′ also continuously increases.

Further, the phase shifter 200b shown in FIG.
More specifically has a configuration in which the variable resistor 34a is replaced with the FET 34b, and variably controls the gate voltage to continuously change the channel resistance, thereby changing the phase shift amount of the entire phase shifter 200b. It is changing. Note that FIG.
Similar to the phase shifters 100a and 100b shown in FIG. 5, a variable resistance is formed by a transmission gate in which a p-channel FET and an n-channel FET are connected in parallel to obtain a stable ON resistance regardless of the voltage level of the input signal. You may do it.

FIG. 9 is a diagram showing the structure of a phase shifter in which the amount of phase shift can be continuously changed by using a variable inductor. FIG. 9A shows a phase advance type phase shifter. 100c
However, FIG. 3B shows each of the delay type phase shifters 200c.

The phase shifter 100c shown in FIG.
It has a configuration in which the inductor 14 of the phase shifter 100 shown in FIG. 2 is replaced with the variable inductor 14a, and the phase shift amount of the entire phase shifter 100c is also continuously changed by continuously changing the inductance of the variable inductor 14a. Change.
That is, when the inductance of the variable inductor 14a increases, the potential of the connection point between the resistor 12 and the variable inductor 14a moves clockwise on the circumference shown in FIG. 2, and accordingly, the phase shift of the entire phase shifter 100c. Amount φ
Also becomes smaller continuously. On the contrary, the variable inductor 14a
When the inductance of the phase shifter 100c decreases, the potential at the connection point of the resistor 12 and the variable inductor 14a moves counterclockwise on the circumference shown in FIG.
The overall phase shift amount φ also continuously increases.

Similarly, the phase shifter 200c shown in FIG.
Has a configuration in which the inductor 32 of the phase shifter 200 shown in FIG. 4 is replaced with a variable inductor 32a, and the amount of phase shift of the entire phase shifter 200c is changed by continuously changing the inductance of the variable inductor 32a. Also changes continuously. That is, when the inductance of the variable inductor 32a increases, the resistance 34 and the variable inductor 32a increase.
Since the potential at the connection point of ∘ moves clockwise on the circumference shown in FIG. 5, the phase shift amount φ ′ of the entire phase shifter 200c continuously increases accordingly. On the contrary, when the inductance of the variable inductor 32a decreases, the potential at the connection point between the resistor 34 and the variable inductor 32a moves counterclockwise on the circumference shown in FIG. The phase shift amount φ ′ also continuously decreases.

As shown in FIGS. 7 to 9, the phase shift amount of the entire phase shifter is continuously changed by changing the element constant of either the resistor or the inductor forming the series circuit. However, it is also possible to change the element constants of both the resistance and the inductor that form the series circuit.

FIG. 10 is a diagram showing the structure of a phase shifter in which a series circuit is composed of a variable resistor and a variable inductor.
FIG. 1A shows the structure of a phase shifter 100d in which a series circuit is composed of a variable resistor 12a and a variable inductor 14a. The phase shifter 100d can continuously reduce the phase shift amount φ by reducing the resistance value of the variable resistor 12a or increasing the inductance of the variable inductor 14a. On the contrary, the phase shifter 100d is configured by increasing the resistance value of the variable resistor 12a or by changing the variable inductor 14a.
It is possible to continuously increase the phase shift amount φ by reducing the inductance of.

Further, FIG. 9B shows the structure of a phase shifter 200d in which a series circuit is composed of a variable inductor 32a and a variable resistor 34a. This phase shifter 200d
Can reduce the phase shift amount φ ′ continuously by reducing the inductance of the variable inductor 32a or by increasing the resistance value of the variable resistor 34a. On the contrary, the phase shifter 200d can continuously increase the phase shift amount φ'by increasing the inductance of the variable inductor 32a or decreasing the resistance value of the variable resistor 34a.

FIG. 11 shows the variable inductor 14a described above.
It is a figure which shows an example of (or variable inductor 32a), and the outline of the planar structure formed on the semiconductor substrate is shown.

The variable inductor 14a shown in the figure has a spiral inductor conductor 112 formed on a semiconductor substrate 110, a control conductor 114 formed so as to surround the outer circumference thereof, and the inductor conductor 112 and the control conductor. The insulating magnetic body 118 is formed so as to cover both the conductors 114.

A variable voltage power supply 116 is connected to both ends of the control conductor 114 to apply a variable bias voltage to the control conductor 114.
By variably controlling the bias voltage applied by, the bias current flowing through the control conductor 114 can be changed.

For the semiconductor substrate 110, for example, an n-type silicon substrate (n-Si substrate) or another semiconductor material (for example, an amorphous material such as germanium or amorphous silicon) is used. The inductor conductor 112 is formed by spirally forming a metal thin film such as aluminum or gold or a semiconductor material such as polysilicon.

In addition to the variable inductor 14a, the semiconductor substrate 110 shown in FIG. 11 includes the phase shifter 1 shown in FIG.
00 (or other circuit of the phase shifter 200 shown in FIG. 4 in the case of the variable inductor 32a).

FIG. 12 shows an inductor conductor 112 and a control conductor 114 of the variable inductor 14a shown in FIG.
It is a figure which shows the shape of more in detail.

As shown in the figure, the inductor conductor 112 located on the inner peripheral side is formed in a spiral shape with a predetermined number of turns (for example, about 4 turns), and two terminal electrodes 122 and 124 are provided at both ends thereof. Are connected. Similarly,
The control conductor 114 located on the outer peripheral side is formed in a spiral shape with a predetermined number of turns (for example, about 2 turns),
Two control electrodes 126 and 128 are connected to both ends thereof.

FIG. 13 is an enlarged cross-sectional view taken along the line AA of FIG. 12, showing a cross section of the insulating magnetic body 118 including the inductor conductor 112 and the control conductor 114.

As shown in the figure, the inductor conductor 1 is formed on the surface of the semiconductor substrate 110 via the insulating magnetic film 118a.
12 and a control conductor 114 are formed, and an insulating magnetic film 118b is formed on the surface of the control conductor 114. The insulating magnetic material 118 shown in FIG. 11 is formed by these two magnetic material films 118a and 118b.

For example, various magnetic films such as gamma ferrite and barium ferrite can be used as the magnetic films 118a and 118b. Various kinds of materials and forming methods of these magnetic films can be considered. For example, a method of forming a magnetic film by vacuum deposition of FeO or the like, other molecular beam epitaxy method (MBE method), chemical vapor deposition, etc. There is a method of forming a magnetic film using a phase growth method (CVD method), a sputtering method or the like.

The insulating film 130 is made of a non-magnetic material, and covers the winding portions of the inductor conductor 112 and the control conductor 114. By eliminating the magnetic films 118a and 118b between the winding portions in this way, the leakage flux generated between the winding portions can be minimized, and thus the inductor conductor 112 can be formed.
It is possible to realize the variable inductor 14a having a large inductance by effectively utilizing the magnetic flux generated by.

As described above, the variable inductor 14a (or 32a) shown in FIG.
2 and the control conductor 114 so as to cover the insulating magnetic body 11
8 (magnetic films 118a and 118b) are formed,
By variably controlling the DC bias current flowing through the control conductor 114, the saturation magnetization characteristic of the inductor conductor 112 having the insulating magnetic body 118 as a magnetic path changes, and the inductance of the inductor conductor 112 changes.

Therefore, the inductance itself of the inductor conductor 112 can be directly changed, and the inductor conductor 112 can be formed on the semiconductor substrate 110 by using a thin film forming technique or a semiconductor manufacturing technique, which facilitates the production. Further, since it is possible to form other components such as the phase shifter 100 on the semiconductor substrate 110, it is suitable when integrally forming the entire phase shifter 100 and the like by integration.

The variable inductor 1 shown in FIG.
4a and 32a, as shown in FIG. 14 or FIG.
The inductor conductor 112 and the control conductor 114 are alternately wound around, or the inductor conductor 112 and the control conductor 114 are circulated.
You may make it overlap and form. In either case, there is no difference in that a magnetic flux penetrating the insulating magnetic body 118 located on the inner peripheral portion of the control conductor 114 can be generated, and the insulating magnetic substance when the inductor conductor 112 is energized is the same. By changing the saturation magnetization characteristic of the body 118, the inductance of the inductor conductor 112 can be changed within a certain range.

Further, the variable inductor 1 shown in FIG.
4a has been described by taking the case where the inductor conductor 112 and the like are formed on the semiconductor substrate 110 as an example, but it may be formed on various insulating or conductive substrates such as ceramics. Further, although the insulating material is used as the magnetic films 118a and 118b, a conductive material such as metal powder (MP) may be used. However, when such a conductive magnetic film is used by replacing it with the above-mentioned insulating magnetic film 118a or the like, each winding portion of the inductor conductor 112 or the like is short-circuited and does not function as an inductor conductor. It is necessary to electrically insulate between the conductor and the conductive magnetic film. Examples of this insulating method include a method of oxidizing the inductor conductor 112 and the like to form an insulating oxide film, a method of forming a silicon oxide film or a nitride film by a chemical vapor deposition method, and the like.

In particular, a conductive material such as metal powder has a larger magnetic permeability than an insulating material such as gamma-ferrite, and therefore has an advantage that a large inductance can be secured.

In addition, the variable inductor 1 shown in FIG.
In 4a, both the inductor conductor 112 and the control conductor 114 are entirely covered with the insulating magnetic body 118, but a magnetic path may be formed by only partially covering them.

FIG. 16 is a diagram showing a variable inductor in which the insulating magnetic material 118 is partially formed. As shown in the figure, the insulating magnetic body 18 is formed so as to cover a part of the inductor conductor 112 and the control conductor 114, and the partially formed insulating magnetic body 118 forms a magnetic path. To be done. In this way, when the insulating magnetic body (or the conductive magnetic body may be used) 118 that serves as a magnetic path is partially formed, the inductor path is narrowed due to the narrow magnetic path.
The magnetic flux generated by 12 and the control conductor 114 is easily saturated. Therefore, even when a small bias current is passed through the control conductor 114, the magnetic flux is saturated, and the inductance of the inductor conductor 112 can be changed by variably controlling the small bias current.
Therefore, the structure of the control system can be simplified.

FIG. 17 is a circuit diagram showing a modified example of the inductor 14 and the like used in the phase shifter 100 and the like shown in the above-mentioned first and second embodiments. The entire circuit shown in FIG. 17 is an inductor. It corresponds to 14.

The inductor 14b shown in the figure has an inductor 210 having a predetermined inductance L0, two operational amplifiers 212 and 214, and two resistors 216 and 21.
And 8 are included.

The operational amplifier 212 in the first stage has an output terminal connected to the inverting input terminal, is a non-inverting amplifier with a gain of 1, and mainly functions as a buffer for impedance conversion. Similarly, the output terminal of the second-stage operational amplifier 214 is also connected to the inverting input terminal, and functions as a non-inverting amplifier having a gain of 1. Further, a voltage dividing circuit by resistors 216 and 218 is inserted between these two non-inverting amplifiers.

By thus inserting the voltage dividing circuit, the overall gain including the two non-inverting amplifiers can be freely set between 0 and less than 1.

When the transfer function of the entire circuit of the inductor 14b shown in FIG. 17 excluding the inductor 210 is K, the inductor 14b can be represented by the system diagram shown in FIG. FIG. 19 is a system diagram in which this is converted by the Miller's theorem.

When the impedance Z0 shown in FIG. 18 is used to represent the impedance Z1 shown in FIG.

[Equation 11] Becomes Here, in the case of the inductor 14b shown in FIG. 17, the impedance Z0 = jωL0, and by substituting this into the equation (11),

(Equation 12)

(Equation 13) Becomes The equation (13) shows that the inductance L0 of the inductor 210 in the inductor 14b is apparently 1 / (1-K) times.

Therefore, when the gain of the amplifier is positive, 0 <(1-K) is always maintained in the range of 0 ≦ K <1.
Since .ltoreq.1, it is possible to change the inductance L0 to the larger one.

By the way, the inductor 14 shown in FIG.
The gain of the amplifier in b, that is, the gain K of the amplifier formed by the entire operational amplifiers 212 and 214 is 2
It is determined by the voltage division ratio of the voltage dividing circuit constituted by 16 and the resistor 218, and when the respective resistors are R16 and R18,

[Equation 14] Becomes Substituting equation (14) into equation (13),

(Equation 15) Becomes

Therefore, the apparent inductance L between the two terminals 220 and 222 can be increased by increasing the resistance ratio R18 / R16 of the resistors 216 and 218. For example, when R18 = R16, the inductance L can be doubled from L0 from the equation (15).

As described above, the inductor 14b described above is used.
Can change the voltage dividing ratio of the voltage dividing circuit inserted between the two non-inverting amplifiers to apparently increase the inductance L0 of the inductor 210 that is actually connected. Therefore, when the phase shifter 100 shown in FIG. 1 and the like is formed on the semiconductor substrate, the inductor 210 having a small inductance L0 is formed by forming the conductor in a spiral shape on the semiconductor substrate. In advance, the configuration shown in FIG. 17 can be converted into a large inductance L, which is convenient for integration. In particular, if a large inductance can be secured in this way, the mounting area of the phase shifter 100 and the like can be reduced, and the material cost and the like can be reduced.

In addition to the case where the voltage dividing ratio of the voltage dividing circuit by the resistors 216 and 218 is fixed, these two resistors 21
At least one of 6, 218 is a variable resistor (specifically,
Similar to the phase shifter 100b of the first embodiment, the MOS-FE is used.
This variable resistance can be formed by a T or J-FET, or a transmission gate in which a p-channel MOS and an n-channel MOS are connected in parallel), and this voltage division ratio may be continuously changed. .
In this case, the gain of the entire amplifier shown in FIG. 17 changes, and the inductance L between the terminals 220 and 222 also changes continuously. Therefore, the phase shift amount of the phase shifter 100 or the like using this changes arbitrarily and continuously. Can be changed.

In addition, the inductor 14b shown in FIG.
Since the gain of the entire amplifier including the two operational amplifiers 212 and 214 is set to 1 or less, it may be replaced with an emitter follower circuit or a source follower circuit.

FIG. 20 is a diagram showing a structure of an inductor in which the entire amplifier including the operational amplifiers 212 and 214 is replaced with an emitter follower circuit. The inductor 14c shown in FIG. 9A includes a bipolar transistor 228 whose emitter is connected to two resistors 224 and 226, a voltage dividing point by the two resistors 224 and 226, and a transistor 228.
Of the inductor 210 and a capacitor 230 for blocking a direct current. The capacitor 230 inserted in the one end side of the inductor 210 is for blocking a direct current, and its impedance is extremely small at the operating frequency, that is, is set to a large capacitance so as not to affect the frequency characteristics.

The gain of the above-described emitter follower circuit is determined mainly by the resistance ratio of the two resistors 224 and 226.
Moreover, since the gain is always less than 1, the inductance L0 actually possessed by the inductor 210 can be apparently increased, as can be seen from the equation (15). Moreover, since only one emitter follower circuit is used, the circuit configuration can be simplified and the maximum operating frequency can be set high.

FIG. 20B is a diagram showing a modified example thereof, which is different in that the two resistors 224 and 226 in FIG. 20A are replaced with variable resistors 232. By using the variable resistor 232 in this way, the gain can be changed arbitrarily and continuously, so that the apparent inductance L can be changed arbitrarily and continuously, and the phase shifter 100 and the like can be changed. The amount of phase shift can also be continuously changed.

The inductor 1 shown in FIG.
4c replaces the two resistors 224 and 226 of FIG. 7A with one variable resistor 232. However, at least one of these two resistors 224 and 226 may be configured by a variable resistor.

In FIG. 21, the inductors 14c and 14d shown in FIGS. 20A and 20B are realized by a source follower circuit, and the bipolar transistor 228 is replaced with a FET 234.
FIG. 21A shows FIG. 20A and FIG. 21B shows FIG.
Each corresponds to (B).

FIG. 22 shows the inductor 14 shown in FIG.
It is a figure which shows the modification of c, and shows the structure of the inductor 14e which does not use the capacitor 230 for a direct current blocking. The inductor 14e shown in FIG. 22 includes an npn-type bipolar transistor 236 and a resistor 240 connected to its emitter, a pnp-type bipolar transistor 238 and a resistor 242 connected to its emitter, and
And an inductor 210 having an inductance L0.

A first emitter follower circuit is formed by the one transistor 236 and the resistor 240, and a second emitter follower circuit is formed by the other transistor 238 and the resistor 242, which are connected in cascade. Moreover, the npn-type transistor 236 and p
Since the np-type transistor 238 is used, the base potential of the transistor 236 at one end of the inductor 210 and the emitter potential of the transistor 238 can be set to be substantially the same, and the DC current blocking capacitor 2 can be set.
30 is unnecessary.

By the way, in the above-described inductor 14a and the like, the inductance L0 of the inductor 210 is actually changed to a larger value, but if the gain of the amplifier can be set to be negative as apparent from the equation (13), it is apparent. The upper inductance L can be changed to be smaller than L0.

FIG. 23 is a diagram showing the structure of the inductor 14f in which the apparent inductance is smaller than that of the actual element. The inductor 14f shown in the figure includes an inductor 210 having a predetermined inductance L0 and two operational amplifiers (operational amplifiers) 25.
0,252 and four resistors 254,256,258,2
And 60.

In the first-stage operational amplifier 250, a resistor 256 (whose resistance value is R56) is connected between the output terminal and the inverting input terminal, and the inverting input terminal further has the resistor 254 (this resistance value is R56). Is designated as R54) and is grounded.

Between the voltage E0 applied to the non-inverting input terminal of the first stage operational amplifier 250 and the voltage E1 appearing at the output terminal,

[Equation 16] There is a relationship. The first-stage operational amplifier 250 mainly functions as a buffer that performs impedance conversion, and the gain may be 1. The case of gain 1 is R
56 / R54 = 0, that is, R54 is infinite (resistance 25
4 should be removed) or R56 should be set to 0Ω (it should be directly connected).

The second stage operational amplifier 252 has a resistor 260 (whose resistance value is R60) connected between the output terminal and the inverting input terminal, and the inverting input terminal and the output of the operational amplifier 250 described above. Resistor 25 between terminals
8 (whose resistance is R58) is connected, and the non-inverting input terminal is grounded.

When the voltage appearing at the output terminal of the second operational amplifier 252 is E2, the voltage E2 appears between this voltage E2 and the voltage E1 appearing at the output terminal of the first operational amplifier 250.

[Equation 17] There is a relationship. As described above, the second-stage operational amplifier 252 functions as an inverting amplifier, and the first-stage operational amplifier 2 is used to set its input side to high impedance.
50 are used.

In addition, between the non-inverting input terminal of the first-stage operational amplifier 250 and the output terminal of the second-stage operational amplifier 252, which are connected as described above, the above-mentioned inductor 21 is provided.
0 is connected.

The inductor 14f having such a configuration
The gain of the amplifier at, ie, operational amplifiers 250 and 2
The gain K of the amplifier constituted by 52 as a whole is (1
From equations (6) and (17),

(Equation 18) Becomes Therefore, if this equation (18) is substituted into equation (13),

[Formula 19] Becomes Therefore, the four resistors 254, 256, 25
By setting the resistance value of 8, 260 to a predetermined value,
It can be changed from the maximum L0 to the smaller one.

When the gain of the amplifier by the first stage operational amplifier 250 is 1, that is, when R54 is set to infinity (resistor 254 is removed) or R56 is set to 0Ω as described above, R56 / R54 When = 0, the above equation (19) is simplified and

(Equation 20) Becomes

FIG. 24 is a diagram showing the structure of the inductor 14g from which the resistor 254 connected to the inverting input terminal of the first operational amplifier 250 shown in FIG. 23 is removed. In this case, since the inductance L appearing between the terminals 220 and 222 is expressed by the equation (20), the maximum value L0 can be changed to the smaller value only by changing the ratio of R60 and R58.

Further, the resistors 254, 256, 258, 26
At least one of 0 (inductor 1 shown in FIG.
In the case of 4 g, at least one of the resistors 258 and 260) is formed by a variable resistor.
A variable inductor can be easily formed by forming a variable resistance with a -FET, J-FET, or a transmission gate in which a p-channel FET and an n-channel FET are connected in parallel. Therefore, by using this variable inductor as the variable inductor 14a or the like shown in FIG. 9 or 10, it is possible to form a phase shifter capable of arbitrarily changing the phase shift amount within a certain range.

Since the first stage operational amplifier 250 is used as a buffer for increasing the input impedance as described above, the operational amplifier 250 may be replaced with an emitter follower circuit or a source follower circuit.

FIG. 25 is a diagram showing the structure of the inductor 14h using the emitter follower circuit in the first stage. The inductor 14h shown in the same figure is an emitter follower circuit 2 including a first-stage operational amplifier 250 and two resistors 254 and 256 shown in FIG.
It has a configuration replaced with 62. Capacitor 230
Are used for DC blocking to obtain the proper gate voltage.

FIG. 26 is a diagram showing the configuration of the inductor 14i using the source follower circuit in the first stage. The inductor 14i shown in the figure has a configuration in which the first stage operational amplifier 250 and the two resistors 254 and 256 shown in FIG. 23 are replaced with a source follower circuit 264 composed of an FET and a resistor. The inductor 14h shown in FIG. 25 is the same as the inductor 14h shown in FIG. 25 in that the capacitor 230 for blocking a direct current is provided to obtain an appropriate gate voltage.

The inductors 14h and 14i described above are also included.
Of each of the terminals 22 by changing the resistance ratio of the resistors 258 and 260 connected to the operational amplifier 252.
The point that the apparent inductance L between 0 and 222 can be arbitrarily changed is the same as the inductor 14f and the like shown in FIG. 23 and the like. Therefore, the resistors 258, 26
A variable inductor can be configured by replacing at least one of 0 with a variable resistor formed by a transmission gate in which a MOS-FET, a J-FET, or a p-channel FET and an n-channel FET are connected in parallel. By using the variable inductor as the variable inductor 14a shown in FIG. 9 and the like, it is possible to form a phase shifter capable of arbitrarily changing the phase shift amount within a certain range.

The present invention is not limited to the above embodiments, and various modifications can be made within the scope of the gist of the present invention.

For example, the phase shifter of each of the above-described embodiments can theoretically shift the phase up to 180 ° by one, but when actually shifting the phase by 180 ° or more, it is 2 It suffices to connect and use more than one phase shifter.

In each of the above embodiments, the FE
Although the phase shifter is constructed by using T10 or 30, especially in FIG.
Alternatively, the phase shifter may be constituted by a bipolar transistor.

FIG. 27 shows the FET 10 or the FET 30.
It is a figure which shows the structure of the phase shifter which used the bipolar transistor 40 instead of. FIG. 1A shows a configuration of a phase shifter 100e in which the FET 10 is replaced with a bipolar transistor 40 in the phase advance type phase shifter 100 shown in FIG. Similarly, FIG. 3B shows the configuration of the phase shifter 200e in which the FET 30 is replaced with the bipolar transistor 40 in the phase retarder 200 of the delay type shown in FIG.

In the phase shifters 100e and 200e using the bipolar transistor 40, since a current flows between the base and the emitter when an input signal is input to the base, the voltage (AC voltage) appearing at the emitter and the collector appearing at the collector. It is not exactly the same as the voltage (AC voltage). However, when the current amplification factor is several tens to 100 times, the difference is 1% to several%, which can be practically ignored.

Therefore, the bipolar transistor can be used in the phase shifter 100 and the like, and the bipolar transistor and the FET can be selectively used according to the application. That is, when it is desired to increase the upper limit of the operating frequency, a phase shifter using a bipolar transistor is preferable, and when it is desired to increase the input impedance, the FE is used.
A phase shifter using T is preferred.

Further, although the configuration corresponding to the phase shifter shown in FIG. 1 or 4 has been described with reference to FIG. 27, FIG.
A resistor 12 forming a series circuit in the phase shifter shown in FIG.
Alternatively, 34 can be replaced with a variable resistor using the channel of the FET, or the inductor 14 or 32 can be replaced with the variable inductor shown in FIG. 11 or FIG. 17 to form a phase shifter whose phase shift amount can be changed. Needless to say.

[0162]

As described above, according to the invention of claim 1, a phase difference of about twice the phase angle shifted by the third resistor and the inductor is generated between the input and output signals, which is large. The amount of phase shift can be set. Further, since the input signal and the output signal have the same amplitude, there is no signal attenuation due to the phase shift, and a stable output amplitude can always be obtained even when the frequency of the input / output signal fluctuates.

According to the invention of claim 2, claim 1
A bipolar transistor is used instead of the transistor (field-effect transistor) used in the above, and a phase shifter can be configured based on exactly the same operating principle, and there is no attenuation of the signal amplitude and there is a large phase shift amount. A phaser can be realized. Particularly, when the bipolar transistor is used, the maximum operating frequency can be increased, and conversely, when the field effect transistor is used, the input impedance can be increased.

Further, according to the invention of claim 3, between the source and drain or the emitter and drain of the above-mentioned transistor.
A direct current blocking capacitor is further inserted in the series circuit between the collectors, and it is possible to prevent the source / drain or the emitter / collector from being short-circuited in terms of direct current and not operating.

Further, according to the invention of claim 4, the third resistor in the series circuit is connected to the source side or the emitter side of the transistor, and a voltage having a phase advanced from the input signal is applied across the inductor. By generating it, it is possible to easily realize a phase-advance type phase shifter having a phase shift amount twice this phase advance angle.

According to the fifth aspect of the invention, the inductor in the series circuit is connected to the source side or the emitter side of the transistor, and a voltage delayed in phase from the input signal is applied across the third resistor. By generating it, it is possible to easily realize a delay type phase shifter having a phase shift amount twice this phase delay angle.

According to the sixth aspect of the present invention, the phase shift amount can be easily changed by changing the resistance value of the third resistor forming the second series circuit described above.

According to the seventh aspect of the present invention, the phase shift amount can be easily changed by changing the inductance of the inductor forming the second series circuit.

Further, according to the invention of claim 8, by changing both the resistance value of the third resistor and the inductance of the inductor which form the above-mentioned second series circuit,
The amount of phase shift can be easily changed.

According to the ninth aspect of the invention, the third resistor whose resistance value can be changed is specifically formed by the channel of the field effect transistor, and the third resistor can be easily changed by changing the gate voltage. The resistance value of the resistor 3 can be changed.

According to the invention of claim 10 or 11, there is disclosed a specific example of an inductor with variable inductance, in which a magnetic body is formed so as to cover the inductor conductor and the control conductor. By variably controlling the DC bias current flowing through the control conductor, the saturation magnetization characteristic of the inductor conductor having a magnetic body as a magnetic path changes, and the inductance of the inductor conductor can be changed.

According to the twelfth aspect of the invention, the above-described inductor is configured by connecting an inductor element having a predetermined inductance and an amplifier in parallel, the inductance of the inductor element is L0, and the transfer function of the amplifier is Is K, the inductance seen from the input side of the amplifier is L0 / (1-K). Therefore, the apparent inductance can be made different from the actual inductance L0 in accordance with the transfer function K, that is, the gain of the amplifier, and in reality, the small inductance can be apparently made large. In particular, when the entire phase shifter is integrated on the semiconductor substrate, only an inductor element having a small inductance can be formed on the semiconductor substrate. Therefore, if this small inductance can be converted into a large inductance, it is possible to integrate it. It is particularly convenient, and there is also an effect of cost reduction by reducing the mounting area.

According to the invention of claim 13 or 14, since the apparent inductance is L0 / (1-K), the gain K of the amplifier is set to a negative value to make L0
L0 can be changed to a larger value by setting the gain to a smaller value and the gain of the amplifier between 0 and 1.

According to the fifteenth aspect of the invention, the amplifier according to the twelfth or thirteenth aspect of the invention is configured by an inverting amplifier using an operational amplifier, and the gain of the amplifier can always be negative.

According to the sixteenth aspect of the invention, the input impedance can be increased by inserting a buffer on the input side of the operational amplifier.
According to the invention of claim 8, this buffer can be easily constructed by using an operational amplifier, or by an emitter follower circuit or a source follower circuit.

According to the nineteenth aspect of the present invention, the amplification factor of the inverting amplifier can be varied by configuring the resistance forming the inverting amplifier together with the above-mentioned operational amplifier with a variable resistor. By continuously changing the inductance of, the amount of phase shift of the entire phase shifter can also be continuously changed.

According to the twentieth or twenty-first aspect of the invention, the amplifier according to the twelfth or fourteenth aspect of the invention is specifically configured by two non-inverting amplifiers or an emitter follower circuit, a source follower circuit and a voltage dividing circuit. Therefore, by using the voltage dividing circuit, the overall gain can be set between 0 and 1 and the apparent inductance can be changed to a value larger than L0.

According to the twenty-second aspect of the present invention, the overall gain can be varied by configuring the resistor composing the above voltage dividing circuit by the variable resistor, whereby the apparent inductance is also continuous. The phase shift amount of the entire phase shifter can also be continuously changed.

According to the twenty-third aspect of the present invention, the inductance of the above-mentioned inductor is changed and the resistance value of the third resistor included in the second series circuit is also changed. The phase shift amount can be easily changed by changing the element constant. According to the invention of claim 24, the third variable resistance value can be changed.
Is specifically formed by the channel of the field effect transistor, and the resistance value of the third resistor can be easily changed only by changing the gate voltage.

According to the twenty-fifth aspect of the invention, the entire phase shifter is integrally formed on the semiconductor substrate, so that it is possible to reduce the circuit size and manufacturing cost by integration.

[0181]

[Brief description of drawings]

FIG. 1 is a diagram showing a circuit configuration of a phase advance type phase shifter of a first embodiment.

FIG. 2 is a vector diagram showing the relationship between the input / output voltage of the phase shifter of the present embodiment and the voltage appearing in the inductor or the like.

FIG. 3 is a diagram showing an equivalent circuit of the phase shifter of the first embodiment.

FIG. 4 is a diagram showing a circuit configuration of a delay type phase shifter of a second embodiment.

FIG. 5 is a vector diagram showing the relationship between the input / output voltage of the phase shifter of the present embodiment and the voltage appearing in the inductor or the like.

FIG. 6 is a diagram showing an equivalent circuit of the phase shifter of the second embodiment.

FIG. 7 is a diagram showing a configuration of a phase-advancing type phase shifter in which the amount of phase shift can be continuously changed by using a variable resistor.

FIG. 8 is a diagram showing a configuration of a delay type phase shifter in which the amount of phase shift can be continuously changed by using a variable resistor.

FIG. 9 is a diagram showing a configuration of a phase shifter in which the amount of phase shift can be continuously changed by using a variable inductor.

FIG. 10 is a diagram showing a configuration of a phase shifter in which the amount of phase shift can be continuously changed by using both a variable resistor and a variable inductor.

FIG. 11 is a diagram showing an example of a variable inductor.

12 is a diagram showing in more detail the shapes of the inductor conductor and the control conductor of the variable inductor shown in FIG.

13 is an enlarged cross-sectional view taken along the line AA of FIG.

14 is a diagram showing a modification of the variable inductor shown in FIG.

15 is a diagram showing a modification of the variable inductor shown in FIG.

16 is a diagram showing a modification of the variable inductor shown in FIG.

FIG. 17 is a circuit diagram showing a modified example of the inductor used in the phase shifter shown in the first and second embodiments.

FIG. 18 is a diagram showing the inductor shown in FIG. 17 using a transfer function.

FIG. 19 is a diagram in which the configuration shown in FIG. 18 is converted by the mirror theorem.

20 is a diagram showing a configuration of an inductor in which the entire amplifier including the two operational amplifiers included in FIG. 17 is replaced with an emitter follower circuit.

21 is a diagram showing a configuration in which the inductor shown in FIG. 20 is realized by a source follower circuit.

22 is a diagram showing a modification of the inductor shown in FIG.

FIG. 23 is a diagram showing a configuration of an inductor in which the apparent inductance is smaller than the inductance of an actual element.

24 is a diagram showing a configuration of an inductor from which a resistor connected to the inverting input terminal of the first operational amplifier shown in FIG. 23 is removed.

FIG. 25 is a diagram showing a configuration of an inductor using an emitter follower circuit in the first stage.

FIG. 26 is a diagram showing a configuration of an inductor using a source follower circuit in the first stage.

FIG. 27 is a diagram showing a configuration of a phase shifter using a bipolar transistor instead of an FET.

FIG. 28 is a diagram showing a configuration of a conventional example.

[Explanation of symbols]

 10,30 Differential amplifier 12,16,18,34,36,38 Resistor 14,32 Inductor 20 Input terminal 22 Output terminal 100,200 Phase shifter

Claims (25)

[Claims] 1. A source and a drain are respectively connected to first and second resistors having substantially the same resistance value, and are composed of a transistor having an input signal inputted to a gate, a third resistor and an inductor. And a series circuit connected between the source and drain of the transistor, and a signal appearing at a connection point between the third resistor and the inductor forming the series circuit is extracted as an output signal. And a phase shifter. 2. An emitter and a collector are respectively connected to first and second resistors having substantially the same resistance value, and each of the emitter and collector is composed of a transistor to which an input signal is input, a third resistor and an inductor. And a series circuit connected between the source and drain of the transistor, and a signal appearing at a connection point between the third resistor and the inductor forming the series circuit is extracted as an output signal. And a phase shifter. 3. The phase shifter according to claim 1, further comprising a DC blocking capacitor connected in series to the series circuit. 4. The third resistor in any one of claims 1 to 3, wherein the third resistor in the series circuit is connected to the source side or the emitter side of the transistor, and advances the phase of the input signal by a predetermined angle to output. A phase shifter characterized by: 5. The inductor according to claim 1, wherein the inductor in the series circuit is connected to the source side or the emitter side of the transistor, and the phase of the input signal is delayed by a predetermined angle before being output. And a phase shifter. 6. The phase shifter according to claim 1, wherein the overall phase shift amount is changed by changing the resistance value of the third resistor. 7. The phase shifter according to claim 1, wherein the total phase shift amount is changed by changing the inductance of the inductor. 8. The phase shifter according to claim 1, wherein a total phase shift amount is changed by changing a resistance value of the third resistor and an inductance of the inductor. . 9. The total phase shift amount according to claim 6, wherein the third resistance is formed by a channel of a field effect transistor, and the channel resistance is changed by changing the gate voltage. Characteristic phase shifter. 10. The inductor according to claim 7, wherein the inductor is formed in a spiral shape in a substantially planar shape on a substrate, and is formed on the substrate in a substantially concentric shape with the inductor conductor. And a control conductor through which a predetermined DC bias current flows, and an insulating magnetic body formed so as to cover the inductor conductor and the control conductor. A phase shifter characterized by changing the overall phase shift amount by changing the inductance appearing at both ends of the inductor conductor. 11. The inductor according to claim 7, wherein the inductor is formed in a spiral shape in a substantially planar shape on a substrate, and is formed on the substrate in a substantially concentric shape with the inductor conductor. And a conductive magnetic body formed so as to cover the inductor conductor and the control conductor with an insulating film interposed therebetween. A phase shifter characterized in that a total amount of phase shift is changed by changing a DC bias current to be flowed to change an inductance appearing at both ends of the inductor conductor. 12. The inductor according to claim 1, wherein the inductor includes an amplifier having a predetermined gain and an inductor element connected in parallel between the input and the output of the amplifier. A phase shifter characterized in that the observed inductance is different from the actual inductance of the inductor element. 13. The inductor according to claim 12, wherein the gain of the amplifier is set to a negative value so that the inductance viewed from the input side of the amplifier is smaller than the inductance actually included in the inductor element. Phase shifter. 14. The gain according to claim 12, wherein the gain of the amplifier is set between 0 and 1 so that the inductance seen from the input side of the amplifier is larger than the inductance of the inductor element. And a phase shifter. 15. The first operational amplifier according to claim 12 or 13, wherein the amplifier has a non-inverting input terminal connected to a fixed potential and an output terminal connected to one end of the inductor element, and the inductor element. A fourth resistor inserted between the other end of the first operational amplifier and the inverting input terminal of the first operational amplifier, and a fifth resistor inserted between the output terminal and the inverting input terminal of the first operational amplifier. A phase shifter comprising a resistor, and operating as an inverting amplifier having a negative gain. 16. The input impedance according to claim 15, further comprising a buffer between the other end of the inductor element and the fourth resistor, which is in a stage preceding the amplifier. Phaser. 17. The second operational amplifier according to claim 16, wherein the buffer has a non-inverting input terminal connected to the other end of the inductor element and an output terminal connected to the fourth resistor. A sixth resistor connected between the output terminal and the inverting input terminal of the second operational amplifier, and a phase shifter. 18. The phase shifter according to claim 16, wherein the buffer is composed of an emitter follower circuit or a source follower circuit. 19. The inductor according to claim 15, wherein at least one of the fourth, fifth, and sixth resistors is formed by a variable resistor to change the inductance of the inductor and to shift the entire phase shift. A phase shifter characterized by changing the amount. 20. The amplifier according to claim 12, further comprising: two non-inverting amplifiers arranged in series; and a first voltage dividing circuit connected between the two non-inverting amplifiers. , The overall gain is set between 0 and 1 according to the voltage dividing ratio of the first voltage dividing circuit. 21. The amplifier according to claim 12, wherein the amplifier is composed of an emitter follower circuit having a second voltage dividing circuit connected to the emitter side or a source follower circuit having a second voltage dividing circuit connected to the source side. By taking out the output from the second voltage dividing circuit, the overall gain is set between 0 and 1 according to the voltage dividing ratio of the second voltage dividing circuit. vessel. 22. The voltage dividing circuit according to claim 20, wherein the voltage dividing circuit includes a variable resistor,
A phase shifter characterized in that the inductance of the inductor is varied to change the overall phase shift amount. 23. The total phase shift amount according to claim 19, wherein the resistance value of the third resistor included in the second series circuit is changed and the inductance of the inductor is changed. A phase shifter characterized by changing. 24. The overall phase shift amount according to claim 23, wherein the third resistance is formed by a channel of a field effect transistor, and a channel voltage is changed by changing a gate voltage, thereby changing an entire phase shift amount. Phase shifter. 25. The phase shifter according to claim 1, wherein the constituent parts are integrally formed on a semiconductor substrate.