JP3216686B2 - Phase shifter - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a small-sized, wide-band analog variable phase shifter using a transistor for a high-frequency signal having a large variable phase amount.

[0002]

2. Description of the Related Art An antenna using a large number of element antennas, such as an array antenna, has the feature that various functions can be obtained depending on the type and arrangement of element antennas and the manner of excitation. Its use is expanding along with. In this type of antenna, in order to excite a large number of element antennas with a predetermined amplitude and phase, respectively,
The power supply circuit becomes complicated and large. Therefore, in order to reduce the size and function of the antenna, it is essential to reduce the size, weight and performance of the power supply circuit.

An analog variable phase shifter is used for the above-described antenna, and adjusts the beam direction of the antenna by changing the phase of the high frequency transmitted and received by the array antenna. In view of the demand for the above-described power supply circuit, it is desirable that such an analog variable phase shifter has a variable phase amount as wide as possible while having a precise phase adjustment function, and has low power consumption and small size.

FIG. 17 is a circuit diagram of a phase shifter according to a first conventional example, which is a reflection type phase shifter using a 90 ゜ 3 dB hybrid. This phase shifter utilizes the characteristic that the junction capacitance changes when a reverse bias is applied to the voltage variable capacitance element. The high-frequency signal input from the input terminal 1 is divided into a coupling terminal 3 and a passing terminal 4 by a 90 ゜ 3 dB hybrid 8 and reflected by varactor diodes 7a and 7b, which are voltage variable capacitance elements, respectively, and again a 90 再 び 3 dB hybrid. 8 and output from the output terminal 2. Here, a bias circuit (not shown) is connected to the terminal 5, and a reverse bias is applied to the varactor diodes 7a and 7b via the resistors 6b and 6a to change the junction capacitance of the varactor diodes 7a and 7b. This changes the output phase.

FIG. 18 is a circuit diagram of a second conventional phase shifter, which is an infinite phase shifter using a vector modulation method. This phase shifter produces four vectors 90 ° out of phase by combining 90 ° 3dB hybrids 8, 8 and 180 ° 3dB hybrid 9. Then, a variable gain amplifier 1 using a dual gate FET (FET: field effect transistor) or the like for each vector
2,12. . . The magnitude of the amplitude is controlled by the common-mode power combiners 11, 11. . . In these variable gain amplifiers 12
Are combined to obtain an arbitrary phase amount from 0 ° to 360 °. For example, for a phase change in the first quadrant (0 ° to 90 °), the vector A
(0 °) and the vector B (90 °) may be combined.
This can be achieved by controlling the amplitude gain by turning on the dual-gate FETs FA and FB and by pinching off the remaining two dual-gate FETs FC and FD.

[0006]

In the phase shifter according to the first conventional example, the phase is changed only by the change in the capacitance of the voltage variable capacitance element such as the varactor diodes 7a and 7b. There was a problem. Also, since a large 90 ° hybrid 8 or the like is used, it is difficult to reduce the size of the circuit. Furthermore, since a single-stage phase shifter cannot provide a sufficient phase change, it is necessary to use a multi-stage configuration in order to set a required phase amount with high accuracy, which makes it difficult to reduce the size.

In the phase shifter according to the second conventional example, the number of components is increased. Further, in order to finely adjust the amplitude and phase, the first gate 13 and the second gate 14 of the dual gate FET constituting the variable gain amplifier 12 are used.
It is necessary to control the voltage applied to each of them in a complicated manner, and it has been difficult to reduce the size and power consumption. The present invention has been made in view of the above points, and an object thereof is to provide an analog variable phase shifter for a high-frequency signal having a small and wide band and a large variable phase amount even in a high-frequency band equal to or higher than a microwave band. It is in.

[0008]

[0009]

[0010]

[Means for Solving the Problems] To solve the above problems,
In order, an invention according to claim 1, wherein a first transistor having a first electrode connected to ground, a second electrode connected to the input terminal,
A second transistor having a first electrode grounded, a second electrode connected to a third electrode of the first transistor, and a third electrode connected to an output terminal; and a third electrode of the first transistor, A variable inductor provided between a connection point of a second electrode of the second transistor and ground; a first electrode connected to the input terminal; a second electrode grounded; and a third electrode connected to the output terminal. And a third transistor connected to the terminal , wherein the first to third transistors are FE
T or HEMT (High Electron Mobility Transi
stor), the first electrode of the transistor is the gate electrode
Pole, the second electrode is the source electrode, and the third electrode is the drain electrode
And the first to third transistors are bi-directional.
In the case of a polar transistor, the first electrode of the transistor
Is the base electrode, the second electrode is the emitter electrode, and the third electrode is
It is a collector electrode.

According to a second aspect of the present invention, there is provided a first transistor having a first electrode grounded and a second electrode connected to an input terminal, a first electrode grounded, and a second electrode connected to the first electrode.
A second transistor having a third electrode connected to an output terminal, a third electrode of the first transistor, and a second electrode of the second transistor.
A variable inductor provided between a connection point of the electrodes and ground; a third transistor having a first electrode connected to the input terminal and a second electrode grounded;
A fourth transistor connected to a third electrode of the third transistor, a second electrode connected to the output terminal, and a third electrode grounded; and a third electrode of the third transistor and a fourth transistor connected to the fourth transistor. comprising a connection point of the first electrode, and a resistive element provided between the ground, the first to fourth
Transistor is FET or HEMT (High Elec
tron Mobility Transistor), the transistor
The first electrode is a gate electrode, the second electrode is a source electrode, the third electrode
The pole is a drain electrode, and the first to fourth electrodes
If the transistor is a bipolar transistor, the transistor
The first electrode of the transistor is the base electrode and the second electrode is the emitter
The electrode and the third electrode are collector electrodes.
You.

According to a third aspect of the present invention, there is provided a first transistor in which a first electrode is connected to an input terminal and a second electrode is grounded, a first electrode is grounded, and a second electrode is connected to the first transistor.
A second transistor having a third electrode connected to an output terminal, a third electrode of the first transistor, and a second electrode of the second transistor.
A variable inductor provided between a connection point of the electrodes and ground, a third electrode connected to the input terminal, a third electrode connected to the input terminal, and a third electrode connected to the output terminal; A first transistor and a third transistor.
The FET is FET or HEMT (High Electron Mob
ility Transistor), the first electrode of the transistor
Is the gate electrode, the second electrode is the source electrode, and the third electrode is the drain.
An in-electrode, and the first to third transistors
If the star is a bipolar transistor, the transistor
The first electrode is a base electrode, the second electrode is an emitter electrode,
The three electrodes are collector electrodes.

According to a fourth aspect of the present invention, there is provided a first transistor in which a first electrode is connected to an input terminal and a second electrode is grounded, a first electrode is grounded, and a second electrode is the first transistor.
A second transistor having a third electrode connected to an output terminal, a third electrode of the first transistor, and a second electrode of the second transistor.
A variable inductor provided between a connection point of the electrodes and ground; a third transistor having a first electrode grounded, a second electrode connected to the input terminal;
A fourth transistor connected to a third electrode of the third transistor, a second electrode connected to the output terminal, and a third electrode grounded; and a third electrode of the third transistor and a fourth transistor connected to the fourth transistor. comprising a connection point of the first electrode, and a resistive element provided between the ground, the first to fourth
Transistor is FET or HEMT (High Elec
tron Mobility Transistor), the transistor
The first electrode is a gate electrode, the second electrode is a source electrode, the third electrode
The pole is a drain electrode, and the first to fourth electrodes
If the transistor is a bipolar transistor, the transistor
The first electrode of the transistor is the base electrode and the second electrode is the emitter
The electrode and the third electrode are collector electrodes.
You.

Further, an invention according to claim 5, wherein, in the invention described in any one of claims 1 a <br/> stone 4, characterized in that connected in parallel to the variable capacitor to said variable inductor And According to a sixth aspect of the present invention, in the first or second aspect of the present invention, the mutual conductance of the first transistor is matched to an input impedance viewed from the input terminal.

According to a seventh aspect of the present invention, in the third or fourth aspect of the invention, the transconductance of the third transistor is matched with the input impedance as viewed from the input terminal. It is characterized by having made it. According to an eighth aspect of the present invention, in the second or fourth aspect , a mutual conductance of the fourth transistor is matched to an output impedance viewed from the output terminal.

In the present invention, the first to fourth transistors are FETs or HEMTs (High
In the case of an electron mobility transistor, the first electrode of the transistor is a gate electrode, the second electrode is a source electrode, and the third electrode is a drain electrode. When the first to fourth transistors are bipolar transistors, the first electrode of the transistor is a base electrode, the second electrode is an emitter electrode, and the third electrode is a collector electrode.

[0017]

[0018]

[0019]

SUMMARY OF] According to the invention of claims 1 to 4, wherein, with the above arrangement has the same transfer function and the secondary of the total frequency passing circuit and phase characteristics, yet (3) admittance given by equation Is configured. Therefore, by varying the inductance value of the variable inductor, it is possible to obtain a phase characteristic in which the amplitude is constant and only the phase continuously changes over a wide range. This makes it possible to provide a phase shifter that uses a transistor and has a large variable phase amount in a wide band and can operate even in a high frequency range such as a microwave band. In addition, since the circuit can be configured only with the transistor, the resistance element, and the inductor, the size of the phase shifter can be reduced.

According to the fifth aspect of the present invention, a variable capacitor is connected in parallel to the variable inductor so that the capacitance value of the variable capacitor can be varied. Thereby, a wider range of variable phase amount can be obtained. According to the sixth or seventh aspect of the present invention, the first or third terminal is connected to the input terminal and the first electrode is grounded.
Are set appropriately with respect to the input impedance. As a result, active matching of the input becomes possible, and as a result, a passive matching element becomes unnecessary, and the size of the phase shifter can be reduced.

According to the eighth aspect of the invention, the transconductance of the fourth transistor connected to the output terminal and having the third electrode grounded is set appropriately with respect to the output impedance. . As a result, active matching of the output becomes possible, and as a result, a passive matching element becomes unnecessary, and the size of the phase shifter can be further reduced.

[0022]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS First, the principle of operation of a phase shifter according to the present invention will be briefly described. The voltage transfer function H (s) of a general second-order all-frequency pass circuit is as follows: input voltage V1, output voltage V2, central angular frequency ω0, Q factor Q, s = jω
(Where ω is the angular frequency), it is expressed by the following equation.

(Equation 5) At this time, the phase characteristic φ (ω) of the circuit is

(Equation 6) It is represented by

Here, when Y21 in the admittance matrix Y of this circuit is represented by the following equation, the central angular frequency ω
Even if 0 changes, a frequency characteristic in which only the phase changes with a constant amplitude is obtained. G is the gain of the circuit.

(Equation 7) Therefore, a circuit in which Y21 of the admittance matrix is given by equation (3) is provided, a parallel circuit of a capacitor and an inductor for determining the central angular frequency ω0 is provided, and at least one of the capacitor and the inductor is configured by a voltage control element or the like. By making the capacitance value or the inductance value variable, only the phase can be continuously and largely changed with a constant amplitude.

[Embodiment 1] Next, a first embodiment of the present invention will be described with reference to the drawings. FIG. 1 is a circuit diagram of a phase shifter 100 according to this embodiment. In this figure,
V1 and V2 are the input and output voltages of the phase shifter 100, Z01 and Z02 are the input and output impedances of the phase shifter 100, C is a DC voltage blocking capacitor, and P is a coil for blocking high frequencies. A point to which a voltage is applied, terminal 16 is an input terminal, terminal 18 is an output terminal, terminals 17 and 19 are ground terminals, and 22 is an inductor having an inductance value Lk.

Reference numerals 31, 35, and 39 denote FETs having Cgs1, Cgs2, and Cgsf as depletion layer capacities between the gate and the source immediately below the gate, and gm1, gm2, and gmf as transconductances, respectively. G.
SD denotes a gate electrode, a source electrode, and a drain electrode of the FET, respectively. On the other hand, terminals 32, 36 and 40 are the drain electrodes of FETs 31, 35 and 39, respectively, and terminals 33 and
Reference numerals 37 and 41 denote source electrodes of FETs 31, 35 and 39, respectively, and terminals 34, 38 and 42 denote FETs 31 and 3, respectively.
5.39 gate electrodes.

This phase shifter 100 has a gate-grounded FET.
31 and the gate-grounded FET 35 are cascaded, and the inductor 22 is shunted between them (that is, one end is FE).
A source-grounded FET 39 is connected in parallel to a circuit inserted between the drain electrode of T31 and the source electrode of the FET 35 and the other end connected to the ground side (as a shunt).

Next, the operation of the phase shifter 100 having the above configuration will be described. First, the admittance matrix Y of the circuit of FIG. 1 is obtained. To simplify the analysis, FET31 and FET3
5. The FETs 39 all have the same electrical characteristics, and each has a gate-source depletion layer capacitance Cgs immediately below the gate.
1, Cgs2, Cgsf and transconductance gm1, g
It is assumed that it is expressed only by m2 and gmf.

Then, the admittance matrix Y is given by the following equation.

(Equation 8) Here, the cutoff frequency fT of the FET = gm1 / (2πCgs)
1) = gm2 / (2πCgs2) = gmf / (2πCgsf) Within the same wafer, such FETs having the same characteristics of fT can be easily configured.

The impedance Z of the inductor 22
Is given by the following equation as s = jω.

(Equation 9) Therefore, from equations (1-1) and (1-2), Y21 of the admittance matrix Y is

(Equation 10) Becomes

Even if the central angular frequency ω0 changes, the condition that the amplitude of Y21 is constant and only the phase changes is as follows from the equations (3) and (1-3).

[Equation 11] Therefore, the transconductance gmf of the FET 39 is set to (1
By setting the value of the expression -4), and configuring the inductor 22 with a voltage control element or the like to vary the inductance value Lk, only the phase can be changed with a constant amplitude. Further, by setting the transconductance gm1 to be the reciprocal of the input impedance Z01, input matching can be achieved using only active elements.

Here, the gate width of the FET 31 is 400
μm FET (mutual conductance gm1 = 80 ms,
Depletion layer capacitance Cgs1 = 0.64 pF, cut-off frequency fT = 2
0 GHz) as an FET 35 with a gate width of 50 μm.
ET (transconductance gm2 = 10 ms, depletion layer capacitance Cgs2 = 0.08 pF, cutoff frequency fT = 20 GH
z) is a FET having a gate width of 200 μm as the FET 39
(Mutual conductance gmf = 40 mS, depletion layer capacitance Cg
FIG. 2 shows the frequency characteristics of the phase shifter using sf = 0.32 pF and cutoff frequency fT = 20 GHz.

In FIG. 2, S11, S21 and S22 correspond to Y11, Y21 and Y22 of the admittance matrix Y, respectively.
The lower left diagram shows the phase characteristics of the phase shifter, the upper right diagram shows the state of input matching, and the lower right figure shows the state of output matching. Further, in this figure, the inductance value Lk of the inductor 22 is changed to 1 nH, 3 nH, and 5 nH.

As is clear from the figure, the frequency f = 10G
In the vicinity of Hz, a phase change of S21 of 90 ° or more with a constant amplitude is obtained. In addition, it can be seen that S11 is about -3 dB and the input is matched. From equation (1-3), since Y21 is proportional to the transconductance gmf,
By increasing the transconductance gmf, S21 can have a gain. Here, the above-described impedance matching is possible without completely adjusting the value of the transconductance gm1 to the input impedance Z01.
Therefore, it is possible to satisfy the expression (1-4) and to vary the gain by changing the transconductance gmf. This is the same in the following embodiments.
From the above, it can be seen that the phase shifter 100 according to this embodiment operates well.

Second Embodiment Next, a second embodiment of the present invention will be described with reference to the drawings. FIG. 3 is a circuit diagram of the phase shifter 200 according to this embodiment. Here, the same components and signals as those in FIG. 1 are denoted by the same reference numerals, and description thereof will be omitted. In this figure, 43 is an FET,
It has Cgsa as the depletion layer capacitance between the gate and the source, and gma as the transconductance. Also, terminal 44
45 and 46 are drain, source and gate electrodes of the FET 43, respectively. Reference numeral 23 denotes a resistor having a resistance value RS.

This phase shifter 200 has a gate-grounded FET.
31 and a gate-grounded FET 35 in cascade, an inductor 22 is inserted in a shunt between them, and a source-grounded FET 39 and a drain-grounded FET 43 are cascaded in parallel with this cascade-connected circuit. A circuit in which a resistor 23 is inserted into a shunt is connected between them.

Next, the operation of the phase shifter 200 having the above configuration will be described. First, the admittance matrix Y of the circuit of FIG. 3 is obtained. To simplify the analysis, FET31 and FET3
5. The FET 39 and the FET 43 all have the same electrical characteristics, and are each expressed only by the gate-source depletion layer capacitance Cgs1, Cgs2, Cgsf, Cgsa and the mutual conductance gm1, gm2, gmf, gma immediately below the gate. And

Then, the admittance matrix Y is given by the following equation.

(Equation 12) Here, the cutoff frequency fT of the FET = gm1 / (2πCgs)
1) = gm2 / (2πCgsf) = gmf / (2πCgsf) =
gma / (2πCgsa). Within the same wafer, such FETs having the same characteristics of fT can be easily configured.

The condition required for the resistance value Rs of the resistor 23 in order that Y22 of the admittance matrix Y has a frequency characteristic in which the amplitude is constant is given by the following equation.

(Equation 13) At this time, the expression (2-1) is as follows.

[Equation 14]

The impedance Z of the inductor 22
Is given by the following equation as s = jω.

(Equation 15) Therefore, from equations (2-3) and (2-4), Y21 of the admittance matrix Y is

(Equation 16) Becomes

The condition that the amplitude of Y21 is constant and only the phase changes even when the center frequency changes is as follows.

[Equation 17] Therefore, the resistance value Rs of the resistor 23 is set to the value of the equation (2-2), and the mutual conductance gmf of the FET 39 is set to (2−2).
By setting the value of the expression (6), the inductor 22 is constituted by a voltage control element or the like, and the inductance value Lk is made variable, it is possible to change only the phase with a constant amplitude. Further, by setting the transconductance gm1 to be the reciprocal of the input impedance Z01, input matching can be achieved only with active elements. Also, the mutual conductance gm
By setting a as the reciprocal of the output impedance Z02, output matching can be achieved only with active elements.

Here, the gate width of the FET 31 is 400
μm FET (mutual conductance gm1 = 80 ms,
Depletion layer capacitance Cgs1 = 0.64 pF, cut-off frequency fT = 2
0 GHz) as an FET 35 with a gate width of 50 μm.
ET (mutual conductance gm2 = 10 ms, depletion layer capacitance Cgs2 = 0.08 pF, cutoff frequency fT = 20 GH
z) is a FET having a gate width of 200 μm as the FET 39
(Mutual conductance gmf = 40 mS, depletion layer capacitance Cg
sf = 0.32 pF, cutoff frequency fT = 20 GHz)
An FET having a gate width of 50 μm (mutual conductance gma = 10 mS, depletion layer capacitance Cgsa = 0.
FIG. 4 shows the frequency characteristics of the phase shifter using the phase shifter (08 pF, cutoff frequency fT = 20 GHz). From the above, the resistance 2
It is assumed that the resistance value of Rs = 1 / gma = 100Ω.

In FIG. 4, the inductance value Lk of the inductor 22 is changed to 1 nH, 3 nH, and 5 nH. As is clear from the figure, at a frequency of about f = 10 GHz,
A phase change of S21 of 90 ° or more with a constant amplitude is obtained. In addition, S11 is about -3 dB and S22 is -9 dB or less, which indicates that both input and output are matched. (2-
From equation (5), since Y21 is proportional to the transconductance gmf, S21 can have a gain by increasing the transconductance gmf. From the above, it can be seen that the phase shifter 200 according to this embodiment operates well.

[Third Embodiment] Next, a third embodiment of the present invention will be described with reference to the drawings. FIG. 5 is a circuit diagram of a phase shifter according to this embodiment. Here, the same components and signals as those in FIG. 1 are denoted by the same reference numerals, and description thereof will be omitted. In the phase shifter 300, a common-source FET 31 and a common-gate FET 35 are cascaded, and a common-gate FET 39 is connected in parallel to a circuit in which the inductor 22 is inserted in a shunt.

Next, the operation of the phase shifter 300 having the above configuration will be described. First, the admittance matrix Y of the circuit of FIG. To simplify the analysis, FET31 and FET3
5. The FETs 39 all have the same electrical characteristics, and each has a gate-source depletion layer capacitance Cgs immediately below the gate.
1, Cgs2, Cgsf and transconductance gm1, gm
2. It is assumed that it is expressed only by gmf.

Then, the admittance matrix Y is given by the following equation.

(Equation 18) Here, the cutoff frequency fT of the FET = gm1 / (2πCgs)
1) = gm2 / (2πCgs2) = gmf / (2πCgsf) Within the same wafer, such FETs having the same characteristics of fT can be easily formed.

The impedance Z of the inductor 22
Is given by the following equation as s = jω.

[Equation 19] Therefore, from equations (3-1) and (3-2), Y21 of the admittance matrix Y is

(Equation 20) Becomes

Even if the central angular frequency changes, the condition that the amplitude of Y21 is constant and only the phase changes is as follows.

(Equation 21) Therefore, the transconductance gmf of the FET 39 is set to (3
By setting the value of the expression -4), and configuring the inductor 22 with a voltage control element or the like to vary the inductance value Lk, only the phase can be changed with a constant amplitude. Further, by making the transconductance gmf the reciprocal of the input impedance Z01, input matching can be achieved with only active elements.

Here, the gate width of the FET 31 is 400
μm FET (mutual conductance gm1 = 80 ms,
Depletion layer capacitance Cgs1 = 0.64 pF, cut-off frequency fT = 2
0 GHz) as an FET 35 with a gate width of 50 μm.
ET (mutual conductance gm2 = 10 ms, depletion layer capacitance Cgs2 = 0.08 pF, cutoff frequency fT = 20 GH
z) is a FET having a gate width of 200 μm as the FET 39
(Mutual conductance gmf = 40 mS, depletion layer capacitance Cg
FIG. 6 shows the frequency characteristics of the phase shifter using sf = 0.32 pF and cutoff frequency fT = 20 GHz.

In this figure, the inductance value Lk of the inductor 22 is changed to 1 nH, 3 nH, and 5 nH. As is apparent from the figure, around the frequency f = 10 GHz, a phase change of S21 of 90 ° or more with a constant amplitude is obtained. In addition, it can be seen that S11 is about -3 dB and the input is matched. From equation (3-3), since Y21 is proportional to the mutual conductance gmf, the gain can be given to S21 by increasing the mutual conductance gmf. From the above, it can be seen that the phase shifter 300 of this embodiment operates well.

Fourth Embodiment Next, a fourth embodiment of the present invention will be described with reference to the drawings. FIG. 7 is a circuit diagram of a phase shifter 400 according to this embodiment. Here, the same components and signals as those in FIG. 3 are denoted by the same reference numerals, and description thereof will be omitted. This phase shifter 400 is a source-grounded FET
31 and a gate-grounded FET 35 in cascade, and a circuit in which the inductor 22 is inserted in the shunt between the circuit and the cascade-connected circuit. A circuit in which the resistor 23 is inserted into the shunt is connected between the two.

Next, the operation of the phase shifter 400 having the above configuration will be described. First, the admittance matrix Y of the circuit of FIG. 7 is obtained. To simplify the analysis, FET31 and FET3
5. The FET 39 and the FET 43 all have the same electrical characteristics, and are each expressed only by the gate-source depletion layer capacitance Cgs1, Cgs2, Cgsf, Cgsa and the mutual conductance gm1, gm2, gmf, gma immediately below the gate. And

Then, the admittance matrix Y is given by the following equation.

(Equation 22) Here, the cutoff frequency fT of the FET = gm1 / (2πCgs)
1) = gm2 / (2πCgsf) = gmf / (2πCgsf) =
gma / (2πCgsa). In the same wafer,
It is possible to easily configure FETs having such characteristics of equal fT.

The condition required for the resistance value Rs of the resistor 23 in order that Y22 of the admittance matrix Y has a frequency characteristic in which the amplitude is constant is given by the following equation.

(Equation 23) At this time, the expression (4-1) becomes the following expression.

(Equation 24)

The impedance Z of the inductor 22
Is given by the following equation as s = jω.

(Equation 25) Therefore, from equations (4-3) and (4-4), Y21 of the admittance matrix Y is

(Equation 26) Becomes

Even if the center angular frequency changes, the condition that the amplitude of Y21 is constant and only the phase changes is as follows.

[Equation 27] Therefore, the resistance Rs of the resistor 23 is set to the value of the equation (4-2), and the mutual conductance gmf of the FET 39 is set to the value of (4−4).
By setting the value of the expression (6), the inductor 22 is constituted by a voltage control element or the like, and the inductance value Lk is made variable, it is possible to change only the phase with a constant amplitude. Further, by setting the transconductance gmf to be the reciprocal of the input impedance Z01, input matching can be achieved with only active elements. Also, the mutual conductance gm
By making a the reciprocal of the output impedance Z02,
Output matching can be achieved only with active elements.

Here, the gate width of the FET 31 is 400
μm FET (mutual conductance gm1 = 80 ms,
Depletion layer capacitance Cgs1 = 0.64 pF, cut-off frequency fT = 2
0 GHz) as an FET 35 with a gate width of 50 μm.
ET (mutual conductance gm2 = 10 ms, depletion layer capacitance Cgs2 = 0.08 pF, cutoff frequency fT = 20 GH
z) is a FET having a gate width of 200 μm as the FET 39
(Mutual conductance gmf = 40 mS, depletion layer capacitance Cg
sf = 0.32 pF, cutoff frequency fT = 20 GHz)
FET having a gate width of 50 μm (transconductance gma = 10 ms, depletion layer capacitance Cgsa = 0.0) as ET43
FIG. 8 shows the frequency characteristics of a phase shifter using 8 pF and a cutoff frequency fT = 20 GHz. From the above, the resistance 23
Is set to Rs = 1 / gma = 100Ω.

In this figure, the inductance value Lk of the inductor 22 is changed to 1 nH, 3 nH, and 5 nH.
As is apparent from the figure, around the frequency f = 10 GHz, a phase change of S21 of 90 ° or more with a constant amplitude is obtained. In addition, S11 is about -3 dB and S22 is -9 dB or less, which indicates that both input and output are matched. In addition,
From equation (4-5), since Y21 is proportional to the transconductance gmf, the gain of S21 can be provided by increasing the transconductance gmf. From the above,
It can be seen that the phase shifter 400 according to this embodiment operates well.

Fifth Embodiment Next, a fifth embodiment of the present invention will be described with reference to the drawings. FIG. 9 is a circuit diagram of a phase shifter according to this embodiment. Here, the same components and signals as those in FIG. 1 are denoted by the same reference numerals, and description thereof will be omitted. In this figure, reference numeral 21 denotes a capacitor having a capacitance Ck. This phase shifter 500 is obtained by inserting a capacitor 21 in a shunt with respect to the inductor 22 of the phase shifter 100 in the first embodiment.

Next, the operation of the phase shifter 500 having the above configuration will be described. Compared with the first embodiment, Y21 of the admittance matrix Y of the phase shifter 500 has a capacity Cgs2 of (Ck + Cgs).
It is replaced by 2) and is expressed by the following equation.

[Equation 28] Therefore, the transconductance gmf of the FET 39 is set to (1
-4) By setting the value to the expression, by configuring the capacitor 21 or the inductor 22 with a voltage control element or the like and making the capacitance value Ck or the inductance value Lk variable, it is possible to change only the phase with a constant amplitude. . Further, by setting the transconductance gm1 to be the reciprocal of the input impedance Z01, input matching can be achieved using only active elements.

Here, the gate width of the FET 31 is 400
μm FET (mutual conductance gm1 = 80 ms,
Depletion layer capacitance Cgs1 = 0.8 pF, cut-off frequency fT = 20
GHz) as the FET 35 with an FE having a gate width of 50 μm.
T (transconductance gm2 = 10 ms, depletion layer capacitance Cgs2 = 0.08 pF, cutoff frequency fT = 20 GHz)
Is replaced by an FET 39 having a gate width of 200 μm (transconductance gmf = 40 mS, depletion layer capacitance Cgsf).
FIG. 10 shows the frequency characteristics of the phase shifter using (= 0.32 pF, cutoff frequency fT = 20 GHz).

In this figure, the inductance value Lk of the inductor 22 is set to 3 nH, and the capacitance value Ck of the capacitor 21 is changed to 0.5 pF, 1 pF, and 2 pF. As is apparent from the drawing, a phase change of S21 of 180 ° or more with a constant amplitude is obtained around the frequency f = 2.5 GHz. In addition, it can be seen that S11 is equal to or less than -3 dB and the input is matched. From equation (5-1), since Y21 is proportional to the mutual conductance gmf, the gain of S21 can be provided by increasing the mutual conductance gmf. As described above, the phase shifter 5 according to this embodiment
00 works well.

Embodiment 6 Next, a sixth embodiment of the present invention will be described with reference to the drawings. FIG. 11 is a circuit diagram of a phase shifter 600 according to this embodiment. Here, the same components and signals as those in FIG. 3 or FIG. 9 are denoted by the same reference numerals, and description thereof will be omitted. This phase shifter 600 has a second
In this embodiment, the capacitor 21 is inserted into the shunt with respect to the inductor 22 of the phase shifter 200.

Next, the operation of the phase shifter 600 having the above configuration will be described. Compared to the second embodiment, Y21 of the admittance matrix Y of the phase shifter 600 has a capacity Cgs2 of (Ck + Cgs).
It is replaced by 2) and is expressed by the following equation.

(Equation 29) Therefore, the resistance value Rs of the resistor 23 is set to the value of the equation (2-2), the mutual conductance gmf of the FET 39 is set to the value of the equation (2-6), and the capacitor 21 or the inductor 22 is connected to a voltage control element or the like. By making the capacitance value Ck or the inductance value Lk variable, only the phase can be changed with a constant amplitude.

Further, by making the transconductance gm1 the reciprocal of the input impedance Z01, input matching can be achieved only with active elements. Further, by making the transconductance gma the reciprocal of the output impedance Z02, output matching can be achieved only with active elements.

Here, the gate width of the FET 31 is 400
μm FET (mutual conductance gm1 = 80 ms,
Depletion layer capacitance Cgs1 = 0.8 pF, cut-off frequency fT = 20G
Hz) is replaced by an FET 35 having a gate width of 50 μm.
(Mutual conductance gm2 = 10 mS, depletion layer capacitance Cg
s2 = 0.08 pF, cut-off frequency fT = 20 GHz)
An FET having a gate width of 200 μm (mutual conductance gmf = 40 mS, depletion layer capacitance Cgsf =
0.32 pF, cut-off frequency fT = 20 GHz)
An FET 43 having a gate width of 50 μm (mutual conductance gma = 10 ms, depletion layer capacitance Cgsa = 0.08 p)
FIG. 12 shows the frequency characteristics of a phase shifter using F, a cutoff frequency fT = 20 GHz). From the above, the resistance value Rs of the resistor 23 is set to 1 / gma = 100Ω.

In this figure, the inductance value Lk of the inductor 22 is set to 3 nH, and the capacitance value Ck of the capacitor 21 is changed to 0.5 pF, 1 pF, and 2 pF. As is apparent from the drawing, a phase change of S21 of 180 ° or more with a constant amplitude is obtained around the frequency f = 2.5 GHz. In addition, S11 is -3 dB or less, and S22 is -9 dB or less, and it can be seen that both inputs and outputs match. From equation (6-1), Y21 is the transconductance gm
Since S21 is proportional to f, the gain can be given to S21 by increasing the transconductance gmf. From the above, it can be seen that the phase shifter 600 according to this embodiment operates well.

[Embodiment 7] Next, a seventh embodiment of the present invention will be described with reference to the drawings. FIG. 13 is a circuit diagram of a phase shifter 700 according to this embodiment. Here, the same components and signals as those in FIG. 5 or FIG. 9 are denoted by the same reference numerals, and description thereof will be omitted. This phase shifter 700 has a third
In this embodiment, the capacitor 21 is inserted in the shunt with respect to the inductor 22 of the phase shifter 300 in the third embodiment.

Next, the operation of the phase shifter 700 having the above configuration will be described. Compared to the third embodiment, Y21 of the admittance matrix Y is obtained by replacing the capacitance Cgs2 with (Ck + Cgs2), and is represented by the following equation.

[Equation 30]

Therefore, the mutual conductance gmf of the FET 39 is set to the value of the equation (3-4), and the capacitor 21 or the inductor 22 is constituted by a voltage control element or the like so that the capacitance value Ck or the inductance value Lk is variable. Thus, only the phase can be changed with a constant amplitude. Further, by setting the transconductance gmf to be the reciprocal of the input impedance Z01, input matching can be achieved only with active elements.

Here, the gate width of the FET 31 is 400
μm FET (mutual conductance gm1 = 80 ms,
Depletion layer capacitance Cgs1 = 0.8 pF, cut-off frequency fT = 20
GHz) as the FET 35 with an FE having a gate width of 50 μm.
T (transconductance gm2 = 10 ms, depletion layer capacitance Cgs2 = 0.08 pF, cutoff frequency fT = 20 GHz)
Is replaced by an FET 39 having a gate width of 200 μm (transconductance gmf = 40 mS, depletion layer capacitance Cgsf).
FIG. 14 shows the frequency characteristics of the phase shifter using (= 0.32 pF, cutoff frequency fT = 20 GHz).

In this figure, the inductance value Lk of the inductor 22 is set to 3 nH, and the capacitance value Ck of the capacitor 21 is changed to 0.5 pF, 1 pF, and 2 pF. As is apparent from the drawing, a phase change of S21 of 180 ° or more with a constant amplitude is obtained around the frequency f = 2.5 GHz. In addition, it can be seen that S11 is equal to or less than -7 dB and the input is matched. From equation (7-1), since Y21 is proportional to the mutual conductance gmf, a gain can be given to S21 by increasing the mutual conductance gmf. As described above, the phase shifter 7 according to this embodiment
00 works well.

Embodiment 8 Next, an eighth embodiment of the present invention will be described with reference to the drawings. FIG. 15 is a circuit diagram of a phase shifter 800 according to this embodiment. Here, the same components and signals as those in FIG. 7 or FIG. 9 are denoted by the same reference numerals, and description thereof will be omitted. This phase shifter 800 has a fourth
In this embodiment, the capacitor 21 is inserted in the shunt with respect to the inductor 22 of the phase shifter 400.

Next, the operation of the phase shifter 800 having the above configuration will be described. Compared with the fourth embodiment, Y21 of the admittance matrix Y is obtained by replacing the capacitance Cgs2 with (Ck + Cgs2), and is represented by the following equation.

(Equation 31)

Therefore, the resistance Rs of the resistor 23 is set to (4-
2) The value of equation (4) is set, and the transconductance gmf of the FET 39 is set to the value of equation (4-6).
Alternatively, by configuring the inductor 22 with a voltage control element or the like and making the capacitance value Ck or the inductance value Lk variable, only the phase can be changed with a constant amplitude.

Further, by setting the transconductance gmf to be the reciprocal of the input impedance Z01, input matching can be achieved only with active elements. Further, by making the transconductance gma the reciprocal of the output impedance Z02, output matching can be achieved only with active elements.

Here, the gate width of the FET 31 is 400
μm FET (transconductance gm1 = 80 ms,
Depletion layer capacitance Cgs1 = 0.8 pF, cut-off frequency fT = 20G
Hz) is replaced by an FET 35 having a gate width of 50 μm.
(Mutual conductance gm2 = 10 mS, depletion layer capacitance C
gs2 = 0.08 pF, cutoff frequency fT = 20 GHz)
An FET 39 having a gate width of 200 μm (transconductance gmf = 40 mS, depletion layer capacitance Cgsf =
0.32 pF, cut-off frequency fT = 20 GHz)
An FET 43 having a gate width of 50 μm (mutual conductance gma = 10 ms, depletion layer capacitance Cgsa = 0.08 p)
F, frequency characteristics of a phase shifter using a cutoff frequency fT = 20 GHz) are shown in FIG. From the above, it is assumed that the resistance value of the resistor 23 is Rs = 1 / gma = 100Ω.

In this figure, the inductance value Lk of the inductor 22 is set to 3 nH, and the capacitance value Ck of the capacitor 21 is changed to 0.5 pF, 1 pF, and 2 pF. As is apparent from the drawing, a phase change of S21 of 180 ° or more with a constant amplitude is obtained around the frequency f = 2.5 GHz. In addition, S11 is -7 dB or less, and S22 is -9 dB or less, and it can be seen that both inputs and outputs match. From equation (8-1), Y21 is the mutual conductance gm
Since S21 is proportional to f, the gain can be given to S21 by increasing the transconductance gmf. From the above, it can be seen that the phase shifter 800 according to this embodiment operates well.

In the above description, the case where the FET is used as the three-terminal active element has been described. However, the present invention is not limited to this. For example, a bipolar transistor or a HEMT can be used. Further, since the mutual conductance gm changes by changing the voltage at the voltage application point P to the gate of the FET, a voltage-controlled active phase shifter capable of suppressing fluctuations in amplitude and phase can be realized.

[0079]

[0080]

[0081]

As described in the foregoing, according to the invention of claims 1 to 4, wherein, with the above configuration, has a transfer function similar to the second-order all-pass circuit and phase characteristics, yet A circuit having the admittance given by the equation (3) is formed. By varying the inductance value of the variable inductor, a phase characteristic in which the amplitude is constant and only the phase continuously changes over a wide range is obtained. In addition, there is obtained an effect that a phase shifter that can operate in a high frequency range such as a microwave band can be provided in a wide band with a large variable phase amount. Also, the circuit is composed of transistors, resistance elements,
Since the phase shifter can be constituted only by the inductor, the effect that the phase shifter can be downsized can be obtained.

According to the fifth aspect of the present invention, since a variable capacitor is connected in parallel to the variable inductor to vary the capacitance value of the variable capacitor, a wider range of variable phase amount can be obtained. The effect is obtained. According to the invention described in claim 6 or 7 , the first electrode connected to the input terminal and the first electrode is grounded.
Alternatively, since the transconductance of the third transistor is appropriately set with respect to the input impedance, active matching of the input becomes possible. As a result, a passive matching element is not required, and the size of the phase shifter can be reduced. The effect that it becomes possible to achieve is obtained.

According to the eighth aspect of the present invention, the mutual conductance of the fourth transistor connected to the output terminal and having the third electrode grounded is appropriately set with respect to the output impedance. , Active matching of the output becomes possible, and as a result, a passive matching element becomes unnecessary,
An effect is obtained that the phase shifter can be further downsized.

[Brief description of the drawings]

FIG. 1 is a circuit diagram of a phase shifter according to a first embodiment of the present invention.

FIG. 2 is a diagram showing frequency characteristics of the phase shifter according to the first embodiment.

FIG. 3 is a circuit diagram of a phase shifter according to a second embodiment of the present invention.

FIG. 4 is a diagram showing frequency characteristics of the phase shifter according to the second embodiment.

FIG. 5 is a circuit diagram of a phase shifter according to a third embodiment of the present invention.

FIG. 6 is a diagram showing frequency characteristics of the phase shifter according to the third embodiment.

FIG. 7 is a circuit diagram of a phase shifter according to a fourth embodiment of the present invention.

FIG. 8 is a diagram showing frequency characteristics of the phase shifter according to the fourth embodiment.

FIG. 9 is a circuit diagram of a phase shifter according to a fifth embodiment of the present invention.

FIG. 10 is a diagram showing frequency characteristics of the phase shifter according to the fifth embodiment.

FIG. 11 is a circuit diagram of a phase shifter according to a sixth embodiment of the present invention.

FIG. 12 is a diagram showing frequency characteristics of the phase shifter according to the sixth embodiment.

FIG. 13 is a circuit diagram of a phase shifter according to a seventh embodiment of the present invention.

FIG. 14 is a diagram showing frequency characteristics of the phase shifter according to the seventh embodiment.

FIG. 15 is a circuit diagram of a phase shifter according to an eighth embodiment of the present invention.

FIG. 16 is a diagram showing frequency characteristics of the phase shifter according to the eighth embodiment.

FIG. 17 is a circuit diagram of a phase shifter according to a first conventional example.

FIG. 18 is a circuit diagram of a phase shifter according to a second conventional example.

[Explanation of symbols]

P Point at which voltage is applied via a coil that blocks high frequency C Capacitor for blocking DC voltage 1 Input terminal 2 Output terminal 3 Coupling terminal 4 Pass-through terminal 5 Terminal to which bias circuit is connected 6a, 6b, 10, 23 Resistance 7a, 7b Varactor diode 8 90 3 dB hybrid 9 180 ゜ 3 dB hybrid 11 In-phase power combiner 12 Variable gain amplifier using dual gate FET, etc. 13 First gate of dual gate FET 14 Second gate of dual gate FET 16 · 17.18.19 Terminal 21 Capacitor 22 Inductor 31.35.39.43 FET 32.36.40.44 Drain 33.37.41.45 Source 34.38.42.46 Gate

──────────────────────────────────────────────────続 き Continuation of the front page (56) References JP-A-58-137320 (JP, A) 1987 IEICE National Conference on Semiconductors and Materials [Volume 2] October 1987 Published on the 15th, p. 2-11 (58) Field surveyed (Int. Cl. ⁷ , DB name) H03H 11/20

Claims (8)

(57) [Claims] A first transistor having a first electrode grounded and a second electrode connected to an input terminal; a first electrode grounded and a second electrode connected to a third electrode of the first transistor; A second transistor having a third electrode connected to an output terminal; a third transistor provided between a third electrode of the first transistor and a second electrode of the second transistor; and a ground. and a variable inductor, a first electrode connected to said input terminal, a second electrode is grounded, the third electrode and a third transistor connected to said output terminal, said first to third transistors Is FET or H
EMT (High Electron Mobility Transistor)
The first electrode of the transistor is a gate electrode, the second electrode
Is a source electrode, a third electrode is a drain electrode, and the first to third transistors are bipolar.
In the case of a transistor, the first electrode of the transistor is a base.
Electrode, the second electrode is the emitter electrode, and the third electrode is the collector
A phase shifter characterized by being an electrode . 2. A first transistor having a first electrode grounded and a second electrode connected to an input terminal, a first electrode grounded, and a second electrode connected to a third electrode of the first transistor. A second transistor having a third electrode connected to an output terminal; a third transistor provided between a third electrode of the first transistor and a second electrode of the second transistor; and a ground. A variable inductor, a third transistor having a first electrode connected to the input terminal, a second electrode grounded, a first electrode connected to a third electrode of the third transistor, and a second electrode connected to the third transistor. A fourth transistor connected to the output terminal and having a third electrode grounded; a fourth transistor provided between a connection point between the third electrode of the third transistor and the first electrode of the fourth transistor; and ground. ; and a resistive element, the first Stone fourth transistor FET or H
EMT (High Electr on Mobility Transistor)
The first electrode of the transistor is a gate electrode, the second electrode
Is a source electrode, the third electrode is a drain electrode, and the first to fourth transistors are bipolar.
In the case of a transistor, the first electrode of the transistor is a base.
Electrode, the second electrode is the emitter electrode, and the third electrode is the collector
A phase shifter characterized by being an electrode . 3. A first transistor having a first electrode connected to the input terminal and a second electrode grounded, a first electrode grounded, and a second electrode connected to a third electrode of the first transistor. A second transistor having a third electrode connected to an output terminal; a third transistor provided between a third electrode of the first transistor and a second electrode of the second transistor; and a ground. A variable inductor, a third transistor having a first electrode grounded, a second electrode connected to the input terminal, and a third electrode connected to the output terminal ; the first to third transistors Is FET or H
EMT (High Electron Mobility Transistor)
The first electrode of the transistor is a gate electrode, the second electrode
Is a source electrode, a third electrode is a drain electrode, and the first to third transistors are bipolar.
In the case of a transistor, the first electrode of the transistor is a base.
Electrode, the second electrode is the emitter electrode, and the third electrode is the collector
A phase shifter characterized by being an electrode . 4. A first transistor having a first electrode connected to the input terminal and a second electrode grounded, a first electrode grounded, and a second electrode connected to a third electrode of the first transistor. A second transistor having a third electrode connected to an output terminal; a third transistor provided between a third electrode of the first transistor and a second electrode of the second transistor; and a ground. A variable inductor, a third transistor having a first electrode grounded, a second electrode connected to the input terminal, a first electrode connected to a third electrode of the third transistor, and a second electrode connected to the third electrode. A fourth transistor connected to the output terminal and having a third electrode grounded; a fourth transistor provided between a connection point between the third electrode of the third transistor and the first electrode of the fourth transistor; and ground. ; and a resistive element, the first Stone fourth transistor FET or H
EMT (High Electron Mobility Transistor)
The first electrode of the transistor is a gate electrode, the second electrode
Is a source electrode, the third electrode is a drain electrode, and the first to fourth transistors are bipolar.
In the case of a transistor, the first electrode of the transistor is a base.
Electrode, the second electrode is the emitter electrode, and the third electrode is the collector
A phase shifter characterized by being an electrode . 5. claims 1, characterized in that connected in parallel to the variable capacitor to said variable inductor
5. The phase shifter according to any one of 4 . 6. A phase shifter according to claim 1 or 2 transconductance of the first transistor, characterized in that is matched to the input impedance viewed from the input terminal. 7. A phase shifter according to claim 3 or 4 the transconductance of the third transistor, characterized in that is matched to the input impedance viewed from the input terminal. 8. A phase shifter according to claim 2 or 4 the transconductance of the fourth transistor, is characterized in that is matched to the output impedance viewed from the output terminal.