JPH08154016A - Oscillator - Google Patents

Abstract

PURPOSE: To vary an oscillation frequency over a wide range by feeding back the outputs of phase shift circuits by providing a non-inverting circuit for outputting an input signal without changing its phase and two phase shift circuits for respectively shifting the phase of the input signal just for a prescribed amount and for shifting the phase of it totally at 0 deg. at a prescribed frequency. CONSTITUTION: When a prescribed AC signal is inputted to an input terminal 22 at a phase shift circuit 10, a voltage dividing a voltage to be applied to the input terminal 22 with resistors 18 and 20 is applied to the non-inverted terminal of a differential amplifier 12. The resistance values of both the resistors 18 and 20 are set almost equal, and a voltage divided into about 1/2 by a voltage dividing circuit composed of a serial circuit is applied to the inverted input terminal of the amplifier 12. On the other hand, when the input signal is inputted to the input terminal 22, a signal to appear at the connecting point of a variable resistor 16 and an inductance 17 is inputted to the non-inverted input terminal of the differential amplifier 12. Since the input signal is inputted to one terminal of an LR circuit composed of the variable resistor 16 and the inductance 17, the phase of the input signal is impressed to the non-inverted input terminal of the amplifier 12.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an oscillator whose oscillation frequency can be adjusted significantly.

[0002]

2. Description of the Related Art Various oscillating circuits using an active element and a reactance element have been proposed and put into practical use as a sine wave oscillator.

[0003]

As a sine wave oscillator,
The Wien bridge type oscillator shown in FIG. 32 and the bridge T type oscillator shown in FIG. 33 are conventionally known.

As is apparent from FIG. 32, in the Wien bridge type oscillator, the resistance value of the variable resistor Rs in the series circuit composed of the capacitor C and the variable resistor Rs for changing the frequency, the capacitor C and the variable resistor Rp. It is necessary to change the resistance value of the variable resistance Rp of the parallel circuit consisting of the following, but if an error occurs in the resistance value of the variable resistance Rs of the series circuit and the resistance value of the variable resistance Rp of the parallel circuit, the amplifier Since the voltage input to A increases or decreases, the oscillation output fluctuates as a result. When the oscillation output becomes small, the oscillation stops, and when it becomes large, the oscillation output is significantly distorted.

Generally, it is difficult to stabilize the output of the sine wave oscillator so as to reduce the fluctuation, and the stabilizing means adds nonlinearity to the amplitude characteristic of the amplifier, that is, the amplification degree depends on the magnitude of the output. A characteristic that changes is added.

The addition of such a characteristic deteriorates the linearity of the amplifier, which deteriorates the distortion factor of the output waveform, and the stability of the output voltage and the distortion coefficient are in a two-fold trade-off. Have a relationship.

Changing the ratio of the resistance Rs of the series circuit to the variable resistance Rp of the parallel circuit at a constant value is particularly effective when the circuit is integrated and the variable resistance is changed by a voltage control method from the outside. Have difficulty.

The same can be said not only for the Wien bridge type oscillator, but also for the bridge T type oscillator and the phase shift type oscillator shown in FIG.

Further, it is also difficult to form a variable frequency oscillator whose oscillation frequency can be adjusted significantly by an integrated circuit.

Therefore, the present invention was devised to solve such a problem.

[0011]

In order to solve the above-mentioned problems, the oscillator of the present invention is composed of first and second resistors which have almost equal resistance values applied to both ends of an input AC signal. A first series circuit, a second series circuit formed of a third resistor and an inductor to which the AC signal is applied to both ends, and the first and second series circuits forming the first series circuit. A differential amplifier that amplifies and outputs a difference between a potential at a connection point of a resistor and a potential at a connection point of the third resistor and the inductor forming the second series circuit with a predetermined amplification degree. Two phase shift circuits are provided, the output of the latter stage of the two cascaded phase shift circuits is fed back to the input side of the former stage, and the sine wave oscillation output is taken out from either one of the two phase shift circuits. Characterize.

In the oscillator of the present invention, the alternating current signal is applied to both ends of the first series circuit composed of first and second resistors having substantially the same resistance value. A second series circuit configured by a third resistance and an inductor applied to the second series circuit, a potential at a connection point of the first and second resistances forming the first series circuit, and the second series circuit. Phase-shifting circuit including a differential amplifier that amplifies a difference between the third resistance forming the third resistance and the potential at the connection point of the inductor with a predetermined amplification degree, and outputs the phase, and a phase of an input AC signal. And a non-inverting circuit that outputs the same without changing the output of each of the two phase shift circuits and the non-inverting circuit in a cascade connection. While returning to the input side, And wherein the retrieving the sinusoidal oscillation output from one of the circuits.

In the oscillator of the present invention, the alternating current signal is applied to both ends of the first series circuit composed of first and second resistors having substantially equal resistance values, and the alternating current signal is applied to both ends of the oscillator. A second series circuit configured by a third resistance and an inductor applied to the second series circuit, a potential at a connection point of the first and second resistances forming the first series circuit, and the second series circuit. Phase-shifting circuit including a differential amplifier that amplifies a difference between the third resistance forming the third resistance and the potential at the connection point of the inductor with a predetermined amplification degree, and outputs the phase, and a phase of an input AC signal. And a phase inversion circuit that inverts and outputs each of the two phase shift circuits and the phase inversion circuits in cascade connection, and outputs the output of the final stage of the plurality of circuits connected in cascade. This is returned to the input side and And wherein the retrieving the sinusoidal oscillation output from one of the plurality of circuits.

The oscillator of the present invention comprises a non-inverting circuit for outputting an input AC signal in phase, a series connection of two resistors and a series connection of an inductor and a resistor,
A first bridge circuit to which the output of the non-inverting circuit is applied; and a first differential amplifier that obtains a difference between two outputs of the first bridge circuit, and input to the first bridge circuit Phase shift circuit for shifting the phased signal, a second bridge circuit including a series connection of two resistors and a series connection of a resistor and an inductor, to which the output of the first phase shift circuit is applied. And a second differential amplifier that obtains a difference between the two outputs of the second bridge circuit, and a signal input to the second bridge circuit is directed in a direction opposite to that of the first phase shift circuit. And a circuit for returning the output of the second phase shift circuit to the input of the non-inverting circuit.

The oscillator of the present invention comprises a phase inverting circuit that inverts and outputs an input AC signal, a series connection of two resistors and a series connection of an inductor and a resistor, and the output of the phase inverting circuit. A first bridge circuit to which is applied and a first differential amplifier that obtains a difference between two outputs of the first bridge circuit, and phase shifts a signal input to the first bridge circuit. A first phase shift circuit, a series connection of two resistors and a series connection of an inductor and a resistor, and a second phase to which an output of the first phase shift circuit is applied.
And a second differential amplifier that obtains a difference between the two outputs of the second bridge circuit, and a signal input to the second bridge circuit is supplied to the first phase shift circuit. A second phase shift circuit that shifts the phase in the same direction, and a circuit that feeds back the output of the second phase shift circuit to the input of the phase inverting circuit,
It is characterized by including.

[0016]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS An oscillator according to an embodiment of the present invention will be specifically described below with reference to the drawings.

(First Embodiment) FIG. 1 is a circuit diagram showing a configuration of an oscillator according to a first embodiment of the present invention. The oscillator 1 shown in the figure has a non-inverting circuit 50 that outputs the phase of the input signal without changing it, and each phase shifts the phase of the input signal by a predetermined amount, so that a total of 0 ° is obtained at a predetermined frequency.
Phase shift circuits 10 and 30 for performing the phase shift of
Feedback resistor that feeds back the output of 30 to the input side of the non-inverting circuit 50
It is composed of 70 and. The feedback resistor 70 has a finite resistance value from 0Ω. Although the non-inverting circuit 50 functions as a buffer circuit, it may be omitted if only the basic operation of the oscillator is focused on.

FIG. 2 shows the extracted structure of the phase shift circuit 10 at the preceding stage shown in FIG. The phase shift circuit 10 at the front stage shown in the figure includes a differential amplifier 12 that amplifies a differential voltage of two inputs with a predetermined amplification degree (for example, about twice) and outputs the differential voltage.
The variable resistor 16 and the inductor 17 that shift the phase of the signal input to 22 by a predetermined amount and input to the non-inverting input terminal of the differential amplifier 12, and its voltage without changing the phase of the signal input to the input end 22. Divide the level to about 1/2 and differential amplifier 12
It is configured to include resistors 18 and 20 that are input to the inverting input terminal of.

The capacitor 19 inserted between the inductor 17 and the variable resistor 16 is for blocking direct current,
Its impedance is very small at the operating frequency, ie it has a large capacitance. Further, the following description will be made assuming that the connection point between the inductor 17 and the resistor 20 is grounded.

In the phase shift circuit 10 having such a configuration, when a predetermined AC signal is input to the input end 22, the voltage applied to the input end 22 (input A voltage obtained by dividing the voltage Ei) by the resistors 18 and 20 is applied. The resistance values of the resistors 18 and 20 are set to be substantially equal to each other, and the voltage Ei / 2 divided by about 1/2 by the voltage dividing circuit configured by the series circuit of these two resistors 18 and 20 is differential. Applied to the inverting input terminal of amplifier 12.

On the other hand, when an input signal is input to the input terminal 22, the non-inverting input terminal of the differential amplifier 12 is connected to the connection point between the variable resistor 16 and the inductor 17 (more precisely, it is connected to the inductor 17 in series). Although it is the connection point between the capacitor 19 and the variable resistor 16, as described above, this capacitor 19 is for blocking the direct current and does not affect the operation, so it can be omitted when explaining the basic operation.) The signal appearing at is input. L composed of variable resistor 16 and inductor 17
Since the input signal is input to one end of the R circuit (series circuit), the voltage of the signal obtained by shifting the phase of the input signal by a predetermined amount by this LR circuit is applied to the non-inverting input terminal of the differential amplifier 12. It

The differential amplifier 12 thus outputs a signal obtained by amplifying the difference between the voltages applied to the two input terminals by a predetermined amplification degree, for example, about twice.

FIG. 3 is a vector diagram showing the relationship between the input / output voltage of the phase shift circuit 10 and the voltage appearing in the inductor or the like.

As shown in the figure, the voltage VL1 appearing across the inductor 17 and the voltage VR1 appearing across the variable resistor 16 are shown.
Are 90 degrees out of phase with each other, and these are combined (added) in vector form to be the input voltage Ei. Therefore, when the amplitude of the input signal is constant and only the frequency changes, the inductor 17 is moved along the circumference of the semicircle shown in FIG.
The voltage VL1 between both ends of the variable resistor 16 and the voltage VR1 across the variable resistor 16 change.

The voltage applied to the non-inverting input terminal of the differential amplifier 12 (voltage VL1 across the inductor 17) to the voltage applied to the inverting input terminal (voltage Ei /
The difference voltage Eo 'is obtained by subtracting 2) in vector. This differential voltage Eo 'can be represented by a vector whose starting point is the center point of the semi-circle shown in FIG. 3 and whose end point is one point on the circumference where the voltage VL1 and the voltage VR1 intersect, and its magnitude. Is equal to the radius Ei / 2 of the semicircle.
In practice, the differential amplifier 12 amplifies this differential voltage Eo 'by a factor of 2, resulting in the output voltage Eo = Eo' * 2 = Ei.
Therefore, in the phase shift circuit 10 of this embodiment, the amplitude of the input signal is equal to the amplitude of the output signal, and it is understood that signal attenuation does not occur between the input and output signals.

Further, as is clear from FIG. 3, the voltage VL1
And the voltage VR1 intersect at a right angle on the circumference, the input voltage E
The phase difference between i and the voltage VL1 changes from 90 ° to 0 ° as the frequency ω changes from 0 to ∞. And
The phase shift amount φ1 of the entire phase shift circuit 10 is twice that,
It varies from 180 ° to 0 ° depending on the frequency.

Next, the relationship between the input and output voltages described above will be verified quantitatively.

FIG. 4 is an equivalent diagram of the phase shift circuit 10 at the preceding stage, and shows a configuration corresponding to two series circuits provided on the input side of the differential amplifier 12.

Since the input voltage Ei is applied to both ends of the series circuit constituted by the resistors 18 and 20, the resistors 18 and 20 are connected.
Each of the two voltage sources 27, 2 generating a voltage Ei / 2
You can replace it with 8. At this time, the current I flowing in the closed loop of the equivalent circuit shown in FIG.
If the inductance of 17 is L and the resistance value of the variable resistor 16 is R,

[Equation 1] Becomes Here, the potential difference (difference) E between the two points shown in FIG.
If we ask for o ′,

[Equation 2] Becomes By substituting equation (1) into equation (2) above,

(Equation 3) Becomes Also, the output voltage Eo of the phase shift circuit 10 of this embodiment is
Is obtained by doubling the difference Eo ′ described above,

[Equation 4] Becomes Here, L consisting of the variable resistor 16 and the inductor 17
The time constant of the R circuit is T (= L / R).

Substituting s = jω in the equation (4) and transforming it,

(Equation 5) Becomes When the absolute value of the output voltage Eo is calculated from the equation (5),

(Equation 6) Becomes That is, the equation (6) is the phase shift circuit 10 of this embodiment.
Indicates that no matter how the phase between input and output rotates, the amplitude of the output signal is equal to the amplitude of the input signal and is constant.

From the equation (5), the input voltage E of the output voltage Eo is
When the phase shift amount φ1 for i is calculated,

(Equation 7) Becomes From this equation (7), for example, ω is approximately 1 / T (= R
/ L), the phase shift amount φ1 becomes approximately 90 °, and only the phase can be shifted approximately 90 ° without attenuating the amplitude of the input signal. Moreover, by changing the resistance value R of the variable resistor 16, the frequency ω at which the phase shift amount φ1 becomes approximately 90 ° can be changed.

FIG. 5 shows an extracted structure of the phase shift circuit 30 at the latter stage shown in FIG. The phase shift circuit 30 at the subsequent stage shown in the figure includes a differential amplifier 32 that amplifies a differential voltage of two inputs with a predetermined amplification degree (for example, about twice) and outputs the differential voltage.
The inductor 37 and the variable resistor 36 that shift the phase of the signal input to 42 by a predetermined amount and input to the non-inverting input terminal of the differential amplifier 32, and its voltage without changing the phase of the signal input to the input terminal 42. Divide the level to about 1/2 and differential amplifier 32
And resistors 38 and 40 for inputting to the inverting input terminal of the.

The capacitor 39 inserted in series with the inductor 37 is for blocking a direct current, and its impedance is extremely small at the operating frequency, that is, it has a large capacitance.

In the phase shift circuit 30 having such a configuration, when a predetermined AC signal is input to the input end 42, the voltage applied to the input end 42 (input A voltage obtained by dividing the voltage Ei) by the resistors 38 and 40 is applied. The resistance values of the resistors 38 and 40 are set to be substantially equal to each other, and the voltage Ei / 2 divided by about 1/2 by the voltage dividing circuit configured by the series circuit of these two resistors 38 and 40 is differential. Applied to the inverting input terminal of amplifier 32.

On the other hand, when the input signal is input to the input terminal 42, the inductor 37 is connected to the non-inverting input terminal of the differential amplifier 32.
The signal that appears at the connection point between the variable resistor 36 and the variable resistor 36 is input. Since the input signal is input to one end of the LR circuit (series circuit) configured by the inductor 37 and the variable resistor 36, the voltage of the signal obtained by shifting the phase of the input signal by a predetermined amount by the LR circuit is a differential amplifier. Applied to 32 non-inverting input terminals.

The differential amplifier 32 thus outputs a signal obtained by amplifying the difference between the voltages applied to the two input terminals by a predetermined amplification degree, for example, about twice.

FIG. 6 is a vector diagram showing the relationship between the input / output voltage of the phase shift circuit 30 and the voltage appearing in the inductor or the like.

As shown in the figure, the voltage VR2 appearing across the variable resistor 36 and the voltage VL2 appearing across the inductor 37.
Are 90 degrees out of phase with each other, and these are combined (added) in vector form to be the input voltage Ei. Therefore, when the amplitude of the input signal is constant and only the frequency changes, the voltage VR2 across the variable resistor 36 and the voltage VL2 across the inductor 37 change along the circumference of the semicircle shown in FIG.

The voltage applied to the non-inverting input terminal of the differential amplifier 32 (voltage VR2 across the variable resistor 36) to the voltage applied to the inverting input terminal (voltage Ei / 2 across the resistor 40).
Is a vectorial subtraction to obtain the differential voltage Eo '.
This differential voltage Eo 'can be represented by a vector whose starting point is the center point of the semicircle shown in FIG. 6 and whose end point is one point on the circumference where the voltage VR2 and the voltage VL2 intersect.
Its size is equal to the radius Ei / 2 of the semicircle. Actually, the differential amplifier 32 amplifies the differential voltage Eo 'by a factor of 2, and the output voltage Eo = Eo' * 2 = Ei. Therefore, in the phase shift circuit 30 of this embodiment, it is understood that the amplitude of the input signal and the amplitude of the output signal are equal, and no signal attenuation occurs between the input and output signals.

As is clear from FIG. 6, the voltage VR2
And the voltage VL2 intersect at a right angle on the circumference, the input voltage E
The phase difference between i and the voltage VR2 changes from 0 ° to 90 ° as the frequency ω changes from 0 to ∞. And
The phase shift amount φ2 of the entire phase shift circuit 30 is twice that,
It varies from 0 ° to 180 ° depending on the frequency.

Next, the relationship between the input and output voltages described above will be quantitatively verified.

FIG. 7 is a diagram equivalently showing the phase shift circuit 30 in the subsequent stage, and shows a configuration corresponding to two series circuits provided on the input side of the differential amplifier 32.

Since the input voltage Ei is applied to both ends of the series circuit constituted by the resistors 38 and 40, each of the resistors 38 and 40 receives the voltage E as in the case of the phase shift circuit 10 in the preceding stage.
It can be considered by replacing with two voltage sources 27 and 28 that generate i / 2. At this time, assuming that the resistance value of the variable resistor 36 is R and the inductance of the inductor 37 is L, the current I flowing in the closed loop of the equivalent circuit shown in FIG.
It can be represented by a formula.

Here, when the potential difference (difference) Eo 'between the two points shown in FIG. 7 is obtained,

(Equation 8) Becomes By substituting equation (1) into equation (8) above,

[Equation 9] Becomes Also, the output voltage Eo of the phase shift circuit 30 of this embodiment is
Is obtained by doubling the difference Eo ′ described above,

[Equation 10] Becomes Here, similarly to the phase shift circuit 10, the time constant of the LR circuit including the inductor 37 and the variable resistor 36 is T (= L / R).
And

Substituting s = jω in the equation (10) and transforming it,

[Equation 11] Becomes

The equations (10) and (11) described above are different from the equations (4) and (5) shown in the phase shift circuit 10 at the preceding stage only in the reference numerals. Therefore, as for the absolute value of the output voltage Eo, the equation (6) can be applied as it is, and the amplitude of the output signal of the phase shift circuit 30 in the subsequent stage is equal to the input signal, no matter how the phase between the input and the output rotates. It can be seen that the amplitude is equal to and constant.

Further, when the phase shift amount φ2 of the output voltage Eo with respect to the input voltage Ei is calculated from the equation (11),

(Equation 12) Becomes From this equation (12), for example, ω is approximately 1 / T (= R
/ L), the phase shift amount φ2 at a frequency is about 90 °, and only the phase can be shifted by about 90 ° without attenuating the amplitude of the input signal. Moreover, by changing the resistance value R of the variable resistor 36, the frequency ω at which the phase shift amount φ2 becomes approximately 90 ° can be changed.

In this way, the phase is shifted by a predetermined amount in each of the two phase shift circuits 10 and 30. Moreover,
As shown in FIG. 3 and FIG. 6, the relative phase relationships of the input and output voltages in the phase shift circuits 10 and 30 are in opposite directions,
At a certain frequency, the two phase shift circuits 10 and 30 together output a signal having a phase shift amount of 0 °.

The output of the phase shift circuit 30 at the subsequent stage is also connected to the non-inverting circuit provided at the front stage of the phase shift circuit 10 via the feedback resistor 70.
It is fed back to the input side of 50, and the fed-back signal is input to the input end (input end 22 shown in FIG. 2) of the phase shift circuit 10 at the previous stage via the non-inverting circuit 50 that functions as a buffer circuit. It

In the oscillator 1 of this embodiment, such a feedback loop is formed, and by setting the loop gain to 1 or more, the frequency at which the phase shift amount becomes 0 ° when the closed loop makes one round. Sine wave oscillation is performed at.
As a method of setting the loop gain to 1 or more,
There is a method of adjusting the amplification degree of each of the differential amplifiers 12 and 32 in the two phase shift circuits 10 and 30, and adjusting the amplification degree of the non-inverting circuit 50.

FIG. 8 is a system diagram in which the entire two phase shift circuits 10 and 30 and the non-inverting circuit 50 having the above-mentioned configuration are replaced with a circuit having a transfer function K1.
A closed loop is formed by the circuit having 1 and the feedback resistor 70 having a resistance value R0. FIG. 9 is a system diagram in which the system shown in FIG. 8 is converted by the Miller's theorem. When the feedback resistor 70 having a resistance value R0 is converted into an input shunt resistance as shown in FIG.

(Equation 13) Can be represented by

In this equation, considering that K1 is larger than 1, it can be seen that the input shunt resistance Rs becomes a negative resistance.

Assuming that the ideal phase shift circuit (all-pass network) having the transfer function K1 satisfies the condition that the phase shift amount is 0 ° at an arbitrary finite frequency, it is selective at this frequency. A negative resistance will be realized and oscillation will be possible. Actually, the input shunt resistance is in the form of being connected in parallel with the input impedance of the phase shift circuit, and it is necessary that a composite of these is a negative resistance, but the resistance value R0 of the feedback resistance 70 is set low, Since it is extremely easy to set the input impedance of the phase shift circuit to be high, it is theoretically possible to ignore the influence of the input impedance of the phase shift circuit.

By the way, as is clear from the equation (4), the transfer function K2 of the phase shift circuit 10 at the preceding stage is

[Equation 14] As is clear from the equation (10), the phase shift circuit 30
The transfer function K3 of

(Equation 15) Is. However, assuming that the time constants of the LR circuits in the phase shift circuits 10 and 30 are different, they are set to T ₁ and T ₂ , respectively.

Therefore, the overall transfer function K1 when the phase shift circuits 10 and 30 are connected is

[Equation 16] Becomes Here, in order to simplify the calculation, s = jω,
s ² = −ω ² , A = 1 + T ₁ · T ₂ · s ² = 1−T ₁ · T ₂ ·
If ω ² and B = T ₁ + T ₂ are set,

[Equation 17] Becomes In this equation (17), if the phase difference between the input and output of the entire two-stage connection of the phase shift circuits 10 and 30 is 0 °, (17)
Since the imaginary term on the right side of the equation must be 0, the following equation holds.

(1-T ₁ · T ₂ · ω ² ) (T ₁ + T ₂ ) ω = 0 (18) Therefore, 1−T ₁ · T ₂ · ω ² = 0 or ω = 0. Here, when ω = 0, the input signal is DC and the phase difference is 180 °, so that the other condition (1
Ω = 1 / √ (T ₁ · T ₂ ) satisfying −T ₁ · T ₂ · ω ² = 0)
At this time, the phase difference becomes 0 °. At this frequency, the input shunt resistance Rs becomes a negative resistance and simultaneously satisfies the oscillation voltage condition and the frequency condition.

As described above, by combining the two phase shift circuits 10 and 30, the phase shift amount of the signal that goes through the closed loop can be set to 0 ° at a certain frequency, and the loop gain at this time becomes 1 or more. By setting, sine wave oscillation is maintained. Further, the frequency at which the phase shift amount becomes 0 ° is the variable resistor 16 in each phase shift circuit 10, 30 or
Since it can be changed by changing the resistance value of 36, a variable frequency oscillator can be easily realized.

In the oscillator 1 of this embodiment, the inductors 17 and 37 can be formed on the semiconductor substrate by forming a spiral conductor by photolithography or the like. Inductor 17 and
By using 37, it is easy to form the whole oscillator 1 on the semiconductor substrate together with other components (differential amplifier, resistance, etc.) to form an integrated circuit. However, in this case, the inductance of the inductors 17 and 37 is extremely small, and the oscillation frequency is high. From another point of view, the oscillation frequency of the oscillator 1 is proportional to R / L, and the inductance L can be easily reduced by integration or the like, so that it is suitable for increasing the oscillation frequency.

In the oscillator 1 of the first embodiment described above, the phase shift circuit 10 is arranged in the front stage and the phase shift circuit 30 is arranged in the rear stage. Therefore, the oscillator may be configured by arranging the front and rear of these and arranging the phase shift circuit 30 in the front stage and the phase shift circuit 10 in the rear stage, respectively.

(Second Embodiment) The oscillator 1 of the first embodiment described above is configured by combining two phase shift circuits 10 and 30 having different configurations, but by combining two phase shift circuits having the same configuration. You may make it comprise an oscillator.

One phase shift circuit 10 included in the oscillator 1 shown in FIG. 1 has the basic configuration shown in FIG.
The relationship expressed by equation (4) is established between the 10 inputs and outputs. Below, the phase shift circuit 10 having the configuration shown in FIG.
For convenience sake, description will be given by using the fractional sign in the equation (4) as a "-type phase shift circuit". Further, the other phase shift circuit 30 included in the oscillator 1 shown in FIG. 1 has the basic configuration shown in FIG. 5, and the equation (10) is provided between the input and the output of the phase shift circuit 30. The relationship established is established. Hereinafter, the phase shift circuit 30 having the configuration shown in FIG. 5 will be referred to as a “+ type phase shift circuit” for the sake of convenience, using the fractional symbols in the expression (10).

In this way, when each phase shift circuit is classified into two types for convenience, the oscillator 1 of the first embodiment is constructed as a whole by combining two phase shift circuits 10 and 30 of different types. The oscillation operation is performed at the frequency where the phase shift amount is 0 °.

By the way, focusing on the relationship between the entire input and output when a phase inversion circuit for inverting the phase of a signal is connected to one − type phase shift circuit 10, the sign of the fraction “ It is sufficient to invert “−” to “+”, and it can be said that the configuration in which the phase inversion circuit is connected to one − type phase shift circuit is equivalent to one + type phase shift circuit. Similarly, focusing on the relationship between the entire input and output when a phase inversion circuit that inverts the phase of a signal is connected to one + -type phase shift circuit 30, the sign "+" of the fraction in Equation (10). It suffices to invert to "-", and it can be said that the configuration in which the phase inversion circuit is connected to one + type phase shift circuit is equivalent to one-type phase shift circuit.

Therefore, instead of combining two phase shift circuits 10 and 30 of different types in the first embodiment to form an oscillator, two phase shift circuits of the same type and a phase inversion circuit are combined to form an oscillator. be able to.

FIG. 10 is a diagram showing the structure of the oscillator of the second embodiment. The oscillator 1a shown in the figure has a phase inverting circuit 80 for inverting the phase of an input signal, two negative-type phase shifting circuits 10 shown in FIG.
A feedback resistor 70 for feeding back to the input side of 80 is included.

In the oscillator 1a having such a configuration, since the phase is shifted by 180 ° by the two phase shift circuits 10 at a certain frequency and the phase is inverted by the phase inverting circuit 80, the signal at a certain frequency as a whole is obtained. The phase shift amount of is 0 °. For example, assuming that the time constants of the LR circuits in the two phase shift circuits 10 are the same, and letting that value be T, the phase shift in each of the two phase shift circuits 10 at the frequency of ω = 1 / T. 90 °
Becomes Therefore, the phase is inverted by the phase inverting circuit 80, and the phase is shifted by 180 ° by the entire two phase shift circuits 10, so that the signal that the phase completes once and the phase shift amount becomes 0 ° is in the latter stage. It is output from the phase shift circuit 10.

The output of the subsequent phase shift circuit 10 is fed back to the input side of the phase inversion circuit 80 via the feedback resistor 70. By setting the loop gain to 1 or more, the predetermined frequency ω is obtained. The sine wave oscillation that it has is sustained.

Transfer function K of each of the two phase shift circuits 10
21 denotes the time constant of the LR circuit in each phase shift circuit 10 by T
Then, replacing T ₁ with T in equation (14),

[Formula 19] Becomes Therefore, the overall transfer function K11 when these two phase shift circuits 10 are connected in cascade and the phase inversion circuit 80 is further connected in the preceding stage is

(Equation 20) Becomes The right side of the equation (20) is (16) in the first embodiment.
It is equal to the transfer function K1 shown in the equation with T ₁ and T ₂ replaced by T. That is, the equation (20) is equal to the entire transfer function when the two phase shift circuits 10 and 30 and the non-inverting circuit 50 shown in the first embodiment are connected, and in this embodiment, the same type of transfer function is used. Two phase shift circuits 10 and phase inversion circuits 80
It can be seen that the configuration in which and are connected is equivalent to the configuration shown in FIG. 1 in the first embodiment.

Therefore, in the oscillator 1a of the second embodiment, the loop gain of the oscillator 1a is set to 1 or more by adjusting the amplification degree of the differential amplifier 12 or the phase inversion circuit 80 in the two phase shift circuits 10. By setting to, the sine wave oscillation is maintained at the frequency such that the phase shift amount becomes 0 ° when the circuit makes one cycle.

By changing the resistance value R of the variable resistor 16 in each phase shift circuit 10, the amount of phase shift in each phase shift circuit 10 can be changed, so that the two phase shift circuits 10
The frequency of which the total phase shift amount is 180 ° can be changed by the whole of the above, and the frequency variable oscillator 1 can be easily
a can be realized.

Further, in the oscillator 1a of this embodiment,
The inductor 17 can be formed on the semiconductor substrate by forming a spiral conductor by a photolithography method or the like, but by using such an inductor 17, other components (differential component) can be formed. It is also easy to form the entire oscillator 1a together with an amplifier, a resistor, etc.) on a semiconductor substrate to form an integrated circuit. Moreover, when integrated, the oscillation frequency can be easily increased.

(Third Embodiment) In the oscillator 1a of the second embodiment described above, the case where the two-type phase shift circuits 10 are connected has been described. However, the + type phase shift circuits 30 should be connected in two stages. You may make it comprise an oscillator.

FIG. 11 is a diagram showing the structure of the oscillator of the third embodiment. The oscillator 1b shown in the figure has a phase inversion circuit 80 for inverting the phase of an input signal, two + type phase shift circuits 30 shown in FIG.
A feedback resistor 70 for feeding back to the input side of 80 is included.

As described in the first embodiment, +
In each of the two phase shift circuits 30 of the type, the phase shift amount changes from 0 ° to 180 ° as the frequency ω of the input signal changes from 0 to ∞. For example, assuming that the time constants of the LR circuits in the two phase shift circuits 30 are the same, and the value is set as T, the phase shift in each of the two phase shift circuits 30 at the frequency of ω = 1 / T. The amount becomes 90 °.
Therefore, the phase of the two phase shift circuits 30 is 1
Since the phase is inverted by the phase inversion circuit 80 provided in the previous stage while being shifted by 80 °, overall,
A signal in which the phase makes one round and the phase shift amount becomes 0 ° is output from the phase shift circuit 30 in the subsequent stage.

The output of the subsequent phase shift circuit 30 is fed back to the input side of the phase inversion circuit 80 via the feedback resistor 70. By setting the loop gain to 1 or more, a predetermined frequency ω is obtained. The sine wave oscillation that it has is sustained.

By the way, the transfer function K of the two phase shift circuits 30
In the equation 31, when T is the time constant of the LR circuit in each phase shift circuit, T ₂ is replaced with T in the equation (15),

[Equation 21] Becomes This transfer function K31 is the phase shift circuit 10 shown in the equation (19).
The sign "-" of the transfer function K21 of is replaced with "+", and the overall transfer function K12 when two phase shift circuits 30 are connected in cascade and the phase inversion circuit 80 is connected to the preceding stage is:

[Equation 22] Therefore, it becomes the same as the transfer function K11 shown in the equation (20) in the second embodiment.

That is, the configuration in which two phase shift circuits 30 of the same type and the phase inversion circuit 80 are connected in this embodiment is two phase shift circuits of different types in the first embodiment.
It can be said that this is equivalent to a configuration in which the non-inverting circuit 50 is connected to the circuits 10 and 30 and a configuration in which two − type phase shift circuits 10 and the phase inverting circuit 80 are connected in the second embodiment.

Therefore, in the oscillator 1b of the third embodiment, the amplification factor of the differential amplifier 32 or the phase inversion circuit 80 in the two phase shift circuits 30 is adjusted so that the loop gain of the oscillator 1b is 1 or more. By setting to, the sine wave oscillation is maintained at the frequency such that the phase shift amount becomes 0 ° when the circuit makes one cycle.

Further, by changing the resistance value R of the variable resistor 36 in each phase shift circuit 30, the phase shift amount in each phase shift circuit 30 can be changed, so that the two phase shift circuits 30
The frequency of which the total phase shift amount is 180 ° can be changed by the whole of the above, and the frequency variable oscillator 1 can be easily
b can be realized.

Further, in the oscillator 1b of this embodiment,
The inductor 17 can be formed on the semiconductor substrate by forming a spiral conductor by a photolithography method or the like, but by using such an inductor 37, other components (differential component) can be formed. It is easy to form the entire oscillator 1b together with the amplifier and the resistor) on the semiconductor substrate to form an integrated circuit. Moreover, when integrated, the oscillation frequency can be easily increased.

(Other Embodiments) Non-inverting circuit 50 or phase inverting circuit 80 included in the oscillator of each of the above-described embodiments.
Can be easily configured by combining transistors, operational amplifiers, resistors, and the like.

FIG. 12 is a diagram showing a specific example of the non-inverting circuit and the phase inverting circuit configured by using the operational amplifier. Same figure
The non-inverting circuit 50 shown in (A) is configured to include an operational amplifier 52 having an inverting input terminal grounded via a resistor 54 and a resistor 56 connected between the inverting input terminal and the output terminal. And functions as a buffer having a predetermined amplification degree determined by the resistance ratio of the two resistors 54 and 56. When an AC signal is input to the non-inverting input terminal of the operational amplifier 52,
In-phase signals are output from the output terminal of the operational amplifier 52.

The phase inversion circuit 80 shown in FIG.
It includes an operational amplifier 82 in which an input signal is input to the inverting input terminal via a resistor 84 and the non-inverting input terminal is grounded, and a resistor 86 connected between the inverting input terminal and the output terminal of the operational amplifier 82. It is composed of. The phase inversion circuit 80 has a predetermined amplification degree determined by the resistance ratio of the two resistors 84 and 86, and the operational amplifier 82 is connected via the resistor 84.
When an AC signal is input to the inverting input terminal of
The 82-phase output terminal outputs an inverted signal whose phase is inverted.

By the way, the oscillators of the respective embodiments described above are
It is composed of two phase shift circuits and a non-inversion circuit or a phase inversion circuit, and performs a predetermined oscillation operation by setting the total phase shift amount to 0 ° at a predetermined frequency by the whole of a plurality of connected circuits. It is like this. Therefore, focusing only on the amount of phase shift, there is some freedom in the order in which the phase shift circuit and the non-inverting circuit or the phase inverting circuit are connected, and the connection order can be determined as necessary. .

FIG. 13 is a diagram showing a connection state of the two phase shift circuits and the non-inverting circuit 50 when the oscillator is constructed by combining the two phase shift circuits of different types and the non-inverting circuit. In these figures, the feedback impedance element 70a most commonly uses a feedback resistor 70 as shown in FIG. However, the feedback impedance element 70a may be formed by a capacitor or an inductor, or may be formed by combining a resistor, a capacitor or an inductor.

FIG. 13A shows a configuration in which a non-inverting circuit 50 is arranged at the subsequent stage of two phase shift circuits of different types (one is a − type and the other is a + type). . As described above, when the non-inverting circuit 50 is arranged in the subsequent stage, a large output current can be taken out by giving the non-inverting circuit 50 a function of an output buffer.

FIG. 13B shows a configuration in which the non-inverting circuit 50 is arranged in the middle of two phase shift circuits of different types. In this way, when the non-inverting circuit 50 is arranged in the middle, the phase shift circuit 10 or 30 at the front stage and the phase shift circuit 30 at the rear stage are arranged.
Alternatively, 10 mutual interferences can be completely prevented.

FIG. 13C shows a configuration in which the non-inverting circuit 50 is arranged in the preceding stage of two phase shift circuits of different types, which corresponds to the oscillator 1 shown in FIG. In this way, when the non-inverting circuit 50 is arranged in the previous stage, the impedance element on the feedback side for the phase shift circuit 10 or 30 in the previous stage
The influence of 70a can be minimized.

Similarly, FIG. 14 is a diagram showing a connection state of two phase shift circuits and a phase inversion circuit 80 when an oscillator is constructed by combining two phase shift circuits of the same type and a phase inversion circuit. . As described with reference to FIG. 13, the feedback impedance element 70a most commonly uses the feedback resistor 70 as shown in FIG. However, the feedback impedance element 70a may be formed by a capacitor or an inductor, or may be formed by combining a resistor, a capacitor or an inductor.

FIG. 14A shows a configuration in which the phase inversion circuit 80 is arranged at the subsequent stage of two phase shift circuits of the same type. As described above, when the phase inverting circuit 80 is arranged in the subsequent stage, a large output current can be taken out by giving the phase inverting circuit 80 a function as an output buffer.

FIG. 14B shows a configuration in which the phase inversion circuit 80 is arranged between two phase shift circuits of the same type. Thus, when the phase inversion circuit 80 is arranged in the middle, mutual interference between the two phase shift circuits can be completely prevented.

FIG. 14C shows a configuration in which the phase inversion circuit 80 is arranged in the preceding stage of the two phase shift circuits.
It corresponds to the oscillator 1a shown in FIG. 11 and the oscillator 1b shown in FIG. As described above, when the phase inversion circuit 80 is arranged in the preceding stage, the influence of the feedback side impedance element 70a on the preceding phase shift circuit 10 or 30 can be minimized.

The phase shift circuits 10 and 30 shown in each of the above-described embodiments include the variable resistor 16 or 36. These variable resistors 16 and 36 can be specifically realized by using a junction type or MOS type FET.

FIG. 15 is a diagram showing the configuration of the phase shift circuit in the case where the variable resistor 16 or 36 in the two types of phase shift circuits shown in the respective embodiments is replaced with an FET.

FIG. 9A shows a configuration in which the variable resistor 16 is replaced with an FET in the one phase shift circuit 10 shown in FIG. 1 and the like. FIG. 2B shows a configuration in which the variable resistor 36 in the other phase shift circuit 30 shown in FIG.

As described above, the channel formed between the source and drain of the FET is used as a resistor to make the variable resistance variable.
When used in place of 16 or 36, the gate voltage can be variably controlled to arbitrarily change the channel resistance within a certain range to change the amount of phase shift in each phase shift circuit. Therefore, in each oscillator, the frequency at which the amount of phase shift of the signal that goes around once becomes 0 ° can be changed, so that the oscillation frequency of the oscillator can be arbitrarily changed.

In each phase shift circuit shown in FIG. 15, the variable resistance is composed of one FET, that is, a p-channel or n-channel FET.
Alternatively, T and n-channel FETs may be connected in parallel to form one variable resistor, and gate voltages of equal magnitude but different polarities may be applied between the gate and substrate of each FET. When changing the resistance value, the magnitude of the gate voltage may be changed. As described above, by combining the two FETs to form the variable resistance, it is possible to improve the non-linear region of the FETs, so that the distortion of the oscillation output can be reduced.

Further, the phase shift circuit 10 or 30 shown in each of the above-mentioned embodiments changes the resistance value of the variable resistor 16 or 36 connected in series with the inductor 17 or 37 to change the amount of phase shift. Although the overall oscillation frequency is changed by means of, the inductors 17, 37 may be formed by variable inductors, and the oscillation frequency may be changed by changing the inductance.

FIG. 16 is a diagram showing the configuration of the phase shift circuit when the inductor 17 or 37 in the two types of phase shift circuits shown in each embodiment is replaced with a variable inductor.

FIG. 1A shows a configuration in which the variable resistor 16 is replaced with a fixed resistor and the inductor 17 is replaced with a variable inductor 17a in the one phase shift circuit 10 shown in FIG. 1 and the like. FIG. 1B shows a configuration in which the variable resistor 36 is replaced with a fixed resistor and the inductor 37 is replaced with a variable inductor 37a in the other phase shift circuit 30 shown in FIG.

As described above, the inductor 17 or 37 can be replaced with the variable inductor 17a or 37a, and the inductance of the variable inductor 17a or 37a can be arbitrarily changed within a certain range to change the phase shift amount in each phase shift circuit.
Therefore, it is possible to change the frequency at which the amount of phase shift of the signal that goes round in each oscillator becomes 0 °, and it is possible to arbitrarily change the oscillation frequency.

By the way, in FIGS. 16A and 16B described above, only the inductance of the variable inductor 17a or 37a is changed, but the resistance value of the variable resistor 16 or 36 may be changed at the same time. FIG. 16C shows a configuration in which the variable resistor 16 is used and the inductor 17 is replaced with the variable inductor 17a in the one phase shift circuit 10 shown in FIG. 1 and the like. FIG. 2D shows a configuration in which the variable resistor 36 is used and the inductor 37 is replaced with a variable inductor 37a in the other phase shift circuit 30 shown in FIG.

Needless to say, the variable resistance shown in FIGS. 16C and 16D can be formed by utilizing the channel resistance of the FET as shown in FIG. In particular,
When a p-channel FET and an n-channel FET are connected in parallel to form one variable resistor and gate voltages of equal magnitude and different polarities are applied between the base and substrate of each FET, Since the nonlinear region can be improved, the distortion of the oscillation output can be reduced.

As described above, even when the variable resistance and the variable inductor are combined to form the phase shift circuit, the resistance value of the variable resistor and the inductance of the variable inductor are arbitrarily changed within a certain range. The amount of phase shift at can be changed. Therefore, it is possible to change the frequency at which the amount of phase shift of the signal that goes round in each oscillator becomes 0 °, and it is possible to arbitrarily change the oscillation frequency.

In addition to the case where the variable resistance or the variable inductor is used as described above, a plurality of resistors or inductors having different element constants are prepared, and the switch is switched so that one of the plurality of elements can be selected. Alternatively, one or a plurality may be selected. In this case, the element constants can be discontinuously switched by the number of elements to be connected and the connection method (serial connection, parallel connection, or a combination thereof) by switching the switches.
For example, instead of the variable resistor, the resistance value is R, 2R, 4R,
By preparing a plurality of 2 n-th power series resistors and selecting one or an arbitrary plurality of resistors and connecting them in series, it is possible to easily switch the resistance values at even intervals with fewer elements. can do.

FIG. 17 is a diagram showing a specific example of the above-described variable inductor 17a, and shows an outline of a planar structure formed on a semiconductor substrate. The structure of the variable inductor 17a shown in the figure can be directly applied to the variable inductor 37a.

The variable inductor 17a shown in the figure is a spiral inductor conductor 112 formed on a semiconductor substrate 110.
And a control conductor 11 formed so as to circulate the outer periphery thereof.
4 and an insulating magnetic body 118 formed so as to cover both the inductor conductor 112 and the control conductor 114.

The control conductor 114 described above is the control conductor 114.
A variable voltage power supply 116 is connected to both ends of the variable voltage power supply 116 to change the bias current flowing through the control conductor 114 by variably controlling the DC bias voltage applied by the variable voltage power supply 116. be able to.

For the semiconductor substrate 110, for example, an n-type silicon substrate (n-Si substrate) or another semiconductor material (for example, amorphous material such as germanium or amorphous silicon) is used. The inductor conductor 112 is formed by spirally forming a metal thin film such as aluminum or gold or a semiconductor material such as polysilicon.

The semiconductor substrate 110 shown in FIG. 17 is formed with other components of the oscillator shown in FIG. 1 and the like in addition to the variable inductor 17a.

FIG. 18 shows the variable inductor shown in FIG.
It is a figure which shows the shape of the inductor conductor 112 and the control conductor 114 of 17a in more detail.

As shown in the figure, the inductor conductor 112 located on the inner peripheral side has a predetermined number of turns (for example, about 4 turns).
Is formed in a spiral shape, and two terminal electrodes 122 and 124 are connected to both ends thereof. Similarly, the control conductor 114 located on the outer peripheral side is formed in a spiral shape with a predetermined number of turns (for example, about 2 turns), and the control conductor 114 has two ends at both ends.
Two control electrodes 126 and 128 are connected.

FIG. 19 is an enlarged sectional view taken along the line AA of FIG. 18, showing a cross section of the insulating magnetic body 118 including the inductor conductor 112 and the control conductor 114.

As shown in the figure, the inductor conductor 112 and the control conductor 114 are formed on the surface of the semiconductor substrate 110 via the insulating magnetic material film 118a, and the insulating magnetic material film 118b is further formed on the surface thereof. Is coated. The insulating magnetic material 118 shown in FIG. 17 is formed by these two magnetic material films 118a and 118b.

For example, as the magnetic films 118a and 118b,
Various magnetic films such as gamma-ferrite and barium-ferrite can be used. Various kinds of materials and forming methods of these magnetic films can be considered. For example, a method of forming a magnetic film by vacuum vapor deposition of FeO or the like, other molecular beam epitaxy method (MBE method), chemical vapor deposition, etc. There is a method of forming a magnetic film by using a phase growth method (CVD method), a sputtering method or the like.

The insulating film 130 is made of a non-magnetic material, and covers the winding portions of the inductor conductor 112 and the control conductor 114. By eliminating the magnetic films 118a and 118b between the winding portions in this way, the leakage magnetic flux generated between the winding portions can be minimized, so that the magnetic flux generated by the inductor conductor 112 can be effectively used. As a result, the variable inductor 17a having a large inductance can be realized.

As described above, in the variable inductor 17a shown in FIG. 17 and the like, the insulating magnetic material 118 (magnetic material films 118a and 118b) is formed so as to cover the inductor conductor 112 and the control conductor 114, By controlling the DC bias current flowing through the control conductor 114 variably, the above-mentioned insulating magnetic material is used.
The saturation magnetization characteristic of the inductor conductor 112 having the magnetic path 118 changes, and the inductance of the inductor conductor 112 changes.

Therefore, the inductance itself of the inductor conductor 112 can be directly changed, and the inductor conductor 112 can be formed on the semiconductor substrate 110 by using a thin film forming technique or a semiconductor manufacturing technique, which facilitates the production.
Further, since it is possible to form other components such as the oscillator 1 on the semiconductor substrate 110, it is suitable when the whole oscillator of each embodiment is integrally formed by integration.

The variable inductor 17 shown in FIG.
20a or 21, the inductor conductor 112 and the control conductor 114 may be alternately wound, or the inductor conductor 112 and the control conductor 114 may be overlapped with each other. In either case, the control conductor 11
The saturation magnetization characteristic of the insulating magnetic body 118 can be changed by changing the DC bias current flowing through the inductor 4, and the inductance of the inductor conductor 112 can be changed within a certain range.

The variable inductor 17 shown in FIG.
Although a has been described by taking the case where the inductor conductor 112 and the like are formed on the semiconductor substrate 110 as an example, it may be formed on various insulating or conductive substrates such as ceramics.

Although the insulating material is used for the magnetic films 118a and 118b, a conductive material such as metal powder (MP) may be used. However, if such a conductive magnetic film is used by replacing it with the above-mentioned insulating magnetic film 118a or the like, each winding portion of the inductor conductor 112 or the like is short-circuited and does not function as an inductor conductor.
It is necessary to electrically insulate each inductor conductor from the conductive magnetic film. As this insulating method, a method of forming an insulating oxide film by oxidizing the inductor conductor 112 or the like,
There is a method of forming a silicon oxide film or a nitride film by a chemical vapor deposition method or the like.

In particular, a conductive material such as metal powder has a large magnetic permeability as compared with an insulating material such as gamma-ferrite, and therefore has an advantage that a large inductance can be secured.

The variable inductor 17 shown in FIG.
In the case of a, the whole of both the inductor conductor 112 and the control conductor 114 is covered with the insulating magnetic material 118, but only part of them may be covered to form the magnetic path.

FIG. 22 is a diagram showing a variable inductor in which the insulating magnetic material 118 is partially formed. As shown in the figure, the insulating magnetic material 118 is used as the inductor conductor 112 and the control conductor.
It is formed so as to cover a part of 114, and a magnetic path is formed by this partially formed insulating magnetic body 118. In this way, when the insulating magnetic body (or the conductive magnetic body may be used) 118 that serves as a magnetic path is partially formed, the magnetic flux is generated by the inductor conductor 112 and the control conductor 114 due to the narrowed magnetic path. It becomes easy to saturate. Therefore, even when a small bias current is passed through the control conductor 114, the magnetic flux is saturated, and the inductance of the inductor conductor 112 can be changed by variably controlling the small bias current. Therefore, the structure of the control system can be simplified.

The variable inductor 17 shown in FIG.
The conductor a is formed by concentrically winding the inductor conductor 112 and the control conductor 114. These conductors are formed at adjacent positions on the surface of the semiconductor substrate 110, and an insulating or conductive magnetic material is formed between them. You may magnetically couple by the magnetic path formed of the body.

FIG. 23 is a variable inductor in which an inductor conductor and a control conductor are formed side by side at adjacent positions.
It is a top view which shows the outline of 17b.

The variable inductor 17b shown in the figure is a spiral inductor conductor 112 formed on a semiconductor substrate 110.
a, a spiral-shaped control conductor 114a formed at a position adjacent to the inductor conductor 112a, and an insulating magnetic body (or conductive material) formed so as to cover each spiral center of the inductor conductor 112a and the control conductor 114a. Magnetic material) 119
It is comprised including.

Similar to the variable inductor 17a shown in FIG. 17 and the like, a variable voltage power source 116 is connected to the control conductor 114a for applying a variable bias voltage across the control conductor 114a. By variably controlling the applied bias voltage, it is possible to change a predetermined bias current flowing through the control conductor 114a.

FIG. 24 shows the variable inductor shown in FIG.
It is the figure which showed in more detail the shape of the inductor conductor 112a and the control conductor 114a of 17b.

As shown in the figure, the inductor conductor 112a
Is formed in a spiral shape with a predetermined number of turns (for example, about 4 turns), and two terminal electrodes 122 and 124 are provided at both ends thereof.
Is connected. Similarly, the control conductor 114a arranged adjacent to the inductor conductor 112a is formed in a spiral shape with a predetermined number of turns (for example, about 2 turns),
Two control electrodes 126 and 128 are connected to both ends thereof.

FIG. 25 is an enlarged sectional view taken along the line BB of FIG. 24, showing a cross section of the insulating magnetic body 119 including the inductor conductor 112a and the control conductor 114a.

As shown in the figure, an insulating magnetic film 119a and an insulating non-magnetic film 132 are formed on the surface of the semiconductor substrate 110, and the inductor conductor 112a and the control conductor 114a are respectively formed on the surface. Has been formed. Then, an insulating magnetic film 119 is further formed on the surface of the inductor conductor 112a and the control conductor 114a so as to penetrate the central portions thereof.
b is formed by coating. These two magnetic films 119
inductor conductor 112a and control conductor 11 by a and 119b.
An annular magnetic body 119 is formed which serves as a common magnetic path for 4a.

The insulating non-magnetic film shown in FIG.
Reference numeral 132 has a thickness substantially the same as that of the magnetic film 119a, and is for forming the inductor conductor 112a and the control conductor 114a at substantially the same height on their surfaces. Therefore, if a slight step may occur in the inductor conductor 112a and the control conductor 114a, one of the inductor conductor 112a and the control conductor 114a may be directly formed on the semiconductor substrate 110 without forming the nonmagnetic film 132. You may make it form a part.

Further, between the inductor conductor 112a and the control conductor 114a on the surface of the magnetic film 119a,
Similar to the variable inductor 17a shown in FIG.
0 is formed. By partially filling the insulating film 130 and excluding the magnetic films 119a and 119b between the winding portions in this manner, the leakage flux generated between the winding portions can be minimized. Most of the magnetic flux generated by 112a is magnetic film 119a, 119a.
It comes to intersect with the control conductor 114a through b. Therefore, by reducing the leakage magnetic flux, a large inductance can be obtained by effectively utilizing the magnetic flux generated by the inductor conductor 112a.

As described above, the variable inductor 17b described above is used.
Is an annular insulating magnetic material 119 (magnetic material film 119) so as to pass through the spiral centers of the inductor conductor 112a and the control conductor 114a.
a, 119b) are formed. Therefore, by variably controlling the DC bias current flowing through the control conductor 114a, the saturation magnetization characteristic of the inductor conductor 112a having the magnetic body 119 as a magnetic path changes, and the inductor conductor 112a
The inductance of the element also changes.

Further, as described above, the oscillator 1 of each embodiment is
When the above is formed on a semiconductor substrate, it is not possible to secure a very large inductance as the inductor 17 or 37. Therefore, if the small inductance such as the inductor 17 actually formed on the semiconductor substrate can be apparently increased by devising the circuit, the time constant T is set to a large value to reduce the oscillation frequency. It is convenient for you.

FIG. 26 shows the phase shift circuits 10 and 30 shown in FIG.
FIG. 9 is a diagram showing a modified example in which the inductor 17 or 37 used for is not a single element but a circuit, and is an inductance conversion circuit that makes the inductance of an inductor element (inductor conductor) actually formed on a semiconductor substrate look large. Function. The entire circuit shown in FIG. 26 corresponds to the inductor 17 or 37 included in the phase shift circuits 10 and 30.

The inductance conversion circuit 17c shown in FIG.
Is an inductor 210 having a predetermined inductance L0.
And two operational amplifiers 212 and 214 and two resistors 216 and 218
It is comprised including.

The operational amplifier 212 in the first stage is a non-inverting amplifier with a gain of 1 whose output terminal is connected to the inverting input terminal, and mainly functions as a buffer for impedance conversion. Similarly, the output terminal of the second-stage operational amplifier 214 is also connected to the inverting input terminal and functions as a non-inverting amplifier having a gain of 1. Further, a voltage dividing circuit by resistors 216 and 218 is inserted between these two non-inverting amplifiers.

By thus inserting the voltage dividing circuit between them, the gain of the entire amplifier including the two non-inverting amplifiers can be freely set between 0 and 1.

The inductance conversion circuit 17 shown in FIG.
27, assuming that the transfer function of the entire circuit excluding the inductor 210 is K4 in FIG.
It can be represented by the system diagram shown in. FIG. 28 is a system diagram in which this is converted by the Miller's theorem.

If the impedance Z0 shown in FIG. 27 is used to represent the impedance Z1 shown in FIG. 28,

(Equation 23) Becomes Here, in the case of the inductance conversion circuit 17c shown in FIG. 26, the impedance Z0 = jωL0, which is substituted into the equation (23),

[Equation 24]

(Equation 25) Becomes The equation (25) shows that the inductance L0 of the inductor 210 in the inductance conversion circuit 17c is apparently 1 / (1-K4) times.

Therefore, when the gain K4 is positive and lies between 0 and 1, 1 / (1-K4) is always larger than 1, so that the inductance L0 can be changed to a larger value.

By the way, the gain of the amplifier in the inductance conversion circuit 17c shown in FIG. 26, that is, the gain K4 of the amplifier constituted by the entire operational amplifiers 212 and 214.
Is determined by the voltage division ratio of the voltage dividing circuit formed by the resistors 216 and 218. If the respective resistance values are R16 and R18,

(Equation 26) Becomes Substituting this gain K4 into equation (25) and calculating the apparent inductance L,

[Equation 27] Becomes Therefore, the resistance ratio of resistors 216 and 218 is R18 / R16.
Is increased, the apparent inductance L between the two terminals 204 and 206 can be increased. For example, when R18 = R16, the inductance L can be doubled from L0 from the equation (27).

In this way, the above-described inductance conversion circuit 17c changes the voltage division ratio of the voltage dividing circuit inserted between the two non-inverting amplifiers to make the inductance L0 of the inductor 210 actually connected apparent. Can be made bigger. Therefore, in the case where the entire oscillator 1 shown in FIG. 1 and the like is formed on the semiconductor substrate, the inductor 210 having a small inductance L0 is formed on the semiconductor substrate by a spiral conductor or the like. The inductance conversion circuit shown in FIG. 26 can convert to a large inductance L, which is convenient for integration. In particular, if a large inductance can be secured in this way, it becomes easy to reduce the oscillation frequency of the oscillator 1 shown in FIG. 1 to a relatively low frequency range. In addition, by integrating,
It is possible to reduce the mounting area of the entire oscillator and reduce the material cost.

In addition to the case where the voltage division ratio of the voltage dividing circuit by the resistors 216 and 218 is fixed, at least one of these two resistors 216 and 218 is formed by a variable resistor, so that it is specifically a junction type or MOS type. Type FET or p-channel FET and n-channel FET are connected in parallel to form a variable resistance, and this voltage division ratio may be continuously changed. In this case, the operational amplifier 21 shown in FIG.
The gain of the entire amplifier configured including 2, 214 changes,
The inductance L between the terminals 204 and 206 also changes continuously. Therefore, by using this inductance conversion circuit 17c instead of the variable inductor 17a shown in FIG. 16, the phase shift amount in each phase shift circuit can be arbitrarily changed within a certain range. Therefore, it is possible to change the frequency at which the phase shift amount of the signal that makes one round in the oscillator becomes 0 °, and it is possible to arbitrarily change the oscillation frequency of the oscillator described above.

Further, in the inductance conversion circuit 17c shown in FIG. 26, since the gain of the entire amplifier including the two operational amplifiers 212 and 214 is set to 1 or less, the whole is replaced with the emitter follower circuit or the source follower circuit. Good.

FIG. 29 is a diagram showing the structure of an inductance conversion circuit in which the entire amplifier including the operational amplifiers 212 and 214 is replaced with an emitter follower circuit. The inductance conversion circuit 17d shown in FIG.
A bipolar transistor 228 to which 4 and 226 are connected, a voltage dividing point by these two resistors 224 and 226, and the transistor 228.
It is configured to include an inductor 210 connected between the base and a base and a capacitor 230 for blocking a direct current.
Capacitor 230 inserted on one end side of inductor 210
Has an extremely small impedance at the operating frequency, that is, a large capacitance so as not to affect the frequency characteristics.

The gain of the above-described emitter follower circuit is determined mainly by the resistance ratio of the two resistors 224 and 226, and the gain is always less than 1. Therefore, as can be seen from the equation (25), the inductor is actually Inductance L0 of 210
It can be made larger in appearance. Moreover, since only one emitter follower circuit is used, the circuit configuration can be simplified and the maximum operating frequency can be set high.

FIG. 29B is a diagram showing a modification thereof,
The difference is that the two resistors 224 and 226 in FIG. By using the variable resistor 232 in this way, the gain can be changed arbitrarily and continuously, so that the apparent inductance L can also be changed arbitrarily and continuously, and the inductance conversion circuit 17e can be changed. Variable inductor 17 shown in FIG.
By using it instead of a, the phase shift amount in each phase shift circuit can be arbitrarily changed within a certain range. Therefore, it is possible to change the frequency at which the phase shift amount of the signal that makes one round in the oscillator becomes 0 °, and it is possible to arbitrarily change the oscillation frequency of the oscillator described above.

In the inductance conversion circuit 17e shown in FIG. 29B, the two resistors 224 and 226 shown in FIG. 29A are replaced by one variable resistor 232.
At least one of 24 and 226 may be configured by a variable resistor.

FIG. 30 is a circuit in which each of the inductance conversion circuits 17d and 17e shown in FIGS. 29A and 29B is realized by a source follower circuit, and the bipolar transistor 228 is replaced by a FET 234. 30A corresponds to FIG. 29A, and FIG. 30B corresponds to FIG. 29B.

FIG. 31 is a diagram showing a modification of the inductance conversion circuit 17c shown in FIG. The inductance conversion circuit 17f shown in FIG. 31 includes an npn-type bipolar transistor 236 and a resistor 240 connected to its emitter.
A pnp-type bipolar transistor 238, a resistor 242 connected to the emitter thereof, and an inductor 210 having an inductance L0.

One transistor 236 and resistor 240 described above
Form a first emitter follower circuit, and the other transistor 238 and the resistor 242 form a second emitter follower circuit, which are connected in cascade. Moreover,
npn-type transistor 236 and pnp-type transistor 2
Since 38 is used, the base potential of the transistor 236 at one end of the inductor 210 and the emitter potential of the transistor 238 can be set to be substantially the same, and a DC current blocking capacitor is not required.

The present invention is not limited to the above embodiments, and various modifications can be made within the scope of the gist of the present invention.

For example, in the oscillator according to each of the above-mentioned embodiments, the differential amplifiers 12 and 32 in the phase shift circuits 10 and 30 are used to increase the difference between the two.
The loop gain of the oscillator is set to 1 or more by amplifying the difference between the inputs by almost 2 times and using it as the output of each phase shift circuit. However, the amplification degree of the differential amplifiers 12 and 32 is set to other than this. It may be set to a value. For example, in each differential amplifier 12, 32, the difference between the two inputs is not amplified or is amplified with an amplification degree other than 2 and output, and the amplification degree of the non-inverting circuit 50 or the phase inverting circuit 80 is adjusted. The loop gain of the oscillator may be set to 1 or more.

In addition, the oscillator 1 or the like of the above-described embodiment has 2
Although two phase shift circuits are included, when the oscillation frequency is changed, in addition to changing the element constants of at least one of the inductor and the resistor forming the LR circuits included in both phase shift circuits, It is conceivable to change the element constant of at least one of the inductor and the resistor that form the LR circuit included in the phase shift circuit. Further, by fixing all element constants of all resistors and inductors, it is possible to construct an oscillator having a fixed oscillation frequency.

Further, the oscillator of each of the above-described embodiments is provided by one of the two phase shift circuits forming the oscillator, or by the two phase shift circuits and the non-inverting circuit 50 or the phase inverting circuit 80. Although the sine wave signal is taken out from one circuit, the sine wave signal may be taken out from two circuits or three circuits forming the oscillator. In particular, when the time constants of the two phase shift circuits 10 or 30 forming the oscillator are set to be the same, the phase shift amount in each phase shift circuit becomes 90 °, so the phases are 90 ° to each other.
It is possible to take out the shifted two-phase outputs. Further, from the circuits before and after the phase inverting circuit 80 is sandwiched, two-phase outputs whose phases are mutually inverted can be taken out.

[0159]

As is clear from the description based on each of the above embodiments, when the oscillation frequency is high, each element constituting the oscillator of the present invention can be formed by the integrated circuit manufacturing method. The oscillator can be miniaturized as an integrated circuit on a semiconductor wafer, and can be manufactured inexpensively by mass production. Further, the inductor in each phase shift circuit can be converted to a larger one by using the inductance conversion circuit, and the oscillation frequency can be lowered.

In particular, the channel between the source and drain of the FET is used as the variable resistance of the LR circuit in each phase shift circuit, and the resistance of the channel is changed by changing the control voltage applied to the gate of this FET. Then, it is possible to avoid the influence of the inductance and the capacitance of the wiring to which the control voltage is applied, and it is possible to obtain an oscillator having ideal characteristics almost as designed.

Further, in the conventional oscillator utilizing LC resonance, the oscillation frequency ω is 1 / √LC. Therefore, if the capacitance C or the inductance L is changed to adjust the oscillation frequency, the oscillation frequency becomes Although it changes in proportion to the square root of the change amount, in the oscillator of the present invention, the oscillation frequency ω is, for example, R / L, and the oscillation frequency can be changed in proportion to the resistance value R. Significant changes and adjustments are possible. Further, since the inductance L can be easily reduced, it is easy to increase the oscillation frequency, and an oscillator having a high oscillation frequency can be realized.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing the configuration of an oscillator according to a first embodiment of the invention,

FIG. 2 is a diagram showing an extracted configuration of a phase shift circuit in the preceding stage shown in FIG.

FIG. 3 is a vector diagram showing the relationship between the input / output voltage of the preceding phase shift circuit and the voltage appearing in the inductor,

FIG. 4 is an equivalent view of the phase shift circuit shown in FIG.

FIG. 5 is a diagram showing an extracted configuration of a phase shift circuit at a subsequent stage shown in FIG.

FIG. 6 is a vector diagram showing the relationship between the input / output voltage of a subsequent phase shift circuit and the voltage appearing in an inductor or the like;

7 is an equivalent view of the phase shift circuit shown in FIG.

FIG. 8 is a system diagram in which the entire two phase shift circuits and the specific inversion circuit are replaced with a circuit having a transfer function K1;

FIG. 9 is a system diagram obtained by converting the system shown in FIG. 8 according to Miller's theorem,

FIG. 10 is a circuit diagram showing a configuration of an oscillator according to a second embodiment,

FIG. 11 is a circuit diagram showing a configuration of an oscillator according to a third embodiment,

FIG. 12 is a diagram showing a specific example of a non-inverting circuit and a phase inverting circuit;

FIG. 13 is a diagram showing a connection form of a phase shift circuit and a non-inverting circuit;

FIG. 14 is a diagram showing a connection form of a phase shift circuit and a phase inversion circuit;

FIG. 15 is a diagram showing a configuration of a phase shift circuit in which a variable resistance of the phase shift circuit is replaced with an FET,

FIG. 16 is a diagram showing a configuration of a phase shift circuit in which the inductor of the phase shift circuit is replaced with a variable inductor;

FIG. 17 is a diagram showing an example of a variable inductor,

18 is a diagram showing in more detail the shapes of the inductor conductor and the control conductor of the variable inductor shown in FIG.

19 is an enlarged cross-sectional view taken along the line AA of FIG.

20 is a diagram showing a modification of the variable inductor shown in FIG. 17,

21 is a diagram showing a modification of the variable inductor shown in FIG. 17,

22 is a diagram showing a modification of the variable inductor shown in FIG. 17,

FIG. 23 is a view showing another example of the variable inductor,

24 is a diagram showing in more detail the shapes of the inductor conductor and the control conductor of the variable inductor shown in FIG. 23;

25 is an enlarged sectional view taken along line BB of FIG.

FIG. 26 is a diagram showing the configuration of an inductance conversion circuit that apparently increases the inductance that the inductor actually has;

27 is a diagram showing the circuit shown in FIG. 26 using a transfer function,

28 is a diagram in which the configuration shown in FIG. 27 is converted by the Miller's theorem,

29 is a diagram showing the configuration of an inductance conversion circuit in which the entire amplifier including the two operational amplifiers included in FIG. 26 is replaced with an emitter follower circuit;

30 is a diagram showing a configuration in which the circuit of FIG. 29 is realized by a source follower circuit,

FIG. 31 is a diagram showing a modified example of the inductance conversion circuit;

FIG. 32 is a circuit diagram showing an example of a conventional sine wave oscillator,

FIG. 33 is a circuit diagram showing an example of a conventional sine wave oscillator.

[Explanation of symbols]

 1 Oscillator 10, 30 Phase shift circuit 12, 32 Differential amplifier 16, 36 Variable resistor 17, 37 Inductor 18, 20, 38, 40 Resistor 19, 39 Capacitor 50 Non-inverting circuit 70 Feedback resistor 92 Output terminal

Claims (25)

[Claims] 1. A first series circuit composed of a first resistor and a second resistor having substantially the same resistance value applied to both ends of an input AC signal, and a third series circuit to which the AC signal is applied to both ends. Second series circuit formed by the resistor and the inductor, the potential at the connection point of the first and second resistors forming the first series circuit, and the third series forming the second series circuit. A phase difference circuit including a differential amplifier that amplifies the difference between the resistance of the inductor and the potential at the connection point of the inductor with a predetermined amplification degree and outputs the amplified difference, the latter stage of the two phase shift circuits connected in cascade. The oscillator is characterized in that the output of the above is fed back to the input side of the previous stage, and the sine wave oscillation output is taken out from either one of the two phase shift circuits. 2. The oscillator according to claim 1, wherein two-phase outputs are taken out from the two phase shift circuits. 3. A first series circuit composed of first and second resistors having substantially the same resistance value applied to both ends of an input AC signal, and a third series circuit to which the AC signal is applied to both ends. Second series circuit formed by the resistor and the inductor, the potential at the connection point of the first and second resistors forming the first series circuit, and the third series forming the second series circuit. And a phase difference circuit including a differential amplifier that amplifies the difference between the resistance of the inductor and the potential at the connection point of the inductor with a predetermined amplification degree, and outputs the same without changing the phase of the input AC signal. A non-inverting circuit, and each of the two phase shift circuits and the non-inverting circuit are cascade-connected, and the output of the final stage of the plurality of cascade-connected circuits is fed back to the input side of the first stage. , Positive from any of these multiple circuits Oscillator, characterized in that retrieving the wave oscillation output. 4. The oscillator according to claim 3, wherein two-phase outputs are taken out from the two phase shift circuits and the non-inverting circuit. 5. The method according to claim 1, wherein the connection between the third resistor and the inductor forming the second series circuit is reversed in the two phase shift circuits. Characteristic oscillator. 6. A first series circuit composed of a first resistor and a second resistor having substantially equal resistance values applied to both ends of an input AC signal, and a third series circuit to which the AC signal is applied to both ends. Second series circuit formed by the resistor and the inductor, the potential at the connection point of the first and second resistors forming the first series circuit, and the third series forming the second series circuit. Of the phase difference of the input AC signal is output after inverting the phase of the input AC signal. A phase inversion circuit, and each of the two phase shift circuits and the phase inversion circuit are cascaded, and the output of the final stage of the plurality of cascaded circuits is fed back to the input side of the first stage. , One of these multiple circuits Oscillator, characterized in that retrieving the sinusoidal oscillation output. 7. The oscillator according to claim 6, wherein a two-phase output is taken out from the two phase shift circuits and the phase inverting circuit. 8. The method according to claim 6 or 7, wherein the third resistor and the inductor forming the second series circuit are connected in the same manner in the two phase shift circuits. Oscillator. 9. The oscillation frequency according to claim 1, wherein the third resistor included in at least one of the two phase shift circuits is formed by a variable resistor and the resistance value is changed. Oscillator characterized by changing. 10. The oscillator according to claim 9, wherein the variable resistance is formed by a channel of an FET, and a channel voltage is changed by changing a gate voltage. 11. The channel according to claim 9, wherein the variable resistor is formed by connecting a p-channel type FET and an n-channel type FET in parallel, and changing the magnitude of the gate voltage of each FET having a different polarity. An oscillator characterized by changing resistance. 12. The method according to claim 1, wherein the inductance of the inductor included in at least one of the two phase shift circuits is changed.
An oscillator characterized by changing an oscillation frequency. 13. The inductor according to claim 12, wherein the inductor is formed in a spiral shape in a substantially planar shape on a substrate, and is formed on the substrate in a substantially concentric shape with the inductor conductor, A inductor for changing a direct current bias current flowing through the control conductor by including a control conductor through which a predetermined direct current bias current flows, and a magnetic body formed so as to cover the inductor conductor and the control conductor. An oscillator characterized by changing the inductance appearing at both ends of a conductor. 14. The inductor according to claim 12, wherein the inductor has an inductor conductor formed in a substantially planar spiral shape on a substrate, and a substantially planar spiral at a position on the substrate adjacent to the inductor conductor. A control conductor which is formed in a shape and through which a predetermined direct current bias current flows, and a magnetic body which is formed in an annular shape so as to penetrate through each spiral center of the inductor conductor and the control conductor, An oscillator characterized in that a DC bias current flowing through a control conductor is changed to change an inductance appearing at both ends of the inductor conductor. 15. The method according to claim 1, wherein the third resistor included in at least one of the two phase shift circuits has a plurality of resistors having a fixed resistance value, An oscillator characterized in that the oscillation frequency is changed by selectively connecting. 16. The inductor according to claim 1, wherein the inductor included in at least one of the two phase shift circuits has a plurality of inductors having a fixed inductance, and is selectively connected by switching a switch. An oscillator characterized in that the oscillation frequency is changed by doing so. 17. The amplifier according to claim 1, wherein the inductor included in at least one of the two phase shift circuits has a gain set between 0 and 1.
An oscillator characterized in that by replacing with an inductor element connected in parallel between the input and output of the amplifier, the inductance seen from the input side of the amplifier is made larger than the actual inductance of the inductor element. 18. The oscillator according to claim 17, wherein the oscillation frequency is changed by changing the gain of the amplifier to change the inductance viewed from the input side of the amplifier. 19. A first non-inverting circuit configured to output an input AC signal in phase, a series connection of two resistors and a series connection of an inductor and a resistor, to which an output of the non-inverting circuit is applied.
And a first differential amplifier for obtaining a difference between two outputs of the first bridge circuit, and a phase shifter for phase-shifting a signal input to the first bridge circuit. A circuit, a second bridge circuit including two resistors connected in series and a resistor and an inductor connected in series, to which the output of the first phase shift circuit is applied, and two outputs of the second bridge circuit. A second differential amplifier for obtaining the difference between
A second phase shift circuit that shifts the signal input to the bridge circuit in the opposite direction to the first phase shift circuit, and the output of the second phase shift circuit is fed back to the input of the non-inverting circuit. An oscillator, comprising: 20. The resistance value of a resistor connected in series with the inductor of the first phase shift circuit, and / or the resistance value of the resistor connected in series with the inductor of the second phase shift circuit according to claim 19. An oscillator characterized by changing an oscillation frequency by changing a resistance value. 21. The oscillator according to claim 19, wherein each resistance is formed by a channel of an FET and the channel resistance is changed. 22. A first phase inversion circuit for inverting an input AC signal and outputting the inverted AC signal, a series connection of two resistors and a series connection of an inductor and a resistor, to which an output of the phase inversion circuit is applied. And a first differential amplifier for obtaining a difference between two outputs of the first bridge circuit, and a phase shifter for phase-shifting a signal input to the first bridge circuit. A circuit, a second bridge circuit to which the output of the first phase shift circuit is applied, and two outputs of the second bridge circuit, which are composed of a series connection of two resistors and a series connection of an inductor and a resistor. A second differential amplifier for obtaining the difference between
Second phase shift circuit that shifts the signal input to the bridge circuit in the same direction as the first phase shift circuit, and outputs the output of the second phase shift circuit to the input of the phase inversion circuit. An oscillator comprising: a circuit. 23. The resistance value of a resistor connected in series with the inductor of the first phase shift circuit and / or the resistance value of a resistor connected in series with the inductor of the second phase shift circuit according to claim 22. An oscillator characterized by varying the oscillation frequency. 24. The oscillator according to claim 22, wherein each resistance is formed by a channel of the FET and the channel resistance is changed. 25. The oscillator according to claim 1, which is formed as a semiconductor integrated circuit.