JPH01188036A - Discrimination feedback type equalizing system - Google Patents

Abstract

PURPOSE:To guarantee the adaptive operation by selecting a difference signal which is opposite in polarity and whose absolute value is nearly equal to the previous values stored in a memory, summing the selected value with the existing value, applying the revision of coefficient with the sum and controlling the adaptive filter. CONSTITUTION:The minimum value of a probability where a present sample and a sample before JT sec preceding are of opposite in the polarity and the absolute value is nearly equal and is a positive value hot being zero according to the characteristic of the eye pattern of a reception signal of a transmission line code including a binary-code system. Thus, a difference signal (=reception signal including residual inter code interference) is stored in a memory 10 corresponding to the symbol waveform to which each sample belongs and when a sample of opposite polarity and equal absolute value is extracted when it is received and added to the existing sample thereby causing the sum to be zero in the absence of inter-code interference and to be the residual inter-code interference itself in other cases. Thus, the residual inter-code interference is detected accurately, then the sum is used as an error signal to guarantee the adaptive operation of the adaptive filter 5.

Description

DETAILED DESCRIPTION OF THE INVENTION [Field of Industrial Application] The present invention relates to a decision feedback equalization method, and more particularly to a decision feedback equalization method used to eliminate intersymbol interference that occurs during waveform transmission.

[Conventional technology]

A decision feedback equalizer is known as a known technique for removing intersymbol interference that occurs during waveform transmission (IEEE TRANSACTIONS ON COMMUNICATIONS).
COMMUNICATI○NS>Volume 32, No. 3, 198
4, pp. 258-266).

A decision feedback equalizer uses an adaptive filter with tap coefficients that are long enough to be affected by intersymbol interference to generate pseudo intersymbol interference corresponding to the received data sequence, thereby changing the waveform of the transmission path. It operates to suppress intersymbol interference received during transmission. At this time, each coefficient of the adaptive filter is successively modified by correlating the residual intersymbol interference with the determination result of the received signal.

When modifying coefficients in a decision feedback equalizer, if the residual intersymbol interference contained in the difference signal obtained by subtracting the pseudo intersymbol interference from the received signal containing intersymbol interference cannot be detected correctly, the adaptive operation will fail. The problem arises that it becomes impossible. For example, when a binary code such as a biphase code is used as a transmission line code, due to the nature of the binary code, there is no section where the received signal level is zero, and it is not possible to extract only the intersymbol interference independently. This will not be possible and the above problem will occur. Next, a conventional technique for solving this problem will be described.

FIG. 5 is a block diagram showing a conventional example of a decision feedback type equalizer. Here, the circuit shown in FIG. 5 is connected to the transmitting side via a transmission path. For simplicity, the explanation will be based on the assumption that baseband transmission is used.

In FIG. 5, a received signal containing intersymbol interference is supplied from a transmission path to an input terminal 1, and is input to a subtracter 2.

The subtracter 2 generates a difference signal obtained by subtracting the pseudo intersymbol interference from the received signal supplied to the input terminal 1 (=received signal including residual intersymbol interference, where [residual intersymbol interference] = [intersymbol interference] - [ Pseudo intersymbol interference]) is obtained, and the determiner 3. It is supplied to a subtracter 6. The result determined by the determiner 3 becomes a binary data series, which is supplied to the output terminal 4, the AGC 7, and the adaptive filter 5. Adaptive filter 5
The output of is supplied to subtractor 2. The closed loop circuit consisting of the subtracter 6 and the AGC 7 operates to accurately extract only the residual intersymbol interference in the difference signal input to the subtracter 6.

This is achieved by the AGC 7 multiplying the signal supplied from the determiner 3 by a certain constant to generate a received signal that does not contain residual intersymbol interference. The received signal generated by the AGC 7 is subtracted from the difference signal, which is the output of the subtracter 2, in a subtracter 6. The output of subtracter 6 is adaptive.
The signal is supplied to the filter 5 and used for updating coefficients. Subtractor 21 determiner 3. The closed loop circuit consisting of the adaptive filter 5 operates to remove intersymbol interference in the received signal applied to the input terminal f. This is achieved by the adaptive filter 5 generating pseudo intersymbol interference.

Therefore, the adaptive filter 5 will be explained in detail. FIG. 6 is a detailed block diagram of the adaptive filter 5 of FIG. Input signals 106 and 107 in FIG. 6 correspond to the binary data series that is the output signal of the determiner 3 and the output signal of the subtractor 6 in FIG. 5, respectively. Further, the output signal 108 in FIG. 6 corresponds to the output signal of the adaptive filter 5 in FIG. Input signal 106 is input to delay element 1001. Multiplier 101o.

101+,..., l0IR-1 and coefficient generators 102o, 102+-,..., 102R-1
is supplied to

Delay element 100□, 1002°... giving a delay of T seconds.
..., 100 N/R-1 are connected in this order, and each can be realized by a flip-flop.

Here, N and R are positive integers, and R is a divisor of N. Further, the data period of the input signal 106 is T seconds. Delay element 100+ (i=1.2......

The outputs of N/R-1) are each sent to a multiplier 101J.

101J+1. ......, l0IJ, tol and coefficient generator 102J, 102J++ +...,
102 J+R-1. However, 9 multipliers 101o, 1011, --, l0IK where j=iXR
+NR (k=o.

1, ..., R-1), respectively, the coefficient generator 1
02o, 1021.・・・・・・＋NR to 102
After the input data of each coefficient, which is the output of , is multiplied, all the multiplication results are input to an adder 103K and added. R adders 103o, 1031,...
, 103R-1 serve as input contacts of the switch 104. The switch 104 is a multi-contact switch with a cycle of T seconds, and has R additions 8103o, 103+.

. . . , the output of 103R-1 is selected and outputted in this order every T/R seconds to form the output signal 108. Output signal 108 is pseudo intersymbol interference generated every T/R seconds. R is called an interpolation constant (interbolation factor), and is usually an integer of 2 or more in order to eliminate intersymbol interference within a required signal band. On the other hand, switch 104
Switch 105, which operates in synchronization with switch 104, has input and output reversed to that of switch 104. That is, the switch 105 functions to sequentially distribute the input signal 107 to R contacts every T/R seconds. Each contact output of the switch 105 is supplied to a coefficient generator present in a signal path input to the contact corresponding to the switch 104 operating synchronously.

Next, the coefficient generation circuit will be explained in detail.

FIG. 7 shows the coefficient generator 102+(1=o) of FIG.

1, . . . , N-1). The input signal 200 of FIG. 7 is the input signal 106 of FIG. 6 or the delay elements 100+, 1002.

......, corresponds to an output signal of 100 N/R-1.

Furthermore, the input signal 201 in FIG. 7 corresponds to the contact output of the switch 105 in FIG. Furthermore, output signal 203 in FIG. 7 corresponds to the output of coefficient generator 1021 in FIG. In FIG. 7, the input signal 200
and 201 are supplied to a multiplier 204, and the multiplication result becomes one input of an adder 205. The output of the adder 205 is fed back through a delay element 206 of T seconds, and updating of the coefficients performed every T seconds changes the correlation value of the input signals 200 and 201 supplied to the multiplier 204 by one sample. This is achieved by adding to the previous coefficient value.Output signal 2
03 is the coefficient.

[Problem that the invention seeks to solve]

As described above, the pseudo intersymbol interference generated by the adaptive filter 5 of FIG. 5, which has been explained with reference to FIGS. 6 and 7, becomes one input of the subtracter 2.

The subtracter 2 generates a difference signal obtained by subtracting the simulated intersymbol interference from the received signal supplied from the input terminal 1 (=received signal including residual intersymbol interference, where [residual intersymbol interference] = [intersymbol interference]) - [pseudo intersymbol interference]) is obtained, and the determiner 3.
It is supplied to a subtracter 6. On the other hand, the closed loop circuit consisting of the AGC 7 and the subtracter 6 operates to accurately remove only the residual intersymbol interference in the difference signal supplied to the subtracter 6. A
The output signal of the determiner 3 supplied to the GC 7 is multiplied by 1 and input to the subtracter 6. Here, γ is a positive number. AGC7
The signal supplied to the subtracter 6 is subtracted from the difference signal supplied to the subtracter 6, and is fed back to the AGC7 as a control signal. The AGC 7 uses the signal fed back from the subtracter 6 to correct γ so that the output of the subtracter 6 is equal to the residual intersymbol interference. That is, the AGC 7 pseudo-generates a signal other than the residual intersymbol interference included in the difference signal.

The output of the subtracter 6 is sent to the adaptive filter 5 as an error.
and used for coefficient update. Here, in order for the adaptive filter 5 to perform an adaptive operation, the residual intersymbol interference must be correctly supplied to the adaptive filter 5. However, since the difference signal that is the output signal of subtractor 2 also contains signals other than residual intersymbol interference,
If it is assumed that the output signal of the subtracter 2 is directly supplied to the adaptive filter 5, the residual intersymbol interference cannot be accurately obtained. Therefore, the adaptive ability of the adaptive filter 5 will be lost. Therefore, conventionally, as shown in FIG. 5, a subtracter 6. A method has been used in which the adaptive operation of the adaptive filter 5 is guaranteed by adding the AGC 7 and subtracting signals other than pseudo residual intersymbol interference from the difference signal that is the output signal of the subtracter 2. In this method, a received signal containing no intersymbol interference is generated by an AGC 7 using a binary data sequence that is a determination result of a received signal, and a subtracter 6 subtracts it from the difference signal. A residual intersymbol interference component is obtained by the AGC 7 and the subtracter 6, and the adaptive operation of the adaptive filter 5 is guaranteed. However, in the conventional control method, A
GC7 is required, and in order to obtain a sufficient degree of intersymbol interference suppression, complex control is required to maintain the received signal, which does not include intersymbol interference, supplied from AGC7 to the subtracter 6 at a desired level, which requires hardware. The disadvantage was that it was large in size.

An object of the present invention is to provide a decision feedback equalization method that is simple and requires small hardware.

[Means for solving problems]

The present invention provides a decision feedback equalization system that removes intersymbol interference that occurs during waveform transmission by using pseudo intersymbol interference generated by an adaptive filter. After subtracting the inter-interference interference to obtain a difference signal, use the determination result of the difference signal to retrieve data already stored in the memory corresponding to the symbol waveform of the difference signal, and then add it to the difference signal. The present invention is characterized in that residual inter-symbol interference is determined, the difference signal is stored in a memory corresponding to the symbol waveform, and the coefficients of the adaptive filter are updated using the residual inter-symbol interference.

[Effect]

Unlike the conventional method of multiplying the output of the determiner by a constant to generate a received signal that does not include residual intersymbol interference and subtracting the difference signal, the present invention focuses on the characteristics of the eye pattern of the received signal and The structure was designed to accurately extract interference between the two. That is, according to the characteristics of the eye pattern of the received signal of the transmission line code including the binary code system, the current sample value and J
The minimum probability that the sample value T seconds ago (J is a positive integer) has opposite polarity and almost the same absolute value takes a certain positive value that is not zero. Therefore, while the difference signal (=received signal containing residual intersymbol interference) is stored in the memory corresponding to the symbol waveform to which each sample value belongs, it is retrieved when a sample value with opposite polarity and equal absolute value is received. By adding it to the current sample value, the phase becomes zero when there is no intersymbol interference, and - otherwise, it becomes residual intersymbol interference itself. Therefore, residual intersymbol interference can be detected accurately.
If the sum is used as an error signal, adaptive operation of the adaptive filter is guaranteed.

〔Example〕

Next, the present invention will be explained in detail with reference to the drawings. FIG. 1 is a block diagram showing one embodiment of the present invention. In the figure, functional blocks given the same reference numbers as in FIG. 5 have the same functions as in FIG. 5. The difference between Fig. 1 and Fig. 5 is that the delay element (M,
T)8. Switch 9. Memory group 101, 102,...
..., io-.

Selector 11. This part consists of an adder 12, and the other configurations are exactly the same as in FIG. 5. Before explaining this circuit, the overall configuration will be briefly described. A received signal input to input terminal 1 is supplied to subtracter 2 . The difference signal obtained by subtracting the pseudo intersymbol interference generated by the adaptive filter 5 in the subtracter 2 (=received signal containing residual intersymbol interference 3) is supplied to the determiner 3 and the delay element 8. . The output of the determiner 3 is supplied to an output terminal 4, a switch selector 11, and an adaptive filter 5. Adaptive filter 5° adder 2. Delay element 8. Switch 9. Memory groups 10+, 102, -
-, 10ff+, selector 11. A closed loop circuit consisting of an adder 12 realizes adaptive operation of the adaptive filter 5, and a switch and a selector 11 select signals to be supplied to memory groups 10+, 10□, . . . , 10m and signals to be taken out. is selected to control the signal supplied to the adder 12. The configuration of the adaptive filter 5 may be the same as the circuit configuration shown in FIGS. 6 and 7, similar to that explained in FIG. 5.

Next, the relationship between the output of the selector 11 and the residual intersymbol interference in the difference signal that is the output of the subtracter 2 will be explained in detail, but before that, the transmission line codes will be described.

Figure 2 is a diagram showing typical examples of binary codes for explaining transmission line codes, in which (a) shows bi-phase codes, and (b) shows MSK (minimum shift keying).
The pulse waveforms of each code are shown.

As shown in Figure 2(a), in the biphase code “0”
Assign pulse waveforms with inverted polarity to the data of °° and “1゛′. Both pulses have a width of 1 bit T
The polarity is reversed at the center of the second, and the positive and negative values are balanced within one bit. On the other hand,
As shown in FIG. 2(b), four types of pulse waveforms are prepared in the MSK code. That is, two types of pulse waveforms, a "0" degree mode and a "1" mode, each having an inverted polarity, are prepared for the data "0" and "1". tr
The O++ mode and the "1" mode indicate that the polarity of the waveform is positive and negative, respectively. These two types of mode transitions are
This is indicated by the arrow in FIG. 2(b), and the current mode is determined by the mode one symbol before. This MSK code has a property that the polarity always inverts at the boundary of the transmitted symbol waveform.

In addition, in the MSK code, the positive and negative balance is maintained within one symbol for II I II, but “0”
′, the positive and negative values are not balanced. However, the second
As is clear from the direction of the arrow indicating the mode transition in Figure (b), if there is an even number of 0°'s in the continuous data series, the positive and negative values are balanced and the DC component can be almost ignored. The transmission path code shown in FIG. 2 is transmitted through the transmission path and is input to the input terminal 1 of FIG. 1 after receiving intersymbol interference.

3 and 4 respectively show an example of a received signal eye pattern and an example of a received signal waveform pattern corresponding to the transmission line code of FIG. 2. FIGS. 3(a) and 3(b) are eye patterns of a pi-phase code and an MSK code, respectively, corresponding to FIG. 2. As shown in the figure, the received signal eye pattern has high frequency components removed and becomes rounded. Originally, a received signal eye pattern includes intersymbol interference components, but for the sake of simplicity, the eye pattern shown is ideal and does not include intersymbol interference.

Now, the received signal eye of the MSK code shown in Fig. 3(b) is
Pay attention to patterns. According to the characteristics of the received signal eye pattern, the probability that the sample value JT seconds (J is a positive integer) before the current sample value has the same polarity and almost the same absolute value is a non-zero positive value. Take. Therefore, by storing sample values every T seconds in memory and adding them to the sample values when a waveform of opposite polarity is received, the received signal can be canceled out. FIG. 4(b) shows how the received signal is canceled in the case of J=2, and the three waveforms are the symbol waveforms of the current, T seconds ago, and 2T seconds ago from the right. In the example shown in FIG. 4(b), the current waveform and the symbol waveform 2T seconds ago have opposite polarities. Therefore, regarding the sample value, it is easy to see that the current sample value and the sample value 2T seconds ago have opposite polarities and the same absolute value.

Therefore, by extracting a sample value from 2T seconds ago from the memory corresponding to the symbol waveform to which the sample value belongs and adding it to the current sample value, the received signal components are canceled and the output becomes zero. It is clear that this is also true for cases other than J=2.

Now, considering a non-ideal case, the received signal includes residual intersymbol interference components. Considering the residual intersymbol interference component, the current residual intersymbol interference value and JT
Since it is uncorrelated with the value of residual intersymbol interference seconds before, JT
The residual intersymbol interference value before the second can be considered as random noise. The amplitude distribution of the residual intersymbol interference values before JT seconds is symmetric between positive and negative, and the amplitude d! d1 × δ (however, 0 × δ)
The probability that , is not zero, but is a certain positive value. Therefore, it can be seen that the probability that the output signal of the adder 12 includes accurate residual intersymbol interference takes a certain positive value that is not zero. Also,
Generally, the magnitude of residual intersymbol interference is sufficiently small relative to the received signal. Therefore, the waveform shown in FIG. 4(b) can be regarded as the received signal waveform even if it is not ideal.

Next, the memory groups 101 and 102 in FIG.

......, 10. The operation of the switch and selector 11 for controlling input/output signals will be explained. switch 9
is a memory group 101, 102 for storing a received sample value corresponding to a symbol waveform to which the received sample value belongs.
, --, 10va. The eye pattern of the MS K code is a superimposition of four types of waveforms as shown in Figure 3, so m = 4, and for example, the memories 10z, 10□, 103, and 104 are each 00°.
', "01", '10', and '11°.Here,
"01'" means data signal "0" and mode signal °1.
′ represents the symbol waveform defined by . The switch 9 uses the data signal and mode signal supplied from the determiner 3 to
These combinations are °“00°’, “01”, “
"10".

The circuit is switched so that the signals supplied from the delay element 8 at the time of "'11" are stored in the memories 10r, 102, 103, and 104, respectively. In addition, in FIG. 1, the determiner 3 and the switch 9. Although the path connecting the selector 11 and the adaptive filter 5 is shown as one line, when the MSK code is adopted, two paths are shown corresponding to the data signal and the mode signal. Since the determiner 3 cannot determine the received symbol waveform until the end of this symbol waveform is received, and the data signal and mode signal are not determined, the signal supplied to the switch 9 is delayed by the delay element 8 for T seconds. delay. That is, M.S.
For K code, M=1. At the same time, the difference signal supplied to the adder 12 is also delayed by the delay element 8 by T seconds.

In the embodiment shown in FIG. 1, assuming that the interpolation constant explained using FIG. 6 is R=4, there are sample values at four types of phases per one symbol waveform. For this reason, the memory 10. , to2°10q, and 104 each consist of four sub-memories, and each sub-memory corresponds to one symbol waveform in one sample phase.

Conversely, since there is only one memory that corresponds to one symbol waveform in one sample phase, the sample values corresponding to the same symbol waveform in the same sample phase are always updated, and the latest values are stored in the memory. This is R
The same applies to the case of ≠4.

The selector 11 selects a memory group 10□, 10 from which data is taken out corresponding to the symbol waveform to which the received sample value belongs.
2. ......, select from 10ffi.

In the case of an MSK code, the data signal and mode signal supplied from the determiner 3 are used to determine whether these are "OQ IT,"
The circuit is switched so that the data stored in the memories 10+ and 102.103.104 are selected and supplied to the adder 12 when the data is at l"01' and "10°Zllllll+, respectively. In this way, the selector 11 selects the data in the memory corresponding to the symbol waveform of opposite polarity to the symbol waveform determined by the determiner 3, so the received signal is canceled by the adder 12, and the residual symbol interval is accurately calculated. Interference can be extracted.

FIG. 1 ■ Pseudo intersymbol interference generated by the adaptive filter 5 is supplied to the subtracter 2.

Subtractor 2 generates a difference signal obtained by subtracting pseudo intersymbol interference from the received signal that is the input signal of input terminal 1 (=received signal including residual intersymbol interference, residual intersymbol interference = intersymbol interference - pseudo intersymbol interference) is obtained, and the determiner 3. The signal is supplied to the delay element 8. The adder 12 receives the output signal of the delay element 8 and the memory groups 101 and 102.

......, 10. The signals selected by the selector 11 from n are added to cancel the received signals, and only the residual intersymbol interference is supplied to the adaptive filter 5.

The result determined by the determiner 3 is supplied to the adaptive filler 5 and appears at the output terminal 7 at the same time. Adaptive filter 5 uses the output signal of adder 12 to update coefficients.

Note that when the MSK code is adopted as explained above, the adaptive filter 5 The configuration is slightly different from that shown in FIG. That is, it is necessary to prepare two types of tap coefficients corresponding to the different pulse waveforms of "0" and "1" and update them separately, and also to differentiate the coefficients based on the mode signal received from the determiner 3. This is necessary.

Up to now, the present invention has been explained in detail using the MSK code as an example. However, as a transmission line code, for example,
) can be used. What differs between biphase codes and MSK codes is the received signal eye pattern.

FIG. 4(a) shows an example of a continuous symbol waveform of a biphase code. The five consecutive waveforms are symbol waveforms of T seconds after the present, the present, T seconds before, 2T seconds before, and 3T seconds before, in order from the right. In the case of bi-phase codes, the symbol waveforms of interest are different depending on the symbol waveforms of the preceding and following symbols, so
Memory groups 101, 102,...
..., select 101. Figure 4(a) is “01
It is easy to see that the current sample value and the sample value 2T seconds ago have opposite polarities and the same absolute value.

Therefore, a sample value every T/R seconds is stored in a memory corresponding to the symbol waveform to which this sample value belongs and the sample phase, while a symbol waveform with opposite polarity and equal absolute value to the symbol waveform to which the current sample value belongs is stored. Pi by adding the corresponding memory value to the current sample value.
In the case of a phase code as well, the received signal components are canceled and the output becomes zero.

However, in the case of the pi-phase code, the input signals to the switch and selector 11 are only data signals. Since it is impossible to know in advance the symbol waveform T seconds from now, the coefficients are updated by waiting until it is determined that the symbol waveform will be T seconds from now.9 Therefore, in the case of a pi-phase code, M-2. Delay element 8 must provide a delay of 2T seconds.

Considering transmission path codes other than these codes in the same way, the memory groups 101, 102, . . . are stored based on the received signal eye pattern corresponding to FIG.

It is clear that the residual intersymbol interference can be extracted accurately by allocating 10□ and using it to update the coefficients of the adaptive filter 5 after canceling the received signal.

〔Effect of the invention〕

As described in detail above, according to the present invention, for the difference signal, two of the past values stored in the memory are selected which have opposite polarity and almost the same absolute value as the current value, and are compared with the current value. By calculating the sum, the residual intersymbol interference component included in the received signal can be accurately extracted. Therefore, by using the above sum to update the coefficients and control the adaptive filter, adaptive operation is guaranteed. Further, according to the present invention,
By combining delay elements, switches, memory groups, selectors, and adders, the above adaptive operation can be guaranteed, so a decision feedback equalization method that is simple and small in hardware size without requiring complicated control can be provided. Effects can be obtained.

[Brief explanation of the drawing]

FIG. 1 is a block diagram showing one embodiment of the present invention, and FIG. 2 (
FIG. 3(a) is a diagram showing a representative example of a binary code for explaining a transmission line code. (b) shows the received signal eye corresponding to the transmission line code in Figure 2.
Figures 4(a) and 4(b) are diagrams showing examples of received signal waveform patterns corresponding to transmission line codes; Figure 5 is a block diagram showing a conventional example of a decision feedback equalizer; 6 is a detailed block diagram of the adaptive filter 5 of FIG. 5, and FIG. 7 is a detailed block diagram of the coefficient generation circuit of FIG. 6. 1... Input terminal, 2... Subtractor, 3... Determiner,
4... Output terminal, 5... Adaptive filter,
8...Delay element (M, T), 9...Switch, 10
...Memory, 11...Selector, 12...Adder. Agent Patent Attorney Uchihara On No. 21M 0″%%//# $ 3 times (a): + l w7--@ l l
((L) (b) $ 7 圀

Claims (1)

[Claims] In a decision feedback equalization method that uses pseudo intersymbol interference generated by an adaptive filter to remove intersymbol interference that occurs during waveform transmission, the pseudo intersymbol interference is subtracted from the received signal that contains the intersymbol interference. After obtaining the difference signal, the determination result of the difference signal is used to extract the data already stored in the memory corresponding to the symbol waveform of the difference signal, and the data is added to the difference signal to eliminate residual intersymbol interference. The decision feedback equalization method is characterized in that the difference signal is calculated, the difference signal is stored in a memory corresponding to the symbol waveform, and the coefficients of the adaptive filter are updated using the residual intersymbol interference.