JP5499595B2 - Control device for permanent magnet type synchronous motor - Google Patents

Description

  The present invention relates to a so-called position sensorless control technique for operating without using a position detector for detecting a magnetic pole position of a rotor in a control device for a permanent magnet type synchronous motor such as an embedded magnet type synchronous motor. is there.

  In order to reduce the cost of the control device for the permanent magnet type synchronous motor, so-called position sensorless control that operates without using a position detector for detecting the magnetic pole position of the rotor has been put into practical use. Position sensorless control realizes torque control and speed control by calculating the magnetic pole position and speed of the rotor from the information on the terminal voltage of the motor and the current of the armature, and performing current control based on these. .

As one of position sensorless control techniques for a permanent magnet type synchronous motor, a conventional technique described in Non-Patent Document 1 is known. This prior art is a method of controlling a motor by calculating a magnetic flux estimated value using a magnetic flux observer and calculating a magnetic pole position and speed from the magnetic flux estimated value.
Hereinafter, a specific method of position sensorless control using the magnetic flux observer according to the prior art will be described.

In the position sensorless control of the permanent magnet type synchronous motor, since the magnetic pole position cannot be directly detected by the rotor coordinate system (d, q axes), the rotation coordinate system (γ, δ axes) is estimated in the control device, and these The current is controlled on the estimated γ and δ axes.
FIG. 8 shows the definitions of the d and q axes and the γ and δ axes. The N-pole direction of the rotor of the permanent magnet synchronous motor is the d axis, the 90 ° advance direction from the d axis is the q axis, The estimated axis corresponding to the d axis is defined as the γ axis, and the direction advanced by 90 ° from the γ axis is defined as the δ axis.
In FIG. 8,
θ _r : Magnetic pole position (u-phase winding reference)
θ ₁ : Estimated value of magnetic pole position (u-phase winding reference)
θ _err : Position estimation error ω _r : Rotor angular velocity ω ₁ : Speed estimation value ω _err : Speed estimation error Further, the position estimation error θ _err and the speed estimation error ω _err are defined by Equations 5 and 6, respectively.

Here, FIG. 9 is a control block diagram for realizing speed control of the permanent magnet type synchronous motor by position sensorless control using a magnetic flux observer, which is described in Non-Patent Document 1.
First, a method for controlling the speed of the permanent magnet synchronous motor using the magnetic pole position estimated value θ ₁ and the speed estimated value ω ₁ calculated by the position / speed estimator 20 will be described.

A deviation between the speed command value ω ^* and the estimated speed value ω ₁ is calculated by the subtractor 14, and the deviation is amplified by the speed regulator 13 to calculate the torque command value τ ^* . The current command calculator 12 calculates γ and δ-axis current command values i _γ ^* and i _δ ^* for outputting a desired torque from the torque command value τ ^* . The phase current detection values i _u and i _w detected by the u-phase current detector 5u and the w-phase current detector 5w are respectively detected by the current coordinate converter 6 using the magnetic pole position estimated value θ ₁ and the γ and δ-axis current detection values. Coordinates are converted to i _γ and i _δ .

The deviation between the γ-axis current command value i _γ ^* and the detected γ-axis current value i _γ is calculated by the subtractor 11a, and this deviation is amplified by the γ-axis current regulator 10a to obtain the γ-axis voltage command value v _γ ^* . Calculate. Similarly, a deviation between the δ-axis current command value i _δ ^* and the δ-axis current detection value i _δ is calculated by the subtractor 11b, and this deviation is amplified by the δ-axis current regulator 10b to be amplified by the δ-axis voltage command value v. Calculate _δ ^* .
The γ and δ-axis voltage command values v _γ ^* and v _δ ^* are converted into phase voltage command values v _u ^* , v _v ^* , and v _w ^* by the voltage coordinate converter 9 using the magnetic pole position estimated value θ _1. .

The rectifier circuit 3 supplies a DC voltage obtained by rectifying the three-phase AC voltage of the three-phase AC power supply 4 to the power converter 2 such as an inverter. The PWM circuit 8 converts the output voltage of the power converter 2 from the phase voltage command values v _u ^* , v _v ^* , v _w ^* and the input voltage detection value E _dc detected by the voltage detection circuit 7 to the phase voltage command value v. _{u ^{*, v} v} ^*, _v to generate the gate signal for controlling the ^{w *.} The power converter 2 controls the internal semiconductor switching element based on the gate signal, thereby converting the terminal voltage of the permanent magnet synchronous motor (PMSM) 1 into the phase voltage command values v _u ^* , v _v ^* , v _w ^*. And the rotational speed of the electric motor 1 is controlled to the speed command value ω ^* .

Next, a method for calculating the magnetic pole position estimated value θ ₁ and the speed estimated value ω ₁ using the magnetic flux observer in the position / speed estimator 20 will be described. FIG. 10 is a block diagram of the position / velocity estimator 20 in FIG. 9. This block diagram is also described in Non-Patent Document 1.
First, the d and q axis voltage equations of the permanent magnet type synchronous motor are expressed by Equation 7 using an extended magnetic flux (a magnetic flux that induces an extended induced voltage). The expanded magnetic flux amplitude ψ _ex in Equation 7 is as shown in Equation 8.

From the relationship shown in FIG. 8, when the voltage equation of d and q axes in Equation 7 is converted into γ and δ axes and a state equation having γ, δ axis current and γ, δ axis magnetic flux as state variables is derived, Equation 9 Become. Note that w ₁ and w ₂ in Expression 9 are expressed by Expression 10 and Expression 11, respectively.

In Equation 9, since the speed estimation value ω ₁ and the actual speed ω _r are equal in the steady state, the speed estimation error ω _err is approximated to zero. Further, the extended magnetic flux amplitude Ψ _ex is regarded as a constant value. At this time, w ₁ and w ₂ in Equation 9 are zero.
Further, the magnetic flux observer is configured by Equation 12 using γ, δ-axis voltage command values v _γ ^* , v _δ ^* instead of γ, δ-axis voltages v _γ , v _δ in Equation 9.

In the actual calculation of the magnetic flux observer 21 in FIG. 10, first, the right side of Equation 12 is calculated, and the result is integrated to obtain the γ and δ-axis current estimated values i _γest and i _δest and the γ and δ-axis magnetic flux estimated values ψ _γest. , Ψ _δest .
The angle error calculator 22 calculates the angle (angle error estimated value) δ _est of the γ and δ axis magnetic flux estimated values Ψ _γest , Ψ _δest viewed from the γ axis, using Expression 13.

The speed estimator 23 of FIG. 10 calculates the speed estimated value ω ₁ by amplifying the angles (−δ _est ) of the γ and δ axis magnetic flux estimated values Ψ _γest and Ψ _{δest according} to Expression 14.

In addition, the magnetic pole position calculator 24 calculates the magnetic pole position estimated value θ ₁ by integrating the speed estimated value ω ₁ as shown in Equation 15.

  On the other hand, in Non-Patent Document 2 and Non-Patent Document 3, instead of calculating the magnetic flux estimation value as in Non-Patent Document 1, the extended induced voltage is calculated. A control method of an embedded magnet type synchronous motor (salient pole type permanent magnet synchronous motor) for calculating the magnetic pole position and speed is shown.

“Sensorless control of permanent magnet synchronous motor using magnetic flux observer”, 2008 IEEJ Industrial Application Division Conference, No. 1-62, I-299-304 "Sensorless control of salient pole type permanent magnet synchronous motor based on extended induced voltage model", IEEJ Transaction D, Vol. 122, No. 12, 2002, 1088-1096 "Position sensorless control of embedded magnet synchronous motor using extended induced voltage", IEEJ Transactions Vol. 125, No. 9, 2005, pp. 833-838

In Non-Patent Document 1, the magnetic flux observer 21 calculates γ and δ-axis magnetic flux estimated values Ψ _γest and Ψ _δest using an arithmetic expression that approximates the speed estimation error ω _err to zero as shown in Expression 12. However, in reality, the speed estimation error ω _err cannot be ignored, and the γ and δ-axis magnetic flux estimated values Ψ _γest and Ψ _δest calculated by Expression 12 are values including the speed estimation error ω _err as a disturbance. As a result, there is a problem that the stability of the control is lowered.
Specifically, the γ and δ-axis magnetic flux estimated values Ψ _γest and Ψ _δest calculated by the magnetic flux observer 21 have a relationship as shown in Expression 16 with the speed estimation error ω _err .

From Equation 16, the γ and δ-axis magnetic flux estimated values Ψ _γest and Ψ _δest are values including the speed estimation error ω _err as disturbance in addition to the information of the position estimation error θ _err that is originally desired to be extracted.
Next, a problem caused by the fact that the speed estimation error ω _err is included in the γ and δ axis magnetic flux estimated values Ψ _γest and Ψ _δest will be described.
From Equation 13 and Equation 16, when the value of the arc tangent of the magnetic flux estimation value is linearly approximated with respect to the position estimation error and the velocity estimation error in the vicinity of the operating point, Equation 17 is obtained at the operating point in the steady state. Equation 18 represents T _dθerr in Equation 17.

FIG. 11 is a block diagram obtained by modeling the transfer characteristic from the actual value θ _r of the magnetic pole position to the estimated value θ ₁ from Expression 17 and Expression 18. In FIG. 11, reference numeral 25 denotes a part in which the magnetic flux observer 21 and the angle error calculator 22 are modeled, and the first-order lag filter represented by 1 / (1 + sσ) models the arithmetic delay of the magnetic flux observer. is there.
When analyzing the pole arrangement of the transfer function from the actual value θ _r of the magnetic pole position to the estimated value θ _{1, if} there is a speed estimation error ω _err (= differential value of the position estimation error θ _err ), the differential coefficient T _dθerr is The larger the value, the more the pole moves toward the right half plane of the s plane, and the stability of control decreases.

From Equation 18 described above, particularly in the case of an embedded magnet type synchronous motor (L _d ≠ L _q ), the speed is low (ω _r is small) and the load is heavy (i _δ is large), and the d-axis inductance L _d And the q-axis inductance L _q are larger, the value of the differential coefficient T _dθerr is larger, and as a result, the control tends to be unstable.

  Also, in the method using the extended induced voltage shown in Non-Patent Document 2 and Non-Patent Document 3, since the calculation that approximates the speed estimation error to zero is performed as in Non-Patent Document 1, the same as above. Problems can occur.

  Accordingly, an object of the present invention is to provide a control device that can operate a permanent magnet type synchronous motor such as an embedded magnet type synchronous motor in a stable manner even at a low speed and under a heavy load.

In order to solve the above-mentioned problem, the invention according to claim 1 is a control device for a permanent magnet type synchronous motor that does not have a magnetic pole position detector.
The N-pole direction of the rotor of the electric motor is the d-axis, the 90 ° advance direction from the d-axis is the q-axis, the first control estimated axis corresponding to the d-axis is the γ ₁ axis, and the γ ₁ axis is 90 ° Take direction [delta] ₁ axis, the second axis estimated the gamma _2-axis for control corresponding to the d-axis, a 90 ° leading direction from the gamma ₂ axes defined respectively [delta] _2-axis,
Taking the voltage and current of the motor as vectors on the γ ₁ , δ ₁ axis and γ ₂ , δ ₂ axis,
Means for controlling the terminal voltage of the electric motor to match the detected current values of the γ ₁ and δ ₁ axes with current command values;
From the control q-axis inductance setting value corresponding to the q-axis inductance of the motor, the detected current values of the γ ₂ and δ ₂ axes, and the voltage command value, the estimated speed value of the motor and the angle of the γ ₂ axis are obtained. Means for computing;
The q-axis inductance setting value, in order to remove the disturbance by the speed estimation error between the actual speed of the speed estimation value and the motor included in the angle of the gamma _2-axis, indicated by formula 1A according to claim 1 Means for calculating the evaluation function f to be zero or near zero;
It means for calculating the formula 1B according to claim 1 the angular difference between the gamma _2-axis and the d-axis generated constantly by the deviation between the q-axis inductance and the q-axis inductance setting value,
Means for correcting the angle of the γ ₁ axis by subtracting the angle difference from the angle of the γ ₂ axis;
It is equipped with.
As a result, the disturbance due to the speed estimation error (= differential value of the position estimation error) that causes the embedded magnet type synchronous motor to become unstable at low speed and heavy load can be made zero or near zero. The electric motor can be stably operated even at a low speed and a heavy load, and the problems in Non-Patent Documents 1 to 3 can be solved. In addition, if the q-axis inductance setting value is set to a value different from the true value, an angle difference is generated between the d-axis and the control shaft that performs current control, resulting in a torque control error. Thus, by correcting the angle of the control axis using the angle difference, current control can be performed on the control axis that matches the d-axis, and stable control can be realized without degrading torque accuracy. become.

According to a second aspect of the present invention, in the first aspect, the means for calculating the q-axis inductance setting value according to the first aspect is based on the δ _2- axis current according to Formula 1 C (refer to the claims). A set value is calculated, and an angular difference between the d axis and the γ ₂ axis generated by a deviation between the q axis inductance and the q axis inductance set value is calculated as the q axis inductance set value and the δ _2. Means for calculating according to Formula 2 (refer to the claims) according to the shaft current, and controlling the currents of the γ ₁ and δ ₁ axes so that the current amplitude of the motor at the same torque is minimized. To do.
Further, in the invention according to claim 3, the means for calculating the q-axis inductance setting value in claim 1 is configured to calculate the q-axis inductance setting value according to the current of the γ ₁ , δ _1- axis according to Equation 3 thereby calculated by reference to the range), the angular difference between the d-axis and the gamma _2-axis generated by the deviation between the q-axis inductance setting value and the q-axis inductance, the said q-axis inductance setting value Means for calculating according to Equation 4 (refer to the claims) according to the current of δ ₂ axis, and the current of the γ ₁ and δ ₁ axis are set to minimize the current amplitude of the motor at the same torque. To control.

  According to the invention according to claim 2 or claim 3, under the maximum torque control, the disturbance due to the speed estimation error can always be zero regardless of the motor constant such as d and q axis inductance, the speed condition, and the load condition. Therefore, in particular, the embedded magnet type synchronous motor can be operated stably, and the problems in Non-Patent Documents 1 to 3 can be solved.

  According to the present invention, a permanent magnet type synchronous motor such as an embedded magnet type synchronous motor can be stably operated even at a low speed and a heavy load regardless of the motor constant.

It is a figure which shows the definition of the coordinate axis in embodiment of this invention. It is a block diagram which shows the position sensorless control method in Example 1 of this invention. FIG. 3 is a block diagram of a position / speed estimator in FIG. 2. It is the block diagram which modeled the transfer characteristic of the magnetic pole position in Example 1 of this invention. It is the block diagram which deform | transformed a part of FIG. It is a block diagram which shows the position sensorless control method in Example 2 of this invention. It is a block diagram which shows the position sensorless control method in Example 3 of this invention. It is a figure which shows the definition of d, q axis | shaft and (gamma), (delta) axis. It is a block diagram which shows the position sensorless control method in a nonpatent literature 1. FIG. 10 is a block diagram of the position / velocity estimator in FIG. 9. It is the block diagram which modeled the transfer characteristic of the magnetic pole position in nonpatent literature 1.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings. This embodiment relates to a control device that calculates a magnetic flux estimated value by a magnetic flux observer in a position / velocity estimator as in Non-Patent Document 1, and estimates the magnetic pole position and speed from this magnetic flux estimated value. It is.

First, FIG. 1 shows the definition of coordinate axes in this embodiment. As shown in the figure, the N-pole direction of the rotor of the permanent magnet synchronous motor is the d-axis, the 90 ° advance direction from this d-axis is the q-axis, and the first control estimated axis corresponding to the d-axis is the γ ₁ axis. the gamma 90 ° advances the direction [delta] ₁ axis from _one axis, the second estimated-axis gamma _2-axis for control corresponding to the d-axis, a 90 ° leading direction from the gamma _2-axis is defined as [delta] ₂ axis.
In FIG. 1,
θ _r : magnetic pole position (u-phase winding reference),
θ _γδ1 : Angle of γ ₁ axis (u- _{phase winding} reference),
_θ γδ2: γ _2-axis angle (u phase winding reference),
θ _err : Angular difference between the d-axis and the γ ₂ axis (the value in the steady state is θ _err0 ),
ω _r : angular velocity of the rotor,
ω ₁ : Speed estimation value,
ω _err : Speed estimation error.
Further, the position estimation error θ _err and the speed estimation error ω _err are defined by Expression 19 and Expression 20, respectively.

  Next, FIG. 2 is a control block diagram for realizing speed control of a permanent magnet type synchronous motor by position sensorless control using a magnetic flux observer as Example 1 according to claim 1. Hereinafter, the first embodiment will be described focusing on the differences between FIG. 2 and FIG. 9.

That is, in the first embodiment shown in FIG. 2, the voltage coordinate converter 30 and the current coordinate converter 31 are provided in addition to the voltage coordinate converter 9 and the current coordinate converter 6, and the voltage used in the position / speed estimator 40, Γ ₂ , δ _2- axis voltage command values v _γ2 ^* , v _δ2 ^* and current obtained by converting the coordinates of the current by the voltage coordinate converter 30 and the current coordinate converter 31 using the γ _2- axis angle θ _γδ2 , respectively. The detected values i _γ2 and i _δ2 are used.
In FIG. 2, current detection values obtained by coordinate conversion by the current coordinate converter 6 using the angle θ _γδ1 of the γ ₁ axis are i _γ1 , i _δ1 , current command values for these are i _γ1 ^* , i _δ1 ^* , and the above The voltage command values converted by the voltage coordinate converter 9 using the angle θ _γδ1 are denoted as v _γ1 ^* and v _δ1 ^* .

In the first embodiment, the q-axis inductance setting device 32 is provided, and the output q-axis inductance setting value L _q ′ is used in place of the true value L _q in the magnetic flux observer in the position / speed estimator 40. ing.
Furthermore, the angular difference setter 33 and the subtractor 34 is provided, the angle difference setting unit 33, occurs between the d-axis and the gamma _2-axis by a deviation between the q-axis inductance L _q and q-axis inductance setting value L _{q '} it sets the angle difference theta _ERR0 to, the subtractor 34 subtracts the angular difference theta _ERR0 from the angle theta _Ganmaderuta2 of gamma ₂ axes calculates the angle theta _Ganmaderuta1 of gamma ₁ axis voltage coordinate the angle theta _Ganmaderuta1 The current is controlled by sending it to the converter 9 and the current coordinate converter 6.

FIG. 3 shows the configuration of the position / velocity estimator 40 in FIG. 2, and specific calculation contents in the position / velocity estimator 40 will be described below.
In the flux observer 41, q-axis inductance setting value _{L q} 'and gamma _2, [delta] _2-axis voltage, using a current, gamma _2, [delta] _2-axis current estimated value _{i γ2est,} i δ2est and flux estimation value [psi _Ganma2est, Ψ _δ2est is calculated by Equation 21.

In the actual calculation of the magnetic flux observer 41, first, the right side of Equation 21 is calculated, and this result is integrated to obtain the current estimated values i _γ2est and i _{δ2est of} the γ ₂ and δ ₂ axes and the estimated magnetic flux values Ψ _γ2est and Ψ _δ2est . Ask.
The angle error calculator 42 calculates the angle δ _est of the magnetic flux estimated values Ψ _γ2est and Ψ _δ2est of the γ ₂ and δ ₂ axes as viewed from the γ ₂ axis, using Expression 22.

The speed estimator 43 amplifies the angle δ _est as shown in Expression 23 to calculate the speed estimated value ω ₁ , and the magnetic pole position calculator 44 integrates the speed estimated value ω ₁ as shown in Expression 24 to obtain γ _The biaxial angle θ _γδ2 is calculated.

Next, a specific operation of the q-axis inductance setting unit 32 in FIG. 2 will be described.
First, to explain the principles, as set by the q-axis inductance setting value L _{q 'to} a value different from the true value L _q, modeling the transfer characteristic from the actual value theta _r of the magnetic pole position to estimate theta _Ganmaderuta2 A block diagram is derived.
If you set the q-axis inductance setting value _{L q} 'to a value different from the true value _{L q,} the flux estimate _Ψ γ2est, Ψ δ2est is represented by formula 25.

From Equation 25, the estimated magnetic flux values Ψ _γ2est and Ψ _δ2est are nonlinear functions of the position estimation error θ _err , the speed estimation error ω _err, and the speed estimation value ω ₁ .
Therefore, linearly approximating the value of arc tangent of the magnetic flux estimated values Ψ _γ2est and Ψ _δ2est obtained by Expression 22 and Expression 25 in the vicinity of the operating point with respect to the position estimation error θ _err , the speed estimation error ω _err, and the speed estimation value ω ₁ , Expressions 26 to 30 are obtained at operating points in the steady state.

A control block diagram modeling transfer characteristics from the actual value θ _r of the magnetic pole position to the estimated value θ _γδ2 from Expression 26 to Expression 30 is as shown in FIG.
Here, if FIG. 4 is modified using the functions of Equations 19 and 20, FIG. 5 is obtained. 4 and 5, 45 and 45 a are portions where the magnetic flux observer 41 and the angle error calculator 42 in FIG. 3 are modeled.
From FIG. 5, the q-axis inductance setting value L _{q 'coefficients} of the differential at the time of setting the value different true value L _q becomes the form of B + C, its value is q-axis inductance setting value L _q' using equations 31.

  Based on Expression 31, the evaluation function f is defined by Expression 32.

Here, since the δ _2- axis magnetic flux in Formula 25 is controlled to be zero in the steady state, the relationship shown in Formula 33 exists between the q-axis inductance setting value L _q ′ and the angle difference θ _err0 .

By selecting the q-axis inductance setting value L _q ′ so that the value of the evaluation function f is zero or close to zero from the equations 32 and 33, it is included in the angle of the γ ₂ axis that causes the control to become unstable. The disturbance due to the speed estimation error (= differential value of the position estimation error) can be removed.
As a specific configuration method of the q-axis inductance setting unit 32, a q-axis inductance setting value L that makes the value of the evaluation function f zero or near zero according to the conditions of γ ₂ and δ _two- axis currents i _γ2 and i _{δ2. What} is necessary is just to calculate _q ′ in advance and set a table value of this in the q-axis inductance setting device 32.
For simplicity, the q-axis inductance set value L _q ′ may be a fixed value. The value of L _{q 'in} this case, gamma ₂ of the heavy _load, [delta] _2-axis current i _.gamma.2, by calculating the condition of i _.delta.2, it is possible to improve the stability of the heavy load, which is a problem.

Next, a specific operation of the angle difference setting unit 33 in FIG. 2 will be described.
The deviation between the q-axis inductance L _q and q-axis inductance setting value L _{q ',} stationary angular difference between the d-axis and the gamma _2-axis is generated. If current control is performed on a control axis that has an angular difference with respect to the d-axis, a torque control error occurs. In this embodiment, the control axis that performs current control is corrected.

Specifically, since there is the relationship of the above-described equation 33 between the q-axis inductance setting value L _q ′ and the angle difference θ _err0 , it corresponds to the above L _q ′ (a _tabulated value or a fixed value). The angle difference θ _err0 is calculated in advance, and a table value or a fixed value is set by the angle setting unit 33. Then, from the angle theta _Ganmaderuta2 of gamma _2-axis by subtracting the angle difference theta _ERR0 calculates the angle theta _Ganmaderuta1 of gamma _1-axis, gamma _1, the current control is performed on the [delta] ₁ axis.
As a result, current control on the control axis coinciding with the d-axis becomes possible, and stable control can be performed without deteriorating torque accuracy.

Next, a second embodiment of the present invention according to claim 2 will be described. FIG. 6 is a control block diagram according to the second embodiment.
The difference between the second embodiment and the first embodiment is that, in particular, the current command calculator 12 sets γ ₁ and δ _1- axis current command values i _γ1 ^* and i _δ1 ^* to the current amplitude of the permanent magnet type synchronous motor 1 at the same torque. The q-axis inductance setter 35 calculates the q-axis inductance set value L _q ′ online according to the δ _2- axis current i _δ2 , and the angle difference setter 36 calculates the angular difference. This is a point where θ _err0 is calculated online according to the δ _2- axis current i _δ2 and the q-axis inductance set value L _q ′.

Next, specific calculation contents of the q-axis inductance setting unit 35 and the angle difference setting unit 36 in the second embodiment will be described.
When current control is performed so that the current amplitude of the permanent magnet type synchronous motor 1 at the same torque is minimized, the d and q axis currents have a relationship as shown in Expression 34.

From the relationship shown in FIG. 8, when the d and q axis currents of Expression 34 are expressed using γ ₂ and δ ₂ axis currents, Expression 35 is obtained.

When the q-axis inductance setting value L _q ′ in Expression 32 is deleted using Expression 33 and Expression 35, the evaluation function f becomes Expression 36.

From Equation 36, if the γ _2- axis current is controlled to zero, the evaluation function f can be made zero.
From Expression 33, the angle difference θ _err0 for making the γ _2- axis current zero becomes Expression 37. Formula 37 is the same as Formula 2 in Claim 2 and Formula 4 in Claim 3.

Substituting Equation 37 into Equation 32 to obtain the q-axis inductance setting value L _q ′ for making the evaluation function f zero, Equation 38 is obtained. This mathematical formula 38 is the same as the mathematical formula 1 C in claim 2.

In the q-axis inductance setting unit 35 of FIG. 6, L _q ′ is calculated online according to the δ _2- axis current i _{δ2 using} Equation 38. Further, the angle difference setting unit 36 _calculates θ _err0 online according to the δ _2- axis current i _δ2 and the q-axis inductance setting value L _q ′ according to Expression 37.
Thereby, it is possible to always stably operate the embedded magnet type synchronous motor in particular under the maximum torque control regardless of the motor constant and the load condition.

Next, a third embodiment of the present invention according to claim 3 will be described. FIG. 7 is a control block diagram according to the third embodiment.
The third embodiment is different from the first embodiment, in particular, that the current command calculator 12 changes the current amplitude of the permanent magnet type synchronous motor 1 to γ ₁ and δ _1- axis current command values i _γ1 ^* and i _δ1 ^* at the same torque. Is calculated such that the q-axis inductance setter 37 calculates the q-axis inductance set value L _q ′ online according to γ ₁ and δ _1- axis currents i _γ1 and i _δ1 , and the angle difference setting. In this case, the angle difference θ _err0 is calculated on-line in accordance with the δ _2- axis current i _δ2 and the d-axis inductance set value L _q ′. That is, the second embodiment is the same as the second embodiment except for the operation of the q-axis inductance setting unit 37.

Here, specific calculation contents of the q-axis inductance setting unit 37 will be described.
Similar to the second embodiment, when the γ _2- axis current is controlled to be zero in order to make the evaluation function f zero, there is a relationship of Formula 39 between the δ _2- axis current and the γ ₁ , δ _1- axis current. .

From Equations 38 and 39, the q-axis inductance setting value L _q ′ for making the evaluation function f zero can also be calculated by Equation 40. Formula 40 is the same as Formula 3 in claim 3.

In the q-axis inductance setting device 37 of FIG. 7, L _q ′ is calculated online according to γ ₁ and δ _1- axis currents i _γ1 and i _δ1 according to Equation 40. Thereby, it is possible to always stably operate the embedded magnet type synchronous motor in particular under the maximum torque control regardless of the motor constant and the load condition.
In FIG. 7, γ ₁ and δ _1- axis current detection values are used as input quantities of the q-axis inductance setting device 37, but γ ₁ and δ _1- axis currents are used instead of γ ₁ and δ _1- axis current detection values. A command value may be used.

1: Permanent magnet synchronous motor 2: Power converter 3: Rectifier circuit 4: Three-phase AC power supply 5u: u-phase current detector 5w: w-phase current detector 6, 31: Current coordinate converter 7: Voltage detection circuit 8 : PWM circuit 9, 30: Voltage coordinate converter 10a: γ-axis current regulator 10b: δ-axis current regulator 11a, 11b, 14, 34: Subtractor 12: Current command calculator 13: Speed regulator 20, 40: Position / speed estimator 32, 35, 37: q-axis inductance setter 33, 36: Angle difference setter 41: Magnetic flux observer 42: Angle error calculator 43: Speed estimator 44: Magnetic pole position calculator 45, 45a: Magnetic flux The modeled part of the observer and angle error calculator

Claims (3)

In a control device for a permanent magnet type synchronous motor having no magnetic pole position detector,
The N-pole direction of the rotor of the electric motor is the d-axis, the 90 ° advance direction from the d-axis is the q-axis, the first control estimated axis corresponding to the d-axis is the γ ₁ axis, and the γ ₁ axis is 90 ° Take direction [delta] ₁ axis, the second axis estimated the gamma _2-axis for control corresponding to the d-axis, a 90 ° leading direction from the gamma ₂ axes defined respectively [delta] _2-axis,
Taking the voltage and current of the motor as vectors on the γ ₁ , δ ₁ axis and γ ₂ , δ ₂ axis,
Means for controlling the terminal voltage of the electric motor to match the detected current values of the γ ₁ and δ ₁ axes with current command values;
From the control q-axis inductance setting value corresponding to the q-axis inductance of the motor, the detected current values of the γ ₂ and δ ₂ axes, and the voltage command value, the estimated speed value of the motor and the angle of the γ ₂ axis are obtained. Means for calculating;
In order to eliminate the disturbance due to the speed estimation error between the speed estimated value included in the angle of the γ ₂ axis and the actual speed of the motor, the q-axis inductance setting value is evaluated by the following formula 1A. Means for calculating the function f to be zero or near zero;
It means for calculating by the following formula 1B the angular difference between the gamma _2-axis and the d-axis generated constantly by the deviation between the q-axis inductance and the q-axis inductance setting value,
Means for correcting the angle of the γ ₁ axis by subtracting the angle difference from the angle of the γ ₂ axis;
A control device for a permanent magnet type synchronous motor.

L _q : q-axis inductance, L _q ′: q-axis inductance setting value,
L _d : d-axis inductance, i _γ2 : γ _2- axis current detection value, i _δ2 : δ _2- axis current detection value,
theta _ERR0: angular difference which regularly occur between the d-axis and the gamma _2-axis by a deviation between the _q-axis inductance and q-axis inductance setting value,
Ψ _ex0 : Expanded magnetic flux amplitude in steady state

Ψ _m : permanent magnet magnetic flux In the control device for the permanent magnet type synchronous motor according to claim 1,
It means for computing the q-axis inductance setting value, the computing the q-axis inductance setting value according to Equation 1 C below in accordance with the current of the [delta] ₂ axis,
An angle difference between the d-axis and the γ ₂ axis generated by a deviation between the q-axis inductance and the q-axis inductance setting value is determined according to the q-axis inductance setting value and the current of the δ ₂ axis. Means for calculating by the following formula 2;
A control apparatus for a permanent magnet type synchronous motor, wherein the currents of the γ ₁ and δ ₁ axes are controlled so that the current amplitude of the motor at the same torque is minimized.

In the control device for the permanent magnet type synchronous motor according to claim 1,
The means for calculating the q-axis inductance setting value calculates the q-axis inductance setting value according to the following Equation 3 according to the currents of the γ ₁ and δ _1- axis,
An angle difference between the d-axis and the γ ₂ axis generated by a deviation between the q-axis inductance and the q-axis inductance setting value is determined according to the q-axis inductance setting value and the current of the δ ₂ axis. Means for calculating according to the following equation 4;
A control apparatus for a permanent magnet type synchronous motor, wherein the currents of the γ ₁ and δ ₁ axes are controlled so that the current amplitude of the motor at the same torque is minimized.

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