JP5047021B2 - Electric motor drive device and air conditioner - Google Patents

Description

  The present invention relates to an electric motor driving device that drives an electric motor at a variable speed and an air conditioner using the electric motor driving device.

  A conventional motor drive device rectifies an AC power supply, smoothes the rectified DC power with a smoothing capacitor, and supplies power to the motor with an inverter. In such a configuration, since a smoothing capacitor is always required, this smoothing capacitor has been a factor in increasing the size and cost. However, it is known that the elimination of the smoothing capacitor causes the rectified DC voltage to pulsate in synchronism with the AC power supply, and adversely affect the motor, such as torque pulsation and efficiency deterioration.

  Therefore, there is a technique for advancing the phase of the motor in order to reduce the adverse effect on the motor due to pulsation when this smoothing capacitor is not required (see, for example, Patent Document 1).

  Some motors are controlled in advance so that the torque pulsates at a cycle twice that of the power source (see, for example, Patent Document 2).

  Furthermore, in the case of a three-phase AC power supply, since the pulsation of the DC voltage is smaller than that of the single-phase AC power supply, some DC voltage pulsations are compensated by instantaneously detecting the DC voltage (see, for example, Patent Document 3). .

  Further, a pulsation caused by resonance between a small-capacitance capacitor having a reduced smoothing capacitor and an inductance such as a power source impedance is compensated by controlling an inverter and an electric motor (for example, see Patent Document 4), and a second load and a switching circuit are provided. Some are provided and suppressed (see, for example, Patent Document 5).

Japanese Patent Laid-Open No. 10-150795 (page 5-7, FIG. 1) JP 2002-51589 A JP-A-6-153534 (FIG. 2) JP 2007-181358 A JP 2007-288971 A

  When the capacity of the smoothing capacitor is reduced, a film capacitor without an electrolytic solution can be used as the smoothing capacitor, and failure due to leakage of electrolytic solution in a typical electrolytic capacitor of a large capacity capacitor can be prevented. On the other hand, since the capacitance of the capacitor is reduced, a number of applications have been filed as conventional techniques for solving the adverse effect on the motor caused by the pulsation of the DC voltage that is the voltage across the capacitor. [0009] As shown in [0009], there is a problem that is opposite to the suppression of the resonance phenomenon with the power source impedance and the like.

The present invention has been made to solve the above-described problems. The first object is to drive an electric motor with a small-capacitance capacitor, and even with this small-capacitance capacitor, a current caused by resonance with a power source impedance or the like. An electric motor drive device and an air conditioner that can suppress pulsation are obtained.
The second object is to obtain an electric motor drive device and an air conditioner that prevent an excessive DC voltage from being generated at both ends of a small-capacitance capacitor when the electric motor is stopped.

Electric motor drive device according to the present invention includes: a rectifier for rectifying an AC voltage from an AC power supply, a smoothing capacitor to produce a DC voltage output of the rectifier smoothing to converts the DC voltage of the smoothing capacitor into an AC voltage Power conversion means applied to the motor, a phase current detector for detecting the phase current of the motor, and a control means for controlling the AC voltage applied from the power conversion means to the motor, the control means detects the phase current A coordinate conversion unit that converts the motor current detected by the generator into a biaxial current, calculates the sum of squares of the biaxial current, and performs a Fourier transform on the calculated square sum of the biaxial currents at the phase angle of the AC power supply Then, the binary DC amount is extracted, the binary DC amount is integrated and added, and the fluctuation component of the sum of squares of the biaxial current is used as the frequency compensation amount together with the frequency component of the power supply resonance extracted from the DC current. Add to motor rotation speed command value And a frequency compensation unit for outputting Te, based on the output of the frequency compensator, so that the current effective value flowing through the motor becomes constant, to drive the motor by modulating the frequency outputted from the power conversion means It is a thing.

  Further, the control means is configured to stop the electric motor after setting the rotational speed of the motor to a predetermined rotational speed or less when stopping the motor.

  According to the present invention, since the motor is driven by modulating the frequency output from the power conversion means so that the effective current value flowing through the motor is constant, a small-capacitance capacitor is connected between the output terminals of the rectifier. Even if it is provided, it is possible to suppress current pulsation due to resonance with the power source impedance, etc., which makes it possible to control the input current to a current waveform on a complete rectangular wave, greatly increasing the fifth harmonic. Can be reduced.

  In addition, when stopping the motor, the motor was stopped after the motor rotation speed was reduced to a predetermined rotation speed or less, so even if a small-capacitance capacitor was provided between the output terminals of the rectifier, an excessive amount would occur at both ends. Can significantly reduce the voltage.

Embodiment 1 FIG.
FIG. 1 is a circuit block diagram showing an electric motor drive apparatus according to Embodiment 1 of the present invention.
In FIG. 1, the rectifier 2 rectifies when an AC voltage from the AC power source 1 is applied, the reactor 3 smoothes the current rectified by the rectifier 2, and the smoothing capacitor 4 smoothes the voltage rectified by the rectifier 2. To do. The inverter main circuit 5 is a power conversion unit that converts the DC voltage smoothed by the smoothing capacitor 4 into an AC based on the control of the control unit 10 described later, and applies the DC voltage to the motor 5 for driving. The phase current detectors 7 a and 7 b detect the phase current flowing through the electric motor 5, and the DC current detector 8 detects the output current of the rectifier 2.

  The control means 10 described above is based on the coordinate conversion unit 11 that converts the motor current detected by the phase current detectors 7 a and 7 b into a biaxial current, and the inverter frequency command from the biaxial current and the frequency compensation unit 13. An inverter frequency command is generated from the output voltage calculation unit 12 that calculates the output voltage, the biaxial current from the coordinate conversion unit 11, the DC current detected by the DC current detector 8, and the rotational speed command ω * of the motor 5. A PWM signal is generated based on the frequency compensation unit 13, the integrator 14 that generates the phase angle in the coordinate conversion unit 11 from the inverter frequency command of the frequency compensation unit 13, and the output voltage calculated in the output voltage calculation unit 12. The PWM generator 15 is configured.

  In this embodiment, the capacity of the smoothing capacitor 4 is significantly smaller than the capacity required for driving the conventional inverter (hereinafter referred to as “conventional capacity”), and is reduced or extremely reduced (hereinafter simply referred to as “reducing the capacity”). "). The behavior of the DC voltage, which is the voltage across the smoothing capacitor 4, is determined by the capacity of the smoothing capacitor 4. The operation waveform is shown in FIG. The waveform of the DC voltage at the conventional capacity is as shown in FIG. 2A. In particular, when the AC power supply 1 is a three-phase AC, the AC application with a phase difference of 120 degrees is obtained, as shown in FIG. Therefore, the output voltage of the electric motor 5 is controlled to a substantially constant value.

  On the other hand, FIG. 2B shows a waveform comparing the behavior of the DC voltage when the capacity of the smoothing capacitor 4 is reduced. Since the DC voltage pulsates due to the reduction in the capacity of the smoothing capacitor 4 and pulsates at 6 times the frequency of the AC power supply 1, when the motor 5 is controlled similarly to the DC voltage shown in FIG. The output torque pulsates due to the pulsation. Therefore, as shown in Patent Documents 3 and 5, correction is made so as not to be affected by the pulsation of the DC voltage.

  The present embodiment is different from the conventional capacity used for conventional motor control in that the capacity to such an extent that the DC voltage pulsates in synchronization with the power supply voltage is set as the capacity of the smoothing capacitor 4. . Here, the conventional capacity is about 2000 to 7000 uF, and the capacity for reducing the capacity is 10 to 50 uF. In addition, although it describes as 1/100 or less in patent document 2, the conventional capacity | capacitance changes according to the capacity | capacitance of the electric motor 5, The value described as the above-mentioned 2000-7000uF and 10-50uF, or the conventional It is added that even if it deviates from 1/100 of the capacity, the effect of the present invention is not impaired.

Further, if the conventional capacity is C, the inductance of the motor 5 is L, the rated current is I, and the allowable DC voltage pulsation amount (variation in FIG. 2A) is V, the energy conservation law is established. 1 / 2CV ² = 1 / 2LI ² , and an approximate value can be calculated from this equation. For example, when an electric motor 5 having an inductance L = 0.5 mH and a rated current I = 30 A is driven at an allowable pulsation voltage = 10 V (FIG. 2A), the capacitor capacity C = 4500 uF, but the allowable pulsation voltage is It is clear from the estimated value that when the voltage increases to 150 V (FIG. 2B), the capacity of the smoothing capacitor 4 can be reduced by allowing the capacitor capacity C = 20 uF and the increase of DC voltage pulsation.

  As for the calculation of this approximate value, the smoothing capacitor 4 is charged by the charge supplied from the AC power supply 1, so that the above calculation method does not represent a complete physical phenomenon, and the allowable pulsation voltage is only used. It should be added that the calculation is a rough calculation showing the relationship between the capacitance of the smoothing capacitor 4.

  First, the case where the capacity | capacitance of the smoothing capacitor 4 is reduced is demonstrated using FIG. The virtual load 9 shown in FIG. 3 simulates the electric motor 5 and the inverter main circuit 6, and the same reference numerals are given to the same parts as those in FIG.

  The current iz consumed by the virtual load 9 varies depending on the operation of the electric motor 5, and the current iz is supplied from the charge stored in the smoothing capacitor 4 and the charge supplied from the AC power supply 1. Therefore, the current is flows from the AC power supply 1 when the DC voltage of the smoothing capacitor 4 becomes equal to or less than the instantaneous value of the AC power supply 1. As shown in FIG. 2 (a), the DC voltage rises only when the current is flows from the AC power supply 1. However, if the conventional capacitor is a smoothing capacitor, the DC voltage is input from the AC power supply 1 before the DC voltage is significantly reduced. The current is flows and becomes a substantially constant DC voltage.

  Therefore, when the input current is flowing from the AC power source 1 flows, the rectified current ib flows to the output side of the rectifier 2. This current ib coincides with the current flowing through the reactor 3. The current ib flows to the virtual load 9 and the smoothing capacitor 4, and the amount of the current ib flowing to the smoothing capacitor 4 charges the smoothing capacitor 4 and the DC voltage rises. When the AC power supply 1 is a three-phase AC, charging is generated 6 times as many times as the power supply cycle when viewed on the output side of the rectifier 2, so in the case of the conventional capacity, as shown in FIG. A phase current flows, and ib ≠ iz.

  On the other hand, in the case of the small-capacity smoothing capacitor 4, electric charges are consumed by the current iz in the virtual load 9, and the DC voltage is lowered. It is generally known that the electric charge Q stored in the smoothing capacitor 4 is expressed by Q = CV from the capacitor capacitance C and the voltage V across the capacitor. The electric charge Q consumed by the virtual load 9 depends on the load state of the electric motor 5 and does not depend on the capacitor capacity C. Therefore, when the smoothing capacitor 4 has a small capacity, the consumed charge Q is not changed. Therefore, the voltage V across the capacitor increases as the capacitor capacity C is smaller, and the drop in the DC voltage becomes larger than that in the conventional capacity. .

  When the smoothing capacitor 4 cannot hold the charge Q consumed by the virtual load 9, the current iz consumed by the virtual load 9 and the current ib flowing through the reactor 3 coincide with each other. Similar waveform between output terminals. Between the output terminals of the rectifier 2, a superimposed waveform of three-phase AC half-wave rectification as shown by the dotted line in FIG. 2B is output, so that the waveform shown by the solid line in FIG. Only the peak voltage appears.

  Here, even if the smoothing capacitor 4 is not charged / discharged, if it is held as a DC voltage having a superimposed waveform of half-wave rectification from the AC power supply 1 and is input to the inverter main circuit 6, For driving, there is no problem other than the pulsation of the DC voltage. Furthermore, if the current iz flowing through the virtual load 9 coincides with the current ib flowing through the reactor 3, it is synonymous with controlling the current ib flowing through the reactor 3 by controlling the electric motor 5 by the inverter main circuit 6, in other words, By controlling the electric motor 5, the input current is flowing from the AC power source 1 is controlled.

  Therefore, when the current iz flowing through the virtual load 9 is set to a constant value and the current ib flowing through the reactor 3 is also constant, the input current is has a current waveform as shown in FIG. Although a 120-degree section current flows and the rectangular section has a non-flow section of 60-degree section, the fifth harmonic current can be reduced from the current shown in FIG. In balanced three-phase systems, harmonic currents that are integral multiples of 3 do not theoretically occur, so the order of the minimum harmonic is fifth, and the harmonic current of the fifth component is generally a problem.

  In the present embodiment, the smoothing capacitor 4 is a small-capacitance capacitor, and the purpose is to suppress the fifth harmonic current by making the input current is a rectangular wave shape by using the current flowing through the motor 5. . Further, the state of so-called iz = ib where there is no charging / discharging current to the smoothing capacitor 4 means that the power supply resonance with the power supply impedance by the smoothing capacitor 4 does not occur. Pulsation suppression can also be realized.

The current iz flowing through the virtual load 9 shown in FIG. 3 is equal to the current input to the inverter main circuit 6 in FIG. Therefore, the relationship between the current iz input to the inverter main circuit 6 and the phase current flowing through the motor 5 will be described.
The DC voltage is Vdc, the voltages applied to the motor 5 are Vu, Vv, and Vw, and the phase currents that flow through the motor 5 are Iu, Iv, and Iw. Assuming that the efficiency of the inverter main circuit 6 is η, the input power Pin, the output power Pout, and the efficiency η of the inverter main circuit 6 have a relationship as shown in Expression 1.
Pout = η × Pin 1

In addition, since the input / output power of the inverter main circuit 6 is a relationship between the inner products of the voltage and the current, the relationship is as shown in Equation 2.
Pout = Vu · Iu + Vv · Iv + Vw · Iw
Pin = Vdc · iz 2

  In general, the voltage applied to the motor 5 and the phase current are controlled so as to be in a three-phase equilibrium. When the three-phase equilibrium condition is established, the equations 1 and 2 are expressed as the following equation 3. The

Therefore, the input current iz to the inverter main circuit 6 is related to the phase current of the electric motor 5 by the equation (4).
iz ² ∝Iu ² + Iv ² + Iw ² ... 4

  Therefore, if the motor 5 is controlled so that the square root of the sum of squares of the phase currents flowing through the motor 5, that is, the effective current value of the motor 5, becomes constant, the current iz input to the inverter main circuit 6 becomes a constant value. .

  The input current iz to the inverter main circuit 6 is a combination of the current flowing from the smoothing capacitor 4 and the current ib flowing through the reactor 3, and the current is controlled so that the current flowing through the smoothing capacitor 4 = 0, that is, iz = ib. For example, LC resonance with the smoothing capacitor 4, the reactor 3, and the power supply impedance does not occur. This means that the current does not flow through the smoothing capacitor 4 means that there is no smoothing capacitor 4, that is, the smoothing capacitor 4 is 0 uF, and that it has the ultimate small capacity.

Therefore, details of the control will be described with reference to FIG.
In order to control the electric motor 5 from the phase current detectors 7 a and 7 b that detect the phase current flowing through the electric motor 5, rotation coordinate axis conversion is performed by the coordinate conversion unit 11. Assuming that the biaxial currents converted by the coordinate conversion unit 11 are iγ and iδ, the output voltage calculation unit 12 acts to calculate the output voltages Vγ and Vδ applied to the motor 5 using iγ and iδ. Here, there is no problem even if the output voltage calculation unit 12 is a general vector control or a configuration in which current feedback is provided for V / f control.

  Vγ and Vδ output from the output voltage calculation unit 12 are generated by the PWM generation unit 15 into PWM signals for operating the inverter main circuit 6. The rotation angle phase at this time is preferably configured to have the same angle as that of the coordinate conversion unit 11, but there is no problem even if the delay due to the control calculation is taken into consideration. The inverter control is constructed such that the PWM signal is input to the inverter main circuit 6 and the electric motor 5 is driven.

  The rotation angle phase is output by the integrator 14, and the rotation angle phase is generated by integrating the output of the frequency compensation unit 13 input to the integrator 14. Since the coordinate conversion unit 11 and the PWM generation unit 15 operate according to the rotation angle phase, the speed of the electric motor 5 can be instantaneously changed by the rotation phase angle or the output of the frequency compensation unit 13. The point of the present invention is to instantaneously control the speed of the electric motor 5.

Next, the operation of the frequency compensation unit 13 will be described. The biaxial currents iγ and iδ converted by the coordinate conversion unit 11 and the rotational speed command value ω * and the DC current i _dc detected by the DC current detector 8 are input to the frequency compensation unit 13. The frequency compensation unit 13 uses these manipulated variables to control the phase current of the electric motor 5 and the current ib flowing through the reactor 3 to be constant values.

FIG. 5 shows an example of the frequency compensation unit 13. As described above, the biaxial currents iγ and iδ, the rotational speed command value ω *, and the direct current i _dc are input to FIG. Since the biaxial currents iγ and iδ and the phase current of the electric motor 5 have a relationship as shown in the following formula 5, the biaxial currents iγ and iδ are added to the square sum calculation unit 20 that calculates the square sum of the biaxial currents. Enter.

iγ ² + iδ ² = Iu ² + Iv ² + Iw ² ... 5

Next, the motor 5 is controlled so that (iγ ² + iδ ² ) is constant. At this time, as described above, since the current is controlled to be constant by instantaneously controlling the speed of the electric motor 5, the electric current is passed through the high-pass filter 21 for removing the DC component. Thereby, only the fluctuation component of the sum of squares of the biaxial current can be extracted. The gain multiplication unit 22 multiplies the extracted square sum of the biaxial currents by the gain k ₀ to obtain the speed compensation amount of the motor.

  When the capacitor capacity is small, the DC voltage pulsates at a frequency six times that of the AC power supply 1. Due to this pulsation, the energy input to the electric motor 5 also pulsates. In general, since the speed-controlled inverter is controlled so that the speed of the electric motor 5 becomes constant, all pulsation components of energy affect the output torque of the electric motor 5. This is because the output energy of the electric motor 5 is a product of speed and torque. Since the output torque of the electric motor 5 pulsates, the effective value of the phase current of the electric motor 5 pulsates.

  Therefore, when the current of the electric motor 5 is increased by the speed compensation amount, the speed is also increased, and the pulsation of energy input to the electric motor 5 is received at the speed of the electric motor 5, and the pulsation is not transmitted to the torque. The pulsation of the phase current of the electric motor 5 is suppressed.

  Furthermore, even if only the phase current of the electric motor 5 is made constant, the current ib flowing through the reactor 3 is not made constant. The input / output current to the capacitor 4 leads to power supply resonance with the power supply impedance.

Therefore, as shown in FIG. 5, the detected direct current i _dc is passed through a band pass filter 23 defining a predetermined band, and the component of the power supply resonance contained in the direct current i _dc is extracted. The power supply resonance frequency with the power supply impedance can be calculated with a small-capacity smoothing capacitor 4, the reactor 3, and a power supply impedance not shown. However, the power supply impedance varies depending on the environment in which this product is installed. By using the band-pass filter 23 having the above, problems relating to the installation environment can be solved.

For passing only the frequency component power resonance increases the band-pass filter 23, is multiplied by a gain k ₁ by the gain multiplication unit 24 to the output, which is also added to the speed command as a speed compensation amount. As a result, no power supply resonance occurs, and the input / output current to the smoothing capacitor 4 becomes zero. Therefore, the current ib flowing through the reactor 3 is also constant, and the input current has a rectangular waveform.

  As described above, the output is not corrected according to the pulsation of the voltage, but the effective value of the phase current flowing through the electric motor 5 and the current ib flowing through the reactor 3 are made constant so that the same action as the action of suppressing the power supply resonance is achieved. Control of the electric motor 5 corresponding to a small-capacitance capacitor becomes possible, and the electric motor 5 can be driven stably.

Next, another example of the frequency compensation unit 13 will be described with reference to FIG. FIG. 6 is a control block diagram showing another example of the frequency compensation unit in the electric motor drive device of the first exemplary embodiment.
The frequency compensator 13 shown in FIG. 6 has the same input / output except that the phase angle θ of the AC power supply 1 is required in addition to the input shown in FIG. 5, and the high-pass filter 21 shown in FIG. , Instead of extracting the fluctuation component of the square sum of i δ, the fluctuation component is extracted by the Fourier transform unit 26 and the integrator 27.

  The fluctuation of the square sum of the biaxial currents iγ and iδ is mainly composed of a pulsation of a frequency component that is six times that of the AC power supply 1 appearing in the DC voltage. Therefore, suppressing the fluctuation of the square sum of the biaxial currents iγ and iδ, that is, suppressing the pulsation of the frequency component that is six times that of the AC power supply 1, becomes a substantially constant value.

Therefore, a phase angle θ having a frequency six times that of the AC power supply 1 is generated from the sum of squares of the biaxial currents iγ and iδ, and Fourier transformation is performed at the phase angle θ. In the Fourier transform, it is known that a binary DC amount of (iγ ² + iδ ² ) × sin θ and (iγ ² + iδ ² ) × cos θ becomes a frequency component at θ. Therefore, a DC amount is extracted from these binary values by a low-pass filter and output. This output value is multiplied by a gain k ₂ by a gain multiplier 29 and integrated by an integrator 27 to obtain output values Gsin and Gcos. Thereby, it is possible to detect a stable DC amount with little steady error.

  Next, in the conversion unit 28, the output values Gsin and Gcos of the integrator 27 are again multiplied by sin and sin and cos, respectively, as Gsin × sin θ and Gcos × cos θ, and add these two values. The added value, Gsin × sin θ + Gcos × cos θ, is the frequency compensation amount in FIG. This frequency compensation amount is a compensation amount obtained by specifying a frequency component that is six times that of the AC power source 1 and extracting a fluctuation component. The motor is not affected by a DC voltage that pulsates with a frequency component that is six times that of the AC power source 1. 5 can be driven with a constant phase current.

  Even in the frequency compensation unit 13 as shown in FIG. 6, the electric motor 5 can be stably driven even under a DC voltage pulsation by the small-capacity smoothing capacitor 4, and the power source resonance current can be suppressed at the same time. it can.

Furthermore, another example of the frequency compensation unit 13 will be described with reference to FIG. FIG. 7 is a control block diagram showing another example of the frequency compensation unit in the electric motor drive device of the first exemplary embodiment.
5 and 6, the pulsating component is extracted, and the speed compensation amount is calculated so as to control the speed so as to cancel it. However, in FIG. 7, the phase current effective value of the motor 5 and the reactor 3 flow. The direct current i _dc is directly controlled so as to be in a proportional relationship with a constant value.

The detected DC current i _dc is multiplied by the gain k ₃ by the gain multiplier 30 to match the DC level with the effective value of the phase current. The gain k ₃ may be stored for each number of rotations of the electric motor 5, may be calculated from the mutual DC amount, and can be configured in any way. If the DC amount levels match, the difference between the effective value of the phase current and the DC current i _dc becomes a pulsating component to be removed. Therefore, the high-pass filter 31 is used to remove even a minute amount of DC. And the gain compensation unit 32 multiplies the gain k ₄ by the gain k ₄ to obtain the speed compensation amount.

As described above, according to the present embodiment, if the pulsation component included in the effective value of the phase current of the electric motor 5 and the pulsation component included in the direct current i _dc are removed, the effective value of the phase current ∝ the inverter main circuit 6 From the relationship of the input current iz to iz, iz = ib, and the input current (= current ib flowing through the reactor 3) can be controlled to be constant. As a result, the input current can be controlled to a complete rectangular wave-like current waveform while controlling the electric motor 5, and the fifth harmonic can be greatly reduced.

  Further, the input / output current to the smoothing capacitor 4 is suppressed, and the LC resonance between the AC power supply 1 and the smoothing capacitor 4 is reduced. As a result, it is not necessary to configure a control in which the DC voltage is stabilized and acts reversely, and the drive control of the electric motor 5 can be realized. Further, the control system can be configured by simple v / f control or simple vector control instead of advanced vector control having a current controller.

  Furthermore, by making the phase current of the motor 5 and the input current ib flowing through the reactor 3 constant, the vibration suppression due to load-side fluctuations and the vibration suppression due to power supply resonance do not simultaneously and counteract, realizable.

  Furthermore, when the stator of the motor 5 is a concentrated winding in which the magnetic flux is easily distorted, the cogging torque due to the concentrated winding is likely to be generated as a pulsation 6 times the frequency of the AC power supply 1, but the phase current of the motor 5 is controlled at a constant current. Thus, the phase current ripple of the electric motor 5 can be suppressed, and the electromagnetic noise generated from the electric motor 5 due to the small capacity capacitor can be suppressed to the same level as the conventional capacity.

  In addition, the control configuration of the present embodiment can be changed to a film capacitor that does not use an electrolytic solution instead of an electrolytic capacitor that uses an electrolytic solution that has been used in the past. There is no need to worry about the decrease, and a highly reliable motor drive device can be provided for a long period of time. Further, it is possible to reduce the size of the product by reducing the size of the smoothing capacitor 4.

  Furthermore, even when the electric motor 5 is mounted on a compressor of an air conditioner or the like, even in an application where inertia of inertia is small with respect to the capacity of the electric motor 5, stable driving is achieved because the current is controlled to be constant. Control can be realized.

Embodiment 2. FIG.
FIG. 8 is a circuit block diagram showing an electric motor drive device according to Embodiment 2 of the present invention. In addition, the same code | symbol is attached | subjected to the same or equivalent part as Embodiment 1 demonstrated in FIG. 1, and description is abbreviate | omitted.
The electric motor drive device shown in FIG. 8 is provided with charge consuming means between the input terminals of the inverter main circuit 6. This charge consuming means comprises a switching element 40 and a braking resistor 41 connected in series between input terminals. The switching element 40 is a switch for suppressing a voltage increase due to excessive charging of the smoothing capacitor 4, and the braking resistor 41 is a resistor for limiting a current flowing through the switching element 40.

  The control means 10 includes a stop command unit 42 for generating a stop command for stopping the operation of the electric motor 5 and the inverter main circuit 6 based on an off command from the outside, and a rotation speed based on the stop command from the stop command unit 42. A deceleration unit 43 that lowers the command and decelerates, and a stop signal generation unit 44 that blocks the PWM signal of the inverter main circuit 6 based on the stop command from the stop command unit 42 and outputs an ON signal of the switching element 40. Is provided.

  If the motor 5 is stopped while the motor 5 is rotating at a high speed, the diode of the inverter main circuit 6 is equivalent to the charge stored in the inductance component of the motor 5, in other words, the phase current flowing through the motor 5. Is regenerated to the smoothing capacitor 4. If the smoothing capacitor has a conventional capacity, the voltage does not increase greatly. However, if the smoothing capacitor 4 has a small capacity, the voltage across the both ends up significantly.

  For example, the amount of charge Q that is regenerated is 3 mC. If the smoothing capacitor has a conventional capacity, for example, 3000 uF, the increasing voltage V becomes 1 V because V = Q / C, and hardly increases. On the other hand, when the smoothing capacitor 4 has a small capacity of 10 uF, it can be seen that the voltage is 300 V, which is a large voltage increase.

  Therefore, when the motor 5 is stopped by an off signal from the outside, the motor 5 is not stopped instantaneously, but the stop command unit 42 instructs the deceleration unit 43 to decelerate regardless of the rotation speed command, The speed reduction unit 43 operates so that the rotation speed of the electric motor 5 is reduced. Then, the stop signal generation unit 44 stops the PWM signal output from the inverter main circuit 6 when the rotation speed of the electric motor 5 becomes equal to or lower than the predetermined rotation speed, and stops driving of the electric motor 5.

  As described above, since the electric motor 5 is decelerated and then stopped, an excessive voltage generated at both ends of the smoothing capacitor 4 can be significantly suppressed even if the smoothing capacitor 4 has a small capacity.

  In addition to stopping the electric motor 5 described above, the electric motor 5 may have to be stopped immediately in order to protect the inverter main circuit 6 when an abnormal current flows. In this case, the stop command unit 42 notifies the stop signal generation unit 44 of an abnormal stop when an off signal due to an abnormal stop is input. At this time, the stop signal generation unit 44 operates so as to turn on all the switching elements on the upper side or the lower side of the inverter main circuit 6 and turn off all the opposite sides. As a result, the current flowing in the motor 5 circulates in the motor 5 via the inverter main circuit 6, so that the rise of the voltage across the smoothing capacitor 4 is suppressed without being regenerated to the smoothing capacitor 4. Can solve problems such as damage.

  Moreover, the electric charge which raises the both-ends voltage of the smoothing capacitor 4 is stored not only in the electric motor 5, but between the reactor 3, the alternating current power supply 1, and the rectifier 2, and the power supply impedance which is not described. In order to consume the energy stored in the reactor 3 and the inductance of the power supply impedance, the stop signal generator 44 operates to turn on the switching element 40. When the switching element 40 is turned on, the smoothing capacitor 4 is short-circuited via the braking resistor 41, so that charge is consumed by the braking resistor 41. Then, when the predetermined DC voltage is reached, the stop command unit 42 commands the stop signal generation unit 44 to turn off the switching element 40.

  A flowchart of this operation is shown in FIG. With such an operation configuration, even a small-capacity smoothing capacitor 4 can suppress an increase in the voltage across the smoothing capacitor 4, and can be suppressed below the maximum rated voltage of the inverter main circuit 6.

  Further, when all of the electric charge for increasing the voltage at the braking resistor 41 is consumed, the loss at the braking resistor 41 increases and the resistance increases. Therefore, as shown in FIG. 10, the switching element 50 and the diode 51 are connected in parallel to the reactor 3 in the reverse direction so that the energy stored in the reactor 3 is consumed using the resistance component of the reactor 3. You may do it. When configured in this way, the energy consumed by the braking resistor 41 is only the energy of the power supply impedance (not shown), and therefore the size of the braking resistor 41 can be suppressed.

  Next, as described in the first embodiment, it will be described that the voltage increase at the time of abnormal stop can be suppressed by making the input current constant.

  When there is no resonance with the reactor 3 or the power supply impedance, the rectangular wave current is ideally as shown in FIG. 4B, but when the input current is pulsating due to the power supply resonance or the like, FIG. The waveform is as shown. A difference occurs in the amount of energy flowing into the smoothing capacitor 4 between the case of abnormal stop at the time of (a) and the case of abnormal stop at the time of (b) shown in FIG. Since the input current is pulsating, it is impossible to delay the abnormal stop and delay the stop until the current pulsation decreases. When the abnormal stop occurs, it is stopped immediately even at the time of (a). There is a need.

  The reason for the difference in the amount of energy flowing into the capacitor 4 is that, as shown in FIG. 11, a series of LCs only with the instantaneous current value flowing through the reactor 3 and the power source impedance at the time of abnormal stop as the initial value. This can be explained by considering the transient response of the circuit. When the initial value of the current flowing through L is large, the energy of L is large, the energy stored in the capacitor C is also large according to the law of conservation of energy, and the voltage across the capacitor 4 increases as the capacitance is small. Therefore, the current flowing through the braking resistor 41 and the switching elements 40 and 50 becomes smaller when the instantaneous current at which the abnormal stop is smaller, and the current rating of the switching elements 40 and 50 and the allowable loss of the braking resistor 41 can be reduced.

  In this embodiment, since the input current id is controlled to be constant regardless of pulsation such as power supply resonance, the amount of energy related to the increase in the capacitor voltage at the time of abnormal stop is calculated from the expected value of the input current. I can guess. Thereby, commercialization can be realized without using the meaninglessly large braking resistance 41 and the switching elements 40 and 50 having a large current capacity.

  As described above, since the input current is controlled to be constant, the maximum rated current and permissible loss of the added parts can be reduced in order to suppress the DC voltage increase during an abnormal stop that stops the motor 5 quickly. The cost can be reduced and the size of the parts can be reduced.

  Further, the pulsation of the input current or the current flowing through the reactor 3 is not limited to the power supply resonance described above. The inverter main circuit 6 has a switching state called a zero voltage vector. The zero voltage vector is a switching state in which all the upper switching elements of the inverter main circuit 6 are turned on or all the lower switching elements are turned on. At the time of this zero voltage vector, the current flowing in the motor 5 circulates in the motor 5 as described in the suppression of the voltage increase described above. Therefore, the current iz = 0 input to the inverter main circuit 6 is obtained.

  This zero voltage vector depends on the PWM switching frequency of the inverter main circuit 6, and is in a higher frequency range than, for example, power resonance such as 4 to 10 kHz. On the other hand, the reactor 3 and the power source impedance (not shown) have no path for current to circulate instantaneously at the time of zero voltage vector. This means that, except for the zero voltage vector, even if control is performed so that iz = ib, ib = 0 cannot be achieved when iz = 0 instantaneously.

  Therefore, the input current, in other words, the current flowing through the reactor 3 pulsates even at the PWM switching frequency or twice the switching frequency. This method of suppressing current pulsation by the zero voltage vector can be solved by using the switching element 40 and the braking resistor 41 described above.

  This is because there is no place of current at the time of the zero voltage vector, and the current flows into the smoothing capacitor 4. Therefore, if the switching element 40 is turned on / off only during the zero voltage vector period, the current flowing in the reactor 3 is increased. Since the flow continues through the braking resistor 41 and the switching element 40, the pulsation can be suppressed by suppressing the inflow to the smoothing capacitor 4. Also, for securing the current path at the time of the zero voltage vector, the same effect as described above can be obtained even if the path of the switching element 50 and the diode 51 is used.

However, if the switching element 40 is turned on during the entire period of the zero voltage vector, the charge may be consumed excessively. Therefore, according to the pulsation of the DC current i _dc detected by the DC current detector 8, the switching element 40 Configure to turn on and off.

  Even during the time of the short-circuit prevention time that prevents the upper and lower switching elements from being turned on at the same time, the same operation as that at the time of the zero voltage vector occurs, so the operation permission period of the switching element 40 is expanded until the short-circuit prevention time period. There is no problem.

  Furthermore, when the electric motor 5 is mounted on the compressor of the air conditioner, the electric motor 5 is only in the power running operation and there is no regenerative operation, so that the switching element 40 and the switching element 50 do not need to operate during the steady operation. Therefore, it is sufficient to consider only the current capacity at the time of an abnormal stop, and when applied to a model without regenerative operation such as an air conditioner, it can be realized by a small switching element 40, a braking resistor 41, or the like.

  Examples of utilization of the present invention include an air conditioner equipped with an inverter that drives a compressor, an electric water heater and a dehumidifier, a refrigerator and a freezer, a showcase, a vacuum cleaner and a washing machine, a washing dryer, a hand dryer, and a fan And equipment equipped with fan motors such as ventilation fans.

It is a circuit block diagram which shows the electric motor drive device which concerns on Embodiment 1 of this invention. FIG. 3 is a waveform diagram of a DC voltage for explaining the first embodiment. FIG. 5 is an equivalent circuit diagram for explaining the operation of the inverter main circuit in the electric motor drive device of the first embodiment. FIG. 3 is a waveform diagram of an input current for explaining the first embodiment. FIG. 3 is a control block diagram illustrating a frequency compensation unit in the electric motor drive device of the first embodiment. FIG. 6 is a control block diagram illustrating another example of the frequency compensation unit in the electric motor drive device according to the first embodiment. FIG. 6 is a control block diagram illustrating another example of the frequency compensation unit in the electric motor drive device according to the first embodiment. It is a circuit block diagram which shows the electric motor drive device which concerns on Embodiment 2 of this invention. 10 is a flowchart showing the operation of the second embodiment. FIG. 10 is a circuit block diagram illustrating another example of the electric motor drive device according to the second embodiment. FIG. 10 is a waveform diagram of an input current in the second embodiment.

Explanation of symbols

1 AC power source, 2 rectifier, 3 reactor, 4 capacitor, 5 motor, 6 inverter main circuit, 7a, 7b phase current detector, 8 DC current detector, 10 control means,
11 coordinate conversion unit, 12 output voltage calculation unit, 13 frequency compensation unit, 14 integrator,
15 PWM generator, 41 Braking resistor, 42 Stop command unit, 43 Decelerator, 44 Stop signal generator.

Claims (8)

A rectifier for rectifying an AC voltage from an AC power supply,
A smoothing capacitor for smoothing the voltage rectified by the rectifier;
Power conversion means for converting a DC voltage of the smoothing capacitor into an AC voltage and applying it to the motor;
A phase current detector for detecting the phase current of the motor;
Control means for controlling the AC voltage applied to the electric motor from the power conversion means,
The control means includes
A coordinate conversion unit for converting the electric current of the motor detected by the phase current detector into a biaxial current;
The sum of squares of the two-axis current is calculated, and the calculated sum of squares of the two-axis current is Fourier-transformed at the phase angle of the AC power source to extract a binary DC amount, and the binary DC amount is integrated. And a frequency compensation unit for adding and outputting to the motor rotation speed command value together with the frequency component of the power supply resonance extracted from the DC current, using the fluctuation component of the sum of squares of the biaxial current as the frequency compensation amount. And
An electric motor driving apparatus for driving an electric motor by modulating a frequency output from the power conversion means so that an effective current value flowing through the electric motor becomes constant based on an output of the frequency compensator . A rectifier for rectifying an AC voltage from an AC power supply,
A smoothing capacitor for smoothing the voltage rectified by the rectifier;
Power conversion means for converting a DC voltage of the smoothing capacitor into an AC voltage and applying it to the motor;
A phase current detector for detecting the phase current of the motor;
Control means for controlling the AC voltage applied to the electric motor from the power conversion means,
The control means includes
A coordinate conversion unit for converting the electric current of the motor detected by the phase current detector into a biaxial current;
The sum of squares of the two-axis currents is calculated, the difference between the calculated sum of the squares of the two-axis currents and the direct current is calculated, the fluctuation component is extracted, and this is used as the speed compensation amount to the rotation speed command value of the motor. A frequency compensation unit for adding and outputting,
Based on the output of the frequency compensator, the motor is driven by modulating the frequency output from the power conversion means so that the effective current value flowing through the motor and the current value flowing from the rectifier are constant. An electric motor drive device. 3. The motor drive device according to claim 1, wherein when the motor is stopped, the control unit stops the motor after setting the rotation speed of the motor to a predetermined rotation speed or less. 4. 3. The motor driving device according to claim 1, wherein when the motor is stopped, the motor is stopped after the current flowing through the motor is consumed by the resistance component of the motor. Charge consuming means connected between input terminals of the power conversion means;
Wherein, when stopping the motor, the motor driving apparatus according to any one of claims 2 to 4, characterized in that the pulsation of the current flowing from the rectifier to control the charge dissipation means so as to suppress . A reactor inserted on the anode side of the rectifier;
A switching element connected in parallel to the reactor in a reverse direction,
6. The electric motor drive device according to claim 5 , wherein, when stopping the electric motor, the control means stops the electric motor after turning off the switching element. The motor drive device according to any one of claims 1 to 6 , wherein a film capacitor is connected between output terminals of the rectifier. An air conditioner characterized by comprising any one of the motor drive system of claim 1 to 7.

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