JP4747252B2 - AC-AC direct power converter controller - Google Patents

Description

  The present invention provides an AC-AC direct power converter that directly converts a multi-phase AC voltage into a multi-phase AC voltage having an arbitrary magnitude and frequency using a semiconductor switching element without using a large energy buffer such as a capacitor. In particular, the present invention relates to a control device for an AC-AC direct power converter characterized by output voltage harmonic reduction, switching frequency reduction, input power factor adjustment, and noise suppression.

FIG. 4 is a simplified model of a matrix converter that is an example of an AC-AC direct power converter. In an actual matrix converter, an LC filter for suppressing harmonic current associated with switching is required on the input side. However, in order to facilitate the explanation, this LC filter is ignored in FIG. Simulated by phase current sources i _uv , i _vw , i _wu . Note that S _ru , S _su , S _tu , S _rv , S _sv , S _tv , S _rw , S _sw , and S _tw are AC switches capable of passing current bidirectionally.
Now, assuming that the power source is a symmetrical three-phase power source, the line voltage effective value is E, the power source angular frequency is ω, and the power source phase angle is θ, the phase voltages _er , _es , and _et are given by Equations 1 and 2. .

Here, it is assumed that approximates the power supply voltage _e r in the switching period _2T, e s, _{e t} and the load current _{i uv, i vw, i wu} as constant values, respectively. The power supply voltage is _er >0> _es > _et (0 ≦ θ ≦ π / 6), and the output phase voltage command value is v _u ^* > v _v ^* > v _w ^* (line voltage command value v _uv ^* , V _vw ^* , v _wu ^* ).
The number of commutations during the switching half cycle T is 6 times when all nine AC switches are provided with a conduction period, but the number of commutations is reduced to 4 times and the power factor is controlled to 1. , a conventional technique for controlling the voltage command value as the output voltage as an average value of the switching half-period T, explaining the method of determining the on-duty D _ru to D _tw of nine AC switch S _ru to S _tw below.

As the conventional method 1, there is a method in which the r-phase having the maximum power supply voltage absolute value and the u-phase having the maximum output voltage continue to be conducted over the switching half cycle T. Each on-duty _D ru _{to D tw} at this time is given by Equation 3.

FIG. 17 (a) shows the on-duty (switching pattern) and the output line voltage showing the operation of the conventional method 1. In all the line voltage waveforms, the zero voltage and the maximum line voltage of the power source e are shown. _tr exists.
A switching method corresponding to the conventional method 1 is described in Non-Patent Document 1.

In the conventional system 1, the conventional system 2 described in Non-Patent Document 2 is known as a system that does not use the e _tr that is the maximum line voltage of the power supply when the output voltage command value is low.
The maximum value of the output voltage of the conventional method 2 is ½ of the power supply voltage. In the conventional method 2, the duties D _{A to} D _E for the combinations of the pulse patterns are defined by using the phase voltage effective value E _m = E / √3, and are given by Equation 4, respectively.

FIG. 17B shows an on-duty (switching pattern) and output line voltage showing the operation of the conventional method 2, and does not use the maximum line voltage e _tr of the power supply.

Akio Ishiguro, Takeshi Furuhashi, Muneaki Ishida, Shigeru Okuma, "Output Voltage Control Method of PWM Controlled Cycloconverter Based on Instantaneous Input Line Voltage", IEEJ Transactions D, Vol. 111, 1991, pp. 201-207 Hidenori Hara, Eiji Yamamoto, Mitsuhiko Zenya, Toshikazu Tsuji, Tsuneo Kume, “One Way to Improve Voltage of Matrix Converter in Low Voltage Region”, 2004 IEEJ Conference on Industrial Applications, I-48, pp.313- 316

  In the conventional method 1, the maximum line voltage of the power supply voltage and the zero voltage are used regardless of the magnitude of the output voltage command value, so that the voltage ripple in the switching cycle increases. As the voltage ripple increases, the noise generated by the converter increases, which may cause malfunction of other devices, which is not preferable. Moreover, there is a problem in that a new filter for removing noise causes an increase in cost and an increase in volume.

  According to the conventional method 2, the voltage ripple can be reduced, but the voltage in the range in which PWM control is possible is as low as ½ of the power supply voltage. Therefore, in order to obtain a high output voltage, it is necessary to switch the control. However, when the control is switched, sequence processing increases, which may complicate the arithmetic device. Further, when the control is not switched, the output voltage is out of the range in which PWM control is possible, and the output voltage is distorted. When the electric motor is connected to the load, distortion of the output voltage causes torque pulsation, which is not preferable because it causes an increase in loss of the electric motor or overheating.

  Further, both conventional methods 1 and 2 are control methods when the power source power factor is 1. When the power source power factor 1 control is performed in this way, the current advances from the AC power source by the LC filter on the input side of the converter, and the power source power factor actually advances and becomes the power factor. Due to the increase in current and loss due to the advance current, there is a problem that the input reactor is increased in size and cost.

  The present invention has been made to solve the above-described drawbacks of the prior art, and its object is to reduce output voltage ripple in the switching cycle of an AC-AC direct power converter and reduce generated noise. Another object of the present invention is to provide an inexpensive control device that does not require complicated control switching.

In order to solve the above-described problem, according to the first aspect of the present invention, each phase of the AC power supply and each phase on the output side are directly connected by a bidirectional AC switch, and the AC power supply is turned on and off by the AC switch. In an AC-AC direct power converter that directly converts voltage to AC voltage of any magnitude and frequency,
Means for determining the maximum voltage phase, the intermediate voltage phase and the minimum voltage phase when the power supply voltage of each phase is input;
Means for determining the maximum voltage output phase, the intermediate voltage output phase, and the minimum voltage output phase when the output voltage command value of each phase is input;
Regardless of the magnitude of the output voltage, wherein the maximum voltage phase of the maximum voltage output phase power supply side AC switches and connected to the intermediate voltage phase creates a switching pattern that is not connected to the minimum voltage phase, the minimum voltage means for creating a switching pattern by connecting the AC switch output phase to the intermediate voltage phase and the minimum voltage phase of the power supply side is not connected to the maximum voltage phase, in which with a.

According to a second aspect of the present invention, there is provided an AC-AC direct power converter control device according to the first aspect of the present invention.
Means for detecting a power supply voltage, means for calculating an output voltage command value, and means for calculating an on-duty of the AC switch from the power supply voltage and the output voltage command value.

According to a third aspect of the present invention, there is provided an AC-AC direct power converter control device according to the second aspect of the present invention.
Means for calculating a power supply current command value based on the power supply voltage is provided, and this power supply current command value is used for the calculation of the on-duty.

According to a fourth aspect of the present invention, there is provided an AC-AC direct power converter control device according to the second aspect of the present invention.
Means for detecting a power supply voltage phase angle based on the power supply voltage; and means for calculating a power supply current command value from the power supply voltage phase angle and a power supply power factor command value. It is used for calculation.

According to a fifth aspect of the present invention, there is provided an AC-AC direct power converter control device according to the second aspect of the present invention.
Means for calculating a cosine value and a sine value of a power supply voltage phase angle based on the power supply voltage, and means for calculating a power supply current command value from these cosine value, sine value and power supply power factor command value. A current command value is used for the calculation of the on-duty.

According to a sixth aspect of the present invention, in the control device for an AC-AC direct power converter according to the fourth or fifth aspect,
Means for adjusting the power source power factor command value according to the output current or output voltage of the power converter is provided.

According to the present invention, continuous control without switching from a low voltage to a high voltage is possible, the advance current of the input side LC filter is compensated by adjusting the power factor, and the number of times of switching is reduced. Loss can be reduced, and common mode noise and harmonic current can be suppressed.
This simplifies the configuration of the control device itself, reduces costs by reducing the size of input filters, heat dissipation fins, noise filters, etc., and improves reliability by controlling the temperature of the load device as the load becomes more efficient Can be made.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings. In the drawings referred to in the following description, the same parts as those in the other drawings are denoted by the same reference numerals.

  FIG. 1 is a block diagram showing the configuration of the first embodiment of the present invention, and corresponds to the first and second aspects of the present invention. In the following description, a matrix converter that performs three-phase to three-phase direct conversion is used as an AC-AC direct power converter. In the main circuit of FIG. 1, 10 is a three-phase AC power source, 20 is a matrix converter, 21 is an input filter composed of a reactor L and a capacitor C, and 30 is a load.

The control device 40A according to the first embodiment includes a power supply voltage detection means 41 connected to the three-phase AC power supply 10, an output voltage command calculation means 42, and an ON switch of the matrix converter 20 based on the power supply voltage and the output voltage command value. And an on-duty calculation means 400 for calculating the duty, and the PWM pulses G _ru , G _su , G _tu , G _rv , G _sv , G _tv , G _rw , G _sw , G _tw of each AC switch as a final output. Is output.

In the power supply voltage detecting means 41, for example, three-phase power supply terminals r, s, divided by the resistance or the like star connection each phase power supply voltage input from t, the phase voltages e _r of the power supply _10, e _s, e _t is detected.
FIG. 2 is a block diagram of the output voltage command calculation means 42 in FIG. 1, and is for the case where the induction motor as the load 30 is subjected to variable speed control. 2, the derived primary angular frequency command value corresponding to the speed command value of the motor ω _L ^*, V / by f table 421 a line voltage command value V _L ^* corresponding to the primary angular frequency command value omega _L ^* obtain. Further, the primary angular frequency command value ω _L ^* is input to the integrator 422, and the electrical angle θ _L ^* of the induction motor is obtained by time integration. Based on the line voltage command value V _L ^* and the electrical angle θ _L ^* , the three-phase oscillator 423 gives the output phase voltage command values v _u ^* , v _v ^* , and v _w ^* according to Equation 5.

FIG. 3 is a block diagram showing the configuration of the on-duty calculation means 400 in FIG.
First, as the basic operation of the first embodiment, regardless of the magnitude of the output voltage, the voltage of the maximum output phase is pulse-width modulated from the voltage of the maximum and intermediate phases of the power supply, and the voltage of the minimum output phase is the power supply. For the AC switch of the phase that outputs the maximum voltage, change the switching pattern so that it is connected to the maximum voltage phase and the intermediate voltage phase on the power supply side. The point that the switching pattern is determined so as to be connected to the intermediate voltage phase and the minimum voltage phase on the power supply side for the AC switch of the phase that determines and outputs the minimum voltage will be described.

When controlling the matrix converter 20 of FIG. 4 described above, the output line voltages v _uv , v _vw , v _wu and the power source power factor cos _ψ are controlled according to the specified values, respectively, according to Equation 6 in the switching half cycle T. and determines the nine switches _S ru _{to S tw} on-duty _D ru _{to D tw} and switching time _T tu _{through T tw.}

  As an on-duty condition, Equation 7 must be established from the continuity of the load current of the matrix converter 20.

For example, in the u phase, only one of the switches S _ru , S _su , S _tu must always be turned on.
In considering the switching pattern, the switching frequency is sufficiently higher than the frequency of the power source and the load voltage, and on the premise of the inductive load, the power source voltages _er , _es , _et and the load current i in the switching period 2T. _uv , i _vw , and i _wu are approximated as constant values, respectively. In addition, the power supply voltage is _{e r> e s> e t} , the output voltage command value ^{_{v u *> v v *>}} v w * that described in the original conditions.

FIG. 5 shows the on-duty (switching pattern) and the output line voltage in this embodiment. Respectively the conduction period of each phase switch by on-duty D _ru to D _tw, the half period of the second half and the first half period of which a pattern of folding.
This switching pattern indicates that, for the u-phase AC switch of the maximum output voltage command v _u ^* , the maximum voltage e _r (r phase that is the maximum voltage phase) and the intermediate voltage e _s (s phase that is the intermediate voltage phase). Only for the minimum voltage e _t (t phase which is the minimum voltage phase), and for the w-phase AC switch of the minimum output voltage command value v _w ^* , the intermediate voltage e _{s of the} power source and a PWM pattern not connected to the maximum voltage e _r connected only to the minimum voltage e _t, the use of such a PWM pattern, when a high output voltage of FIG. (a), low in FIG (b) There is no change in the output voltage. That is, regardless of the magnitude of the output voltage, the switching pattern is determined so that the AC switch of the phase that outputs the maximum voltage is connected to the maximum voltage phase and the intermediate voltage phase on the power supply side, and the minimum voltage is output. The switching pattern is determined so that the AC switch of the phase to be connected is connected to the intermediate voltage phase and the minimum voltage phase on the power supply side.
The number of commutations in the switching half cycle T can be reduced to four times as in the conventional methods 1 and 2, since the switches S _tu and S _rw are not conducted in this embodiment (D _tu = 0, D _rw = 0). Yes. In addition, since there is no direct switching between the maximum voltage and the minimum voltage, switching loss and noise can be reduced, and an intermediate voltage is used in all phases, thus suppressing potential changes at the load neutral point. it can.

In the present embodiment, as shown in FIG. 5, the line voltage can be configured with voltage pulses having the same sign as the line voltage command value (for example, for the line voltage v _uv , the command value v _uv ^* the same polarity voltage pulse _e rs of _constructed line voltages _{v uv} by _{e st),} further, the line voltage waveform _{v wu} absolute value of the command value is maximized is shown in the diagram (a) Thus, when the output voltage is high, zero voltage is not output, and when the output voltage is low as shown in FIG. 5B, the maximum voltage pulse e _tr is not output, so that output voltage harmonics are suppressed. be able to.

Next, the on-duty calculation means 400 in FIG. 1 will be described.
FIG. 6 shows a connection relationship in consideration of only the load current i _wu between the u phase having the maximum output voltage command value and the w phase having the minimum value. At this time, as shown in FIG. 5, the u-phase switches S _ru and S _su and the w-phase switches S _sw and S _tw are respectively conducted to generate a pattern of the switching half cycle T.

In FIG. 6, when the load-side impedance viewed from the power source is considered as the resistance R _wu in order to realize the power source power factor 1, the average power source currents i _rwu , i _suu , i _thu in the switching half cycle T are obtained by Equation 8. (Because the symbol “-” cannot be added in the specification text, the symbol “-” is added above the current i and power P in each equation to indicate the average value).

Thus, the average power P _wu of the power source in the switching half cycle T is given by Equation 9.

Since the average P _{wu in} Equation 9 is equal to the power consumption of the load, the current source i _wu between w and u is expressed by Equation 10.

  The u phase, which is the maximum output voltage phase, is connected only to the r phase of the maximum power supply voltage phase and the s phase of the intermediate power supply voltage phase, and is not connected to the t phase of the minimum power supply voltage phase.

Further, r-phase current _{i RWU} Since only _{-i wu} flows when turning on the switch _{S ru,} from Equation 8 and 10, the on-duty _{D ru} is as Equation 12.

From Equation 7, the on-duty D _su is obtained by Equation 13.

The w-phase, the r-phase of the maximum power supply voltage phase is not connected, also, t-phase current _{i tWU,} since the flow is only _{i wu} when turning on the switch _{S tw,} the on-duty _{D rw, D tw,} D _sw is obtained by Expression 14 to Expression 16, respectively.

Next, consider the v-phase switching pattern of the intermediate voltage phase.
FIG. 7 shows a connection relationship in consideration of only the load current i _uv between the u phase having the maximum output voltage command value and the v phase having the intermediate value. In FIG. 7, when the load-side impedance viewed from the power source is considered as the resistance R _uv in order to realize the power source power factor 1, the power source currents i _rub , i _suv , i _tuv are expressed by Expression 17.

Further, since the power source average power P _uv in the switching half cycle T is equal to the load power, the current source i _uv is given by Equation 18.

In FIG. 7, t-phase current _{i tuv,} only when turning on the switch _{S tv,} since flows _{-i uv,} on-duty _{D tv} is obtained by equation 19.

FIG. 8 shows a connection relationship in consideration of only the load current _ivw between the v-phase whose output voltage command value is an intermediate value and the w-phase of the minimum phase. In FIG. 8, when the load side impedance viewed from the power source is considered as the resistance R _vw in order to realize the power source power factor 1, the power source currents i _rvw , i _svw , and i _tvw are expressed by Equation 20.

Further, since the power source average power P _vw in the switching half cycle T is equal to the load power, the current source i _vw is given by Equation 21.

In FIG. 8, r-phase current _{i rvw} Since only _{i vw} flows when turning on the switch _{S rv,} on-duty _{D rv} is obtained by equation 22.

From the equations 7, 19, and 22, the on-duty D _sv is obtained by the equation 23.

In the above, the power supply voltage to derive the _{e r> e s> e t} , the output voltage command value ^{_{v u *> v v *>}} v w * on-duty of each switch in the case of.
The operation of the on-duty calculation means 400 when a switching pattern is generated in an arbitrary magnitude relationship between input and output voltages will be described below.

In on-duty operation means shown in FIG. 3, in order to obtain the information of the supply voltage magnitude relation, the phase / line between transformation blocks 401 of the power supply voltage, power supply voltage e _r, e _s, between the lines from e _t voltage e _rs , E _st , e _tr are obtained by Equation 24.

In the power supply magnitude determination block 402, the power supply mode m _s (1-6) is determined from the sign of the line voltage according to Table 1, and the maximum power supply voltage e ₁ , intermediate voltage e ₂ , and minimum voltage e ₃ are obtained. Here, the subscripts 1, 2, and 3 mean the maximum, middle, and minimum of the power supply voltage, respectively.

The output of the power magnitude discrimination block 402 is a maximum voltage e ₁ , an intermediate voltage e ₂ , a minimum voltage e _3, and 3-bit signals r ₁ , s ₁ , r ₂ s ₃ representing the magnitude relationship. Signal _r 1, _{s 1,} respectively _e r, becomes "1" only when _{e s} is the maximum voltage, the signal _r 2 _{s 3} are _{e r} only when the intermediate voltage or _{e s} is the minimum voltage " This is a signal that becomes 1 ″.

The output voltage phase / line conversion block 403 in FIG. 3 converts the line voltage command values v _uv ^* , v _vw ^* , and v _wu ^* from the output voltage command values v _u ^* , v _v ^* , and v _w ^* to Equation 25. Ask for.

The output level decision block 404, according to Table 2, to determine the output from the sign of the line voltage command value voltage mode _m L (1 to 6), the maximum voltage _v ^a of the output phase voltage command value ^*, the intermediate voltage _v ^{b *} To obtain the minimum voltage v _c ^* . Here, the subscripts a, b, and c mean the maximum, middle, and minimum of the output voltage, respectively.

Further, line voltage command values v _ab ^* , v _bc ^* and v _ca ^* are calculated by Equation 26 by the voltage command phase / line conversion block 405.

In the on-duty calculation block 406 of FIG. 3, the output voltage command is _generated using the maximum voltage e ₁ , the intermediate voltage e ₂ , the minimum voltage e _3, and the line voltage command values v _ab ^* , v _bc ^* , v _ca ^*. On-duty D of the switch in which the maximum voltage phase a phase, the intermediate voltage phase b phase, and the minimum voltage phase c phase of the value are connected to the maximum voltage phase 1 phase, the intermediate voltage phase 2 phase, and the minimum voltage phase 3 phase of the power supply voltage, respectively respect _1a to D _3c, calculates the equation 27.

In phase decision block 407, in accordance with Table 3, the maximum voltage phase a phase of the output voltage instruction value from the output voltage mode m _L received from the output level decision block 404, the intermediate voltage phase b phase, the minimum voltage phase c phase is u-phase, respectively, It is determined whether it corresponds to the v phase or the w phase, and the on-duty of Expression 27 is output to the PWM generation unit of the corresponding phase.
For example, when m _L = 1, since the output intermediate voltage phase b phase is v phase according to Table 3, D _1v = D _1b and D _12v = D _12b and D _1v and D _12v are compared with the v phase. Output to the circuit block 410.

  In FIG. 3, each of the phase comparison circuit blocks 409 to 411 and the PWM generation blocks 412 to 414 for generating the PWM pulse is the same circuit for all three phases, and a specific PWM generation principle diagram is shown in FIG.

FIG. 9 is a principle diagram of PWM generation for the v phase, and is when the output phase voltage command is an intermediate voltage (v _b ^* = v _v ^* ).
In FIG. _6A , the triangular wave carrier changing between 0 and 1 is compared with the on-duty D _1v , D _12v (= D _1v + D _2v ), and signals Q _1v , Q _12v are obtained. The switching circuit G _1v , G _2v , G _3v is obtained from the signals Q _1v , Q _12v by the logic circuit shown in FIG. 5B, and further _divided into r, s, t phases to generate the PWM pulse G _rv. , G _sv , G _tv are generated. In this distribution, the signal r ₁ (e ₁ = _er signal), the signal s ₁ (e ₁ = _es signal), the signal r ₂ s ₃ (e ₂ = _er or e ₃ = e) obtained from Table 1 are used. _{The s} signal is used to determine whether the power supply r and s phases are the maximum voltage phase, intermediate voltage phase, and minimum voltage phase, respectively, and PWM pulses G _rv and G _sv are generated. The power supply t-phase PWM pulse G _tv is obtained as logic that turns on when both G _rv and G _sv are in the off state.

  The power supply voltage detection means 41 or the output voltage command calculation means 42 in the present embodiment is merely an example, and the power supply voltage detection means 41 may detect a line voltage in addition to detecting the phase voltage of the power supply 10. Alternatively, it can be realized by assuming that the amplitude of the power supply voltage is constant and detecting the positive / negative of the power supply voltage. Further, the output voltage command calculation means 42 can be realized by vector control in addition to V / f control.

Next, FIG. 10 is a block diagram showing a second embodiment of the present invention, and corresponds to the invention of claim 3. Control device 40B according to this embodiment is obtained by adding the source current command computing means 43 to the control device 40A of FIG. 1, the computing means 43 the supply voltage e _r, e _s, the power current command value from e _{t i r ^*,} i ^{s *,} calculates the _i ^{t *,} and outputs it to the on-duty computation means 400.

Hereinafter, a case where the power source power factor is controlled to 1 will be described.
Source current command value ^{_{i r *, i s *,}} i t * is a value proportional to the power supply voltage _{e r, e s, e t} , given by Equation 28 the proportionality constant as the 1 / _{R s.} Incidentally, R _s is the matrix converter of the impedance viewed from the power source (resistor).

11, * source current command value i r, i s *, is a block diagram of the on-duty operation unit 400 in the case of adding the i t * as a new input.
In the matrix converter simplified model shown in FIG. 4, the power supply voltage is e r> e s> e t , the output voltage command value v u *> v v *> v and w *, the power supply voltage e r in the switching period 2T , e s, approximating e t and the load current i uv, i vw, i wu as constant values, respectively. Then, to derive a calculation formula for the on-duty D ru to D tw of the switches S ru to S tw.

From Equation 28, the power supply voltage e r, e s, e t is * source current command value i r, i s *, represented by equation 29 using i t *.

By substituting Equation 29 into Equation 11～16,19,22,23, each on-duty _D ru _{to D tw} can be obtained by equation 30.

Thus, the on-duty _D ru _{to D tw} of nine AC switch _S ru _{to S tw} of the matrix converter 20 it has been determined. It can be seen that these on-duty calculation formulas do not include the proportionality constant 1 / R _s in Formula 28 and do not depend on R _s . Accordingly, the actual supply current _{i r, i s, i t} is * source current command value ^{_{i r, i s *, i}} t * each become proportional to the current in the phase (input power factor) of the power supply current command value The magnitude of the power supply current is determined by the load current.

In the above description, the power supply voltage to derive the _{e r> e s> e t} , the output voltage command value ^{_{v u *> v v *>}} v w * on-duty of each switch in the case of.
The operation of the on-duty calculation means 400 when a switching pattern is generated in an arbitrary magnitude relationship between input and output voltages will be described below.

In the power supply level decision block 402 of the on-duty operation means in FIG. 11, already supply voltage _e r according to Table 1 as described with reference to FIG. _3, e s, the maximum voltage _{e 1} of _{e t,} intermediate voltage _{e 2,} the minimum to determine the voltage _{e 3.} In Figure 11, the power source current command value _i ^r * to the power level decision block ^{_{402, i s *, i t}} * are newly input, according to Table 1, the maximum voltage _{e 1} of the power supply voltage, an intermediate voltage _e 2, _i ¹ * and minimum voltage _{e 3} a current command value of each same _phase, ⁱ 2 _*, and outputs it as _i ^{3 *.} For _example, if ^{_{e 1 = e r, i 1}} * = a _i ^{r *} regardless of the magnitude relation between the current command value.

In the calculation by the blocks 403 to 405 related to the output voltage in FIG. 11, as already described in the explanation of FIG. 3, the line voltage command value v _ab is calculated from the output voltage command values v _u ^* , v _v ^* , v _w ^{*. *} , V _bc ^* , v _ca ^* are obtained.

Further, in the on-duty calculation block 406, the maximum voltage e ₁ , intermediate voltage e ₂ , minimum voltage e ₃ , power source current command values i ₁ ^* , i ₂ ^* , i ₃ ^* and line voltage command value v _ab ^*. , V _bc ^* , v _ca ^* , the maximum voltage phase a phase, intermediate voltage phase b phase, and minimum voltage phase c phase of the output voltage command value are respectively the maximum voltage 1 phase, intermediate voltage 2 phase, minimum of the power supply voltage regard the on-duty _D 1a _{to D 3c} which is connected to the voltage 3-phase, it calculates the equation 31.

  The on-duty calculation block 406 outputs the on-duty of Expression 31 to the phase determination block 407. The configuration and operation of each block until the PWM pulse after the phase determination block 407 is output is as described in the explanation of FIG.

FIG. 12 is a block diagram showing the configuration of the third embodiment of the present invention, and corresponds to the invention of claim 4.
In the control device 40C according to this embodiment, the power supply voltage phase detection means 44 is added to the control device 40B of FIG. 10, and the power supply current command calculation means 43 is supplied from the power supply voltage phase angle θ and the power supply power factor command value ψ ^*. current command value ^{_i r} _{*, i} s _*, it is obtained so as to calculate the _i ^{t *.}
In FIG. 12, the components other than the power supply voltage phase detection means 44 and the power supply current command calculation means 43 are the same as those in FIG.

FIG. 13 shows an example of the configuration of the power supply voltage phase detection means 44 when a PLL (Phase Locked Loop) is used.
13, the waveform shaping circuit 441, the power supply voltage _{e r,} e s, detects the zero cross of _{e t,} and generates a signal theta ₀ of 3 pulses in the power supply 1 cycle changes every [pi / 3. Further, a signal θ _p corresponding to θ ₀ is generated from the detected power phase angle value θ. The phase comparator 444 detects the phase angle error Δθ between the signals θ ₀ and θ _p and adjusts the frequency of the output pulse p of the oscillator 442 so that the phase angle error Δθ is reduced. By counting the output pulse p of the oscillator 442 by the counter 443, the output value of the counter 443 becomes the power supply phase angle θ.

Next, the operation of the power supply current command calculation means 43 in FIG. 12 will be described. From the power supply phase angle θ and the power supply power factor angle command value ψ ^* , the power supply current command values i _r ^* , i _s ^* , and i _t ^* are calculated by Equation 32.

What is 1 the amplitude of the current command value in Equation 32, the actual supply current _i r, as already _mentioned, i s, _{i t} is the power current command value ^{_{i r *, i s *,}} i t * , respectively This is because the current becomes proportional and does not depend on the amplitude of the power supply current command value. Therefore, the power source power factor corresponding to the power source power factor angle command value ψ ^* can be realized by the proportional relationship between the actual current and the current command value.

FIG. 14 is a block diagram showing the configuration of the fourth embodiment of the present invention, and corresponds to the invention of claim 5.
The control device 40D in this embodiment is obtained by adding a cosine / sine calculation means 45 for calculating a cosine value and a sine value of the power supply voltage phase angle θ to the control device 40B of FIG. the cosine value cos [theta], the sine value sinθ and power factor angle command value [psi ^* from the power source current command value ⁱ _{r ^*,} i s ^_*, adapted to calculate a i _t ^*.
14 is the same as FIG. 10 and FIG. 12 except for the cosine / sine calculation means 45 and the power supply current command calculation means 43, and different parts will be described below.

In the cosine-sine calculating means 45 in FIG. 14, the power supply voltage e _r, e _s, e _t is symmetrical three-phase voltage equation 1, the power supply voltage stationary coordinates (alpha-beta coordinate) transformation as Equation 33 The two-phase power supply voltages e _α and e _β are obtained.

  From Equation 33, cos θ and sin θ are obtained by Equation 34, respectively.

In the power supply current command calculation means 43 in FIG. 14, the power supply current command values i _r ^* , i _s ^* , and i _t ^* are calculated from cos θ, sin θ and power supply power factor angle command value ψ ^* in Expression 34, respectively. Calculate by

FIG. 15 is a block diagram showing a fifth embodiment of the present invention, and corresponds to the invention of claim 6.
The control device 40E in this embodiment is configured by adding a load current detecting means 46 and a power factor angle command calculating means 47 to the control device 40C of FIG. 12, and according to the load current, a power source power factor angle command value ψ. ^* Is now adjusted.

In each of the above embodiments, the explanation is made by ignoring the LC filter 21 on the input side of the matrix converter 20, but the actually controlled power supply current command value is the converter input current after passing through the LC filter 21. is there.
In the following, a method for controlling the power source power factor to 1 by the power factor angle command calculation means 47 will be described assuming a light load where the phase advance current flowing into the LC filter 21 cannot be ignored.

FIG. 16A shows a one-phase equivalent circuit of the matrix converter input section, and FIG. 16B shows a vector diagram thereof. In the following description, a “.” (Dot) indicating a vector cannot be added in the specification text, and thus will be omitted.
In FIG. 16A, the current command value I _m ^* flowing into the matrix converter is represented by a current source. When I _m ^* = 0, the capacitor current I _C of the LC filter 21 is given by Equation 38.

Usually, since the resonance frequency of the LC filter is sufficiently higher than the power supply frequency, a relationship of ωL−1 / ωC <0 is obtained, and I _q > 0, so that the capacitor current I _C becomes a leading current. As shown in FIG. 16 (b), and the power supply current _{I s} only valid current component _{I p} required by the matrix converter 20 as in Equation 39.

At this time, in order to realize the power source power factor 1, the input current command value I _m ^* of the matrix converter 20 may be given as in Expression 40.

That is, the power source power factor 1 can be realized by giving the power factor angle command value ψ ^* shown in FIG.
In the control device 40E of FIG. 15, since the calculation of the line voltage effective value E by the power supply voltage phase detection means 44, the load current detection means 46, and the power factor angle command calculation means 47 are the same as those in the above embodiments, Below, a different part is demonstrated.

The value of the line voltage effective value E can also be held as a constant in the controller if the power supply to be used is defined. Further, when the effective value E is not necessarily constant value, the power supply voltage phase detection means 44, the power supply voltage e _r, e _s, the power supply voltage of the two-phase formula 33 using a _{e t} e _α, e β Therefore, the effective value E can be calculated by Equation 41.

Load current detection means 46 in FIG. 15, the load current _i u, the _{i v} detected by using a Hall CT, to obtain the remaining w-phase current _{i w} using Equation 42.

Next, the operation of the power factor angle command calculating means 47 in FIG. 15 will be described. The average output power P _{out in} the switching half cycle T of the matrix converter can be obtained from Equation 43 using the load currents i _u , i _v , i _w and the output voltage command values v _u ^* , v _v ^* , v _w ^*. it can.

If the loss is ignored, the input / output power of the matrix converter is equal, so that the effective current _Ip on the power supply side can be obtained by Equation 44.

In addition, when considering the loss, an effective current corresponding to the loss may be added to the right side of Equation 44 to cope with it. On the other hand, from the relationship between the line voltage effective value E and the equation 38, the reactive current I _q flowing through the capacitor is obtained by the equation 45.

The power factor angle command value ψ ^* is given by Expression 46 from the vector diagram of FIG. 16B using the effective current I _p of Expression 44 and the reactive current I _q of Expression 45.

Or, the power factor angle command calculating means 42, cos ^* according to Equation 47 instead of the power factor angle command value [psi ^*, calculates the Sinpusai ^*, the supply current command value means 43 in FIG. 15, these cosψ ^{*, sinψ The} power supply current command values i _r ^* , i _s ^* , and i _t ^* can also be calculated by using Equations 35 to 37 using ^* .

In this embodiment, the power factor angle command value ψ ^* is calculated based on the load current (output current of the matrix converter 20) i _u , i _v , i _w, but based on the output voltage of the matrix converter 20. The power factor angle command value ψ ^* may be adjusted.
Further, the idea of adjusting the power factor angle command value ψ ^* according to the output current and output voltage as in the fifth embodiment is also applicable to the control device 40D of the fourth embodiment shown in FIG. .

It is a block diagram which shows the structure of 1st Embodiment of this invention. It is a block diagram which shows the structure of the output voltage calculating means in FIG. It is a block diagram which shows the structure of the on-duty calculating means in FIG. It is a figure which shows the simple model of a matrix converter. It is a figure which shows the on-duty (switching pattern) and output line voltage in 1st Embodiment, Fig.5 (a) is an operation example at the time of a high output voltage, FIG.5 (b) is an operation example at the time of a low output voltage. is there. It is a connection diagram of the maximum and minimum output voltage phases. It is a connection diagram of the maximum / intermediate output voltage phase. It is a connection diagram of an intermediate and minimum output voltage phase. FIG. 9A is a diagram illustrating the principle of generating a PWM pattern. FIG. 9A is a waveform diagram of a triangular wave carrier and a switching signal, and FIG. 9B is a circuit diagram of a switching signal and a PWM pulse generating circuit. It is a block diagram which shows the structure of 2nd Embodiment of this invention. It is a block diagram which shows the structure of the on-duty calculating means which added the power supply current command value. It is a block diagram which shows the structure of 3rd Embodiment of this invention. It is a block diagram of the power supply voltage phase angle detection means using PLL. It is a block diagram which shows the structure of 4th Embodiment of this invention. It is a block diagram which shows the structure of 5th Embodiment of this invention. FIG. 16A is an equivalent circuit diagram for one phase of the input portion of the matrix converter, and FIG. 16B is a vector diagram. It is a figure which shows the on-duty (switching pattern) and output line voltage in a prior art, FIG. 17 (a) is an operation example of the conventional system 1, FIG.17 (b) is an operation example of the conventional system 2 (only low voltage output). .

Explanation of symbols

10: Three-phase AC power supply 20: Matrix converter 21: Input filter 30: Load 40A, 40B, 40C, 40D, 40E: Control device 400: On-duty calculation means 41: Power supply voltage detection means 42: Output voltage command calculation means 421: V / f table 422: integrator 423: three-phase oscillator 43: power supply current command calculation means 44: power supply voltage phase detection means 45: cosine / sine calculation means 46: load current detection means 47: power factor angle command calculation means

Claims (6)

Each phase of the AC power supply and each phase on the output side are directly connected by a bidirectional AC switch, and the AC power supply voltage is directly converted into an AC voltage of an arbitrary magnitude and frequency by the on / off operation of the AC switch. -In AC direct power converter,
Means for determining the maximum voltage phase, the intermediate voltage phase and the minimum voltage phase when the power supply voltage of each phase is input;
Means for determining the maximum voltage output phase, the intermediate voltage output phase, and the minimum voltage output phase when the output voltage command value of each phase is input;
Regardless of the magnitude of the output voltage, wherein the maximum voltage phase of the maximum voltage output phase power supply side AC switches and connected to the intermediate voltage phase creates a switching pattern that is not connected to the minimum voltage phase, the minimum voltage It means for creating a switching pattern by connecting the AC switch output phase to the intermediate voltage phase and the minimum voltage phase of the power supply side is not connected to the maximum voltage phase,
AC characterized by comprising a - AC direct power converter controller. In the control apparatus of the AC-AC direct power converter according to claim 1,
AC-, comprising: means for detecting a power supply voltage; means for calculating an output voltage command value; and means for calculating an on-duty of the AC switch from the power supply voltage and the output voltage command value. Control device for AC direct power converter. In the control apparatus for an AC-AC direct power converter according to claim 2,
A control device for an AC-AC direct power converter, comprising means for calculating a power supply current command value based on the power supply voltage, and using the power supply current command value for the calculation of the on-duty. In the control apparatus for an AC-AC direct power converter according to claim 2,
Means for detecting a power supply voltage phase angle based on the power supply voltage; and means for calculating a power supply current command value from the power supply voltage phase angle and a power supply power factor command value. A control device for an AC-AC direct power converter, characterized by being used for calculation. In the control apparatus for an AC-AC direct power converter according to claim 2,
Means for calculating a cosine value and a sine value of a power supply voltage phase angle based on the power supply voltage, and means for calculating a power supply current command value from these cosine value, sine value and power supply power factor command value. A control apparatus for an AC-AC direct power converter, wherein a current command value is used for the calculation of the on-duty. In the control apparatus for an AC-AC direct power converter according to claim 4 or 5,
An AC-AC direct power converter control device comprising means for adjusting the power source power factor command value in accordance with an output current or an output voltage of the power converter.

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