JP3637897B2 - Synchronous motor drive device, inverter device, and synchronous motor control method - Google Patents

Description

【００５７】
数式１において、Ｖｄは埋込磁石型同期電動機（ＩＰＭＳＭ）のｄ軸電圧、Ｖｑは埋込磁石型同期電動機（ＩＰＭＳＭ）のｑ軸電圧、Ｒは埋込磁石型同期電動機（ＩＰＭＳＭ）の１相あたりの抵抗値、Ｌｄは埋込磁石型同期電動機（ＩＰＭＳＭ）のｄ軸インダクタンス、Ｌｑは埋込磁石型同期電動機（ＩＰＭＳＭ）のｑ軸インダクタンス、Φｆは埋込磁石型同期電動機の誘起電圧定数、ｉｄは埋込磁石型同期電動機のｄ軸電流、ｉｑは埋込磁石型同期電動機のｑ軸電流、ωは埋込磁石型同期電動機の回転周波数、ｐは時間ｔによる微分演算子、を表している。また、埋込磁石型同期電動機の極対数をＰｍとすると、埋込磁石型同期電動機（ＩＰＭＳＭ）の出力トルクτｍは一般的に数式２で与えられる。
【００５８】
【数２】

【００５９】
また、ｄ軸電流ｉｄ、ｑ軸電流ｉｑ、および電流の大きさＩａは数式３のように定義される。
【００６０】
【数３】

【００６１】
数式３を数式２に代入するとトルクτｍは数式４のように表される。
【００６２】
【数４】

【００６３】
ここで、出力トルクが最大となる（あるいは電流が最小となる）電流位相角をβmとし、βｍを求める手順について以下に説明する。数式４を電流位相角βで偏微分して＝０とおくことによって、最大トルクが得られる（あるいは電流が最小となる）電流位相角βｍが数式５のように得られる。
【００６４】
【数５】

【００６５】
ただし、βmの演算においては、数式５のように出力トルクが最大となるように演算しなくても、出力トルクが所定値以上となるように以下に示す数式６のような近似式を用いても良い。
【００６６】
【数６】

【００６７】
ここで、数式６は、数式５をマクローリン級数展開して２次の項まで求めたものである。この近似式６を用いると出力トルクは最大とならない（電流は最小とならない）が、出力トルクを所定値以上に（電流値を所定値以下に）することができるので、高出力で低電流の装置を得ることができる。また、βｍの演算が近似式であり簡略化されているため、マイコンでの演算負荷を低減させることができるので、演算時間の低減ができ、さらには、低コストのマイコンでの対応が可能となり、マイコンの選定の自由度が広がる。また、βｍの演算が近似式であり簡略化されているので、収束性も改善されるので、制御において追従性のよい装置や制御方法が得られる。ただし、制御精度を向上させる必要がある場合は、マクローリン級数の３項まで用いるなどして近似式の精度を向上させるか、あるいは数式５をそのまま使用すればよい。
【００６８】
ここで、本実施の形態では、座標系としてｄ−ｑ軸ではなく、γ―δ軸を使用しているので、図４を用いて、ｄ−ｑ軸座標系とγ−δ座標系の関係について説明する。本実施の形態では、同期電動機を回転子位置検出のための位置センサを用いずに駆動する場合、実際には埋込磁石（ＩＰＭ：Ｉｎｔｅｒｉｏｒ Ｐｅｒｍａｎｅｎｔ Ｍａｇｎｅｔ）回転子のｄ−ｑ軸を検出することが困難であることから、制御上の軸として図４で示すようなγ−δ軸を用いるようにしている。
【００６９】
図４は同期電動機（特に埋込磁石型同期電動機（ＩＰＭＳＭ））のある運転条件でのベクトル線図である。図４において、横軸はγ軸、縦軸はδ軸を表している。また、ｄ軸およびｑ軸は、γ軸およびδ軸に対してそれぞれΔθだけ回転した角度位置に位置している。ここで、ｉγはＩＰＭＳＭのγ軸電流、ｉδはＩＰＭＳＭのδ軸電流、ＩａはＩＰＭＳＭに流れる電流の大きさであり、ｉγとｉδの合成ベクトルである。また、Ｉａはｑ軸に対してβ（電流位相角）だけ角度がずれている。このように、電流位相角βは、ｑ軸と電流ベクトルＩａとのなす角である。
【００７０】
ΦＦは回転子磁石により生じる磁束ベクトルであり、その大きさは誘起電圧定数Φｆであり、ｄ軸上に位置する。また、Ｌｑ・ｉδは固定子磁束のδ軸成分、Ｌｄ・ｉγは固定子磁束のγ軸成分、Φ０は固定子磁束と回転子磁束の合成磁束であり、回転子磁束（ΦＦ）と固定子磁束（Ｌｑ・ｉδ、Ｌｄ・ｉγ）の合成磁束ベクトルである。また、ω・Φ０は合成磁束により生じる電圧ベクトル、Ｒ・Ｉａはモータの抵抗により生じる電圧効果分を補償する電圧ベクトル、ＶａはＩＰＭＳＭに印加される電圧であり、ω・Φ０とＲ・Ｉａの合成ベクトルである。ここで、制御上の軸として、γ−δ軸を用いずにそのままｄ-ｑ軸を用いてもよく、どちらを用いても制御可能であるが、本実施の形態では電流成分が検出しやすいγ-δ軸を用いた場合について説明する。
【００７１】
ここで、本実施の形態では、制御軸としてγ-δ軸を用いており、制御の簡素化のためｄ−ｑ軸とγ―δ軸との位相差ΔθをΔθ≒０と近似して、各軸のインダクタンスをＬγ≒Ｌｄ、Ｌδ≒Ｌｑとして取り扱っている。
【００７２】
本実施の形態においては、電流が最小となる電流位相角にて駆動するための制御量として励磁電流を制御するようにしているので、この励磁電流について以下に説明する。図４に示したように、制御軸として定義されたγ-δ軸（あるいはＩＰＭＳＭの回転子位置を基準にしたｄ-ｑ軸）において、γ軸（あるいはｄ軸）に対応する電流ｉγ（あるいはｉｄ）が一般的に励磁電流と呼ばれ、この励磁電流により発生する固定子側の磁束（Ｌｄ・ｉγ＋Ｌｑ・ｉδ）と回転子磁石の磁束（ΦＦ）とにより合成磁束ベクトル（Φ０）が生じる。
【００７３】
一方、δ軸（あるいはｑ軸）に対応する電流は一般的に電流成分のトルクに関する成分(トルク電流)と呼ばれ、モータの出力トルクに応じた大きさとなる。
【００７４】
ここで、電流ベクトルＩａは励磁電流ｉγ(またはｉｄ)とトルク電流ｉδ(またはｉｑ)の合成ベクトルである。ＩＰＭＳＭの運転状態（所定回転数、所定負荷トルク）に応じて出力電圧ベクトルＶａを変化させると、励磁電流ｉγが変化（δ軸電流ｉδはトルク成分のためほとんど変化しない）し、この励磁電流ｉγが変化することにより電流位相角βが変化する。図４に示したように励磁電流ｉγ（またはｉｄ）とトルク電流ｉδ（またはｉｑ）の合成ベクトルが電流ベクトルＩａであるため、励磁電流ｉγ（またはｉｄ）とトルク電流ｉδ（またはｉｑ）を制御して、出力電圧ベクトルＶａを制御すれば電流ベクトルＩａの電流位相角βを変化させることができる。
【００７５】
本実施の形態では速度制御器を持たない、つまりトルク電流指令値（ｉδ＊）を有さない場合の制御においても、電流が所定値以下（あるいは最小）となる電流位相角が得られるように励磁電流ｉγのみを制御することによって出力電圧ベクトルＶａを制御して、電流が所定値以下（電流最小）となるように制御している。電流が最小（あるいは電流が所定値以下）となる電流位相角βｍは、ｄ−ｑ軸座標系のｑ軸から電流ベクトルＩａへのなす角である。しかし、速度制御器を持たない制御方式あるいは位置センサを持たない制御方式においては、実際のｄ−ｑ軸の位置が分からないため、本実施の形態では、制御軸（γ−δ軸）とモータ軸（ｄ−ｑ軸）との位相差Δθの影響を無視できるように、制御に用いる変数として電圧ベクトルの大きさＶａ、電圧指令値Ｖａ＊、電流ベクトルの大きさＩａを用いるようにしている。
【００７６】
本実施の形態では、同期電動機の駆動手段に速度制御ループや電流制御ループを持たない制御方式においても、電流最小制御を実現可能とするため、電流位相制御手段１４にて出力される励磁電流指令値ｉγ＊を以下のようにして求めている。
【００７７】
出力電圧演算手段１５が出力電圧信号（たとえば出力電圧指令値演算手段１０の前回の出力値であるＶγ＊およびＶδ＊など）を用いて次式（数式７）によりインバータの出力する電圧の大きさＶａを求める。
【００７８】
【数７】

【００７９】
ここで、出力電圧信号にインバータ主回路３が出力する出力電圧（Ｖｕ、Ｖｖ、Ｖｗ）を使用してもよく、その場合は、別途電圧検出手段（図示せず）をインバータ主回路３の出力側に設ければよい。本実施の形態では、電圧検出手段などを不要とするため出力電圧指令値演算手段１０の前回の出力値であるＶγ＊およびＶδ＊を使用するようにしているので、電圧検出手段などが不要で構成が簡単であり、しかも低コストありながら、低電流で高効率な装置が得られる。
【００８０】
また、電流演算手段２１では、駆動負荷である同期電動機５を流れる電流を３相２相座標変換を行なって得られるγ軸電流ｉγおよびδ軸電流ｉδを用いて、電流の大きさＩａを数式８にて演算する。
【００８１】
【数８】

【００８２】
ここで、数式８では、制御軸（γ−δ軸）上にて電流ベクトルの大きさＩａを演算しているが、制御軸（γ−δ軸）とモータ軸（ｄ−ｑ軸）とは、位相差Δθだけずれているだけで、電流ベクトルの大きさＩａは同一である。したがって、本実施の形態のように電流ベクトルの大きさＩａを制御に用いると、制御軸（γ−δ軸）とモータ軸（ｄ−ｑ軸）の位相差Δθを考慮しなくても良くなるので、速度制御ループや電流制御ループを持たない制御方式においても、制御可能となる。電流位相演算手段２２では、数式８を使用して数式６（あるいは数式５）より電流が所定値以下（あるいは最小）となる電流位相角βｍを演算する。電圧指令値演算手段１６では、ｄ−ｑ軸座標系の演算式を用いて、電流が所定値以下（あるいは最小）となる電流位相角βｍで駆動負荷５を駆動するための電圧指令値Ｖａ＊を以下のようにして求める。数式３を数式１に代入すればＶｄ、Ｖｑが求まるから、その合力の大きさＶａはＶｄとＶｑの２乗和の平方根（数式９）で表される。
【００８３】
【数９】

【００８４】
ただし、制御軸（γ−δ軸）とモータ軸（ｄ−ｑ軸）とは、Δθの位相差だけずれているが、制御軸（γ−δ軸）上の電圧ベクトルの大きさＶａとモータ軸（ｄ−ｑ軸）上の電圧ベクトルの大きさＶａは同じなので、座標軸の置き換えが可能である。すなわち、制御上、実際に得られる座標系はγ−δ軸座標系なので、モータ軸（ｄ−ｑ軸）上の電圧ベクトルの大きさを表す数式９を制御軸（γ−δ軸）上の電圧ベクトルの大きさＶａを表す数式７に置き換えて考えることができる。
【００８５】
そして、この数式７に数式８で求めたＩａおよび数式６（あるいは数式５）で求めた電流位相角βｍを代入して電流位相角βがβｍのときの電圧指令値Ｖａ＊が数式１０のように求まる。ここでも、制御軸（γ−δ軸）とモータ軸（ｄ−ｑ軸）との位相差Δθを考慮しないて良いように、指令値を電圧ベクトルの大きさＶａ＊で指令するようにしている。
【００８６】
【数１０】

【００８７】
ここで、上述したように、Ｉａは同期電動機５に流入するγ-δ軸での電流の大きさ、βmは出力トルクが所定値以上（あるいは最大）となる電流位相角、すなわち電流が所定値以下（最小）となる電流位相角である。
【００８８】
ここで、数式１０の電圧指令値Ｖａ＊の演算時において、数式５あるいは数式６を使用して電流が最小（電流が所定値以下）となる電流位相角βmまで求めると演算負荷が大きくなり、演算負荷が増大し、演算時間が長くなる場合がある。この場合は、βｍまで求める必要はなく、sinβmを求めることができれば良い。そのためには、数式５（あるいは数式６）を変形してsinβmを求め、数式８で得られたＩａ及び一次周波数演算手段９により求まる一次周波数ω１をωとして数式１０に代入すれば、電圧指令値Ｖａ＊を演算することが可能であり、βｍまで求める場合に比べてｓｉｎβｍまでしか求めないので、演算数を減少させることができ、演算時間の短縮が図れる。ここで、ｓｉｎβｍ、ｃｏｓβｍは、たとえば数式６より数式１１のように求めることができる。
【００８９】
【数１１】

【００９０】
ここで、Ｖａ＊がωとＩａとの関数であることを利用して、ωとＩａまたはωとＩａ^２の関係を有する数値テーブルを予め電圧指令値演算手段１６や別途設けた記憶手段などに記憶するようにしておいてもよい。
【００９１】
このようにテーブルを有するようにすれば、平方根などの演算時間のかかる演算の低減および演算数の低減を行なうことが可能となり、マイコンの選定の自由度が大きくなり、低コスト化が図れる。また、演算式を用いるかテーブルを用いるかの判断は使用するマイコンの性能に応じて選択すればよい。
【００９２】
基準励磁電流演算手段１８においては、数値テーブル、あるいは周波数ωとの関数、あるいは予め与えられた一定値の定数等により、励磁電流指令値ｉγ＊の基準値となるｉγ０を求める。励磁電流演算手段１７では基準励磁電流指令値演算手段１８により得られた基準励磁電流指令値ｉγ０、出力電圧演算手段１５により得られた出力電圧の大きさＶａ、および電圧指令値演算手段１６により得られた電圧指令値Ｖａ＊を用いて以下に示す数式１２により励磁電流指令値ｉγ＊を求める。ここで、数式１２におけるＫｈｅは予め実験などにより与えられる励磁電流の制御ゲインである。
【００９３】
【数１２】

【００９４】
数式１２における励磁電流指令値ｉγ＊の演算は、電圧指令値Ｖａ＊と出力電圧の大きさＶａとの差に比例ゲインＫｈｅを乗じて演算する比例制御方式の一例である。比例制御方式では電圧指令値Ｖａ＊と出力電圧の大きさＶａとの差分に予め与えられた制御ゲインを乗じているだけなので、電圧指令値Ｖａ＊に対して定常的な誤差を生じるため、より精度よく電流位相角βの制御を行いたい場合は、積分制御を用いればよい。積分制御を用いた制御方式の一例を数式１３に示す。
【００９５】
【数１３】

【００９６】
ここで、ｉγ＊_ｐｒｖは、電流位相制御手段１４における前回の制御周期での励磁電流指令値の出力値であり、この前回の制御周期での励磁電流指令値の出力値を基準励磁電流指令値としてｉγ０の代わりに使用することにより電圧指令値Ｖａ＊と出力電圧の大きさＶａとの差分を極力０に近づけることができるようになるため、電圧指令値Ｖａ＊に対する誤差が小さくでき、演算精度が向上する。ここで、数式１２で示したような比例制御方式では精度よく電流位相角βを電流最小位相角βｍへ追従させることが困難ではあるが、速度制御ループや電流制御ループを持たない制御方式や過変調領域での制御を行なう場合には、数式１３で示したような積分制御よりも数式１２で示したような比例制御のほうがより安定した制御性が得られる。
【００９７】
電流位相制御手段１４では以上の数式１〜数式１３を使用することによって、励磁電流指令値ｉγ＊を演算する。そして、この電流位相制御手段１４にて演算された励磁電流指令値ｉγ＊は同期電動機の制御の安定性を得るためにローパスフィルタなどのフィルタ手段１９によりフィルタリングされ、フィルタリングされた励磁電流指令値ｉγ＊が出力電圧指令値演算手段１０において演算に用いられる。ここで、フィルタ手段１９は設けなくても良いが、制御の安定性を得るためにはフィルタ手段１９は設けた方が良い。
【００９８】
以上のように、同期電動機５を電流が所定値以下（あるいは最小）となる電流位相角にて駆動するために電流位相角を制御する方法として、本実施の形態では、同期電動機に流れる電流を検出する電流検出手段６ａ、６ｂと、電流検出手段６ａ、６ｂにより得られた電流信号および同期電動機５に印加される電圧の電圧信号に基づいて同期電動機５の出力トルクが所定値以上となるように同期電動機５に流れる電流の電流位相角βを制御するための励磁電流指令値ｉγ＊を出力する電流位相制御手段１４と、電流位相手段１４により出力された励磁電流指令値ｉγ＊をフィルタリングするフィルタ手段１９と、電流検出手段６ａ、６ｂにより検出された電流の変動量より周波数指令値ωｍ＊に対する補償量ωｄを求め、補償量ωｄにより補償された周波数指令値ω１を出力する周波数演算手段９と、補償された周波数指令値ω１、およびフィルタリングされた励磁電流指令値ｉγ＊に基づいて同期電動機５を駆動するための出力電圧指令値Ｖγ＊、Ｖδ＊を求める出力電圧指令値演算手段１０と、を備えたので、電流が所定値以下となる電流位相角の励磁電流指令値および補償された周波数指令値に基づいて出力電圧指令値を演算するため、周波数変動や負荷変動により同期電動機の運転条件が変化した場合においても、その条件に応じて電流が所定値以下（あるいは最小）となる電流位相角へ電流を制御することが可能であり、運転条件の変化に追従して精度よく電流値を所定値以下（あるいは最小）にすることができ、高効率な電動機駆動装置を得ることができる。また、フィルタ手段１９を設けているので、演算毎の励磁電流指令値ｉγ＊のばらつきが小さくなり、安定した制御が行なえる。
【００９９】
また、本実施の形態では、電流検出手段６ａ、６ｂにより得られた電流信号に基づいて同期電動機５の出力トルクが所定値以上となるような電流位相角βｍを求める電流位相演算手段２２と、電流位相演算手段２２により得られた電流位相角βｍおよび周波数指令値演算手段９により得られた前回の周波数指令値である一次周波数ω１とに基づいて電圧指令値Ｖａ＊を演算する電圧指令値演算手段１６と、同期電動機５に印加される電圧の電圧信号に基づいて電圧の大きさを演算する電圧演算手段１５と、電圧指令値演算手段１６にて得られた電圧指令値Ｖａ＊、および電圧演算手段１５より得られた電圧の大きさＶａとに基づいて同期電動機５を駆動するための励磁電流指令値ｉγ＊を求める励磁電流指令値演算手段１７と、によって電流位相制御手段１４を構成して、電流が所定値以下（あるいは最小）となるような電流位相角βｍへ電流位相角βを制御するようにしたため、電流が所定値以下（あるいは最小）となる電流位相角βｍでの出力電圧と実際の出力電圧を求める簡単な構成でありながら、運転条件ごとに励磁電流指令値の数値テーブルや関数を持つ必要がないため、インバータ制御手段に用いられるマイコンのメモリ使用量を増加させることがなく、メモリ容量が小さい安価なマイコンが使用でき、また、簡単な制御回路で対応できるため、信頼性が高く安価な同期電動機駆動装置を得ることができる。
【０１００】
また、本実施の形態では、直流を供給する直流電源４と、スイッチング素子をオン、オフ動作させることにより直流電源４よりの直流を交流に変換するインバータ主回路３と、インバータ主回路３内のスイッチング素子のオン、オフ動作を制御するための電圧指令値を出力するインバータ制御手段２と、インバータ制御手段２により制御されるインバータ主回路３により出力される電圧が印加されることにより制御される駆動負荷５と、を備え、インバータ制御手段２を、駆動負荷を流れる電流に基づいて駆動負荷の出力トルクが所定値以上となるような電流位相角βｍにて駆動負荷５を駆動するための励磁電流指令値ｉγ＊を演算する電流位相制御手段２２と、電流位相制御手段２２より得られた励磁電流指令値ｉγ＊をフィルタリングするフィルタ手段１９と、駆動負荷５を流れる電流の変動量より周波数指令値ωｍ＊を補償して一次周波数ω１として出力する周波数演算手段９と、フィルタ手段１９によりフィルタリングされた励磁電流指令値ｉγ＊と周波数演算手段９により得られた一次周波数ω１とに基づいて駆動負荷５を駆動するための電圧指令値Ｖγ＊、Ｖδ＊を演算する出力電圧指令値演算手段１０と、によってインバータ装置を構成しているので、電流位相制御手段と周波数演算手段を設けるだけの簡単な構成でありながら、駆動負荷の運転条件が変化しても、その条件での出力トルクが所定値以上となるように制御できる。また、出力トルクに応じて電流値が所定値以下となる電流位相角で電流を制御でき、低電流で高効率な同期電動機駆動装置が実現できる。また、電流値を小さくできるのでスイッチング素子の定格容量を小さくでき、低コストなインバータ装置を提供できる。また、フィルタ手段１９を設けているので、励磁電流指令値ｉγ＊のばらつきが少なくなり、駆動負荷を安定して制御できる信頼性の高いインバータ装置が得られる。
【０１０１】
さらに、駆動負荷５を流れる電流により出力トルクが所定値以上となる電流位相角βｍを求める電流位相演算手段２２と、電流位相演算手段２２により求められた電流位相角βｍが得られるような電圧指令値Ｖａ＊を演算する電圧指令値演算手段１６と、出力電圧指令値演算手段１０により求められた前回の出力電圧指令値Ｖγ＊、Ｖδ＊を基にして電圧の大きさＶａを演算する電圧演算手段１５と、電圧演算手段１５より得られた電圧の大きさＶａ、および電圧指令値演算手段１６により求められた電圧指令値Ｖａ＊とに基づいて駆動負荷５を駆動するための励磁電流指令値ｉγ＊を求める励磁電流指令値演算手段１７と、によって電流位相制御手段１４を構成したので、駆動負荷に流れる電流値が所定値以下となるような電流位相角への追従性がよく、短時間で効率の良い状態に制御できる。また、電流値が所定値以下となる電流位相角付近での収束性および安定性が良いので、制御性がよく低電流で高効率なインバータ装置が実現できる。また、励磁電流指令値のみを制御するため、制御に用いる値として電圧ベクトルの大きさＶａ、Ｖａ＊や電流ベクトルの大きさＩａを用いることで制御軸（γ−δ軸）とモータ軸（ｄ−ｑ軸）との位相差を考慮する必要がなく、さらに、速度制御器や電流制御器を持たない場合でも、低電流で高効率なインバータ装置が実現できる。
【０１０２】
また、本実施の形態では、周波数変動や負荷変動により同期電動機の運転条件が変化した場合においても、その条件に応じて出力トルクが所定値以上（あるいは最大）となる電流位相角へ電流を制御することが可能となるので、高出力なインバータ装置や同期電動機駆動装置が得られる。
【０１０３】
また、常に電流が所定値以下（あるいは最小）となるように電流位相角を制御するので、電流最小位相角への追従性がよく、また、電流最小位相角付近での収束性および安定性が良いため、制御性のよいインバータ装置や同期電動機駆動装置が得られる。また、励磁電流指令値のみを制御するため、制御に用いる値として電圧ベクトルの大きさＶａ、Ｖａ＊や電流ベクトルの大きさＩａを用いることで制御軸（γ−δ軸）とモータ軸（ｄ−ｑ軸）との位相差を考慮する必要がなく、さらに、速度制御器や電流制御器を持たない場合でも、電流が所定値以下（あるいは最小）となる電流位相制御が実現できるインバータ装置や同期電動機駆動装置が得られる。
【０１０４】
また、本実施の形態では、回転子位置を検出するための位置センサを用いない位置センサレスで電動機を駆動する場合であるため、位置センサが不要で低コストで信頼性が高く、かつ、電流が所定値以下（あるいは最小）となる電流位相制御が実現できるので、高効率なインバータ装置や同期電動機駆動装置が得られる。また、位置センサが不要なので、位置センサを取り付けることができない圧縮機などに使用される同期電動機においても高出力で高効率な高効率な制御の行なえるインバータ装置や同期電動機駆動装置を実現できる。
【０１０５】
また、本実施の形態では、出力電圧演算手段１５にて用いる電圧信号は出力電圧指令値演算手段１０より得られる前回の出力電圧指令値Ｖγ＊およびＶδ＊でなくともよく、実際にインバータ主回路３から出力される電圧値を電圧検出手段（図示せず）を設けて検出しても良い旨を説明したが、この場合は、電圧検出手段（図示せず）を設けてインバータ主回路３から出力される電圧値を検出した後に、３相２相変換手段を用いてγ-δ軸上の電圧値Ｖγ、Ｖδに変換し、この変換した電圧値Ｖγ、Ｖδを用いればよい。実際の電圧値を検出して使用することで、より精度よく駆動負荷である同期電動機５を制御することができ、高効率で信頼性の高い装置を得ることができる。
【０１０６】
また、図１において、インバータ制御手段２内の電流位相制御手段１４での演算周期と電流位相制御手段１４以外での演算処理の周期を異なる周期としてもよい。インバータ制御手段２内の電流位相制御手段１４内での演算は、電流位相制御手段１４以外の演算処理に比べて早い応答性を必要としないため、電流位相制御手段１４による演算周期（励磁電流制御周期）を電流位相制御手段１４以外の制御周期（出力電圧制御周期）よりも大きくしてもよい。すなわち、電流位相制御手段１４での演算処理周期（励磁電流制御周期）を電流位相制御手段１４以外の演算処理周期（出力電圧制御周期）の数回〜数百回に１回程度にすればよい。このようにすることでインバータ制御手段２の演算処理をマイコン等で実現する場合、マイコンでの処理負荷の低減が可能となり、低コストなインバータ制御手段が得られ、さらには低コストな電動機駆動装置を得ることができる。
【０１０７】
次に、図５を用いて本発明の実施の形態1における同期電動機の制御方法の一例を説明する。図５は本実施の形態を表す同期電動機を所定値以下の電流で高効率駆動するための制御方法を示すフローチャート図である。図において、ＳＴ１は同期電動機に流れる電流を検出し、２相座標系の電流値ｉγ、ｉδを演算する電流検出・演算ステップ、ＳＴ２は電流検出・演算ステップＳＴ１により得られた電流信号を基に電流が所定値以下（あるいは最小）となる電流位相角βｍを電流位相演算手段２２にて求める電流位相演算ステップ、ＳＴ３は、周波数演算手段９により得られた前回の補償された周波数指令値である一次周波数ω１、および電流位相演算ステップＳＴ２により得られた電流位相角βｍとを基にして電圧指令値Ｖａ＊を求める電圧指令値演算ステップ、ＳＴ４は同期電動機に印加される電圧の大きさＶａを演算する電圧演算ステップ、ＳＴ５は基準励磁電流指令値ｉγ０を求める基準励磁電流演算ステップである。
【０１０８】
ＳＴ６は実際の電流位相角を、電圧指令値演算ステップＳＴ３にて得られた電圧指令値Ｖａ＊、および電圧演算ステップＳＴ２にて得られた電圧の大きさＶａを基にして、電流が所定値以下（あるいは最小）となる電流位相角βｍで動機電動機５を駆動するための励磁電流指令値ｉγ＊を求める励磁電流指令値演算ステップ、ＳＴ７はＳＴ６にて得られた励磁電流指令値ｉγ＊をフィルタ手段１９を通過させることによりフィルタリングするフィルタステップ、ＳＴ８はフィルタステップＳＴ７により得られたフィルタリングされた励磁電流指令値ｉγ＊に基づき同期電動機を駆動するための出力電圧指令値Ｖγ＊、Ｖδ＊を求める出力電圧指令値演算ステップである。
【０１０９】
まず、ＳＴ１にて駆動負荷である同期電動機５の少なくとも２相の電流を検出し、２相座標系の電流値に変換する。すなわち、電流検出手段６ａ、６ｂにより同期電動機５に流れる２相分の電流（たとえばＵ相電流ｉｕ、Ｖ相電流ｉｖ）を検出し、３相２相変換手段７により２相座標系の電流（たとえばγ軸電流ｉγ、δ軸電流ｉδ）に変換し、Ｉａを数式８にて演算する。
【０１１０】
ＳＴ２では、ＳＴ１で得られたγ軸電流ｉγ、δ軸電流ｉδを用いて電流値が所定値以下（あるいは最小）となる電流位相角βｍを電流位相演算手段２２にて数式６（あるいは数式５）を用いて演算する。（ここで、数式６や数式５のように電流位相角βｍまで求めなくとも数式１１のようにｓｉｎβｍ、ｃｏｓβｍまで求めればよいので、数式１１にてｓｉｎβｍ、ｃｏｓβｍまで求めて、演算負荷の低減を図ってもよい。）
【０１１１】
ＳＴ３では、周波数演算手段９により得られた前回の補償された周波数指令値である一次周波数ω１、およびＳＴ２により得られた電流位相角βｍとを基にして電圧指令値演算手段１６にて数式１０を使用して電圧指令値Ｖａ＊を求める。
【０１１２】
ＳＴ４では図１に示した出力電圧指令値演算手段１０などにより得られた電圧指令値より駆動負荷に印加される電圧値を電圧演算手段１５にて演算する。すなわち、ＳＴ４では、出力電圧指令値演算手段１０にて演算された前回の電圧指令値Ｖγ＊、Ｖδ＊を用いて同期電動機５を駆動するための電圧の大きさＶａを電圧演算手段１５にて数式７を使用して演算する。ここで、電圧指令値Ｖγ＊、Ｖδ＊は、電圧検出手段（図示せず）などを別途設けて駆動負荷５に印加される電圧を検出して、その電圧信号を使用してもよい。そうすると、常に最新の電圧情報が利用できるので、木目細かな信頼性の高い制御が行なえる同期電動機の制御方法が得られる。
【０１１３】
ＳＴ５では、周波数指令値（たとえば周波数演算手段９により得られた前回の補償された周波数指令値である一次周波数ω１）を基にして,基準励磁電流演算手段１８にて基準励磁電流指令値ｉγ０を求める。
【０１１４】
ＳＴ６では、ＳＴ３により得られた電圧指令値Ｖａ＊、ＳＴ４により得られた電圧の大きさＶａ、およびＳＴ５により得られた基準励磁電流指令値ｉγ＊とを基にして電流位相角βｍで駆動負荷５を駆動するための励磁電流指令値ｉγ＊を励磁電流指令値演算手段１７にて数式１２（あるいは数式１３）を使用して求める。そして、ＳＴ７では、ＳＴ６にて得られた励磁電流指令値ｉγ＊をローパスフィルタなどのフィルタ手段１９を通過させて励磁電流指令値ｉγ＊の高周波成分を除去して演算周期毎のばらつきを小さくして平均化させ、演算の安定性および制御の安定性を向上させている。ここで、ローパスフィルタなどのフィルタ手段１９を設けないと励磁電流指令値ｉγ＊を平均化させることができないので、ばらつきが大きくなり精度よく安定した制御ができない。
【０１１５】
ＳＴ８では、ＳＴ７により得られたフィルタ手段１９によりフィルタリングされた励磁電流指令値ｉγ＊に基づいて駆動負荷５を駆動するための出力電圧指令値Ｖγ＊、Ｖδ＊を出力電圧指令値演算手段１０にて求める。ここで、出力電圧指令値演算ステップＳＴ８で用いられる演算式は同期電動機の制御方式によって異なるので、Ｖγ＊、Ｖδ＊が演算できればどのような演算式でもよい。
【０１１６】
たとえば、電圧指令値演算手段１０の出力であるγ軸電圧指令値Ｖγ＊、δ軸電圧指令値Ｖδ＊を演算する数式の一例を数式１４に示す。
【０１１７】
【数１４】

【０１１８】
数式１４は、たとえば一次磁束制御による同期電動機の位置センサレス制御方式を用いた場合の演算式を表している。ここで、数式１４において、Φγ＊は一次磁束制御における一次磁束指令値、Φγerrは一次磁束誤差、Kγはγ軸制御ゲイン、Kδはδ軸制御ゲインである。
【０１１９】
以上のように、本実施の形態では、同期電動機に流れる電流値を基にして同期電動機の出力トルクが所定値以上となるような電流位相角を求める電流位相演算ステップＳＴ３と、同期電動機５に印加される電圧の大きさを演算する電圧演算ステップＳＴ４と、基準励磁電流指令値を求める基準励磁電流演算ステップＳＴ５と、電圧指令値演算演算ステップＳＴ３にて得られた電圧指令値Ｖａ＊、電圧演算ステップＳＴ４により得られた電圧の大きさＶａ、および基準励磁電流指令値演算ステップにより得られた基準励磁電流指令値ｉγ０とを基にして同期電動機５を駆動するための励磁電流指令値ｉγ＊を求める励磁電流指令値演算ステップＳＴ６と、励磁電流指令値演算ステップＳＴ６により得られた励磁電流指令値ｉγ＊をフィルタ手段１９にて高周波成分を除去して演算周期毎のばらつきを抑制するためのフィルタリングを行なうフィルタステップと、を備えたので、簡単な制御でありながら高出力、低電流で効率よく安定して制御の行なえる同期電動機の制御方法を得ることができる。
【０１２０】
また、回転周波数や負荷変動により同期電動機の運転条件が変化した場合においても、その条件に応じた出力トルクが所定値以上となる低電流の電流位相角で電流を制御でき、低電流でかつ高効率で同期電動機を駆動できる。また、低電流の位相への追従性がよく、低電流の位相付近での収束性および安定性が良いので、制御性が改善できる。また、座標軸を考慮する必要がなく励磁電流指令値のみを制御する簡単な制御であるため、速度制御器や電流制御器を持たない制御手段においても低電流、高効率で同期電動機を精度よく駆動できる制御方法が得られる。
【０１２１】
ここで、図５において、電圧演算ステップＳＴ４は、励磁電流指令値演算ステップＳＴ６よりも前のステップにて行なわれればどこで行なっても良く、ＳＴ１〜ＳＴ５までのどこで行なわれてもよい。また、出力電圧指令値演算ステップＳＴ８は励磁電流の制御のみでなく、回転周波数指令値や負荷トルクに応じた同期電動機駆動を行うための出力電圧指令値も求めるステップである。図５において、電流検出・演算ステップＳＴ１から励磁電流指令値演算ステップＳＴ６までの処理周期を処理周期Ａとし、インバータ制御手段２内の出力電圧指令値演算ステップＳＴ７以降の処理周期を処理周期Ｂとすると、処理周期Ａと処理周期Ｂは、処理を行うマイコン等の処理負荷低減のため異なる周期に設定してもよい。この場合、処理周期Ａは処理周期Ｂに比べて応答性を必要としないため、処理周期Ａを処理周期Ｂよりも長くした方がマイコンの負荷軽減が行なえるため、低コストのマイコンが使用でき、マイコンの選定の自由度が広がり、簡単な制御でありながら低コストで信頼性の高い電動機制御方法が得られる。
【０１２２】
また、実施の形態１にて説明した、電流が所定値以下（あるいは最小）となる電流位相角βmを求めずに電流最小の高効率制御を行う方法を図６を用いて説明する。図６は本発明の実施の形態１を表す別のインバータ装置および同期電動機駆動装置の構成を示す図である。図において、図１と同等部分は同一の符号を付して説明は省略する。
【０１２３】
図６において、１４０は電流位相制御手段であり、出力電圧の大きさＶａを求める出力電圧演算手段１５、２相３相座標変換手段７により得られたγ軸電流ｉγ、δ軸電流ｉδをもとにして電流の大きさＩａを演算する電流演算手段２１、電流値が所定値以下（あるいは最小）となる電流位相角βｍをｓｉｎβｍとして演算し、この電流位相角βｍにて駆動負荷５を駆動するための電圧指令値Ｖａ＊を求める電圧指令値演算手段１６０、電圧指令値Ｖａ＊と出力電圧の大きさＶａの誤差を演算する電圧誤差演算手段２２５、基準となる励磁電流指令値ｉγ０を求める基準励磁電流指令値演算手段１８、駆動負荷を駆動するための励磁電流指令値ｉγ＊を求める励磁電流指令値演算手段１７０、により構成される。
【０１２４】
したがって、図６に示す構成の装置では、電流検出手段６ａ、６ｂにより検出された電流値あるいは電流信号、および同期電動機５に印加される電圧信号（たとえば出力電圧信号Ｖγ＊、Ｖδ＊やその大きさＶａ）に基づいて同期電動機５の出力トルクが所定値以上となるような電流位相角βｍで駆動するための電圧指令値Ｖａ＊を求める電圧指令値演算手段１６０と、電圧指令値Ｖａ＊と電圧信号Ｖａとを比較して電圧誤差を求める電圧誤差演算手段２２５と、電圧誤差演算手段２２５より得られる電圧誤差から同期電動機５の出力トルクが所定値以上となる電流位相角βｍで駆動するための励磁電流指令値ｉγ＊を演算する励磁電流指令値演算手段１７０と、によって、電流位相制御手段１４０を構成しており、電圧誤差を演算する電圧誤差演算手段２２５を有するため、演算式が複雑化せず簡略化した演算式にて簡単に演算でき、演算時間の短縮化が行える。また、回転周波数や負荷変動により同期電動機の運転条件が変化した場合においても、その条件に応じた出力トルクが所定値以上となる低電流の電流位相角で電流を制御でき、低電流で高効率な同期電動機駆動装置が実現できる。
【０１２５】
また、電流値が所定値以下となる電流位相角βｍへの追従性がよく、この電流位相角付近での収束性および安定性の良い低電流で高効率な同期電動機駆動装置が実現できる。また、座標軸を考慮する必要がなく励磁電流指令値のみを制御するため、速度制御器や電流制御器を持たない制御手段においても良好な低電流で高効率な同期電動機駆動装置が実現できる。
【０１２６】
次に動作をフローチャート図を用いて説明する。図７は本実施の形態の同期電動機の別の制御方法を示すフローチャート図である。図において、ＳＴ１１は同期電動機に流れる電流を検出し、電流の大きさを求める電流検出・演算ステップ、ＳＴ１２は電流検出・演算ステップＳＴ１１で得られた電流信号および同期電動機の回転周波数より電流が所定値以下（あるいは最小）となる電流位相角で駆動負荷を駆動するための出力電圧指令値Ｖａ＊を求める電圧指令値演算ステップである。
【０１２７】
ＳＴ１３は同期電動機に印加される電圧の大きさＶａを演算する電圧演算ステップ、ＳＴ１４は電圧指令値演算ステップＳＴ１２より得られた電圧指令値Ｖａ＊と電圧演算ステップＳＴ１３で得られた電圧の大きさＶａとの差（Ｖａ＊−Ｖａ）を求める電圧誤差演算ステップ、ＳＴ１５は基準励磁電流指令値ｉγ０を演算する基準励磁電流指令値演算ステップ、ＳＴ１６は基準励磁電流指令値演算ステップＳＴ１５により得られた基準励磁電流指令値ｉγ０、および電圧誤差演算ステップＳＴ１４により得られた電圧誤差（Ｖａ＊−Ｖａ）とを基にして励磁電流指令値ｉγ＊を求める励磁電流指令値演算ステップ、ＳＴ１７は励磁電流指令値演算ステップＳＴ１６により得られた励磁電流指令値ｉγ＊をフィルタ手段１９により高周波成分を除去するフィルタリングを行なうフィルタステップ、ＳＴ１８はフィルタステップＳＴ１７によりフィルタリングされた励磁電流指令値ｉγ＊に基づいて同期電動機５を駆動するための出力電圧指令値Ｖγ＊、Ｖδ＊を求める出力電圧指令値演算ステップである。
【０１２８】
まず、ＳＴ１１にて駆動負荷である同期電動機５の少なくとも２相の電流を検出し、２相座標系の電流値に変換し、その大きさを求める。すなわち、電流検出手段６ａ、６ｂにより同期電動機５に流れる２相分の電流（たとえばＵ相電流ｉｕ、Ｖ相電流ｉｖ）を検出し、３相２相変換手段７により２相座標系の電流（たとえばγ軸電流ｉγ、δ軸電流ｉδ）に変換し、大きさＩａを数式８にて演算する。
【０１２９】
ＳＴ１２では、ＳＴ１１で得られたγ−δ軸電流ｉγ、ｉδ、および電流の大きさＩａを用いて電流値が所定値以下（あるいは最小）となる電流位相角βｍを数式１１にてｓｉｎβｍ、ｃｏｓβｍを求め、電圧指令値演算手段１６０にて数式１０を用いて電圧指令値Ｖａ＊を演算する。
【０１３０】
そして、ＳＴ１３にて出力電圧指令値演算手段１０などにより得られた電圧指令値より駆動負荷に印加される電圧値を電圧演算手段１５にて演算する。すなわち、ＳＴ１３では、出力電圧指令値演算手段１０にて演算された前回の電圧指令値Ｖγ＊、Ｖδ＊を用いて同期電動機５を駆動するための電圧の大きさＶａを電圧演算手段１５にて数式７を使用して演算する。ここで、ＳＴ１３では、電圧指令値として駆動負荷５に印加される電圧を電圧検出手段などを別途設けて検出して、その電圧信号を使用してもよい。そうすると、常に最新の電圧情報が利用できるので、木目細かな信頼性の高い制御が行なえる装置が得られる。
【０１３１】
ＳＴ１４ではＳＴ１２で得られた電圧指令値Ｖａ＊およびＳＴ１３にて得られた電圧の大きさＶａより電圧誤差演算手段２２５にて電圧誤差（Ｖａ＊−Ｖａ）を求める。そして、ＳＴ１５にて、周波数演算手段９の前回の出力値である補償された周波数指令値（一次周波数）ω１を本にして基準励磁電流指令値ｉγ０を演算する。ＳＴ１６では、ＳＴ１５にて得られた基準励磁電流指令値ｉγ０、ＳＴ１４にて得られた電圧誤差（Ｖａ＊−Ｖａ）とを基にして電流位相角がβｍとなる電流値で駆動負荷５を駆動するための励磁電流指令値ｉγ＊を励磁電流指令値演算手段１７０にて数式１２（あるいは数式１３）を使用して求める。ここで、本実施の形態では、ＳＴ１４にて電圧誤差（Ｖａ＊−Ｖａ）を直接演算により求めているので、ＳＴ１６での励磁電流指令値ｉγ＊の演算が簡略化され、演算の高速化が行える。
【０１３２】
ＳＴ１７では、ＳＴ１６により求められた励磁電流指令値ｉγ＊をフィルタ手段１９にて高周波成分を除去し、演算周期毎のばらつきを抑制するためのフィルタリングを行なう。ＳＴ１８では、ＳＴ１７にてフィルタリングされた励磁電流指令値ｉγ＊に基づいて駆動負荷５を駆動するための出力電圧指令値Ｖγ＊、Ｖδ＊を出力電圧指令値演算手段１０にて求める。ここで、出力電圧指令値演算ステップＳＴ１８で用いられる演算式は同期電動機の制御方式によって異なるので、Ｖγ＊、Ｖδ＊が演算できればどのような演算式でもよく、たとえば数式１４が使用される。
【０１３３】
以上のように、本実施の形態では、同期電動機に流れる電流値を基にして同期電動機の出力トルクが所定値以上となるような電流位相角で同期電動機を駆動するための電圧指令値を求める電圧指令値演算ステップＳＴ１２と、電圧指令値演算ステップＳＴ１２にて得られた電圧指令値Ｖａ＊と、同期電動機に印加される電圧値の大きさＶａを演算する電圧演算ステップＳＴ１３と、電圧指令値演算ステップＳＴ１２により得られた電圧指令値Ｖａ＊と電圧演算ステップＳＴ１３により得られた電圧の大きさＶＡとを比較して電圧誤差（Ｖａ＊−ＶＡ）を求める電圧誤差演算ステップＳＴ１４と、基準励磁電流指令値ｉγ０を求める基準励磁電流指令値演算ステップと、電圧誤差演算ステップＳＴ１４にて得られた電圧誤差（Ｖａ＊−ＶＡ）、および基準励磁電流演算ステップＳＴ１５により得られた基準励磁電流指令値ｉγ０とを基にして同期電動機の出力トルクが所定値以上となるような電流位相角βｍで同期電動機５を駆動するための励磁電流指令値ｉγ＊を演算する励磁電流指令演算ステップＳＴ１６と、励磁電流指令値演算ステップＳＴ１６により得られた励磁電流指令値ｉγ＊をフィルタリングするフィルタステップＳＴ１７と、フィルタステップＳＴ１７によりフィルタリングされた励磁電流指令値ｉγ＊を基にして同期電動機を駆動するための出力電圧指令値Ｖγ＊、Ｖδ＊を求める出力電圧指令値演算ステップＳＴ１８と、を備えたので、電圧誤差を求める簡単な制御でありながら制御性がよく、高出力・低電流で効率よく同期電動機を駆動できる同期電動機の制御方法を得ることができる。また、フィルタステップを有するので、同期電動機を安定した状態で制御できる信頼性の高い同期電動機の制御方法を得ることができる。
【０１３４】
また、同期電動機に流れる電流に基づいて得られる周波数補償量により補償された周波数指令値（一次周波数）ω１と、励磁電流指令値演算ステップＳＴ６、ＳＴ１６により得られた励磁電流指令値ｉγ＊と、に基づいて同期電動機を駆動するための出力電圧指令値Ｖγ＊、Ｖδ＊を求める出力電圧指令値演算ステップＳＴ１８を備えたので、電流が所定値以下となる電流位相角βｍでの励磁電流指令値ｉγ＊および補償された周波数指令値ω１に基づいて出力電圧指令値Ｖγ＊、Ｖδ＊を演算するため、回転周波数や負荷変動により同期電動機の運転条件が変化した場合においても、精度良く追従でき、安定した制御の行なえる同期電動機の制御方法を得ることができる。
【０１３５】
また、回転周波数や負荷変動により同期電動機の運転条件が変化した場合においても、その条件に応じて電流値が所定値以下となる低電流の電流位相角で電流を制御でき、低電流でかつ高効率で同期電動機を駆動できる。また、低電流の位相への追従性がよく、低電流の位相付近での収束性および安定性が良いので、制御性が改善できる。また、座標軸を考慮する必要がなく励磁電流指令値のみを制御するため、速度制御器や電流制御器を持たない制御手段においても低電流、高効率で同期電動機を駆動できる制御方法が得られる。
【０１３６】
ここで、図７において、電圧演算ステップＳＴ１３は、電圧誤差演算ステップＳＴ１４より前のステップであればどこで行なってもよい。また、出力電圧指令値演算ステップＳＴ１８は励磁電流の制御のみでなく、回転周波数指令値や負荷トルクに応じた同期電動機駆動を行うための出力電圧指令値も求めるステップである。図７において、電流検出・演算ステップＳＴ１１から励磁電流指令値演算ステップＳＴ１６までの処理周期を処理周期Ｘとし、インバータ制御手段２内のフィルタステップＳＴ１７以降の処理周期を処理周期Ｙとすると、処理周期Ｘと処理周期Ｙは、処理を行うマイコン等の処理負荷低減のため異なる周期に設定してもよい。この場合、処理周期Ｘは処理周期Ｙに比べて応答性を必要としないため、処理周期Ｘを処理周期Ｙよりも長くした方がマイコンの負荷軽減が行なえるため、低コストのマイコンが使用でき、マイコンの選定の自由度が広がり、簡単な制御でありながら低コストで信頼性の高い電動機制御方法が得られる。
【０１３７】
本発明の実施の形態１に示したインバータ装置や同期電動機駆動装置や同期電動機制御方法は、同期電動機駆動装置（あるいはインバータ装置）と駆動負荷である同期電動機が１：１で接続された場合の関係を一例として示したものであり、このような場合においては、駆動負荷である同期電動機５内に本実施の形態の同期電動機駆動装置および制御方法を同期電動機に一体に組み込むことで、駆動装置や制御方法を備えた高効率で信頼性の高い同期電動機を得ることができる。
【０１３８】
実施の形態２．
本発明の実施の形態２における同期電動機駆動装置について以下に説明する。図８は本発明の実施の形態２を表すインバータ装置および同期電動機駆動装置の構成を示す図である。図において、実施の形態１の図１〜図７と同等部分は同一の符号を付して説明は省略する。図において、１は同期電動機駆動装置であり、複数のスイッチング素子から構成され、直流を交流に変換するインバータ主回路３、インバータ主回路３の複数のスイッチング素子をオン・オフ制御するためのＰＷＭ信号を出力するインバータ制御手段２により構成される。
【０１３９】
また、４はインバータ主回路３に接続された直流電源、５はインバータ主回路３により駆動される駆動負荷である同期電動機であり、同期電動機駆動装置１、直流電源４、駆動負荷５によりインバータ装置が構成される。インバータ制御手段２が出力するＰＷＭ信号を基にインバータ主回路３内のスイッチング素子がオン・オフ動作し、直流電源４の直流電圧が３相交流に変換されて駆動負荷である同期電動機５に印加されることによって、インバータ装置が動作する。
【０１４０】
６ａは駆動負荷である同期電動機５に流入する電流のうちの一相の電流（たとえばＵ相電流ｉｕ）を検出する電流検出手段、６ｂは電流検出手段６ａで検出される電流（Ｕ相電流ｉｕ）とは異なる相の電流（たとえばＶ相電流ｉｖ）を検出する電流検出手段である。この電流検出手段６ａ、６ｂは検出した電流を電流値あるいは電流信号に変換して出力する。
【０１４１】
ここで、インバータ制御手段２は、以下に示す３相２相変換手段７、回転子位置・速度推定手段２０、電流位相制御手段１４５、周波数比較手段２１０、速度制御手段２２０、γ軸電流比較手段２３０、δ軸電流比較手段２４０、電流制御手段２５０、２相３相変換手段１１、ＰＷＭ信号発生手段１２、フィルタ手段１９とにより構成されている。
【０１４２】
３相２相変換手段７は電流検出手段６ａ、６ｂにより検出された２相分の電流値（たとえばＵ相電流ｉｕ、Ｖ相電流ｉｖ）あるいは電流信号を電気角位相θを用いて励磁電流成分（γ軸電流ｉγ）およびトルク電流成分（δ軸電流ｉδ）で表わされるγ-δ軸の電流値あるいは電流信号に変換する。
【０１４３】
回転子位置・速度推定手段２０は３相２相変換手段７より得られたγ-δ軸電流ｉγ、ｉδおよび前回の出力電圧指令値Ｖγ＊、Ｖδ＊とから駆動負荷である同期電動機５の回転子位値θおよび回転速度(回転周波数)ωｍを推定する。周波数比較手段２１０は外部から与えられる周波数指令値ωｍ＊と回転子位置・速度推定手段２０により得られた回転子回転周波数ωｍとを比較し周波数の差（ωｍ＊−ωｍ）を演算する。速度制御手段２２０は周波数比較手段２１０の出力（ωｍ＊−ωｍ）を基にδ軸電流指令値ｉδ＊を求める。
【０１４４】
γ軸電流比較手段２３０は、電流位相制御手段１４５の出力である励磁電流指令値（γ軸電流指令値ともいう。）ｉγ＊をフィルタ手段１９によりフィルタリングした励磁電流指令値ｉγ＊と座標変換手段７により得られたγ軸電流ｉγとを比較し、γ軸電流差（ｉγ＊−ｉγ）を演算する。δ軸電流比較手段２４０は、速度制御手段２２０により得られたδ軸電流指令値ｉδ＊と座標変換手段７により得られたδ軸電流ｉδとを比較し、δ軸電流差（ｉδ＊−ｉδ）を演算する。電流制御手段２５０は、γ軸電流比較手段２３０の出力値（ｉγ＊−ｉγ）およびδ軸電流比較手段２４０の出力値（ｉδ＊−ｉδ）とを基にして出力電圧指令値Ｖγ＊、Ｖδ＊を求める。
【０１４５】
電流位相制御手段１４５は、出力電圧の大きさを求める出力電圧演算手段１５、電流が所定値以下（あるいは最小）となる電圧指令値を求める電圧指令値演算手段１６５、電流が所定値以下（あるいは最小）となるように駆動負荷を駆動するための励磁電流指令値を求める励磁電流指令値演算手段１７５、基準となる励磁電流指令値を求める基準励磁電流指令値演算手段１８とにより構成され、励磁電流指令値演算手段１７５より励磁電流指令値ｉγ＊が出力される。
【０１４６】
また、１９は実施の形態１で説明したフィルタ手段と同等であり、電流位相制御手段１４５の出力である励磁電流指令値ｉγ＊をローパスフィルタなどにより高周波成分を除去して演算周期毎のばらつきを抑制するためのフィルタリングを行なうフィルタ手段である。本実施の形態では、実施の形態１でも説明したように、フィルタ手段１９を設けて高周波成分を除去するフィルタリングを行なうようにしているので、出力電圧指令値Ｖγ＊、Ｖδ＊の演算の安定性および演算精度を向上させることができ、励磁電流指令値ｉγ＊の変動による同期電動機５の動作の不安定を回避でき、制御性がよく、信頼性の高い装置を得ることができる。
【０１４７】
１１は電流制御手段２５０により得られたγ-δ軸電圧指令値Ｖγ＊、Ｖδ＊を回転子位置・速度推定手段２０により得られた電気角位相θを用いて３相電圧指令値Ｖｕ＊、Ｖｖ＊、Ｖｗ＊に変換する２相３相変換手段、１２は２相３相変換手段１１より得られる３相電圧指令値Ｖｕ＊、Ｖｖ＊、Ｖｗ＊をインバータ主回路３内のスイッチング素子をオン・オフ制御するためのＰＷＭ信号を発生するＰＷＭ信号発生手段である。
【０１４８】
次に動作について図８を用いて説明する。図において、同期電動機駆動装置１はインバータの駆動負荷である同期電動機５に流入する相電流のうち２相分の電流（たとえば、Ｕ相電流ｉｕとＶ相電流ｉｖ）を電流検出手段６ａ、６ｂより検出する。電流検出手段６ａ、６ｂにより検出した２相分の電流（例えばＵ相電流ｉｕおよびＶ相電流ｉｖ）あるいは２相分の電流信号を用いて、インバータ制御手段２は、同期電動機５を駆動するためにインバータ主回路３が出力する電圧（あるいは電圧信号）を演算により求め、インバータ主回路３内のスイッチング素子をオン・オフ制御するためのＰＷＭ信号を出力する。
【０１４９】
すなわち、インバータ制御手段２においては、以下に示す動作にてＰＷＭ信号を出力する。電流検出手段６ａ、６ｂにより検出された相電流ｉｕ、ｉｖより電気角位相θを用いて３相２相変換手段７よりγ-δ軸電流ｉγ、ｉδを求める。回転子位置・速度推定手段２０は３相２相変換手段７により得られたγ-δ軸電流ｉγ、ｉδおよび電流制御手段２５０の前回の出力値であるγ-δ軸電圧指令値Ｖγ＊、Ｖδ＊とにより同期電動機５の現在の回転子位置（置電気角位相）θおよび速度（回転周波数）ωｍを推測する。
【０１５０】
周波数比較手段２１０は、外部より与えられる回転周波数指令値ωm＊と回転子位置・速度推定手段２０の出力である回転周波数ωmとを比較し、その差Δωmを求める。速度制御手段２２０では、回転周波数の差Δωmより比例積分制御（速度制御）によりトルク電流指令値（δ軸電流指令値）ｉδ＊を求める。δ軸電流比較手段２４０ではトルク電流指令値ｉδ＊とδ軸電流ｉδとを比較し、その差（δ軸電流差）Δｉδを求める。
【０１５１】
電流位相制御手段１４５では、２相３相座標変換手段７により得られたγ軸電流ｉγ、δ軸電流ｉδ、および電流制御手段２５０の前回の出力であるＶγ＊、Ｖδ＊を基にして同期電動機を電流が所定値以下（あるいは最小）となる電流位相角βｍにて駆動するための励磁電流指令値ｉγ＊を出力する。以下に電圧指令値演算手段１６５、電圧演算手段１５、基準励磁電流指令値演算手段１８、励磁電流指令値演算手段１７５によって構成される電流位相制御手段１４５について説明する。
【０１５２】
電圧指令値演算手段１６５は、２相３相座標変換手段７により得られたγ軸電流ｉγ、δ軸電流ｉδ、および回転子位置・速度推定手段２０より得られた推定速度ωｍを基にして電流が所定値以下（あるいは最小）となる電流位相角βｍを演算し、この電流位相角βｍで同期電動機を駆動するための電圧指令値Ｖａ＊を演算する。電圧演算手段１５は電流制御手段２５０の前回の出力値である電圧指令値Ｖγ＊、Ｖδ＊をもとにしてその大きさＶａを演算する。また、基準励磁電流指令値演算手段１８では、回転子位置・速度推定手段２０より得られた推定速度ωｍを基にして基準励磁電流指令値ｉγ０を演算する。そして、励磁電流指令値演算手段１７５では、電圧指令値Ｖａ＊、電圧の大きさＶａ（あるいは電圧指令値Ｖａ＊と出力電圧指令値の電圧誤差（Ｖａ＊−Ｖａ））および基準励磁電流指令値演算手段１８より得られる基準励磁電流指令値ｉγ０と、を基にして同期電動機を電流が所定値以下（あるいは最小）で駆動するための励磁電流指令値ｉγ＊を出力する。このようにして電流位相制御手段１４５は、励磁電流指令値演算手段１７５により励磁電流指令値ｉγ＊を出力する。
【０１５３】
フィルタ手段１９は、電流位相制御手段１４５の出力である励磁電流指令値ｉγ＊の高周波成分を除去して演算周期毎のばらつきを抑制して安定した制御を行なうために励磁電流指令値ｉγ＊をローパスフィルタなどにより高周波成分を除去するフィルタリングを行なう。ここで、同期電動機の回転周波数および負荷トルクがほぼ変化しない定常運転状態であれば、励磁電流指令値ｉγ＊はほとんど変化しない。このような場合にフィルタ手段１９としてローパスフィルタを使用する場合は、ローパスフィルタの時定数を励磁電流制御周期（電流位相制御手段１４５での演算周期）に比べて十分大きな値とすれば良い。たとえば、ローパスフィルタの時定数は励磁電流制御周期（電流位相制御手段１４５での演算周期）や励磁電流演算以外の演算周期（たとえば電流位相制御手段１４５での演算周期を除いた電流制御手段２５０での演算周期など）を考慮して設計すればよく、励磁電流制御周期が１０ｍｓｅｃ程度の場合にはローパスフィルタの時定数を１ｓｅｃ程度に設定すればよい。
【０１５４】
γ軸電流比較手段２３０では、フィルタ手段１９によりフィルタリングされたγ軸電流指令値（励磁電流指令値）ｉγ＊と３相２相変換手段７により得られたγ軸電流値ｉγとを比較しその差（γ軸電流差）Δｉγを求める。電流制御手段２５０では、γ軸電流差Δｉγおよびδ軸電流差Δｉδより比例積分制御（電流制御）により出力電圧指令値Ｖγ＊、Ｖδ＊を求める。
【０１５５】
２相３相変換手段１１ではγ-δ軸電圧指令値Ｖγ＊、Ｖδ＊を電気角位相θを用いて３相座標系の電圧指令値Ｖｕ＊、Ｖｖ＊、Ｖｗ＊に変換する。ＰＷＭ信号発生手段１２は３相電圧指令値Ｖｕ＊、Ｖｖ＊、Ｖｗ＊よりインバータ主回路３内のスイッチング素子をオン・オフ制御するためのＰＷＭ信号を発生させる。このＰＷＭ信号に基づいてインバータ主回路３内のスイッチング素子がオン・オフ動作し、直流電源４の直流電圧が３相交流に変換され駆動負荷である同期電動機５に印加され、同期電動機５が駆動される。
【０１５６】
本実施の形態におけるインバータ装置および同期電動機駆動装置は、回転子位置を検出する位置センサを用いずに駆動する装置の別の例であり、実施の形態１のインバータ装置および同期電動機駆動装置とは異なっており、駆動負荷である同期電動機５をベクトル制御する場合の制御ブロック図の一例を表したものである。実施の形態1と異なる点は、回転子位置の推定を行なう回転子位置・速度推定手段２０と速度制御手段２２０と電流制御手段２５０を備えている点である。本実施の形態のように、速度制御器や電流制御器を備えた駆動装置であっても、同期電動機５をトルクが所定値以上（あるいは最大）の状態、すなわち電流が所定値以下（あるいは最小）の状態で高効率駆動することが可能である。電流が所定値以下（あるいは最小）となる高効率駆動の方法（電流位相制御方法）においては、実施の形態１の図５や図７にて説明した方法と同様の制御方法により制御すれば、簡単な制御でありながら低コスト・高効率・高出力で信頼性の高い電動機制御方法を得ることができる。
【０１５７】
また、本実施の形態では、同期電動機に流れる電流を検出する電流検出手段６ａ、６ｂと、電流検出手段６ａ、６ｂにより得られた電流信号および同期電動機５に印加される電圧の電圧信号に基づいて同期電動機５の出力トルクが所定値以上（あるいは最大）となるように同期電動機５に流れる電流の電流位相角を制御するための励磁電流指令値ｉγ＊を出力する電流位相制御手段１４５と、電流位相制御手段１４５により得られた励磁電流指令値ｉγ＊をフィルタリングするフィルタ手段１９と、電流検出手段６ａ、６ｂにより検出された電流あるいは電流信号より得られた推定速度ωｍによりトルク電流指令値ｉδ＊を演算する速度制御手段２２０と、速度制御手段２２０より得られたトルク電流指令値ｉδ＊およびフィルタ手段１９によりフィルタリングされた励磁電流指令値ｉγ＊とに基づいて同期電動機を出力トルクが所定値以上（あるいは最大）となるような電流位相角βｍにて駆動するための出力電圧指令値Ｖγ＊、Ｖδ＊を求める電流制御手段２５０と、を備えたので、トルク電流も制御するため速度制御性がよく、また、トルク制御が可能なため騒音、振動を抑制できる。また、フィルタ手段１９を備えているので、励磁電流指令値ｉγ＊の演算周期毎のばらつきを抑制でき、安定した制御の行える動機電動機駆動装置やインバータ装置が得られる。
【０１５８】
また、本実施の形態や実施の形態１においては、駆動負荷である同期電動機５を駆動する場合、起動から所定の回転数（低速回転）に到達するまでの間に電流位相制御手段１４、１４０、１４５を実行すると、制御実行時の同期電動機５の出力トルクの変動等により同期運転を維持できず脱調する場合がある。このため、同期電動機５が所定の回転数以上に到達するまでは、電流位相制御手段１４、１４０、１４５を実行しないようにすれば、脱調を回避することができ信頼性の高い装置を得ることができる。
【０１５９】
また、本実施の形態や実施の形態１においては、ＰＷＭ信号が過変調となる領域においては、過変調により出力電圧の制御が困難となるため、電流位相制御手段１４、１４０、１４５を実行する効果がほとんどなくなる。このため、過変調領域では、電流位相制御手段１４、１４０、１４５を実行しないようにして制御の簡素化を図っても良い。特に励磁電流の制御などに積分制御を用いている場合には、過変調時に積分器の動作が異常となるため、過変調領域での電流位相制御は行わないほうが安定した制御が行なえ、高効率で信頼性の高い装置を得ることができる。
【０１６０】
また、本実施の形態や実施の形態１においては、起動時や加速・減速運転時などの非定常運転状態にて電流位相制御手段１４、１４０、１４５を実行させると、制御が不安定になったり、あるいは脱調するような場合には、非定常運転状態（起動時や加減速時など）は基準励磁電流指令値演算手段１８より得られる基準励磁電流指令値ｉγ０を強めに設定し、この強めに設定されたｉγ０を励磁電流指令値ｉγ＊として（ｉγ＊＝ｉγ０として）使用すれば、制御が不安定になったり、あるいは脱調するようなことがなくなり、安定した制御の行なえる信頼性の高い装置を得ることができる（強め励磁制御）。（ここで、強め励磁とは、一般的に合成磁束ベクトルΦ０が回転子磁石による磁束ベクトルΦＦを強めるように励磁電流を流す場合（ｉｄ＞０の場合）をいい、逆に合成磁束ベクトルが回転子磁石による磁束ベクトルΦＦを弱めるように励磁電流を流す場合（ｉｄ＜０の場合）を弱め励磁という。）
【０１６１】
また、本実施の形態や実施の形態１において説明したインバータ装置や同期電動機駆動装置は、駆動負荷である同期電動機５の回転子位置を検出する位置センサを用いないセンサレス駆動方式の例であり、位置センサが不要のため低コストで信頼性の高い装置が得られる。また、本実施の形態にて説明した制御方式は、回転子位置を検出する駆動方式（位置センサ有り）においても実施可能であり、同等の効果を得ることができる。
【０１６２】
また、本実施の形態や実施の形態１において説明したインバータ装置や同期電動機駆動装置においては、電流が所定値以下（あるいは最小）となる高効率制御を行うための制御量（励磁電流指令値）の演算において、Ｒ（ＩＰＭＳＭの１相あたりの抵抗値）、Ｌｄ（ＩＰＭＳＭのｄ軸インダクタンス）、Ｌｑ（ＩＰＭＳＭのｑ軸インダクタンス）、Φｆ（ＩＰＭＳＭの誘起電圧定数）のような電動機の特性を示す電動機定数を用いて演算を行っている。このため電動機の発熱や周囲環境の変化などにより電動機の温度が変化すると、電動機定数が変化するため、実際の電動機定数と演算に用いている電動機定数との値に誤差が生じる場合がある。この誤差が生じると電流最小の高効率制御を行なっている場合は、電流最小となる動作ポイントが実際の電流最小ポイントからずれて、電流最小ポイントでの制御が困難となる。
【０１６３】
このような電動機定数の変化に対しても、電流最小の高効率駆動を実現するためには、たとえば実施の形態１で説明した出力電圧指令値演算手段１０や実施の形態２で説明した電流制御手段２５０内などに駆動負荷５に流れる電流値と電圧値などより電動機定数を測定する電動機定数測定手段や制御ゲインや周波数補償ゲインなどの制御定数を変更する制御定数変更手段などを設けるなどしてオンラインまたはオフラインにて、電動機定数測定手段にて電動機定数を測定して、電動機定数を温度などの周囲環境の変化に対して補正して、この補正された電動機定数に応じて制御定数変更手段にて制御定数を変更するようにすればよい。このようにすることで、電動機定数を電動機の特性の変化（温度変化など）に追従させて制御定数も変更することができるので、電動機の温度変化など周囲環境の変化により電動機定数に変動が生じた場合であっても、出力トルクが最大で電流最小の高効率・高出力制御が実現可能となり、高効率なインバータ装置や電動機駆動装置を得ることができる。
【０１６４】
また、本実施の形態や実施の形態１においては、同期電動機５として埋込磁石型同期電動機（ＩＰＭＳＭ）の例を示したが、同期電動機であればどのような電動機であっても実施の形態１や実施の形態２で説明した効果と同等の効果を得ることができる。例えば永久磁石型同期電動機（ＰＭＳＭ）、ブラシレスDCモータ（ＢＬＤＣＭ）などの永久磁石を有する同期電動機であれば、実施の形態１や実施の形態２と同等の数式を使用することにより実現できる。特に永久磁石型同期電動機（ＰＭＳＭ）において、表面磁石型同期電動機（ＳＰＭＳＭ）では、数式１〜数式１３中のｄ軸インダクタンスＬｄ、ｑ軸インダクタンスＬｑをＬｄ＝Ｌｑとすればよく、また、リラクタンスモータ（ＲＭ）、シンクロナスリラクタンスモータ（ＳｙＲＭ）、スイッチドリラクタンスモータ（ＳＲＭ）などの永久磁石を有さない同期電動機であれば数式１〜数式１３中の誘起電圧定数ΦｆをΦｆ＝０として演算すればよい。
【０１６５】
【発明の効果】
本発明の請求項１係る同期電動機駆動装置は、同期電動機に流れる電流を検出する電流検出手段と、電流検出手段により得られた電流信号および同期電動機に印加される電圧の電圧信号に基づいて同期電動機の出力トルクが所定値以上となるように同期電動機に流れる電流の電流位相角を制御するための励磁電流指令値を出力する電流位相制御手段と、電流位相制御手段により出力された励磁電流指令値をフィルタリングするフィルタ手段と、電流検出手段により検出された電流の変動量より周波数指令値に対する補償量を求め、補償量により補償された周波数指令値を出力する周波数演算手段と、周波数演算手段により補償された周波数指令値およびフィルタ手段によりフィルタリングされた励磁電流指令値に基づいて同期電動機を駆動するための出力電圧指令値を求める出力電圧指令値演算手段と、を備えたので、電流が所定値以下となる電流位相角での励磁電流指令値および補償された周波数指令値に基づいて出力電圧指令値を演算するため、回転周波数や負荷変動により同期電動機の運転条件が変化した場合においても、その条件での出力トルクに応じて電流値が所定値以下となる電流位相角で電流を制御でき、低電流で高効率な同期電動機駆動装置が実現できる。また、フィルタ手段１９を設けているので、演算毎の励磁電流指令値ｉγ＊のばらつきが小さくなり、安定した制御が行なえる。
【０１６６】
本発明の請求項２係る同期電動機駆動装置は、電流検出手段により得られた電流信号に基づいて同期電動機の出力トルクが所定値以上となるような電流位相角を求める電流位相演算手段と、電流位相演算手段により得られた電流位相角および周波数指令値とに基づいて電圧指令値を演算する電圧指令値演算手段と、同期電動機に印加される電圧の電圧信号より電圧の大きさを求める電圧演算手段と、電圧指令値演算手段により得られた電圧指令値および電圧演算手段により得られた電圧の大きさとに基づいて同期電動機を駆動するための励磁電流指令値を求める励磁電流指令値演算手段と、によって電流位相制御手段を構成したので、電流が所定値以下（あるいは最小）となる電流位相角での出力電圧と実際の出力電圧の差を求める簡単な構成でありながら、運転条件ごとに励磁電流指令値の数値テーブルや関数を持つ必要がないため、インバータ制御手段に用いられるマイコンのメモリ使用量を増加させることがなく、メモリ容量が小さい安価なマイコンが使用でき、また、簡単な制御回路で対応できるため、信頼性が高く安価な同期電動機駆動装置を得ることができる。
【０１６７】
本発明の請求項３係る同期電動機駆動装置は、電流検出手段により検出された電流信号および同期電動機に印加される電圧の電圧信号に基づいて同期電動機の出力トルクが所定値以上となるような電流位相角で駆動するための電圧指令値を求める電圧指令値演算手段と、同期電動機に印加される電圧の電圧信号より電圧の大きさを求める電圧演算手段と、電圧指令値演算手段により得られた電圧指令値と電圧演算手段により得られた電圧の大きさとを比較して電圧誤差を求める電圧誤差演算手段と、電圧誤差演算手段より得られる電圧誤差から同期電動機の出力トルクが所定値以上となる電流位相角で駆動するための励磁電流指令値を演算する励磁電流指令値演算手段と、によって、電流位相制御手段を構成したので、回転周波数や負荷変動により同期電動機の運転条件が変化した場合においても、その条件に応じた出力トルクが所定値以上となる低電流の電流位相角で電流を制御でき、低電流で高効率な同期電動機駆動装置が実現できる。また、電流値が所定値以下となる電流位相角への追従性がよく、この電流位相角付近での収束性および安定性の良い低電流で高効率な同期電動機駆動装置が実現できる。また、励磁電流指令値のみを制御するため、速度制御器や電流制御器を持たない制御手段においても良好な低電流で高効率な同期電動機駆動装置が実現できる。
【０１６８】
本発明の請求項４係る同期電動機駆動装置は、出力電圧指令値演算手段より得られる出力電圧指令値を電圧信号に使用するようにしたので、簡単な構成でありながら低電流で高効率な同期電動機駆動装置が実現できる。
【０１６９】
本発明の請求項５に記載の同期電動機駆動装置は、同期電動機に流れる電流を検出する電流検出手段と、電流検出手段により得られた電流信号および同期電動機に印加される電圧の電圧信号に基づいて同期電動機の出力トルクが所定値以上となるように同期電動機に流れる電流の電流位相角を制御するための励磁電流指令値を出力する電流位相制御手段と、電流位相制御手段により得られた励磁電流指令値をフィルタリングするフィルタ手段と、電流検出手段により検出された電流より得られた推定速度によりトルク電流指令値を演算する速度制御手段と、速度制御手段より得られたトルク電流指令値およびフィルタ手段によりフィルタリングされた励磁電流指令値とに基づいて同期電動機を出力トルクが所定値以上となるような電流位相角にて駆動するための出力電圧指令値を求める電流制御手段と、を備えたので、トルク電流も制御するため速度制御性がよく、また、トルク制御が可能なため騒音、振動を抑制できる。
【０１７０】
本発明の請求項６係る同期電動機駆動装置は、電流位相制御手段により制御する励磁電流制御周期を、同期電動機駆動のための出力電圧指令値を演算する出力電圧制御周期と異なるようにしたので、マイコンの処理負荷を低減できるため、安価なマイコンが使用でき、マイコンの選定の自由度が広がる。また、簡単な構成でありながら低電流で高効率な同期電動機駆動装置が実現できる。
【０１７１】
本発明の請求項７係る同期電動機駆動装置は、起動から所定回転数までの低速回転時には電流位相制御手段による励磁電流制御を行わないようにしたので、低速回転時に脱調を起こしにくい信頼性の高い高効率な同期電動機駆動装置を実現できる。
【０１７２】
本発明の請求項８係る同期電動機駆動装置は、ＰＷＭ変調が過変調となる領域では、電流位相制御手段による励磁電流制御を行わないようにしたので、過変調時に制御が不安定になることを抑制でき、信頼性の高い高効率な同期電動機駆動装置を実現できる。
【０１７３】
本発明の請求項９係る同期電動駆動装置は、起動時や加速・減速運転時には、電流位相制御手段を使用せずに所定の励磁電流指令値を用いて同期電動機を駆動し、所定の運転条件に達したのちは電流位相制御手段により求められる励磁電流指令値を使用するように切り替えるようにしたので、同期電動機の起動時や過渡時における制御の安定性を確保でき、信頼性の高い高効率な同期電動機駆動装置を実現できる。
【０１７４】
本発明の請求項１０に係る同期電動機駆動装置は、同期電動機の電動機定数を測定する電動機定数測定手段と、電動機定数測定手段により測定された電動機定数を基に同期電動機を制御する制御定数を変更する制御定数変更手段と、を備え、同期電動機の周囲環境の変化に応じて同期電動機を駆動制御するようにしたので、電動機の温度変化による特性の変化や、電動機を使用する環境の変化によりモータ定数などの電動機特性が変化した場合においても、自動的にモータ定数の変化に適応することが可能となり、低電流で高効率な同期電動機駆動装置を実現できる。
【０１７５】
本発明の請求項１１係る同期電動機駆動装置は、同期電動機のロータ位置を検出するための位置センサを用いないセンサレスで駆動するようにしたので、位置センサを取り付けることができない同期電動機においても高出力で高効率な制御の行なえる同期電動機駆動装置を実現できる。
【０１７６】
本発明の請求項１２に記載のインバータ装置は、請求項１乃至請求項１０のいずれかに記載の同期電動機駆動装置を備えたので、高出力で高効率なインバータ装置が得られる。
【０１７７】
本発明の請求項１３に係るインバータ装置は、スイッチング素子をオン、オフ動作させることにより直流電源よりの直流を交流に変換するインバータ主回路と、インバータ主回路内のスイッチング素子のオン、オフ動作を制御するための電圧指令値を出力するインバータ制御手段と、インバータ制御手段により制御されるインバータ主回路により出力される電圧が印加されることにより制御される駆動負荷と、を備え、インバータ制御手段を、駆動負荷を流れる電流に基づいて駆動負荷の出力トルクが所定値以上となるような電流位相角にて駆動負荷を駆動するための励磁電流指令値を演算する電流位相制御手段と、電流位相制御手段より得られた励磁電流指令値をフィルタリングするフィルタ手段と、駆動負荷を流れる電流の変動量より周波数指令値を補償して一次周波数として出力する周波数演算手段と、フィルタリングされた励磁電流指令値と一次周波数とに基づいて駆動負荷を駆動するための電圧指令値を演算する出力電圧指令値演算手段と、によって構成したので、駆動負荷の運転条件が変化しても、その条件での出力トルクが所定値以上となるように制御できる。また、出力トルクに応じて電流値が所定値以下となる電流位相角で電流を制御でき、低電流で高効率な同期電動機駆動装置が実現できる。また、フィルタ手段１９を設けているので、励磁電流指令値ｉγ＊のばらつきが少なくなり、駆動負荷を安定して制御できる信頼性の高いインバータ装置が得られる。
【０１７８】
本発明の請求項１４に係るインバータ装置は、駆動負荷を流れる電流により出力トルクが所定値以上となる電流位相角を求める電流位相演算手段と、電流位相演算手段により求められた電流位相角が得られるような電圧指令値を演算する電圧指令値演算手段と、出力電圧指令値演算手段により求められた出力電圧指令値および電圧指令値演算手段により求められた電圧指令値とに基づいて駆動負荷を駆動するための励磁電流指令値を求める励磁電流指令値演算手段と、によって電流位相制御手段を構成したので、駆動負荷に流れる電流値が所定値以下となるような電流位相角への追従性がよく、また、電流値が所定値以下となる電流位相角付近での収束性および安定性が良いので、制御性がよく低電流で高効率なインバータ装置が実現できる。また、励磁電流指令値のみを制御するため、速度制御器や電流制御器を持たない場合でも、低電流で高効率なインバータ装置が実現できる。
【０１７９】
本発明の請求項１５に係る同期電動機駆動装置の制御方法は、同期電動機に流れる電流値を基にして同期電動機の出力トルクが所定値以上となるような電流位相角を求める電流位相演算ステップと、電流位相演算ステップにて得られた電流位相角および同期電動機に印加される電圧値とを基にして同期電動機を駆動するための励磁電流指令値を求める励磁電流指令値演算ステップと、励磁電流指令値演算ステップにて得られた励磁電流指令値の高周波成分を除去するフィルタリングを行なうフィルタステップと、フィルタステップにてフィルタリングされた励磁電流指令値を基にして同期電動機を駆動するための出力電圧指令値を求める出力電圧指令値演算ステップと、を備えたので、簡単な制御でありながら高出力・低電流で効率よく安定して制御の行なえる同期電動機の制御方法を得ることができる。また、励磁電流指令値のみを制御する簡単な制御のため、速度制御器や電流制御器を持たない制御手段においても低電流、高効率で同期電動機を精度よく駆動できる制御方法が得られる。
【０１８０】
本発明の請求項１６に係る同期電動機駆動装置の制御方法は、同期電動機に流れる電流値を基にして同期電動機の出力トルクが所定値以上となるような電流位相角で同期電動機を駆動するための電圧指令値を求める電圧指令値演算ステップと、同期電動機に印加される電圧の大きさを演算する電圧演算ステップと、電圧演算ステップにて得られた電圧の大きさと電圧指令値演算ステップにて得られた電圧指令値とを比較して電圧誤差を求める電圧誤差演算ステップと、電圧誤差演算ステップにて得られた電圧誤差を基にして同期電動機の出力トルクが所定値以上となるような電流位相角で同期電動機を駆動するための励磁電流指令値を演算する励磁電流指令演算ステップと、励磁電流指令値演算ステップにより得られた励磁電流指令値の高周波成分を除去するフィルタリングを行なうフィルタステップと、フィルタステップにてフィルタリングされた励磁電流指令値を基にして同期電動機を駆動するための出力電圧指令値を求める出力電圧指令値演算ステップと、を備えたので、簡単な制御でありながら制御性がよく、低電流で同期電動機を駆動できる。また、回転周波数や負荷変動により同期電動機の運転条件が変化した場合においても、その条件に応じて電流が所定値以下となる低電流の電流位相角で電流を制御でき、低電流でかつ高効率で同期電動機を駆動できる。また、低電流の位相への追従性がよく、低電流の位相付近での収束性および安定性が良いので、制御性が改善できる。また、フィルタステップを有するので、同期電動機を安定した状態で制御できる信頼性の高い同期電動機の制御方法を得ることができる。
【０１８１】
本発明の請求項１７に係る同期電動機駆動装置の制御方法は、同期電動機に流れる電流に基づいて得られる周波数補償量により補償された周波数指令値と、励磁電流指令値演算ステップにより得られた励磁電流指令値と、に基づいて同期電動機を駆動するための出力電圧指令値を求める出力電圧指令値演算ステップを備えたので、電流が所定値以下となる電流位相角での励磁電流指令値および補償された周波数指令値に基づいて出力電圧指令値を演算するため、回転周波数や負荷変動により同期電動機の運転条件が変化した場合においても、精度良く追従でき、安定した制御がおこなえる。
【図面の簡単な説明】
【図１】 本発明の実施の形態1を表す同期電動機駆動装置の構成の一例を示す図である。
【図２】 本発明の実施の形態1における同期電動機の電流位相角と発生トルクとの関係を説明する図である。
【図３】 本発明の実施の形態1における同期電動機の電流位相角と電流との関係を出力トルクの変化に対して説明する図である。
【図４】 本発明の実施の形態1における一般的な同期電動機のベクトル線図を示す図である。
【図５】 本発明の実施の形態１における同期電動機駆動装置の制御方法を説明するフローチャート図である。
【図６】 本発明の実施の形態１を表す同期電動機駆動装置の別の構成の一例を示す図である。
【図７】 本発明の実施の形態１における同期電動機駆動装置の別の制御方法を説明するフローチャート図である。
【図８】 本発明の実施の形態２を表す同期電動機駆動装置の構成の一例を示す図である。
【図９】 従来の同期電動機駆動装置の構成の一例を示す図である。
【符号の説明】
１ 同期電動機駆動装置、２ インバータ制御手段、３ インバータ主回路、４ 直流電源、５ 同期電動機、６ａ、６ｂ 電流検出手段、７ ３相２相変換手段、８ 周波数補償器、９ 一次周波数演算手段、１０ 出力電圧指令値演算手段、１１ ２相３相座標変換手段、１２ PWM信号発生手段、１３ 電気角位相演算手段、１４ 電流位相制御手段、１５ 出力電圧演算手段、１６ 電圧指令値演算手段、１７ 励磁電流指令演算手段、１８ 基準励磁電流指令演算手段、１９ フィルタ手段、２０ 回転子位置・速度推定手段、２１ 電流演算手段、２２ 電流位相演算手段、１４０、１４５ 電流位相制御手段、１６０、１６５ 電圧指令値演算手段、１７０、１７５ 励磁電流指令値演算手段、２１０ 周波数比較手段、２２０ 速度制御手段、２２５ 電圧誤差演算手段、２３０ γ軸電流比較手段、２４０ δ軸電流比較手段、２５０ 電流制御手段。[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a synchronous motor driving device that drives a synchronous motor, an inverter device, and a method for controlling the synchronous motor.
[0002]
[Prior art]
FIG. 9 is a diagram showing a configuration of a general conventional synchronous motor driving apparatus. In the figure, 1 is a synchronous motor drive device, 2 is an inverter control means, 3 is an inverter main circuit composed of a plurality of switching elements, 4 is a DC power source connected to the inverter main circuit, and 5 is a synchronous motor.
[0003]
6a is a current detection means for detecting one-phase current (for example, U-phase current iu) out of the current flowing into the synchronous motor 5, and 6b is for detecting a current of a phase different from the current detection means 6a (for example, V-phase current iv). Current detecting means 7 for performing the current value detected by the current detecting means 6a, 6b using the electrical angle phase and representing the exciting current component (γ-axis current) and torque current component (δ-axis current) γ-δ-axis current 3 is a frequency compensator for obtaining a rotational frequency compensation amount ωd from the δ-axis current obtained from the 3-phase 2-phase conversion unit 7, and 9 is a frequency compensation amount ωd of the frequency compensator 8. Primary frequency computing means for obtaining a primary frequency ω1 from the rotational frequency command value ωm *, 13 an electrical angle phase computing means for integrating the primary frequency ω1 to obtain an electrical angle phase θ, and 50 for excitation from the rotational frequency command value ωm *. Current command value iγ * Is Mel excitation current command value calculating means.
[0004]
10 is synchronized with the γ-δ axis current obtained from the three-phase to two-phase conversion means 7, the primary frequency ω 1 obtained from the primary frequency calculation means 9, and the excitation current command value iγ * obtained from the excitation current calculation means 50. Output voltage command value calculation means 11 for calculating voltage command values Vγ * and Vδ * for driving the electric motor 5, 11 is a γ-δ axis voltage command value Vγ *, Vδ * obtained by the output voltage command value calculation means 10. Is converted into a three-phase voltage command value Vu *, Vv *, Vw * using the electrical angle phase θ, and 12 is a three-phase voltage command value Vu * obtained from the two-phase three-phase conversion means. This is a PWM signal generating means for generating a PWM signal for on / off control of the switching elements in the inverter main circuit 3 by Vv * and Vw *.
[0005]
The operations of the conventional synchronous motor drive device 1 and the synchronous motor 5 configured as described above will be described with reference to FIG. In the figure, the synchronous motor drive device 1 detects currents of two phases (for example, iu, iv) among the phase currents flowing into the synchronous motor 5 by the current detection means 6a, 6b. Using the detected two-phase currents, for example, the U-phase current iu and the V-phase current iv, the inverter control means 2 calculates the voltage output from the inverter main circuit 3 to drive the synchronous motor 5, and calculates the inverter A PWM signal for ON / OFF control of the switching element in the main circuit 3 is output.
[0006]
The inverter control means 2 outputs a PWM signal by the operation described below. From the phase currents iu and iv detected by the current detectors 6a and 6b, the γ-δ axis currents iγ and iδ are obtained from the three-phase / two-phase converter 7 using the electrical angle phase θ. The frequency compensator 8 obtains the frequency compensation amount ωd from the δ-axis current iδ. The primary frequency calculation means 9 obtains the primary frequency ω1 by taking the difference between the rotational frequency command value ωm * and the frequency compensation amount ωd. The excitation current calculation means 50 obtains an excitation current command value (γ-axis current command value) iγ * from the rotation frequency command value ωm *.
[0007]
The output voltage command value calculation means 10 obtains voltage command values Vγ * and Vδ * for driving the synchronous motor 5 from the primary frequency ω1, the excitation current command value iγ *, and the γ-δ axis currents iγ and iδ. The two-phase / three-phase conversion means 11 converts the γ-δ axis voltage command values Vγ * and Vδ * into voltage command values Vu *, Vv * and Vw * in a three-phase coordinate system using the electrical angle phase θ. The PWM signal generating means 12 generates a PWM signal for on / off control of the switching elements in the inverter main circuit 3 from the three-phase voltage command values Vu *, Vv *, Vw *. Based on this PWM signal, the switching element in the inverter main circuit 3 is turned on / off, and the DC voltage of the DC power supply 4 is converted into three-phase AC and applied to the synchronous motor 5.
[0008]
Here, in the conventional synchronous motor driving device, the excitation current calculation means 50 uses the data table corresponding to the rotation frequency command value ωm * or the function representing the relationship with the rotation frequency command value ωm * to determine the excitation current command value iγ *. Is required. The excitation current command value iγ * is determined based on a value obtained in advance by a theoretical formula or experiment in accordance with the rotation frequency command value ωm *.
[0009]
An example of a conventional synchronous motor driving device is disclosed in Japanese Patent Laid-Open No. 2000-209886. In the synchronous motor driving device disclosed in Japanese Patent Laid-Open No. 2000-209886, the energization phase of voltage is set in a direction in which the current decreases from the previous energization phase information and current information with respect to the rotor position of the synchronous motor. An example of approaching the current phase angle at which the current is minimized while changing with time is disclosed.
[0010]
An example of a conventional synchronous motor driving device is disclosed in Japanese Patent Laid-Open No. 2001-8486. In the synchronous motor driving device disclosed in Japanese Patent Application Laid-Open No. 2001-8486, in a device including a speed control device that performs speed control, when the load torque is constant, the excitation current id is changed to An example is disclosed in which the excitation current id is changed in the direction in which the current is reduced from the current information, and the phase approaches the phase where the current is minimized.
[0011]
An example of a conventional synchronous motor driving apparatus is disclosed in Japanese Patent Laid-Open No. 2001-86782. In the synchronous motor driving device disclosed in Japanese Patent Application Laid-Open No. 2001-86782, in which the speed is controlled by the speed control device, the motor constants of the d-axis inductance, the q-axis inductance, the torque constant, and the number of pole pairs of the brushless DC motor are set. The current phase angle β at which the generated torque is maximized is calculated, and the excitation current command value id * torque current command value iq *, which is the output of the speed controller, is set so that the current phase angle β at which the generated torque is maximized. An example of performing current control so that the actual current becomes the current command value is disclosed.
[0012]
[Problems to be solved by the invention]
In the conventional synchronous motor driving device, the excitation current calculation means 50 obtains the excitation current command value from a data table corresponding to the rotation frequency command value or a function representing the relationship with the rotation frequency command value. This excitation current command value is determined to be a value that can achieve the maximum efficiency at the rated load torque according to the rotation frequency command value (a value for realizing the current phase angle that minimizes the current). The maximum efficiency can be achieved only under the assumed rated load torque condition, but if there is a difference between the rated load torque condition and the actual load torque condition, the current phase angle will deviate from the maximum efficiency point. It was driven in bad condition.
[0013]
In addition, when the synchronous motor is loaded with a compressor of an air conditioner, a condition in which the load torque fluctuates, such as cooling operation or heating operation, occurs even at the same rotational frequency. For this reason, since there is only one excitation current command value under the same rotational frequency condition, it has been difficult to realize high-efficiency operation with a minimum current according to the load torque. In addition, if an excitation current command value condition is provided for each operating condition, the amount of memory used for a microprocessor or the like increases, which causes an increase in the cost of the control device. Also in this case, when the actual load torque condition deviates from the assumed rated load torque condition, maximum efficiency operation cannot be realized.
[0014]
In addition, in Japanese Patent Laid-Open No. 2000-209886 and Japanese Patent Laid-Open No. 2001-8486, the energization phase that minimizes the current is not obtained in the control process, and the current phase angle that reaches the minimum current is reached. It took time. In addition, since this method does not converge very well to the energization phase at which the current is minimized, the energization phase slightly changes in the vicinity of the energization phase at which the current minimum phase angle is reached, resulting in fluctuations in the motor current and stable operation. There were cases where it was not possible. In Japanese Patent Laid-Open No. 2001-8486, since the control is performed only when the load torque is constant, it cannot be used for a load in which the load torque constantly fluctuates.
[0015]
Further, in the synchronous motor driving device disclosed in Japanese Patent Application Laid-Open No. 2001-86782, the current that generates the maximum torque using the motor constants of the d-axis inductance, the q-axis inductance, the torque constant, and the number of pole pairs of the brushless DC motor. The phase angle β is calculated, the excitation current command value id * and the torque current command value iq *, which are the outputs of the speed controller, are calculated so that the current phase angle β is the maximum generated torque, and the actual current is the current command. The current is controlled so as to be a value, aiming at high efficiency. For this reason, a speed controller for obtaining id * and iq * is required, and this method cannot be used without a speed controller. Further, since the filter means is not provided, the variation of the excitation current command value for each calculation cycle becomes large, the operation of the synchronous motor becomes unstable, and the controllability is poor.
[0016]
The present invention has been made to solve the above-described problems, and an object thereof is to obtain a high-power inverter device or synchronous motor drive device. It is another object of the present invention to obtain an efficient inverter device or synchronous motor drive device. It is another object of the present invention to obtain an inverter device or a synchronous motor drive device with good controllability. Another object of the present invention is to obtain an inverter device or a synchronous motor drive device that can be controlled in an efficient state in a short time. It is another object of the present invention to obtain a low-cost inverter device or synchronous motor drive device. It is another object of the present invention to obtain a highly efficient inverter device or synchronous motor drive device that can be driven under the condition that the current becomes small even if the rotational frequency and load torque vary. It is another object of the present invention to obtain a method for controlling a synchronous motor that is simple and efficient.
[0017]
[Means for Solving the Problems]
The synchronous motor drive device according to claim 1 of the present invention is based on a current detection means for detecting a current flowing through the synchronous motor, a current signal obtained by the current detection means and a voltage signal of a voltage applied to the synchronous motor. Current phase control means for outputting an excitation current command value for controlling the current phase angle of the current flowing through the synchronous motor so that the output torque of the synchronous motor becomes a predetermined value or more, and the excitation output by the current phase control means A filter means for filtering the current command value, a frequency calculation means for obtaining a compensation amount for the frequency command value from the fluctuation amount of the current detected by the current detection means, and outputting a frequency command value compensated by the compensation amount, and a frequency calculation The synchronous motor is driven based on the frequency command value compensated by the means and the excitation current command value filtered by the filter means. , An output voltage command value calculating means for calculating an output voltage command value for those equipped with.
[0018]
The synchronous motor drive device according to claim 2 of the present invention includes current phase calculation means for obtaining a current phase angle such that the output torque of the synchronous motor becomes a predetermined value or more based on the current signal obtained by the current detection means. The voltage command value calculating means for calculating the voltage command value based on the current phase angle and the frequency command value obtained by the current phase calculating means, and the voltage magnitude is obtained from the voltage signal of the voltage applied to the synchronous motor. Excitation current command value calculation for obtaining an excitation current command value for driving the synchronous motor based on the voltage calculation means, the voltage command value obtained by the voltage command value calculation means, and the magnitude of the voltage obtained by the voltage calculation means And current phase control means.
[0019]
In the synchronous motor drive device according to claim 3 of the present invention, the output torque of the synchronous motor becomes a predetermined value or more based on the current signal detected by the current detection means and the voltage signal of the voltage applied to the synchronous motor. Voltage command value calculating means for obtaining a voltage command value for driving at a current phase angle, voltage calculating means for obtaining a voltage magnitude from a voltage signal of a voltage applied to the synchronous motor, and voltage command value calculating means. Voltage error calculation means for comparing the voltage command value obtained and the voltage magnitude obtained by the voltage calculation means to obtain a voltage error, and the output torque of the synchronous motor is equal to or greater than a predetermined value from the voltage error obtained from the voltage error calculation means The current phase control means is composed of excitation current command value calculation means for calculating an excitation current command value for driving at a current phase angle.
[0020]
According to a fourth aspect of the present invention, an output voltage command value obtained from an output voltage command value calculation means is used for a voltage signal.
[0021]
The synchronous motor drive device according to claim 5 of the present invention is based on a current detection means for detecting a current flowing through the synchronous motor, a current signal obtained by the current detection means, and a voltage signal of a voltage applied to the synchronous motor. Current phase control means for outputting an excitation current command value for controlling the current phase angle of the current flowing through the synchronous motor so that the output torque of the synchronous motor becomes equal to or greater than a predetermined value, and the excitation obtained by the current phase control means Filter means for filtering the current command value, speed control means for calculating the torque current command value based on the estimated speed obtained from the current detected by the current detection means, torque current command value and filter obtained from the speed control means Current phase angle at which the output torque of the synchronous motor exceeds the predetermined value based on the excitation current command value filtered by the means , A current control means for calculating an output voltage command value for driving Te is obtained with a.
[0022]
In the synchronous motor drive device according to claim 6 of the present invention, the excitation current control cycle controlled by the current phase control means is different from the output voltage control cycle for calculating the output voltage command value for driving the synchronous motor. Is.
[0023]
According to a seventh aspect of the present invention, there is provided a synchronous motor drive device that does not perform excitation current control by the current phase control means during low-speed rotation from startup to a predetermined rotational speed.
[0024]
The synchronous electric drive device according to claim 8 of the present invention is such that the excitation current control by the current phase control means is not performed in the region where the PWM modulation is overmodulated.
[0025]
The synchronous motor drive device according to claim 9 of the present invention drives the synchronous motor using a predetermined excitation current command value without using the current phase control means at the time of start-up or acceleration / deceleration operation, After reaching the operating condition, the excitation current command value obtained by the current phase control means is switched to use.
[0026]
The synchronous motor drive device according to claim 10 of the present invention includes a motor constant measuring means for measuring a motor constant of the synchronous motor, and a control constant for controlling the synchronous motor based on the motor constant measured by the motor constant measuring means. Control constant changing means for changing, and the synchronous motor is driven and controlled in accordance with a change in the surrounding environment of the synchronous motor.
[0027]
A synchronous motor drive device according to an eleventh aspect of the present invention is driven without a sensor without using a position sensor for detecting the rotor position of the synchronous motor.
[0028]
An inverter device according to a twelfth aspect of the present invention includes the synchronous motor driving device according to any one of the first to eleventh aspects.
[0029]
According to a thirteenth aspect of the present invention, there is provided an inverter main circuit that converts a direct current from a direct current power source into an alternating current by turning on and off the switching element, and an on / off operation of the switching element in the inverter main circuit. An inverter control means for outputting a voltage command value for controlling the inverter, and a drive load controlled by applying a voltage output by an inverter main circuit controlled by the inverter control means. A current phase control means for calculating an excitation current command value for driving the drive load at a current phase angle such that the output torque of the drive load becomes a predetermined value or more based on the current flowing through the drive load; and a current phase Filter means for filtering the excitation current command value obtained from the control means, and fluctuation amount of current flowing through the drive load Frequency calculation means for compensating the wave number command value and outputting it as a primary frequency, and output voltage command value calculation means for calculating a voltage command value for driving the drive load based on the filtered excitation current command value and the primary frequency It is constituted by.
[0030]
According to a fourteenth aspect of the present invention, there is provided an inverter device comprising: a current phase calculating means for obtaining a current phase angle at which an output torque becomes a predetermined value or more by a current flowing through a driving load; and a current phase angle obtained by the current phase calculating means. A drive load based on a voltage command value calculation means for calculating a voltage command value as obtained, an output voltage command value obtained by the output voltage command value calculation means, and a voltage command value obtained by the voltage command value calculation means Current phase control means is constituted by excitation current command value calculation means for obtaining an excitation current command value for driving the current.
[0031]
According to a fifteenth aspect of the present invention, there is provided a control method for a synchronous motor drive device, wherein a current phase calculation step for obtaining a current phase angle such that an output torque of the synchronous motor is equal to or greater than a predetermined value based on a current value flowing through the synchronous motor. An excitation current command value calculation step for obtaining an excitation current command value for driving the synchronous motor based on the current phase angle obtained in the current phase calculation step and the voltage value applied to the synchronous motor; A filter step that performs filtering to remove high frequency components of the excitation current command value obtained in the current command value calculation step, and an output for driving the synchronous motor based on the excitation current command value filtered in the filter step An output voltage command value calculating step for obtaining a voltage command value.
[0032]
According to a sixteenth aspect of the present invention, there is provided a control method for a synchronous motor driving apparatus that drives a synchronous motor at a current phase angle such that an output torque of the synchronous motor becomes a predetermined value or more based on a current value flowing through the synchronous motor. A voltage command value calculating step for obtaining a voltage command value, a voltage calculating step for calculating the magnitude of the voltage applied to the synchronous motor, and the voltage magnitude and voltage command value calculating step obtained in the voltage calculating step. A voltage error calculation step for comparing the voltage command value obtained in this way to obtain a voltage error and a voltage error obtained in the voltage error calculation step based on the voltage error so that the output torque of the synchronous motor is not less than a predetermined value. The excitation current command calculation step for calculating the excitation current command value for driving the synchronous motor with the current phase angle, and the excitation current command value obtained by the excitation current command value calculation step A filter step for performing filtering to remove the wave component, and an output voltage command value calculating step for obtaining an output voltage command value for driving the synchronous motor based on the excitation current command value filtered in the filter step. It is a thing.
[0033]
According to a seventeenth aspect of the present invention, there is provided a control method for a synchronous motor drive device comprising: a frequency command value compensated by a frequency compensation amount obtained based on a current flowing through the synchronous motor; and a filtered excitation obtained by a filter step. An output voltage command value calculation step for obtaining an output voltage command value for driving the synchronous motor based on the current command value is provided.
[0034]
DETAILED DESCRIPTION OF THE INVENTION
Embodiment 1 FIG.
An inverter device and a synchronous motor drive device representing Embodiment 1 of the present invention will be described below. FIG. 1 is a diagram showing a configuration of an inverter device and a synchronous motor drive device representing Embodiment 1 of the present invention. In the figure, reference numeral 1 denotes a synchronous motor drive device, which is composed of a plurality of switching elements, an inverter main circuit 3 for converting direct current to alternating current, and a PWM signal for on / off control of the plurality of switching elements of the inverter main circuit 3. It is comprised by the inverter control means 2 which outputs.
[0035]
Reference numeral 4 denotes a DC power source connected to the inverter main circuit 3, 5 denotes a synchronous motor which is a driving load driven by the inverter main circuit 3, and the inverter device is driven by the synchronous motor driving device 1, the DC power source 4 and the driving load 5. Is configured. The switching element in the inverter main circuit 3 is turned on / off based on the PWM signal output from the inverter control means 2, and the DC voltage of the DC power source 4 is converted into a three-phase AC and applied to the synchronous motor 5 as a driving load. As a result, the inverter device operates.
[0036]
Reference numeral 6a denotes current detection means for detecting one-phase current (for example, U-phase current iu) out of current flowing into the synchronous motor 5 as a driving load, and 6b denotes current detected by the current detection means 6a (U-phase current iu). ) Is a current detection means for detecting a current of a phase different from (). The current detection means 6a and 6b convert the detected current into a current value or a current signal and output it.
[0037]
Here, the inverter control means 2 includes the following three-phase two-phase conversion means 7, frequency compensator 8, primary frequency calculation means 9, output voltage command value calculation means 10, two-phase three-phase conversion means 11, PWM signal generation A means 12, an electrical angle phase calculation means 13, a current phase control means 14, and a filter means 19 are configured.
[0038]
The three-phase / two-phase conversion means 7 uses the current value (for example, iu, iv) or current signal detected by the current detection means 6a and 6b or the current signal using the electrical angle phase θ to generate an excitation current component (γ-axis current iγ) and a torque current component. The current value or current signal of the γ-δ axis represented by (δ-axis current iδ) is converted. The frequency compensator 8 calculates a rotational frequency compensation amount ωd from the δ-axis current obtained by the three-phase / two-phase conversion means 7. The primary frequency calculation means 9 uses the frequency compensation amount ωd obtained by the frequency compensator 8 to compensate the frequency command value ωm * that is a target frequency given from the outside (the primary frequency ω1 that is the compensated frequency command value). ). The electrical angle phase calculating means 13 integrates the primary frequency ω1, which is a compensated frequency command value, to determine the electrical angle phase θ.
[0039]
The current phase control means 14 is the output voltage calculation means 15 for obtaining the magnitude Va of the output voltage based on the voltage signal of the voltage applied to the synchronous motor 5, and the γ axis obtained by the two-phase three-phase coordinate conversion means 7. Current calculation means 21 for calculating the current value Ia based on the current iγ and the δ-axis current iδ, and the current that is less than or equal to (or minimum) the current value based on iγ, iδ, and Ia Current phase calculation means 22 for calculating the phase angle βm, voltage command value calculation means 16 for obtaining the voltage command value Va * based on the current phase angle βm at which the current value is less than or equal to a predetermined value (or the minimum), excitation as a reference A reference excitation current command value calculation means 18 for obtaining a current command value iγ0, and an excitation current command value calculation means 17 for obtaining an excitation current command value iγ * for driving a driving load based on Va *, Va, iγ0. And The filter means 19 is a means for filtering the excitation current command value iγ *, which is the output of the current phase control means 14, to remove high frequency components by a low-pass filter or the like, and to suppress variation for each calculation.
[0040]
In the present embodiment, the excitation current command value iγ * output by the excitation current command value calculating means 17 is filtered by the filter means 19 such as a low-pass filter to remove high frequency components. The stability and calculation accuracy of the calculations of the values Vγ * and Vδ * can be improved, and the synchronous motor 5 can be controlled stably. That is, the output value of the excitation current command value iγ * obtained by the current phase control means 14 may fluctuate greatly every calculation cycle depending on the operating state, and the output voltage calculation means 10 outputs using this iγ *. Since the voltage command values Vγ * and Vδ * are calculated, fluctuations in the excitation current command value iγ * are directly reflected in the output voltage command values Vγ * and Vδ *, and the operation of the synchronous motor 5 becomes unstable.
[0041]
As in the present embodiment, the filter means 19 is provided to remove the fluctuation for each calculation cycle due to the high-frequency component generated in the excitation current command value iγ *, thereby suppressing the fluctuation of the excitation current command value iγ * for each calculation cycle. By doing so, instability of the operation of the synchronous motor 5 due to fluctuations in the excitation current command value iγ * can be avoided, and a highly reliable device with good controllability can be obtained. Here, in a steady operation state in which the rotation frequency and load torque of the synchronous motor do not substantially change, the excitation current command value iγ * hardly changes. When a low-pass filter is used as the filter means 19 in such a case, the time constant of the low-pass filter may be set to a value sufficiently larger than the excitation current control period (the calculation period in the current phase control means 14). For example, the time constant of the low pass filter is the output voltage means 10 excluding the excitation current control period (calculation period in the current phase control means 14) and the calculation period other than the excitation current calculation (for example, the calculation period in the current phase control means 14). For example, when the excitation current control period is about 10 msec, the time constant of the low-pass filter may be set to about 1 sec.
[0042]
Here, if the filter means 19 such as a low-pass filter is not provided, the excitation current command value iγ * cannot be averaged. Therefore, the dispersion becomes large every calculation cycle, and accurate and stable calculation cannot be performed. It becomes large or the convergence is poor and stable control cannot be performed.
[0043]
The output voltage command value calculation means 10 includes γ-δ axis currents iγ and iδ obtained from the three-phase to two-phase conversion means 7, the primary frequency ω 1 obtained from the primary frequency calculation means 9, and the excitation current obtained from the filter means 19. Voltage command values Vγ * and Vδ * for driving the synchronous motor 5 which is a driving load are calculated from the filtered value iγ * of the command value. The electrical angle phase calculation means 13 calculates the electrical angle phase θ based on the primary frequency ω 1 obtained by the primary frequency calculation means 9. The two-phase / three-phase conversion means 11 uses the electrical angle phase θ obtained by the electrical angle phase computation means 13 as the γ-δ axis voltage command values Vγ * and Vδ * obtained by the output voltage command value computation means 10. It converts into phase voltage command value Vu *, Vv *, Vw *. The PWM signal generation means 12 is a PWM for on / off control of switching elements in the inverter main circuit 3 based on the three-phase voltage command values Vu *, Vv *, Vw * obtained from the two-phase / three-phase conversion means 11. Means for generating a signal.
[0044]
Next, the operation will be described with reference to FIG. In the figure, the synchronous motor drive apparatus 1 uses current detection means 6a, 6b for currents for two phases (for example, U-phase current iu and V-phase current iv) out of phase currents flowing into the synchronous motor 5 which is a drive load of the inverter. Detect more. The inverter control means 2 is supplied from the outside by the frequency compensator 8 using the current for two phases (for example, the U-phase current iu and the V-phase current iv) detected by the current detection means 6a and 6b or the current signal for two phases. A compensation amount ωd for compensating the given frequency command value ωm * is calculated, and the frequency command means ωm * is compensated by the compensation amount ωd by the frequency calculation means 9 and output as the primary frequency ω1. Then, the output voltage command value calculation means 10 outputs the output voltage command values (Vγ *, Vδ *) for driving the synchronous motor 5 as the driving load using the compensated frequency command value, which is the primary frequency ω1. ) By calculation. Thereafter, the two-phase / three-phase conversion means 11 converts the two-phase coordinate system output voltage command value (Vγ *, Vδ *) into the three-phase coordinate system output voltage command value (Vu *, Vv *, Vw *). And output as a voltage value or voltage signal. Based on the output voltage command values (Vu *, Vv *, Vw *) of the three-phase coordinate system, the PWM signal generating means 12 outputs a PWM signal for on / off control of the switching elements in the inverter main circuit 3. .
[0045]
That is, the inverter control means 2 outputs a PWM signal by the following operation. Using the phase currents (for example, iu, iv) for two phases detected by the current detection means 6a, 6b and the electrical angle phase θ, the three-phase / two-phase conversion means 7 obtains γ-δ axis currents iγ, iδ. The frequency compensator 8 obtains the frequency compensation amount ωd from the δ-axis current iδ obtained by the three-phase / two-phase conversion means 7. The primary frequency calculating means 9 compensates the frequency command value ωm * by the difference between the rotational frequency command value ωm *, which is a target frequency given from the outside, and the frequency compensation amount ωd obtained by the frequency compensator 8 to compensate the primary frequency ω1. Output as.
[0046]
In the output voltage command value calculation means 10, the excitation current command value iγ obtained by filtering the primary frequency ω 1 obtained by the primary frequency calculation means 9 and the excitation current command value iγ * obtained from the current phase control means 14 by the filter means 19. * Output voltage command values Vγ * and Vδ * for driving the synchronous motor 5 which is a driving load of the inverter are obtained from the γ-δ axis currents iγ and iδ obtained by the three-phase to two-phase conversion means 7.
[0047]
Here, the current phase control means 14 calculates the excitation current command value iγ * as follows. Based on the γ-axis current iγ and the δ-axis current iδ obtained by the two-phase / three-phase coordinate conversion means 7, the current calculation means 21 calculates the magnitude Ia of the current vector. Based on Ia obtained from the current calculation means 21, the current phase angle βm at which the output torque is not less than a predetermined value (or maximum), that is, the current value is not more than a predetermined value (or minimum) is determined by the current phase calculation means 22. Calculate. Then, the voltage command value calculating means 16 is based on the current phase angle βm obtained by the current phase calculating means 22 and the primary frequency ω1 which is the compensated frequency command value which is the previous output value of the frequency calculating means 9. To calculate the voltage command value Va *.
[0048]
The voltage calculation means 15 calculates the magnitude Va of the output voltage command value based on the output voltage command values Vγ * and Vδ * which are the previous output values of the output voltage command value calculation means 10. Further, the reference excitation current command value calculation means 18 obtains a reference excitation current command value iγ0 based on the compensated frequency command value (primary frequency) ω1 which is the previous output value of the frequency calculation means 9. In the excitation current command value calculation means 17, the voltage command value Va * obtained from the voltage command value calculation means 16, the magnitude Va of the output voltage command value obtained from the voltage calculation means 15, and the reference excitation current command value Based on the reference excitation current command value iγ0 obtained from the calculation means 18, the excitation current command value iγ * for driving the synchronous motor 5 at a current phase angle βm at which the current is equal to or less than a predetermined value (or minimum) is calculated. To do. The excitation current command value iγ * is passed through a filter means 19 such as a low-pass filter to remove high frequency components and average to suppress variations in the excitation current command value for each calculation cycle, thereby outputting an output voltage command value calculation means. 10 is output as the excitation current command value iγ * filtered to 10.
[0049]
The electrical angle phase calculation means 13 calculates the electrical angle phase θ based on the compensated frequency command value (primary frequency) ω1 that is the output value of the frequency calculation means 9. In the two-phase / three-phase converter 11, the electrical angle phase θ obtained by the electrical angle phase calculator 13 is used for the γ-δ axis voltage command values Vγ * and Vδ * obtained by the output voltage command value calculator 10. Are converted into voltage command values Vu *, Vv *, Vw * in a three-phase coordinate system. The PWM signal generating means 12 controls on / off of the switching elements in the inverter main circuit 3 based on the three-phase voltage command values Vu *, Vv *, Vw * obtained by the two-phase / three-phase converting means 11. PWM signal is generated. Based on this PWM signal, the switching element in the inverter main circuit 3 is turned on / off, the DC voltage of the DC power supply 4 is converted into a three-phase AC and applied to the synchronous motor 5 as a driving load to drive the driving load. .
[0050]
Here, the relationship between the current phase angle β and the output torque τ in a predetermined operation state when a synchronous motor, particularly an embedded magnet type motor (IPMSM) is used as a driving load will be described with reference to FIG. FIG. 2 is a diagram showing an example of the relationship between the current phase angle β and the output torque τ in the operation state where the IPMSM current is constant. In the figure, the horizontal axis represents the current advance angle (current phase angle) β, and the vertical axis represents the torque. In the figure, the magnet torque generated by the rotor magnet of the IPMSM, the d-axis inductance Ld, and the q-axis inductance Lq. The relationship with respect to the current phase angle β is shown for the reluctance torque generated by the difference between the output torque and the output torque that is the combined torque.
[0051]
As shown in the figure, in the IPMSM, the output torque varies depending on the current phase angle β even under the same current condition. Therefore, the output torque is determined by controlling the current phase angle β to a predetermined phase even under the same current condition. It is possible to output at a value greater than (or maximum). Therefore, under the condition where the load torque is constant, the current value required to obtain the same output torque as the load torque can be reduced to a predetermined value or less (or minimum) by appropriately controlling the current phase angle β. . Also, if the output torque is controlled to be greater than or equal to the predetermined value (or maximum) under the constant current condition, the current value will be less than or equal to the predetermined value (or minimum) under the constant load torque. And a highly efficient apparatus can be obtained.
[0052]
FIG. 3 is a graph showing the relationship between the current phase angle β and the current value when driven by changing the load torque of the IPMSM under a constant rotational frequency condition. In the figure, the horizontal axis represents the current advance angle (current phase angle) β, and the vertical axis represents the current value. In the figure, τn (n = 0, 1, 2, 3, 4) indicates the output torque of the IPMSM, and the magnitude relationship is τ4>τ3>τ2>τ1> τ0.
[0053]
As shown in FIG. 3, even when the IPMSM is driven under the same rotational frequency condition, when the load torque changes (τ0 → τ1 → …… → τ4), the current phase angle β that minimizes the current value is obtained. Will change. However, even when the rotational frequency and load torque change, if the current phase angle β can be controlled to the current phase angle βm that minimizes the current value in accordance with the change of the output torque as shown in FIG. It is possible to drive the IPMSM in a state where the current is minimum, the copper loss of the IPMSM can be minimized, and the conduction loss of the switching element of the inverter main circuit can be minimized, so that high-efficiency operation can be realized.
[0054]
Here, in the present embodiment, the synchronous motor 5 that is a driving load is driven so that the current is equal to or less than a predetermined value (or the current is minimized), and the method will be described below.
[0055]
A voltage equation in dq axis coordinates of a synchronous motor, particularly an embedded magnet type synchronous motor (IPMSM), is generally given by Equation 1. Here, the d-axis and the q-axis are defined as the q-axis, where the magnetic pole direction of the rotor of the electric motor is the d-axis, and the axis having a phase advanced from the d-axis in the rotation direction by a predetermined angle (90 degrees in the case of two poles). doing.
[0056]
[Expression 1]

[0057]
In Equation 1, Vd is the d-axis voltage of the embedded magnet type synchronous motor (IPMSM), Vq is the q axis voltage of the embedded magnet type synchronous motor (IPMSM), and R is one phase of the embedded magnet type synchronous motor (IPMSM). Resistance value, Ld is the d-axis inductance of the embedded magnet type synchronous motor (IPMSM), Lq is the q axis inductance of the embedded magnet type synchronous motor (IPMSM), Φf is an induced voltage constant of the embedded magnet type synchronous motor, id is the d-axis current of the embedded magnet type synchronous motor, iq is the q axis current of the embedded magnet type synchronous motor, ω is the rotational frequency of the embedded magnet type synchronous motor, and p is the differential operator according to time t. Yes. When the number of pole pairs of the embedded magnet type synchronous motor is Pm, the output torque τm of the embedded magnet type synchronous motor (IPMSM) is generally given by Equation 2.
[0058]
[Expression 2]

[0059]
Further, the d-axis current id, the q-axis current iq, and the current magnitude Ia are defined as Equation 3.
[0060]
[Equation 3]

[0061]
When Formula 3 is substituted into Formula 2, torque τm is expressed as Formula 4.
[0062]
[Expression 4]

[0063]
Here, the current phase angle at which the output torque is maximized (or the current is minimized) is βm, and the procedure for obtaining βm will be described below. By partial differentiation of Equation 4 with respect to the current phase angle β and setting it to = 0, the maximum torque can be obtained (or the current can be minimized), and the current phase angle βm can be obtained as shown in Equation 5.
[0064]
[Equation 5]

[0065]
However, in the calculation of βm, an approximate expression such as Expression 6 shown below is used so that the output torque becomes equal to or greater than a predetermined value without calculating the output torque to be maximum as in Expression 5. Also good.
[0066]
[Formula 6]

[0067]
Here, Equation 6 is obtained up to the second order term by expanding the Equation 5 to the Macrolin series. When this approximate expression 6 is used, the output torque does not become maximum (the current does not become minimum), but the output torque can be increased to a predetermined value or more (the current value is set to a predetermined value or less). A device can be obtained. In addition, since the calculation of βm is an approximate expression and simplified, the calculation load on the microcomputer can be reduced, so that the calculation time can be reduced, and furthermore, it can be handled with a low-cost microcomputer. , The freedom of choice of microcomputer is expanded. In addition, since the calculation of βm is an approximate expression and is simplified, the convergence is also improved, so that an apparatus and a control method with good followability in control can be obtained. However, when it is necessary to improve the control accuracy, the accuracy of the approximate expression may be improved by using up to three terms of the Macrolin series, or the expression 5 may be used as it is.
[0068]
Here, in this embodiment, since the γ-δ axis is used as the coordinate system instead of the dq axis, the relationship between the dq axis coordinate system and the γ-δ coordinate system is shown in FIG. Will be described. In the present embodiment, when the synchronous motor is driven without using a position sensor for detecting the rotor position, the dq axes of the interior permanent magnet (IPM) rotor are actually detected. Therefore, a γ-δ axis as shown in FIG. 4 is used as a control axis.
[0069]
FIG. 4 is a vector diagram under certain operating conditions of a synchronous motor (in particular, an embedded magnet type synchronous motor (IPMSM)). In FIG. 4, the horizontal axis represents the γ-axis and the vertical axis represents the δ-axis. Further, the d-axis and the q-axis are located at angular positions rotated by Δθ with respect to the γ-axis and the δ-axis, respectively. Here, iγ is the IPMSM γ-axis current, iδ is the IPMSM δ-axis current, Ia is the magnitude of the current flowing through the IPMSM, and is a combined vector of iγ and iδ. Further, Ia is shifted in angle by β (current phase angle) with respect to the q axis. Thus, the current phase angle β is an angle formed by the q axis and the current vector Ia.
[0070]
ΦF is a magnetic flux vector generated by the rotor magnet, and its magnitude is an induced voltage constant Φf, which is located on the d-axis. Lq · iδ is a δ-axis component of the stator magnetic flux, Ld · iγ is a γ-axis component of the stator magnetic flux, Φ0 is a combined magnetic flux of the stator magnetic flux and the rotor magnetic flux, and the rotor magnetic flux (ΦF) and the stator It is a composite magnetic flux vector of magnetic fluxes (Lq · iδ, Ld · iγ). Further, ω · Φ0 is a voltage vector generated by the combined magnetic flux, R · Ia is a voltage vector that compensates for the voltage effect caused by the motor resistance, Va is a voltage applied to the IPMSM, and ω · Φ0 and R · Ia It is a composite vector. Here, as a control axis, the dq axis may be used as it is without using the γ-δ axis, and either can be used for control, but in this embodiment, the current component is easily detected. A case where the γ-δ axis is used will be described.
[0071]
Here, in the present embodiment, the γ-δ axis is used as the control axis, and in order to simplify the control, the phase difference Δθ between the dq axis and the γ-δ axis is approximated as Δθ≈0, The inductance of each axis is handled as Lγ≈Ld and Lδ≈Lq.
[0072]
In the present embodiment, the excitation current is controlled as a control amount for driving at a current phase angle at which the current is minimized. This excitation current will be described below. As shown in FIG. 4, in the γ-δ axis defined as the control axis (or the dq axis based on the rotor position of the IPMSM), the current iγ corresponding to the γ axis (or d axis) (or id) is generally called an exciting current, and a combined magnetic flux vector (Φ0) is generated by a magnetic flux (Ld · iγ + Lq · iδ) generated by the exciting current and a magnetic flux (ΦF) of the rotor magnet.
[0073]
On the other hand, the current corresponding to the δ-axis (or q-axis) is generally called a component (torque current) relating to the torque of the current component, and has a magnitude corresponding to the output torque of the motor.
[0074]
Here, the current vector Ia is a combined vector of the excitation current iγ (or id) and the torque current iδ (or iq). When the output voltage vector Va is changed according to the operation state of the IPMSM (predetermined rotation speed, predetermined load torque), the excitation current iγ changes (the δ-axis current iδ hardly changes because of the torque component), and this excitation current iγ Changes the current phase angle β. As shown in FIG. 4, since the combined vector of the excitation current iγ (or id) and the torque current iδ (or iq) is the current vector Ia, the excitation current iγ (or id) and the torque current iδ (or iq) are controlled. If the output voltage vector Va is controlled, the current phase angle β of the current vector Ia can be changed.
[0075]
In the present embodiment, even in the control without the speed controller, that is, without the torque current command value (iδ *), the current phase angle at which the current is less than or equal to the predetermined value (or the minimum) is obtained. The output voltage vector Va is controlled by controlling only the excitation current iγ, and the current is controlled to be equal to or less than a predetermined value (current minimum). The current phase angle βm at which the current is minimum (or the current is equal to or less than a predetermined value) is an angle formed from the q axis of the dq axis coordinate system to the current vector Ia. However, in a control method without a speed controller or a control method without a position sensor, the actual position of the dq axis is unknown, so in this embodiment, the control axis (γ-δ axis) and the motor The voltage vector magnitude Va, the voltage command value Va *, and the current vector magnitude Ia are used as variables used for control so that the influence of the phase difference Δθ with respect to the axes (dq axes) can be ignored. .
[0076]
In the present embodiment, the excitation current command output from the current phase control means 14 is provided in order to realize the minimum current control even in a control method in which the drive means of the synchronous motor does not have a speed control loop or a current control loop. The value iγ * is obtained as follows.
[0077]
The output voltage calculation means 15 uses the output voltage signal (for example, Vγ * and Vδ * which are the previous output values of the output voltage command value calculation means 10), and the magnitude of the voltage output from the inverter according to the following expression (Expression 7) Find Va.
[0078]
[Expression 7]

[0079]
Here, the output voltage (Vu, Vv, Vw) output from the inverter main circuit 3 may be used as the output voltage signal. In this case, a separate voltage detection means (not shown) is output from the inverter main circuit 3. What is necessary is just to provide in the side. In this embodiment, since the voltage detection means is not required, the previous output values Vγ * and Vδ * of the output voltage command value calculation means 10 are used, so that the voltage detection means is not required. A device having a simple structure and a low cost, yet having a low current and high efficiency can be obtained.
[0080]
Further, the current calculation means 21 uses the γ-axis current iγ and the δ-axis current iδ obtained by performing the three-phase / two-phase coordinate conversion on the current flowing through the synchronous motor 5 that is a driving load, to calculate the current magnitude Ia. The operation is performed at 8.
[0081]
[Equation 8]

[0082]
Here, in Equation 8, the magnitude Ia of the current vector is calculated on the control axis (γ-δ axis), but the control axis (γ-δ axis) and the motor axis (dq axis) are The current vector magnitude Ia is the same only by the phase difference Δθ. Therefore, when the magnitude Ia of the current vector is used for control as in the present embodiment, it is not necessary to consider the phase difference Δθ between the control axis (γ−δ axis) and the motor axis (dq axis). Therefore, control is possible even in a control system that does not have a speed control loop or a current control loop. The current phase calculation means 22 calculates the current phase angle βm at which the current is less than or equal to a predetermined value (or minimum) from Equation 6 (or Equation 5) using Equation 8. The voltage command value calculation means 16 uses the calculation formula of the dq axis coordinate system, and the voltage command value Va * for driving the drive load 5 at a current phase angle βm at which the current is equal to or less than a predetermined value (or minimum). Is obtained as follows. Since Vd and Vq can be obtained by substituting Equation 3 into Equation 1, the magnitude Va of the resultant force is expressed by the square root of the square sum of Vd and Vq (Equation 9).
[0083]
[Equation 9]

[0084]
However, although the control axis (γ-δ axis) and the motor axis (dq axis) are shifted by the phase difference of Δθ, the magnitude Va of the voltage vector on the control axis (γ-δ axis) and the motor Since the magnitude Va of the voltage vector on the axes (dq axes) is the same, the coordinate axes can be replaced. That is, since the coordinate system actually obtained for control is the γ-δ axis coordinate system, Equation 9 representing the magnitude of the voltage vector on the motor axis (dq axis) is expressed on the control axis (γ−δ axis). It can be considered by replacing it with Equation 7 representing the magnitude Va of the voltage vector.
[0085]
Then, by substituting Ia obtained by Equation 8 and the current phase angle βm obtained by Equation 6 (or Equation 5) into Equation 7, the voltage command value Va * when the current phase angle β is βm is as shown in Equation 10. I want to. Also here, the command value is commanded by the voltage vector magnitude Va * so that the phase difference Δθ between the control shaft (γ-δ axis) and the motor shaft (dq axis) need not be taken into consideration. .
[0086]
[Expression 10]

[0087]
Here, as described above, Ia is the magnitude of the current on the γ-δ axis flowing into the synchronous motor 5, and βm is the current phase angle at which the output torque is greater than or equal to (or maximum), that is, the current is a predetermined value. The current phase angle is the following (minimum).
[0088]
Here, when calculating the voltage command value Va * of Expression 10, if the current phase angle βm at which the current is minimized (the current is equal to or less than a predetermined value) using Expression 5 or 6, the calculation load increases. In some cases, the calculation load increases and the calculation time becomes longer. In this case, it is not necessary to obtain up to βm, and it is sufficient that sinβm can be obtained. For that purpose, if sin βm is obtained by modifying Equation 5 (or Equation 6) and Ia obtained in Equation 8 and the primary frequency ω1 obtained by the primary frequency calculating means 9 are substituted into Equation 10 as ω, the voltage command value Va * can be calculated, and only sin βm is obtained as compared to the case where βm is obtained. Therefore, the number of operations can be reduced, and the calculation time can be shortened. Here, sin βm and cos βm can be obtained from Equation 6 as Equation 11, for example.
[0089]
[Expression 11]

[0090]
Here, using the fact that Va * is a function of ω and Ia, ω and Ia or ω and Ia ² A numerical table having the above relationship may be stored in advance in the voltage command value calculation means 16 or a storage means provided separately.
[0091]
If the table is provided as described above, it is possible to reduce the computation time such as the square root and the number of computations, increase the degree of freedom in selecting the microcomputer, and reduce the cost. Whether to use an arithmetic expression or a table may be selected according to the performance of the microcomputer to be used.
[0092]
The reference excitation current calculation means 18 obtains iγ0 as a reference value of the excitation current command value iγ * by using a numerical table, a function with the frequency ω, or a constant having a constant value given in advance. In the excitation current calculation means 17, the reference excitation current command value iγ0 obtained by the reference excitation current command value calculation means 18, the output voltage magnitude Va obtained by the output voltage calculation means 15, and the voltage command value calculation means 16 are obtained. Using the voltage command value Va * thus obtained, the excitation current command value iγ * is obtained by the following formula 12. Here, Khe in Expression 12 is a control gain of the excitation current given in advance through experiments or the like.
[0093]
[Expression 12]

[0094]
The calculation of the excitation current command value iγ * in Equation 12 is an example of a proportional control method in which the difference between the voltage command value Va * and the output voltage magnitude Va is multiplied by a proportional gain Khe. In the proportional control method, since the difference between the voltage command value Va * and the output voltage magnitude Va is simply multiplied by a control gain given in advance, a steady error occurs with respect to the voltage command value Va *. If it is desired to control the current phase angle β with high accuracy, integral control may be used. An example of a control method using integral control is shown in Equation 13.
[0095]
[Formula 13]

[0096]
Here, iγ * _prv is the output value of the excitation current command value in the previous control cycle in the current phase control means 14, and the output value of the excitation current command value in this previous control cycle is the reference excitation current command value. As a difference between the voltage command value Va * and the output voltage magnitude Va can be made as close to 0 as possible by using it instead of iγ0, the error with respect to the voltage command value Va * can be reduced, and the calculation accuracy can be reduced. Will improve. Here, it is difficult to accurately follow the current phase angle β to the minimum current phase angle βm with the proportional control method shown in Equation 12, but the control method without the speed control loop or the current control loop may be excessive. When control is performed in the modulation region, proportional control as shown in Formula 12 can be more stable than integral control as shown in Formula 13.
[0097]
The current phase control means 14 calculates the excitation current command value iγ * by using the above formulas 1 to 13. The excitation current command value iγ * calculated by the current phase control means 14 is filtered by a filter means 19 such as a low-pass filter in order to obtain the control stability of the synchronous motor, and the filtered excitation current command value iγ. * Is used for calculation in the output voltage command value calculation means 10. Here, the filter means 19 may not be provided, but the filter means 19 is preferably provided in order to obtain control stability.
[0098]
As described above, as a method for controlling the current phase angle in order to drive the synchronous motor 5 at a current phase angle at which the current is equal to or less than a predetermined value (or minimum), in the present embodiment, the current flowing through the synchronous motor is Based on the current detection means 6a, 6b to be detected, the current signal obtained by the current detection means 6a, 6b, and the voltage signal of the voltage applied to the synchronous motor 5, the output torque of the synchronous motor 5 becomes a predetermined value or more. The current phase control means 14 for outputting the excitation current command value iγ * for controlling the current phase angle β of the current flowing through the synchronous motor 5 and the excitation current command value iγ * output by the current phase means 14 are filtered. A compensation amount ωd for the frequency command value ωm * is obtained from the fluctuation amount of the current detected by the filter unit 19 and the current detection units 6a and 6b, and is compensated by the compensation amount ωd. Frequency calculation means 9 for outputting the frequency command value ω1, output voltage command values Vγ * and Vδ for driving the synchronous motor 5 based on the compensated frequency command value ω1 and the filtered excitation current command value iγ *. Output voltage command value calculating means 10 for obtaining *, in order to calculate the output voltage command value based on the excitation current command value and the compensated frequency command value of the current phase angle at which the current is equal to or less than a predetermined value. Even when the operating conditions of the synchronous motor change due to frequency fluctuations or load fluctuations, it is possible to control the current to a current phase angle at which the current is below a predetermined value (or minimum) according to the conditions. The current value can be accurately set to a predetermined value or less (or minimum) following the change in conditions, and a highly efficient electric motor drive device can be obtained. Further, since the filter means 19 is provided, the variation of the excitation current command value iγ * for each calculation is reduced, and stable control can be performed.
[0099]
In the present embodiment, the current phase calculation means 22 for obtaining the current phase angle βm so that the output torque of the synchronous motor 5 becomes a predetermined value or more based on the current signals obtained by the current detection means 6a and 6b, Voltage command value calculation for calculating the voltage command value Va * based on the current phase angle βm obtained by the current phase calculation means 22 and the primary frequency ω1 which is the previous frequency command value obtained by the frequency command value calculation means 9 Means 16, voltage calculation means 15 for calculating the magnitude of the voltage based on the voltage signal of the voltage applied to the synchronous motor 5, voltage command value Va * obtained by the voltage command value calculation means 16, and voltage Excitation current command value calculation means 17 for obtaining an excitation current command value iγ * for driving the synchronous motor 5 based on the voltage magnitude Va obtained from the calculation means 15, Since the control means 14 is configured to control the current phase angle β to the current phase angle βm at which the current is less than or equal to the predetermined value (or minimum), the current phase at which the current is less than or equal to the predetermined value (or minimum). Although it is a simple configuration to obtain the output voltage at the angle βm and the actual output voltage, it is not necessary to have a numerical table or function of the excitation current command value for each operating condition, so the memory of the microcomputer used for the inverter control means is used Since an inexpensive microcomputer with a small memory capacity can be used without increasing the amount, and a simple control circuit can be used, a highly reliable and inexpensive synchronous motor drive device can be obtained.
[0100]
Further, in the present embodiment, a DC power supply 4 that supplies DC, an inverter main circuit 3 that converts DC from the DC power supply 4 by turning on and off the switching elements, and an inverter main circuit 3 Control is performed by applying an inverter control means 2 that outputs a voltage command value for controlling the on / off operation of the switching element and a voltage output by an inverter main circuit 3 controlled by the inverter control means 2. An excitation for driving the drive load 5 at a current phase angle βm such that the output torque of the drive load becomes a predetermined value or more based on the current flowing through the drive load. The current phase control means 22 for calculating the current command value iγ * and the excitation current command value iγ * obtained from the current phase control means 22 are filtered. The filter means 19, the frequency calculation means 9 for compensating the frequency command value ωm * from the fluctuation amount of the current flowing through the drive load 5 and outputting it as the primary frequency ω1, and the excitation current command value iγ * filtered by the filter means 19. Output voltage command value calculation means 10 for calculating voltage command values Vγ * and Vδ * for driving the drive load 5 based on the primary frequency ω1 obtained by the frequency calculation means 9 constitutes an inverter device. Therefore, even if the operating condition of the driving load changes, the output torque under the condition can be controlled to be equal to or higher than a predetermined value even though the current phase control means and the frequency calculation means are simply provided. In addition, the current can be controlled at a current phase angle at which the current value is equal to or less than a predetermined value in accordance with the output torque, and a synchronous motor drive device with low current and high efficiency can be realized. Further, since the current value can be reduced, the rated capacity of the switching element can be reduced, and a low-cost inverter device can be provided. Further, since the filter means 19 is provided, the variation in the excitation current command value iγ * is reduced, and a highly reliable inverter device that can stably control the driving load is obtained.
[0101]
Furthermore, a current phase calculation means 22 for obtaining a current phase angle βm at which the output torque becomes equal to or greater than a predetermined value due to a current flowing through the drive load 5, and a voltage command for obtaining the current phase angle βm obtained by the current phase calculation means 22 Voltage command value calculating means 16 for calculating the value Va *, and voltage calculation for calculating the voltage magnitude Va based on the previous output voltage command values Vγ * and Vδ * obtained by the output voltage command value calculating means 10 Excitation current command value for driving the drive load 5 based on the means 15, the voltage magnitude Va obtained from the voltage calculation means 15 and the voltage command value Va * obtained by the voltage command value calculation means 16 Since the current phase control means 14 is configured by the excitation current command value calculation means 17 for obtaining i γ *, the current phase angle is tracked so that the current value flowing through the drive load is a predetermined value or less. And can be controlled in an efficient state in a short time. Further, since the convergence and stability near the current phase angle at which the current value is equal to or less than the predetermined value are good, an inverter device with good controllability and low current and high efficiency can be realized. Further, since only the excitation current command value is controlled, the control axis (γ-δ axis) and the motor axis (d) are used by using the voltage vector magnitudes Va and Va * and the current vector magnitude Ia as values used for the control. It is not necessary to consider the phase difference with respect to (−q axis), and furthermore, a low-current and high-efficiency inverter device can be realized even without a speed controller or a current controller.
[0102]
In the present embodiment, even when the operating conditions of the synchronous motor change due to frequency fluctuations or load fluctuations, the current is controlled to a current phase angle at which the output torque is equal to or greater than a predetermined value (or maximum) according to the conditions. Therefore, a high-power inverter device and synchronous motor drive device can be obtained.
[0103]
In addition, since the current phase angle is controlled so that the current is always less than or equal to the predetermined value (or minimum), the followability to the minimum current phase angle is good, and the convergence and stability around the minimum current phase angle are good. Therefore, an inverter device and a synchronous motor driving device with good controllability can be obtained. Further, since only the excitation current command value is controlled, the control axis (γ-δ axis) and the motor axis (d) are used by using the voltage vector magnitudes Va and Va * and the current vector magnitude Ia as values used for the control. An inverter device capable of realizing current phase control in which the current is less than or equal to a predetermined value (or minimum) even without a speed controller or current controller. A synchronous motor drive device is obtained.
[0104]
In this embodiment, since the electric motor is driven without a position sensor that does not use a position sensor for detecting the rotor position, the position sensor is unnecessary, the cost is high, and the current is high. Since current phase control that is not more than a predetermined value (or minimum) can be realized, a highly efficient inverter device or synchronous motor drive device can be obtained. Further, since a position sensor is unnecessary, an inverter device and a synchronous motor drive device that can perform high-output and high-efficiency control can be realized even in a synchronous motor used for a compressor or the like to which a position sensor cannot be attached.
[0105]
In the present embodiment, the voltage signal used in the output voltage calculation means 15 does not have to be the previous output voltage command values Vγ * and Vδ * obtained from the output voltage command value calculation means 10, but actually the inverter main circuit In this case, the voltage value output from the inverter main circuit 3 may be detected by providing a voltage detection means (not shown). After the output voltage value is detected, the voltage value Vγ, Vδ on the γ-δ axis is converted using the three-phase / two-phase conversion means, and the converted voltage values Vγ, Vδ may be used. By detecting and using the actual voltage value, the synchronous motor 5 that is a driving load can be controlled more accurately, and a highly efficient and highly reliable device can be obtained.
[0106]
In FIG. 1, the calculation cycle in the current phase control unit 14 in the inverter control unit 2 may be different from the calculation processing cycle other than in the current phase control unit 14. Since the calculation in the current phase control means 14 in the inverter control means 2 does not require quick response compared to the calculation processing other than the current phase control means 14, the calculation cycle (excitation current control) by the current phase control means 14 is not required. (Cycle) may be made larger than the control cycle (output voltage control cycle) other than the current phase control means 14. That is, the calculation processing cycle (excitation current control cycle) in the current phase control unit 14 may be set to about once every several to several hundreds of calculation processing cycles (output voltage control cycle) other than the current phase control unit 14. . In this way, when the arithmetic processing of the inverter control means 2 is realized by a microcomputer or the like, the processing load on the microcomputer can be reduced, and a low-cost inverter control means can be obtained. Can be obtained.
[0107]
Next, an example of the method for controlling the synchronous motor according to Embodiment 1 of the present invention will be described with reference to FIG. FIG. 5 is a flowchart showing a control method for driving the synchronous motor representing the present embodiment with high efficiency at a current equal to or lower than a predetermined value. In the figure, ST1 is a current detection / calculation step for detecting the current flowing through the synchronous motor and calculating the current values iγ and iδ in the two-phase coordinate system, and ST2 is based on the current signal obtained by the current detection / calculation step ST1. The current phase calculation step ST3 for obtaining the current phase angle βm at which the current is less than or equal to the predetermined value (or the minimum) by the current phase calculation means 22, ST3 is the previous compensated frequency command value obtained by the frequency calculation means 9. A voltage command value calculation step for obtaining a voltage command value Va * based on the primary frequency ω1 and the current phase angle βm obtained in the current phase calculation step ST2, ST4 indicates the magnitude Va of the voltage applied to the synchronous motor. A voltage calculation step ST5 is a reference excitation current calculation step for obtaining a reference excitation current command value iγ0.
[0108]
In ST6, the actual current phase angle is determined based on the voltage command value Va * obtained in the voltage command value calculation step ST3 and the voltage magnitude Va obtained in the voltage calculation step ST2. An excitation current command value calculation step for obtaining an excitation current command value iγ * for driving the motive motor 5 at a current phase angle βm which is the following (or minimum), and ST7 is the excitation current command value iγ * obtained in ST6. A filter step for filtering by passing through the filter means 19, ST8 outputs output voltage command values Vγ * and Vδ * for driving the synchronous motor based on the filtered excitation current command value iγ * obtained by the filter step ST7. This is an output voltage command value calculation step to be obtained.
[0109]
First, at ST1, at least two-phase currents of the synchronous motor 5 that is a driving load are detected and converted to current values of a two-phase coordinate system. That is, current detection means 6a and 6b detect currents for two phases (for example, U-phase current iu and V-phase current iv) flowing in synchronous motor 5, and three-phase two-phase conversion means 7 detects currents in a two-phase coordinate system ( For example, γ-axis current iγ and δ-axis current iδ) are converted, and Ia is calculated by Equation 8.
[0110]
In ST2, using the γ-axis current iγ and δ-axis current iδ obtained in ST1, the current phase angle βm at which the current value is less than or equal to a predetermined value (or the minimum) is expressed by Equation 6 (or Equation 5). ) To calculate. (Here, it is only necessary to find sin βm and cos βm as in Equation 11 without obtaining the current phase angle βm as in Equation 6 and Equation 5, so that sin βm and cos βm are obtained in Equation 11 to reduce the computation load. (You may plan.)
[0111]
In ST3, the voltage command value calculation means 16 uses the primary frequency ω1 that is the previous compensated frequency command value obtained by the frequency calculation means 9 and the current phase angle βm obtained in ST2 to obtain the formula 10 Is used to obtain the voltage command value Va *.
[0112]
In ST4, the voltage calculation means 15 calculates the voltage value applied to the drive load from the voltage command value obtained by the output voltage command value calculation means 10 shown in FIG. That is, in ST4, the voltage calculation means 15 sets the voltage magnitude Va for driving the synchronous motor 5 using the previous voltage command values Vγ * and Vδ * calculated by the output voltage command value calculation means 10. Calculation is performed using Equation 7. Here, as the voltage command values Vγ * and Vδ *, a voltage detection means (not shown) or the like may be separately provided to detect a voltage applied to the drive load 5 and use the voltage signal. Then, since the latest voltage information can always be used, a control method for a synchronous motor capable of performing fine and reliable control can be obtained.
[0113]
In ST5, based on the frequency command value (for example, the primary frequency ω1 which is the previous compensated frequency command value obtained by the frequency calculation unit 9), the reference excitation current command value iγ0 is set by the reference excitation current calculation unit 18. Ask.
[0114]
In ST6, based on the voltage command value Va * obtained in ST3, the voltage magnitude Va obtained in ST4, and the reference excitation current command value iγ * obtained in ST5, the drive load is applied at the current phase angle βm. The excitation current command value iγ * for driving 5 is obtained by the excitation current command value calculation means 17 using Equation 12 (or Equation 13). In ST7, the excitation current command value iγ * obtained in ST6 is passed through filter means 19 such as a low-pass filter to remove the high frequency component of the excitation current command value iγ *, thereby reducing the variation for each calculation cycle. To improve the stability of computation and control. Here, if the filter means 19 such as a low-pass filter is not provided, the excitation current command value iγ * cannot be averaged, so that variation becomes large and stable control cannot be performed with high accuracy.
[0115]
In ST8, output voltage command values Vγ * and Vδ * for driving the drive load 5 based on the excitation current command value iγ * filtered by the filter means 19 obtained in ST7 are output to the output voltage command value calculation means 10. Ask. Here, since the calculation formula used in the output voltage command value calculation step ST8 differs depending on the control method of the synchronous motor, any calculation formula may be used as long as Vγ * and Vδ * can be calculated.
[0116]
For example, Formula 14 shows an example of a formula for calculating the γ-axis voltage command value Vγ * and the δ-axis voltage command value Vδ *, which are outputs of the voltage command value calculation means 10.
[0117]
[Expression 14]

[0118]
Formula 14 represents an arithmetic expression in the case of using a position sensorless control method for a synchronous motor by primary magnetic flux control, for example. In Equation 14, Φγ * is a primary magnetic flux command value in primary magnetic flux control, Φγerr is a primary magnetic flux error, Kγ is a γ-axis control gain, and Kδ is a δ-axis control gain.
[0119]
As described above, in the present embodiment, the current phase calculation step ST3 for obtaining a current phase angle such that the output torque of the synchronous motor becomes a predetermined value or more based on the value of the current flowing through the synchronous motor, and the synchronous motor 5 The voltage calculation step ST4 for calculating the magnitude of the applied voltage, the reference excitation current calculation step ST5 for obtaining the reference excitation current command value, and the voltage command value Va * and voltage obtained in the voltage command value calculation calculation step ST3 An excitation current command value iγ * for driving the synchronous motor 5 based on the voltage magnitude Va obtained in the calculation step ST4 and the reference excitation current command value iγ0 obtained in the reference excitation current command value calculation step. The excitation current command value calculation step ST6 for obtaining the excitation current command value iγ * obtained by the excitation current command value calculation step ST6 is used as the filter means 1. And a filter step that performs filtering to remove high-frequency components and suppress variation in each calculation cycle, so that control can be performed efficiently and stably with high output and low current while being simple control. The synchronous motor control method can be obtained.
[0120]
In addition, even when the operating conditions of the synchronous motor change due to rotational frequency and load fluctuations, the current can be controlled with a low current phase angle at which the output torque corresponding to the conditions is equal to or greater than a predetermined value. The synchronous motor can be driven with efficiency. In addition, the followability to the low current phase is good and the convergence and stability in the vicinity of the low current phase are good, so that the controllability can be improved. In addition, since it is a simple control that controls only the excitation current command value without considering the coordinate axis, the synchronous motor can be driven accurately with low current and high efficiency even in a control means that does not have a speed controller or current controller. A possible control method is obtained.
[0121]
Here, in FIG. 5, the voltage calculation step ST4 may be performed anywhere as long as it is performed in a step before the excitation current command value calculation step ST6, and may be performed anywhere from ST1 to ST5. Further, the output voltage command value calculation step ST8 is a step for obtaining not only the excitation current control but also the output voltage command value for driving the synchronous motor according to the rotation frequency command value and the load torque. In FIG. 5, the processing cycle from the current detection / calculation step ST1 to the excitation current command value calculation step ST6 is set as a processing cycle A, and the processing cycle after the output voltage command value calculation step ST7 in the inverter control means 2 is set as a processing cycle B. Then, the processing cycle A and the processing cycle B may be set to different cycles in order to reduce the processing load of a microcomputer that performs processing. In this case, since the processing cycle A does not require responsiveness compared to the processing cycle B, the microcomputer load can be reduced if the processing cycle A is longer than the processing cycle B, so a low-cost microcomputer can be used. Therefore, the degree of freedom in selecting a microcomputer is widened, and a motor control method with a low cost and high reliability can be obtained with simple control.
[0122]
A method for performing high-efficiency control with the minimum current without obtaining the current phase angle βm at which the current is less than or equal to the predetermined value (or the minimum) described in the first embodiment will be described with reference to FIG. FIG. 6 is a diagram showing the configuration of another inverter device and synchronous motor drive device representing Embodiment 1 of the present invention. In the figure, parts equivalent to those in FIG.
[0123]
In FIG. 6, reference numeral 140 denotes current phase control means, which also includes the γ-axis current iγ and the δ-axis current iδ obtained by the output voltage calculation means 15 for obtaining the magnitude Va of the output voltage and the two-phase three-phase coordinate conversion means 7. The current calculation means 21 for calculating the current magnitude Ia calculates the current phase angle βm at which the current value is less than or equal to the predetermined value (or minimum) as sin βm, and drives the drive load 5 with this current phase angle βm. Voltage command value calculation means 160 for obtaining a voltage command value Va * for calculating the voltage, voltage error calculation means 225 for calculating an error between the voltage command value Va * and the output voltage magnitude Va, and a reference excitation current command value iγ0. The reference excitation current command value calculation means 18 and the excitation current command value calculation means 170 for obtaining the excitation current command value iγ * for driving the drive load are configured.
[0124]
Therefore, in the apparatus having the configuration shown in FIG. 6, the current value or current signal detected by the current detection means 6a, 6b and the voltage signal applied to the synchronous motor 5 (for example, the output voltage signals Vγ *, Vδ * and their magnitudes). Voltage command value calculation means 160 for obtaining a voltage command value Va * for driving at a current phase angle βm such that the output torque of the synchronous motor 5 is equal to or greater than a predetermined value based on the voltage Va), and a voltage command value Va * Voltage error calculation means 225 for comparing the voltage signal Va to obtain a voltage error, and driving at a current phase angle βm at which the output torque of the synchronous motor 5 becomes a predetermined value or more from the voltage error obtained from the voltage error calculation means 225 The current phase control means 140 is constituted by the excitation current command value calculation means 170 for calculating the excitation current command value iγ * of the voltage, and the voltage error for calculating the voltage error Since the calculation unit 225 is included, the calculation formula can be easily calculated using a simplified calculation formula without complicating the calculation formula, and the calculation time can be shortened. In addition, even when the operating conditions of a synchronous motor change due to rotational frequency or load fluctuations, the current can be controlled with a low current phase angle at which the output torque corresponding to the conditions is greater than or equal to a predetermined value, resulting in low current and high efficiency. A synchronous motor driving device can be realized.
[0125]
In addition, it is possible to realize a synchronous motor drive device with a low current and high efficiency that has good followability to the current phase angle βm at which the current value is equal to or less than a predetermined value and has good convergence and stability near the current phase angle. In addition, since only the excitation current command value is controlled without considering the coordinate axis, a good low current and high efficiency synchronous motor drive device can be realized even in a control means having no speed controller or current controller.
[0126]
Next, the operation will be described with reference to a flowchart. FIG. 7 is a flowchart showing another control method for the synchronous motor of the present embodiment. In the figure, ST11 is a current detection / calculation step that detects the current flowing through the synchronous motor and determines the magnitude of the current, and ST12 is a current that is determined by the current signal obtained in the current detection / calculation step ST11 and the rotation frequency of the synchronous motor. This is a voltage command value calculation step for obtaining an output voltage command value Va * for driving the drive load at a current phase angle that is less than (or minimum) the value.
[0127]
ST13 is a voltage calculation step for calculating the magnitude Va of the voltage applied to the synchronous motor. ST14 is a voltage command value Va * obtained from the voltage command value calculation step ST12 and the voltage magnitude obtained in the voltage calculation step ST13. The voltage error calculating step for obtaining the difference (Va * −Va) from Va, ST15 is obtained by the reference exciting current command value calculating step for calculating the reference exciting current command value iγ0, and ST16 is obtained by the reference exciting current command value calculating step ST15. An excitation current command value calculation step for obtaining an excitation current command value iγ * based on the reference excitation current command value iγ0 and the voltage error (Va * −Va) obtained in the voltage error calculation step ST14, ST17 is an excitation current command The excitation current command value iγ * obtained in the value calculation step ST16 is subjected to high frequency component removal by the filter means 19. A filter step for performing filtering, ST18 is an output voltage command value calculation step for obtaining output voltage command values Vγ * and Vδ * for driving the synchronous motor 5 based on the excitation current command value iγ * filtered in the filter step ST17. It is.
[0128]
First, in ST11, at least two-phase currents of the synchronous motor 5 that is a driving load are detected, converted into current values in a two-phase coordinate system, and the magnitudes thereof are obtained. That is, current detection means 6a and 6b detect currents for two phases (for example, U-phase current iu and V-phase current iv) flowing in synchronous motor 5, and three-phase two-phase conversion means 7 detects currents in a two-phase coordinate system ( For example, γ-axis current iγ and δ-axis current iδ) are converted, and the magnitude Ia is calculated by Equation 8.
[0129]
In ST12, using the γ-δ axis currents iγ and iδ obtained in ST11 and the current magnitude Ia, the current phase angle βm at which the current value is equal to or less than a predetermined value (or the minimum) is expressed by sin βm, cos βm And the voltage command value calculation means 160 calculates the voltage command value Va * using Equation 10.
[0130]
In ST13, the voltage calculation means 15 calculates the voltage value applied to the drive load from the voltage command value obtained by the output voltage command value calculation means 10 or the like. That is, in ST13, the voltage calculation means 15 sets the voltage magnitude Va for driving the synchronous motor 5 using the previous voltage command values Vγ * and Vδ * calculated by the output voltage command value calculation means 10. Calculation is performed using Equation 7. Here, in ST13, a voltage applied to the drive load 5 as a voltage command value may be detected by separately providing a voltage detection means and the voltage signal may be used. As a result, since the latest voltage information can always be used, a device capable of performing fine and highly reliable control can be obtained.
[0131]
In ST14, a voltage error (Va * −Va) is obtained by the voltage error calculation means 225 from the voltage command value Va * obtained in ST12 and the voltage magnitude Va obtained in ST13. In ST15, a reference excitation current command value iγ0 is calculated using the compensated frequency command value (primary frequency) ω1 which is the previous output value of the frequency calculation means 9 as a book. In ST16, the drive load 5 is driven with a current value at which the current phase angle is βm based on the reference excitation current command value iγ0 obtained in ST15 and the voltage error (Va * −Va) obtained in ST14. The excitation current command value iγ * for performing the calculation is calculated by the excitation current command value calculation means 170 using Formula 12 (or Formula 13). Here, in this embodiment, since the voltage error (Va * −Va) is directly calculated in ST14, the calculation of the excitation current command value iγ * in ST16 is simplified, and the calculation speed is increased. Yes.
[0132]
In ST17, the exciting current command value iγ * obtained in ST16 is filtered by the filter means 19 to remove high frequency components and to suppress variation in each calculation cycle. In ST18, the output voltage command value calculation means 10 obtains output voltage command values Vγ * and Vδ * for driving the drive load 5 based on the excitation current command value iγ * filtered in ST17. Here, since the calculation formula used in the output voltage command value calculation step ST18 differs depending on the control system of the synchronous motor, any calculation formula may be used as long as Vγ * and Vδ * can be calculated. For example, Formula 14 is used.
[0133]
As described above, in the present embodiment, a voltage command value for driving the synchronous motor at a current phase angle such that the output torque of the synchronous motor becomes a predetermined value or more is obtained based on the current value flowing through the synchronous motor. Voltage command value calculation step ST12, voltage command value Va * obtained in voltage command value calculation step ST12, voltage calculation step ST13 for calculating the magnitude Va of the voltage value applied to the synchronous motor, and voltage command value A voltage error calculation step ST14 for comparing the voltage command value Va * obtained in the calculation step ST12 with the voltage magnitude VA obtained in the voltage calculation step ST13 to obtain a voltage error (Va * -VA), and a reference excitation A reference excitation current command value calculation step for obtaining the current command value iγ0, and a voltage error (Va * −VA) obtained in the voltage error calculation step ST14; And the excitation current for driving the synchronous motor 5 at a current phase angle βm such that the output torque of the synchronous motor becomes a predetermined value or more based on the reference excitation current command value iγ0 obtained in the reference excitation current calculation step ST15. Excitation current command calculation step ST16 for calculating command value iγ *, filter step ST17 for filtering excitation current command value iγ * obtained by excitation current command value calculation step ST16, and excitation current command filtered by filter step ST17 The output voltage command value calculation step ST18 for obtaining output voltage command values Vγ * and Vδ * for driving the synchronous motor based on the value iγ * is provided. Control method for synchronous motors that can drive synchronous motors efficiently with high output and low current It is possible to obtain. In addition, since the filter step is included, it is possible to obtain a highly reliable synchronous motor control method capable of controlling the synchronous motor in a stable state.
[0134]
Further, the frequency command value (primary frequency) ω1 compensated by the frequency compensation amount obtained based on the current flowing through the synchronous motor, the excitation current command value iγ * obtained by the excitation current command value calculation steps ST6 and ST16, Since the output voltage command value calculation step ST18 for obtaining the output voltage command values Vγ * and Vδ * for driving the synchronous motor based on the above is provided, the excitation current command value at the current phase angle βm at which the current becomes a predetermined value or less. Since the output voltage command values Vγ * and Vδ * are calculated based on iγ * and the compensated frequency command value ω1, even when the operating conditions of the synchronous motor change due to the rotation frequency or load fluctuation, it is possible to accurately follow. It is possible to obtain a method for controlling a synchronous motor that can perform stable control.
[0135]
Even when the operating conditions of the synchronous motor change due to rotational frequency and load fluctuations, the current can be controlled with a low current phase angle at which the current value is a predetermined value or less according to the conditions. The synchronous motor can be driven with efficiency. In addition, the followability to the low current phase is good and the convergence and stability in the vicinity of the low current phase are good, so that the controllability can be improved. Further, since only the excitation current command value is controlled without considering the coordinate axis, a control method capable of driving the synchronous motor with low current and high efficiency can be obtained even in a control means having no speed controller or current controller.
[0136]
Here, in FIG. 7, the voltage calculation step ST13 may be performed anywhere as long as it is a step before the voltage error calculation step ST14. Further, the output voltage command value calculation step ST18 is a step for obtaining not only the excitation current control but also the output voltage command value for driving the synchronous motor according to the rotation frequency command value and the load torque. In FIG. 7, if the processing cycle from the current detection / calculation step ST11 to the excitation current command value calculation step ST16 is the processing cycle X and the processing cycle after the filter step ST17 in the inverter control means 2 is the processing cycle Y, the processing cycle X and processing cycle Y may be set to different cycles in order to reduce the processing load of a microcomputer or the like that performs processing. In this case, since the processing cycle X does not require responsiveness compared with the processing cycle Y, the load on the microcomputer can be reduced by making the processing cycle X longer than the processing cycle Y, so a low-cost microcomputer can be used. Therefore, the degree of freedom in selecting a microcomputer is widened, and a motor control method with a low cost and high reliability can be obtained with simple control.
[0137]
In the inverter device, the synchronous motor drive device, and the synchronous motor control method shown in the first embodiment of the present invention, the synchronous motor drive device (or the inverter device) and the synchronous motor that is the drive load are connected in a 1: 1 ratio. The relationship is shown as an example, and in such a case, the synchronous motor drive device and the control method of the present embodiment are integrated into the synchronous motor in the synchronous motor 5 as the drive load, so that the drive device And a synchronous motor with high efficiency and high reliability including a control method can be obtained.
[0138]
Embodiment 2. FIG.
A synchronous motor drive apparatus according to Embodiment 2 of the present invention will be described below. FIG. 8 is a diagram showing the configuration of the inverter device and the synchronous motor drive device representing Embodiment 2 of the present invention. In the figure, the same parts as in FIGS. 1 to 7 of the first embodiment are denoted by the same reference numerals, and the description thereof is omitted. In the figure, reference numeral 1 denotes a synchronous motor drive device, which is composed of a plurality of switching elements, an inverter main circuit 3 for converting direct current to alternating current, and a PWM signal for on / off control of the plurality of switching elements of the inverter main circuit 3. It is comprised by the inverter control means 2 which outputs.
[0139]
Reference numeral 4 denotes a DC power source connected to the inverter main circuit 3, 5 denotes a synchronous motor which is a driving load driven by the inverter main circuit 3, and the inverter device is driven by the synchronous motor driving device 1, the DC power source 4 and the driving load 5. Is configured. The switching element in the inverter main circuit 3 is turned on / off based on the PWM signal output from the inverter control means 2, and the DC voltage of the DC power source 4 is converted into a three-phase AC and applied to the synchronous motor 5 as a driving load. As a result, the inverter device operates.
[0140]
Reference numeral 6a denotes current detection means for detecting one-phase current (for example, U-phase current iu) out of current flowing into the synchronous motor 5 as a driving load, and 6b denotes current detected by the current detection means 6a (U-phase current iu). ) Is a current detection means for detecting a current of a phase different from (for example, V-phase current iv). The current detection means 6a and 6b convert the detected current into a current value or a current signal and output it.
[0141]
Here, the inverter control means 2 includes the following three-phase / two-phase conversion means 7, rotor position / speed estimation means 20, current phase control means 145, frequency comparison means 210, speed control means 220, and γ-axis current comparison means. 230, δ-axis current comparison means 240, current control means 250, two-phase / three-phase conversion means 11, PWM signal generation means 12, and filter means 19.
[0142]
The three-phase to two-phase conversion means 7 uses the current value (for example, U-phase current iu, V-phase current iv) or current signal detected by the current detection means 6a and 6b as an excitation current component using the electrical angle phase θ. The current value or current signal of the γ-δ axis expressed by (γ-axis current iγ) and torque current component (δ-axis current iδ) is converted.
[0143]
The rotor position / speed estimation means 20 is based on the γ-δ axis currents iγ and iδ obtained from the three-phase to two-phase conversion means 7 and the previous output voltage command values Vγ * and Vδ *. The rotor position value θ and the rotation speed (rotation frequency) ωm are estimated. The frequency comparison unit 210 compares the frequency command value ωm * given from the outside with the rotor rotation frequency ωm obtained by the rotor position / speed estimation unit 20 and calculates a frequency difference (ωm * −ωm). The speed control unit 220 obtains the δ-axis current command value iδ * based on the output (ωm * −ωm) of the frequency comparison unit 210.
[0144]
The γ-axis current comparison means 230 is an excitation current command value iγ * obtained by filtering the excitation current command value (also referred to as γ-axis current command value) iγ *, which is an output of the current phase control means 145, by the filter means 19 and the coordinate conversion means. 7 is compared with the γ-axis current iγ obtained in step 7, and the γ-axis current difference (iγ * −iγ) is calculated. The δ-axis current comparison unit 240 compares the δ-axis current command value iδ * obtained by the speed control unit 220 with the δ-axis current iδ obtained by the coordinate conversion unit 7 and compares the δ-axis current difference (iδ * −iδ). ) Is calculated. The current control means 250 outputs the output voltage command values Vγ *, Vδ based on the output value (iγ * −iγ) of the γ-axis current comparison means 230 and the output value (iδ * -iδ) of the δ-axis current comparison means 240. * Ask for.
[0145]
The current phase control unit 145 includes an output voltage calculation unit 15 that calculates the magnitude of the output voltage, a voltage command value calculation unit 165 that calculates a voltage command value at which the current is less than or equal to (or minimum), and a current that is less than or equal to the predetermined value (or Excitation current command value calculation means 175 for obtaining an excitation current command value for driving the drive load so as to be minimum), and reference excitation current command value calculation means 18 for obtaining a reference excitation current command value. An excitation current command value iγ * is output from the current command value calculation means 175.
[0146]
Reference numeral 19 is equivalent to the filter means described in the first embodiment, and the excitation current command value iγ *, which is the output of the current phase control means 145, is removed from the high-frequency component by a low-pass filter or the like, and the variation for each calculation cycle is obtained. Filter means for performing filtering for suppression. In the present embodiment, as described in the first embodiment, the filtering means 19 is provided to perform filtering to remove high frequency components, so that the stability of calculation of the output voltage command values Vγ * and Vδ * is improved. In addition, the calculation accuracy can be improved, the instability of the operation of the synchronous motor 5 due to the fluctuation of the excitation current command value iγ * can be avoided, and a controllable and highly reliable device can be obtained.
[0147]
11 is a three-phase voltage command value Vu * obtained by using the electrical angle phase θ obtained by the rotor position / speed estimation means 20 using the γ-δ axis voltage command values Vγ * and Vδ * obtained by the current control means 250. Two-phase three-phase conversion means for converting Vv * and Vw *, and 12 a three-phase voltage command value Vu *, Vv * and Vw * obtained from the two-phase three-phase conversion means 11 as switching elements in the inverter main circuit 3 PWM signal generating means for generating a PWM signal for on / off control.
[0148]
Next, the operation will be described with reference to FIG. In the figure, the synchronous motor drive apparatus 1 uses current detection means 6a, 6b for currents for two phases (for example, U-phase current iu and V-phase current iv) out of phase currents flowing into the synchronous motor 5 which is a drive load of the inverter. Detect more. Inverter control means 2 drives synchronous motor 5 using currents for two phases (for example, U phase current iu and V phase current iv) detected by current detection means 6a and 6b or current signals for two phases. The voltage (or voltage signal) output from the inverter main circuit 3 is obtained by calculation, and a PWM signal for on / off control of the switching elements in the inverter main circuit 3 is output.
[0149]
That is, the inverter control means 2 outputs a PWM signal by the following operation. From the phase currents iu and iv detected by the current detectors 6a and 6b, the γ-δ axis currents iγ and iδ are obtained from the three-phase / two-phase converter 7 using the electrical angle phase θ. The rotor position / speed estimation means 20 includes γ-δ axis currents iγ and iδ obtained by the three-phase to two-phase conversion means 7 and a γ-δ axis voltage command value Vγ * which is the previous output value of the current control means 250. The current rotor position (positioning electrical angle phase) θ and speed (rotational frequency) ωm of the synchronous motor 5 are estimated from Vδ *.
[0150]
The frequency comparison unit 210 compares the rotation frequency command value ωm * given from the outside with the rotation frequency ωm that is the output of the rotor position / speed estimation unit 20 to obtain the difference Δωm. The speed control means 220 obtains a torque current command value (δ-axis current command value) iδ * by proportional integral control (speed control) from the difference Δωm in rotational frequency. The δ-axis current comparison means 240 compares the torque current command value iδ * with the δ-axis current iδ, and obtains the difference (δ-axis current difference) Δiδ.
[0151]
The current phase control means 145 synchronizes on the basis of the γ-axis current iγ and δ-axis current iδ obtained by the two-phase / three-phase coordinate conversion means 7 and the previous outputs Vγ * and Vδ * of the current control means 250. An excitation current command value iγ * for driving the motor at a current phase angle βm at which the current is equal to or less than a predetermined value (or minimum) is output. The current phase control means 145 constituted by the voltage command value calculation means 165, the voltage calculation means 15, the reference excitation current command value calculation means 18, and the excitation current command value calculation means 175 will be described below.
[0152]
The voltage command value calculation means 165 is based on the γ-axis current iγ and δ-axis current iδ obtained by the two-phase / three-phase coordinate conversion means 7 and the estimated speed ωm obtained by the rotor position / speed estimation means 20. A current phase angle βm at which the current is equal to or less than a predetermined value (or minimum) is calculated, and a voltage command value Va * for driving the synchronous motor is calculated at the current phase angle βm. The voltage calculation means 15 calculates the magnitude Va based on the voltage command values Vγ * and Vδ * which are the previous output values of the current control means 250. Further, the reference excitation current command value calculation means 18 calculates a reference excitation current command value iγ0 based on the estimated speed ωm obtained from the rotor position / speed estimation means 20. In the excitation current command value calculation means 175, the voltage command value Va *, the voltage magnitude Va (or the voltage error between the voltage command value Va * and the output voltage command value (Va * −Va)), and the reference excitation current command value are obtained. Based on the reference excitation current command value iγ0 obtained from the calculation means 18, an excitation current command value iγ * for driving the synchronous motor at a current below a predetermined value (or minimum) is output. In this way, the current phase control means 145 outputs the excitation current command value iγ * by the excitation current command value calculation means 175.
[0153]
The filter means 19 removes the high-frequency component of the excitation current command value iγ *, which is the output of the current phase control means 145, to suppress the variation for each calculation cycle and to perform the stable control so as to perform the excitation current command value iγ *. Filtering is performed to remove high-frequency components using a low-pass filter or the like. Here, in a steady operation state in which the rotation frequency and load torque of the synchronous motor do not substantially change, the excitation current command value iγ * hardly changes. When a low-pass filter is used as the filter means 19 in such a case, the time constant of the low-pass filter may be set to a value sufficiently larger than the excitation current control period (the calculation period in the current phase control means 145). For example, the time constant of the low-pass filter is determined by the current control unit 250 excluding the excitation current control cycle (the calculation cycle in the current phase control unit 145) and the calculation cycle other than the excitation current calculation (for example, the calculation cycle in the current phase control unit 145). For example, when the excitation current control period is about 10 msec, the time constant of the low-pass filter may be set to about 1 sec.
[0154]
The γ-axis current comparison means 230 compares the γ-axis current command value (excitation current command value) iγ * filtered by the filter means 19 with the γ-axis current value iγ obtained by the three-phase / two-phase conversion means 7, and The difference (γ-axis current difference) Δiγ is obtained. Current control means 250 obtains output voltage command values Vγ * and Vδ * by proportional-integral control (current control) from γ-axis current difference Δiγ and δ-axis current difference Δiδ.
[0155]
The two-phase / three-phase conversion means 11 converts the γ-δ axis voltage command values Vγ * and Vδ * into voltage command values Vu *, Vv * and Vw * in a three-phase coordinate system using the electrical angle phase θ. The PWM signal generating means 12 generates a PWM signal for on / off control of the switching elements in the inverter main circuit 3 from the three-phase voltage command values Vu *, Vv *, Vw *. Based on this PWM signal, the switching element in the inverter main circuit 3 is turned on / off, the DC voltage of the DC power supply 4 is converted into a three-phase AC and applied to the synchronous motor 5 as a driving load, and the synchronous motor 5 is driven. Is done.
[0156]
The inverter device and the synchronous motor driving device in the present embodiment are another example of a device that drives without using a position sensor that detects the rotor position. What is the inverter device and the synchronous motor driving device in the first embodiment? It is different, and represents an example of a control block diagram in the case of vector control of the synchronous motor 5 that is a driving load. The difference from the first embodiment is that a rotor position / speed estimation means 20, a speed control means 220, and a current control means 250 for estimating the rotor position are provided. As in the present embodiment, even in a drive device including a speed controller and a current controller, the synchronous motor 5 is in a state where the torque is not less than a predetermined value (or maximum), that is, the current is not more than a predetermined value (or minimum). ) With high efficiency. In the high efficiency driving method (current phase control method) in which the current is less than or equal to the predetermined value (or the minimum), if the control is performed by the same control method as the method described in FIGS. 5 and 7 of the first embodiment, It is possible to obtain a highly reliable motor control method with low cost, high efficiency, high output and simple control.
[0157]
In the present embodiment, the current detection means 6a, 6b for detecting the current flowing through the synchronous motor, the current signal obtained by the current detection means 6a, 6b and the voltage signal of the voltage applied to the synchronous motor 5 are used. Current phase control means 145 for outputting an excitation current command value iγ * for controlling the current phase angle of the current flowing through the synchronous motor 5 so that the output torque of the synchronous motor 5 is equal to or greater than a predetermined value (or the maximum); The torque current command value iδ is obtained from the filter means 19 for filtering the excitation current command value iγ * obtained by the current phase control means 145 and the estimated speed ωm obtained from the current or current signal detected by the current detection means 6a and 6b. The speed control means 220 for calculating *, the torque current command value iδ * obtained from the speed control means 220 and the filter means 19 Based on the filtered excitation current command value iγ *, output voltage command values Vγ * and Vδ * for driving the synchronous motor at a current phase angle βm such that the output torque is greater than or equal to a predetermined value (or the maximum). Current control means 250 to be obtained, the torque current is also controlled so that the speed controllability is good, and since the torque control is possible, noise and vibration can be suppressed. In addition, since the filter means 19 is provided, it is possible to obtain a motor drive device and an inverter device that can suppress the variation of the excitation current command value iγ * for each calculation cycle and perform stable control.
[0158]
Further, in the present embodiment and the first embodiment, when driving the synchronous motor 5 that is a driving load, the current phase control means 14 and 140 are from the start until reaching a predetermined rotational speed (low speed rotation). When 145 is executed, there may be a case where the synchronous operation cannot be maintained due to a change in the output torque of the synchronous motor 5 at the time of execution of the control and the step-out occurs. For this reason, if the current phase control means 14, 140, 145 are not executed until the synchronous motor 5 reaches the predetermined number of revolutions or more, step-out can be avoided and a highly reliable device is obtained. be able to.
[0159]
Further, in the present embodiment and the first embodiment, in the region where the PWM signal is overmodulated, it becomes difficult to control the output voltage due to overmodulation, so the current phase control means 14, 140, 145 are executed. Almost no effect. For this reason, in the overmodulation region, the control may be simplified by not executing the current phase control means 14, 140, and 145. Especially when integral control is used for excitation current control, etc., the operation of the integrator becomes abnormal during overmodulation, so stable control can be achieved without current phase control in the overmodulation region, and high efficiency. A highly reliable device can be obtained.
[0160]
Further, in the present embodiment and the first embodiment, if the current phase control means 14, 140, 145 are executed in an unsteady operation state such as startup or acceleration / deceleration operation, the control becomes unstable. In the non-steady operation state (startup, acceleration / deceleration, etc.), the reference excitation current command value iγ0 obtained from the reference excitation current command value calculation means 18 is set to a stronger value. If iγ0 set to a higher value is used as the excitation current command value iγ * (iγ * = iγ0), the control does not become unstable or step out and can be reliably controlled. A highly functional device can be obtained (strong excitation control). (In this case, strong excitation generally refers to a case in which an excitation current flows so that the combined magnetic flux vector Φ0 intensifies the magnetic flux vector ΦF by the rotor magnet (when id> 0), and conversely, the combined magnetic flux vector rotates. (When excitation current is applied so as to weaken the magnetic flux vector ΦF by the child magnet (when id <0), it is called weakening excitation.)
[0161]
In addition, the inverter device and the synchronous motor driving device described in the present embodiment and the first embodiment are examples of a sensorless driving method that does not use a position sensor that detects the rotor position of the synchronous motor 5 that is a driving load. Since a position sensor is not required, a low-cost and highly reliable device can be obtained. The control method described in the present embodiment can also be implemented in a drive method (with a position sensor) for detecting the rotor position, and the same effect can be obtained.
[0162]
Further, in the inverter device and the synchronous motor driving device described in the present embodiment and the first embodiment, a control amount (excitation current command value) for performing high-efficiency control in which the current is equal to or less than a predetermined value (or minimum). In this calculation, the characteristics of the motor such as R (resistance value per phase of IPMSM), Ld (d-axis inductance of IPMSM), Lq (q-axis inductance of IPMSM), and Φf (induced voltage constant of IPMSM) are shown. Calculations are performed using motor constants. For this reason, when the temperature of the motor changes due to the heat generation of the motor or changes in the surrounding environment, the motor constant changes, and therefore an error may occur between the actual motor constant and the motor constant used in the calculation. When this error occurs, when high-efficiency control with the minimum current is performed, the operation point at which the current is minimum shifts from the actual current minimum point, and control at the current minimum point becomes difficult.
[0163]
In order to realize high-efficiency driving with a minimum current even for such a change in the motor constant, for example, the output voltage command value calculation means 10 described in the first embodiment and the current control described in the second embodiment. A motor constant measuring means for measuring a motor constant from a current value and a voltage value flowing through the driving load 5 and a control constant changing means for changing a control constant such as a control gain and a frequency compensation gain are provided in the means 250 or the like. On-line or off-line, the motor constant is measured by the motor constant measuring means, the motor constant is corrected with respect to changes in the surrounding environment such as temperature, and the control constant changing means is changed according to the corrected motor constant. The control constant may be changed. By doing so, the control constant can be changed by following the change of the motor constant (such as temperature change), so that the change of the motor constant occurs due to the change of the surrounding environment such as the temperature change of the motor. Even in such a case, high-efficiency and high-power control with the maximum output torque and the minimum current can be realized, and a highly efficient inverter device and electric motor drive device can be obtained.
[0164]
Further, in the present embodiment and the first embodiment, the example of the embedded magnet type synchronous motor (IPMSM) is shown as the synchronous motor 5, but the embodiment is applicable to any synchronous motor as long as it is a synchronous motor. 1 and the effect equivalent to the effect demonstrated in Embodiment 2 can be acquired. For example, a synchronous motor having a permanent magnet, such as a permanent magnet type synchronous motor (PMSM) or a brushless DC motor (BLDCM), can be realized by using mathematical expressions equivalent to those in the first and second embodiments. In particular, in the permanent magnet type synchronous motor (PMSM), in the surface magnet type synchronous motor (SPMSM), the d-axis inductance Ld and the q-axis inductance Lq in Formulas 1 to 13 may be set to Ld = Lq, and the reluctance motor (RM), synchronous reluctance motor (SyRM), switched reluctance motor (SRM), and other synchronous motors that do not have a permanent magnet, the induced voltage constant Φf in Equations 1 to 13 is calculated as Φf = 0. That's fine.
[0165]
【The invention's effect】
According to a first aspect of the present invention, there is provided a synchronous motor drive device based on a current detection means for detecting a current flowing in a synchronous motor, a current signal obtained by the current detection means and a voltage signal of a voltage applied to the synchronous motor. Current phase control means for outputting an excitation current command value for controlling the current phase angle of the current flowing through the synchronous motor so that the output torque of the motor is equal to or greater than a predetermined value, and the excitation current command output by the current phase control means A filter means for filtering the value, a frequency calculating means for obtaining a compensation amount for the frequency command value from the fluctuation amount of the current detected by the current detecting means, and outputting a frequency command value compensated by the compensation amount, and a frequency computing means The synchronous motor is driven based on the compensated frequency command value and the excitation current command value filtered by the filter means. Output voltage command value calculating means for obtaining an output voltage command value for the output voltage command value based on the excitation current command value and the compensated frequency command value at a current phase angle at which the current is equal to or less than a predetermined value. In order to calculate the value, even when the operating condition of the synchronous motor changes due to the rotation frequency or load fluctuation, the current can be controlled with the current phase angle at which the current value becomes a predetermined value or less according to the output torque under the condition, A synchronous motor drive device with low current and high efficiency can be realized. Further, since the filter means 19 is provided, the variation of the excitation current command value iγ * for each calculation is reduced, and stable control can be performed.
[0166]
The synchronous motor driving device according to claim 2 of the present invention comprises a current phase calculation means for obtaining a current phase angle such that the output torque of the synchronous motor is equal to or greater than a predetermined value based on the current signal obtained by the current detection means, Voltage command value calculation means for calculating a voltage command value based on the current phase angle and frequency command value obtained by the phase calculation means, and voltage calculation for determining the magnitude of the voltage from the voltage signal of the voltage applied to the synchronous motor Excitation current command value calculating means for obtaining an excitation current command value for driving the synchronous motor based on the voltage command value obtained by the voltage command value calculating means and the voltage magnitude obtained by the voltage calculating means Since the current phase control means is configured by the above, a simple configuration for obtaining the difference between the output voltage and the actual output voltage at the current phase angle at which the current is below a predetermined value (or minimum). However, since there is no need to have a numerical table or function for the excitation current command value for each operating condition, an inexpensive microcomputer with a small memory capacity can be used without increasing the memory usage of the microcomputer used for the inverter control means. Since it can be used and can be handled with a simple control circuit, a highly reliable and inexpensive synchronous motor drive device can be obtained.
[0167]
According to a third aspect of the present invention, there is provided a synchronous motor driving apparatus comprising: a current that causes the output torque of the synchronous motor to exceed a predetermined value based on the current signal detected by the current detection means and the voltage signal of the voltage applied to the synchronous motor. Voltage command value calculation means for obtaining a voltage command value for driving at a phase angle, voltage calculation means for obtaining a voltage magnitude from a voltage signal of a voltage applied to the synchronous motor, and voltage command value calculation means The voltage error calculation means for comparing the voltage command value and the magnitude of the voltage obtained by the voltage calculation means to obtain a voltage error, and the output torque of the synchronous motor exceeds a predetermined value from the voltage error obtained from the voltage error calculation means Since the current phase control means is configured by the excitation current command value calculation means for calculating the excitation current command value for driving at the current phase angle, the rotation frequency and load fluctuation Even when the operating conditions of the synchronous motor change, the current can be controlled with a low current phase angle at which the output torque corresponding to that condition is greater than or equal to a predetermined value, realizing a low current and high efficiency synchronous motor drive device it can. In addition, it is possible to realize a synchronous motor drive device with low current and high efficiency that has good followability to a current phase angle at which the current value is equal to or less than a predetermined value, and has good convergence and stability near the current phase angle. Further, since only the excitation current command value is controlled, a good low current and high efficiency synchronous motor drive device can be realized even in a control means having no speed controller or current controller.
[0168]
In the synchronous motor driving device according to claim 4 of the present invention, the output voltage command value obtained from the output voltage command value calculating means is used for the voltage signal. An electric motor drive device can be realized.
[0169]
The synchronous motor drive device according to claim 5 of the present invention is based on a current detection means for detecting a current flowing through the synchronous motor, a current signal obtained by the current detection means, and a voltage signal of a voltage applied to the synchronous motor. Current phase control means for outputting an excitation current command value for controlling the current phase angle of the current flowing through the synchronous motor so that the output torque of the synchronous motor becomes equal to or greater than a predetermined value, and the excitation obtained by the current phase control means Filter means for filtering the current command value, speed control means for calculating the torque current command value based on the estimated speed obtained from the current detected by the current detection means, torque current command value and filter obtained from the speed control means Current phase angle at which the output torque of the synchronous motor exceeds the predetermined value based on the excitation current command value filtered by the means A current control means for calculating an output voltage command value for driving Te, so with a speed controllability for controlling the torque current may also noise due capable torque control, the vibration can be suppressed.
[0170]
In the synchronous motor drive device according to claim 6 of the present invention, the excitation current control cycle controlled by the current phase control means is different from the output voltage control cycle for calculating the output voltage command value for driving the synchronous motor. Since the processing load of the microcomputer can be reduced, an inexpensive microcomputer can be used, and the degree of freedom in selecting the microcomputer is expanded. In addition, a synchronous motor drive device with a low current and high efficiency can be realized with a simple configuration.
[0171]
In the synchronous motor drive device according to the seventh aspect of the present invention, since the excitation current control by the current phase control means is not performed at the time of low speed rotation from the start to the predetermined number of revolutions, the reliability is unlikely to cause step-out at low speed rotation. A highly efficient synchronous motor drive device can be realized.
[0172]
In the synchronous motor drive device according to claim 8 of the present invention, the excitation current control by the current phase control means is not performed in the region where the PWM modulation is overmodulated, so that the control becomes unstable during overmodulation. A highly efficient synchronous motor drive device that can be suppressed and highly reliable can be realized.
[0173]
A synchronous electric drive device according to claim 9 of the present invention drives a synchronous motor using a predetermined excitation current command value without using a current phase control means at the time of start-up or acceleration / deceleration operation, Since the excitation current command value obtained by the current phase control means is switched to use, the stability of control at the start of the synchronous motor and during transients can be secured, and the highly efficient and highly reliable A synchronous motor driving device can be realized.
[0174]
A synchronous motor drive device according to claim 10 of the present invention changes a motor constant measuring means for measuring a motor constant of the synchronous motor and a control constant for controlling the synchronous motor based on the motor constant measured by the motor constant measuring means. Control constant changing means for controlling the drive of the synchronous motor in accordance with changes in the surrounding environment of the synchronous motor, so that the motor can be controlled by a change in characteristics due to a temperature change of the motor or a change in the environment in which the motor is used. Even when motor characteristics such as constants change, it is possible to automatically adapt to changes in motor constants, and to realize a synchronous motor drive device with low current and high efficiency.
[0175]
Since the synchronous motor drive device according to the eleventh aspect of the present invention is driven without a sensor that does not use a position sensor for detecting the rotor position of the synchronous motor, a high output can be achieved even in a synchronous motor to which a position sensor cannot be attached. Thus, a synchronous motor drive device capable of highly efficient control can be realized.
[0176]
Since the inverter device according to claim 12 of the present invention includes the synchronous motor drive device according to any one of claims 1 to 10, an inverter device with high output and high efficiency can be obtained.
[0177]
According to a thirteenth aspect of the present invention, there is provided an inverter main circuit that converts a direct current from a direct current power source into an alternating current by turning on and off the switching element, and an on / off operation of the switching element in the inverter main circuit. Inverter control means for outputting a voltage command value for control, and a drive load controlled by application of a voltage output by an inverter main circuit controlled by the inverter control means, and the inverter control means Current phase control means for calculating an excitation current command value for driving the drive load at a current phase angle such that the output torque of the drive load becomes a predetermined value or more based on the current flowing through the drive load; and current phase control Filter means for filtering the excitation current command value obtained from the means, and the fluctuation amount of the current flowing through the drive load. Frequency calculating means for compensating the numerical command value and outputting it as a primary frequency, and output voltage command value calculating means for calculating a voltage command value for driving the driving load based on the filtered excitation current command value and the primary frequency Therefore, even if the operating condition of the driving load changes, the output torque under the condition can be controlled to be a predetermined value or more. In addition, the current can be controlled at a current phase angle at which the current value is equal to or less than a predetermined value in accordance with the output torque, and a synchronous motor drive device with low current and high efficiency can be realized. Further, since the filter means 19 is provided, the variation in the excitation current command value iγ * is reduced, and a highly reliable inverter device that can stably control the driving load is obtained.
[0178]
According to a fourteenth aspect of the present invention, there is provided an inverter device that obtains a current phase angle that obtains a current phase angle at which an output torque becomes a predetermined value or more by a current flowing through a driving load, and a current phase angle that is obtained by the current phase calculation means. Based on the voltage command value calculation means for calculating the voltage command value, the output voltage command value obtained by the output voltage command value calculation means, and the voltage command value obtained by the voltage command value calculation means. Since the current phase control means is constituted by the excitation current command value calculating means for obtaining the excitation current command value for driving, the followability to the current phase angle is such that the current value flowing through the drive load is a predetermined value or less. In addition, since the convergence and stability near the current phase angle where the current value is equal to or less than the predetermined value are good, it is possible to realize an inverter device with good controllability and low current and high efficiency. Further, since only the excitation current command value is controlled, a low-current and high-efficiency inverter device can be realized even without a speed controller or current controller.
[0179]
According to a fifteenth aspect of the present invention, there is provided a control method for a synchronous motor driving device, comprising: a current phase calculation step for obtaining a current phase angle such that an output torque of the synchronous motor becomes equal to or greater than a predetermined value based on a current value flowing through the synchronous motor; An excitation current command value calculation step for obtaining an excitation current command value for driving the synchronous motor based on the current phase angle obtained in the current phase calculation step and the voltage value applied to the synchronous motor; A filter step for filtering to remove high frequency components of the excitation current command value obtained in the command value calculation step, and an output voltage for driving the synchronous motor based on the excitation current command value filtered in the filter step Equipped with an output voltage command value calculation step to obtain the command value. Control method for controlling the enable synchronous motor Te can be obtained. In addition, because of simple control that controls only the excitation current command value, a control method that can accurately drive the synchronous motor with low current and high efficiency can be obtained even in a control means that does not have a speed controller or current controller.
[0180]
According to a sixteenth aspect of the present invention, there is provided a control method for a synchronous motor driving apparatus for driving a synchronous motor at a current phase angle such that an output torque of the synchronous motor becomes a predetermined value or more based on a current value flowing through the synchronous motor. A voltage command value calculating step for obtaining the voltage command value, a voltage calculating step for calculating the magnitude of the voltage applied to the synchronous motor, a voltage value obtained in the voltage calculating step, and a voltage command value calculating step. A voltage error calculation step that compares the obtained voltage command value to obtain a voltage error, and a current that causes the output torque of the synchronous motor to be a predetermined value or more based on the voltage error obtained in the voltage error calculation step The excitation current command calculation step for calculating the excitation current command value for driving the synchronous motor with the phase angle, and the high frequency of the excitation current command value obtained by the excitation current command value calculation step A filter step for performing filtering to remove components, and an output voltage command value calculation step for obtaining an output voltage command value for driving the synchronous motor based on the excitation current command value filtered in the filter step. Therefore, the controllability is good while being simple control, and the synchronous motor can be driven with a low current. In addition, even when the operating conditions of the synchronous motor change due to rotational frequency and load fluctuations, the current can be controlled with a low current phase angle at which the current is below a predetermined value according to the conditions, resulting in low current and high efficiency. Can drive a synchronous motor. In addition, the followability to the low current phase is good and the convergence and stability in the vicinity of the low current phase are good, so that the controllability can be improved. In addition, since the filter step is included, a highly reliable synchronous motor control method capable of controlling the synchronous motor in a stable state can be obtained.
[0181]
According to a seventeenth aspect of the present invention, there is provided a method for controlling a synchronous motor driving apparatus comprising: a frequency command value compensated by a frequency compensation amount obtained based on a current flowing through the synchronous motor; and an excitation current command value calculating step. Since there is an output voltage command value calculation step for obtaining an output voltage command value for driving the synchronous motor based on the current command value, an excitation current command value and compensation at a current phase angle at which the current is equal to or less than a predetermined value Since the output voltage command value is calculated on the basis of the frequency command value thus set, even when the operating condition of the synchronous motor is changed due to the rotation frequency or load fluctuation, it is possible to accurately follow and perform stable control.
[Brief description of the drawings]
FIG. 1 is a diagram showing an example of a configuration of a synchronous motor driving device representing Embodiment 1 of the present invention.
FIG. 2 is a diagram for explaining a relationship between a current phase angle and a generated torque of the synchronous motor in the first embodiment of the present invention.
FIG. 3 is a diagram for explaining the relationship between the current phase angle and current of the synchronous motor according to Embodiment 1 of the present invention with respect to changes in output torque.
FIG. 4 is a diagram showing a vector diagram of a general synchronous motor in Embodiment 1 of the present invention.
FIG. 5 is a flowchart illustrating a method for controlling the synchronous motor drive device according to the first embodiment of the present invention.
FIG. 6 is a diagram showing an example of another configuration of the synchronous motor driving device representing Embodiment 1 of the present invention.
FIG. 7 is a flowchart illustrating another control method for the synchronous motor drive device according to the first embodiment of the present invention.
FIG. 8 is a diagram showing an example of a configuration of a synchronous motor driving device representing Embodiment 2 of the present invention.
FIG. 9 is a diagram showing an example of a configuration of a conventional synchronous motor driving device.
[Explanation of symbols]
DESCRIPTION OF SYMBOLS 1 Synchronous motor drive device, 2 Inverter control means, 3 Inverter main circuit, 4 DC power supply, 5 Synchronous motor, 6a, 6b Current detection means, 7 3 phase 2 phase conversion means, 8 Frequency compensator, 9 Primary frequency calculation means, 10 output voltage command value calculation means, 11 two-phase three-phase coordinate conversion means, 12 PWM signal generation means, 13 electrical angle phase calculation means, 14 current phase control means, 15 output voltage calculation means, 16 voltage command value calculation means, 17 Excitation current command calculation means, 18 reference excitation current command calculation means, 19 filter means, 20 rotor position / speed estimation means, 21 current calculation means, 22 current phase calculation means, 140, 145 current phase control means, 160, 165 voltage Command value calculation means, 170, 175 Excitation current command value calculation means, 210 Frequency comparison means, 220 Speed control means, 225 Voltage error calculation Stage, 230 γ-axis current comparison means, 240 δ-axis current comparison means, 250 current control means.

Claims (17)

Based on the current detection means for detecting the current flowing through the synchronous motor, the current signal obtained by the current detection means and the voltage signal of the voltage applied to the synchronous motor, the output torque of the synchronous motor becomes a predetermined value or more. Current phase control means for outputting an excitation current command value for controlling the current phase angle of the current flowing through the synchronous motor, and filter means for filtering the excitation current command value output by the current phase control means, A frequency calculation means for obtaining a compensation amount for a frequency command value from a fluctuation amount of the current detected by the current detection means, and outputting a frequency command value compensated by the compensation amount; and a frequency command compensated by the frequency calculation means The synchronous motor is driven based on the value and the excitation current command value filtered by the filter means Synchronous motor drive apparatus comprising: the output voltage command value calculating means for calculating an output voltage command value of the order, the. A current phase calculating means for obtaining a current phase angle such that an output torque of the synchronous motor becomes a predetermined value or more based on a current signal obtained by the current detecting means; and a current phase angle obtained by the current phase calculating means. And a voltage command value calculating means for calculating a voltage command value based on the frequency command value, a voltage calculating means for obtaining a voltage magnitude from a voltage signal of a voltage applied to the synchronous motor, and the voltage command value calculating means Excitation current command value calculation means for obtaining an excitation current command value for driving the synchronous motor based on the voltage command value obtained by the above and the magnitude of the voltage obtained by the voltage calculation means. 2. The synchronous motor drive device according to claim 1, wherein the phase control means is configured. A voltage command value for driving at a current phase angle such that the output torque of the synchronous motor is equal to or greater than a predetermined value based on the current signal detected by the current detection means and the voltage signal of the voltage applied to the synchronous motor Voltage command value calculating means for determining voltage, voltage calculating means for determining the magnitude of the voltage from a voltage signal of the voltage applied to the synchronous motor, the voltage command value obtained by the voltage command value calculating means and the voltage calculation A voltage error calculation means for obtaining a voltage error by comparing the magnitude of the voltage obtained by the means, and a current phase angle at which the output torque of the synchronous motor is equal to or greater than a predetermined value from the voltage error obtained from the voltage error calculation means 2. The current phase control means comprises: an excitation current command value calculating means for calculating an excitation current command value for driving at a current level. Synchronous motor driving device according. 4. The synchronous motor drive device according to claim 1, wherein an output voltage command value obtained from the output voltage command value calculating means is used for the voltage signal. Based on the current detection means for detecting the current flowing through the synchronous motor, the current signal obtained by the current detection means and the voltage signal of the voltage applied to the synchronous motor, the output torque of the synchronous motor becomes a predetermined value or more. Current phase control means for outputting an excitation current command value for controlling the current phase angle of the current flowing through the synchronous motor, and filter means for filtering the excitation current command value obtained by the current phase control means, A speed control means for calculating a torque current command value based on an estimated speed obtained from the current detected by the current detection means; a torque current command value obtained from the speed control means; and the filter means filtered by the filter means. Based on the excitation current command value, the synchronous motor is set to a current phase angle so that the output torque becomes a predetermined value or more. Synchronous motor drive apparatus characterized by comprising a current control means for calculating an output voltage command value for driving. 6. The excitation current control cycle controlled by the current phase control means is made different from an output voltage control cycle for calculating an output voltage command value for driving a synchronous motor. A synchronous motor drive device according to claim 1. The synchronous motor drive device according to any one of claims 1 to 6, wherein excitation current control by the current phase control means is not performed during low-speed rotation from startup to a predetermined rotational speed. 8. The synchronous motor drive device according to claim 1, wherein excitation current control by the current phase control means is not performed in a region where PWM modulation is overmodulated. During startup and acceleration / deceleration operation, the synchronous motor is driven using a predetermined excitation current command value without using the current phase control means, and after reaching a predetermined operation condition, the current phase control means The synchronous motor drive device according to any one of claims 1 to 8, wherein switching is performed so that a required excitation current command value is used. Motor constant measuring means for measuring a motor constant of the synchronous motor, and control constant changing means for changing a control constant for controlling the synchronous motor based on the motor constant measured by the motor constant measuring means, the synchronization The synchronous motor drive device according to any one of claims 1 to 9, wherein the synchronous motor is driven and controlled in accordance with a change in an ambient environment of the motor. The synchronous motor drive device according to any one of claims 1 to 10, wherein the drive is performed without using a position sensor for detecting a rotor position of the synchronous motor. An inverter device comprising the synchronous motor drive device according to any one of claims 1 to 11. A direct current power source for supplying direct current, an inverter main circuit for converting direct current from the direct current power source into alternating current by turning on and off the switching element, and controlling on and off operations of the switching element in the inverter main circuit Inverter control means for outputting a voltage command value for driving, and a drive load controlled by applying a voltage output by the inverter main circuit controlled by the inverter control means, the inverter control means Current phase control means for calculating an excitation current command value for driving the drive load at a current phase angle such that an output torque of the drive load becomes a predetermined value or more based on a current flowing through the drive load; Filter means for filtering the excitation current command value obtained from the current phase control means; and the drive negative Frequency calculation means for compensating the frequency command value from the amount of fluctuation of the current flowing through it and outputting it as a primary frequency, and a voltage command for driving the drive load based on the filtered excitation current command value and the primary frequency An inverter device comprising: an output voltage command value calculating means for calculating a value. Based on the current flowing through the driving load, a current phase calculation means for obtaining a current phase angle at which the output torque becomes a predetermined value or more, and a voltage command value for obtaining the current phase angle obtained by the current phase calculation means Excitation for driving the drive load based on the voltage command value calculating means, the output voltage command value obtained by the output voltage command value computing means and the voltage command value obtained by the voltage command value computing means 14. The inverter device according to claim 13, wherein the current phase control means is constituted by exciting current command value calculation means for obtaining a current command value. A current phase calculation step for obtaining a current phase angle such that an output torque of the synchronous motor is equal to or greater than a predetermined value based on a current value flowing through the synchronous motor; the current phase angle obtained in the current phase calculation step; and An excitation current command value calculation step for obtaining an excitation current command value for driving the synchronous motor based on a voltage value applied to the synchronous motor, and an excitation current command obtained in the excitation current command value calculation step A filter step for performing filtering to remove high-frequency components of the value, and an output voltage command value calculating step for obtaining an output voltage command value for driving the synchronous motor based on the excitation current command value filtered in the filter step And a method for controlling a synchronous motor. A voltage command value calculating step for obtaining a voltage command value for driving the synchronous motor at a current phase angle such that an output torque of the synchronous motor becomes equal to or greater than a predetermined value based on a current value flowing through the synchronous motor; A voltage calculation step for calculating the magnitude of the voltage applied to the electric motor, and a voltage obtained by comparing the voltage magnitude obtained in the voltage computation step with the voltage command value obtained in the voltage command value computation step A voltage error calculating step for obtaining an error, and driving the synchronous motor at a current phase angle such that an output torque of the synchronous motor becomes a predetermined value or more based on the voltage error obtained in the voltage error calculating step. An excitation current command calculation step for calculating the excitation current command value of the current, and a high frequency component of the excitation current command value obtained by the excitation current command value calculation step are removed. A filter step for performing tulling; and an output voltage command value calculation step for obtaining an output voltage command value for driving the synchronous motor based on the excitation current command value filtered in the filter step. A control method of a synchronous motor as a feature. To drive the synchronous motor based on the frequency command value compensated by the frequency compensation amount obtained based on the current flowing through the synchronous motor and the filtered excitation current command value obtained by the filter step 17. The method for controlling a synchronous motor according to claim 15, further comprising an output voltage command value calculation step for obtaining an output voltage command value.

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