JP2003259680A - Synchronous motor driving apparatus, inverter apparatus and control method of synchronous motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide an inverter apparatus or a synchronous motor driving apparatus and a control method of a synchronous motor which have a high output, high efficiency and excellent controllability. <P>SOLUTION: The apparatus is provided with a current phase control means which outputs an excitation current command value for controlling the current phase angle of a current so that output torque is not less than a prescribed value on the basis of a current signal obtained by a current detecting means for detecting a current flowing into a synchronous motor and a voltage signal of a voltage applied to the motor, a filtering means which filters the excitation current command value, a frequency operating means which determines a compensation amount for a frequency command value from a fluctuation amount of the current detected by the current detecting means and outputs a frequency command value compensated by the compensation amount, and an output voltage command value operating means which determines an output voltage command value for driving the motor on the basis of the compensated frequency command value and the filtered excitation current command value. <P>COPYRIGHT: (C)2003,JPO

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a synchronous motor drive device for driving a synchronous motor, an inverter device, and a synchronous motor control method.

[0002]

2. Description of the Related Art FIG. 9 is a diagram showing a structure of a general conventional synchronous motor drive device. In the figure, 1 is a synchronous motor driving device, 2 is inverter control means, 3 is an inverter main circuit composed of a plurality of switching elements, 4 is a DC power source connected to the inverter main circuit, and 5 is a synchronous motor.

Reference numeral 6a denotes a current detecting means for detecting a one-phase current (for example, U-phase current iu) of the current flowing into the synchronous motor 5, and 6b denotes a current for a phase different from the current detecting means 6a (for example, V-phase current iv). ) For detecting current, 7
Is a three-phase two that converts the current value detected by the current detecting means 6a, 6b into the γ-δ axis current representing the exciting current component (γ axis current) and the torque current component (δ axis current) using the electrical angle phase. Phase conversion means 8 is a frequency compensator for obtaining the compensation amount ωd of the rotation frequency from the δ-axis current obtained by the three-phase / two-phase conversion means 7, and 9 is the frequency compensation amount ωd of the frequency compensator 8 and the rotation frequency command value ωm. The primary frequency calculating means for obtaining the primary frequency ω1 from *, 13 is the electrical angle phase calculating means for obtaining the electrical angle phase θ by integrating the primary frequency ω1, and 50 is the exciting current command value iγ * from the rotation frequency command value ωm *. It is an exciting current command value calculating means to be obtained.

Reference numeral 10 denotes γ obtained by the three-phase / two-phase conversion means 7.
-Voltage command values Vγ *, Vδ for driving the synchronous motor 5 from the δ-axis current, the primary frequency ω1 obtained by the primary frequency calculation means 9 and the excitation current command value iγ * obtained by the excitation current calculation means 50. An output voltage command value calculating means for calculating *, 11 is a three-phase voltage command value Vu using the electrical angle phase θ for the γ-δ axis voltage command values Vγ *, Vδ * obtained by the output voltage command value calculating means 10. Two-phase / three-phase conversion means for converting into *, Vv *, Vw *, and 12 is a switching element in the inverter main circuit 3 according to three-phase voltage command values Vu *, Vv *, Vw * obtained from the two-phase three-phase conversion means. Is a PWM signal generating means for generating a PWM signal for controlling ON / OFF of the.

The operation of the conventional synchronous motor drive device 1 and the synchronous motor 5 configured as described above will be described with reference to FIG. In the figure, the synchronous motor drive device 1 detects currents for two phases (for example, iu and iv) among the phase currents flowing into the synchronous motor 5 by the current detection means 6a and 6b. Using the detected currents of two phases, for example, the U-phase current iu and the V-phase current iv, the inverter control unit 2 calculates the voltage output from the inverter main circuit 3 for driving the synchronous motor 5, and the inverter A PWM signal for controlling on / off of the switching element in the main circuit 3 is output.

The inverter control means 2 outputs a PWM signal by the operation described below. Current detecting means 6a, 6
From the phase currents iu and iv detected by b, the electrical angle phase θ
The γ-δ axis current iγ, i from the three-phase / two-phase conversion means 7 using
Find δ. The frequency compensator 8 obtains the frequency compensation amount ωd from the δ-axis current iδ. The primary frequency calculation means 9 obtains the primary frequency ω1 by taking the difference between the rotation frequency command value ωm * and the frequency compensation amount ωd. The exciting current calculation means 50 obtains the exciting current command value (γ-axis current command value) iγ * from the rotation frequency command value ωm *.

The output voltage command value calculating means 10 has a primary frequency ω1, an exciting current command value iγ *, a γ-δ axis current iγ and iδ.
From the voltage command value Vγ for driving the synchronous motor 5
*, Vδ * is calculated. The 2-phase / 3-phase conversion means 11 converts the γ-δ axis voltage command values Vγ *, Vδ * into the voltage command values Vu *, Vv *, Vw * of the three-phase coordinate system using the electrical angle phase θ.
The PWM signal generating means 12 has three-phase voltage command values Vu *, Vv.
A PWM signal for controlling on / off of the switching element in the inverter main circuit 3 is generated from * and Vw *. Based on this PWM signal, the switching element in the inverter main circuit 3 is turned on / off, the DC voltage of the DC power supply 4 is converted into a three-phase AC, and applied to the synchronous motor 5.

Here, in the conventional synchronous motor drive device, the exciting current calculating means 50 causes the rotation frequency command value ω
From the data table corresponding to m * or the function expressing the relationship with the rotation frequency command value ωm *, the exciting current command value iγ *
Is required. The exciting current command value iγ * is determined in advance according to the rotation frequency command value ωm * based on a theoretical formula or a value obtained by an experiment.

An example of a conventional synchronous motor driving device is disclosed in Japanese Patent Laid-Open No. 2000-209886.
In the synchronous motor drive device disclosed in Japanese Unexamined Patent Publication No. 2000-209886, with respect to the rotor position of the synchronous motor, the energization phase of the voltage is determined in the direction in which the current decreases from the previous energization phase information and the current information. An example is disclosed in which the current phase angle is minimized while changing with time.

An example of a conventional synchronous motor drive device is disclosed in Japanese Patent Laid-Open No. 2001-8486. JP 2
In the synchronous motor drive device disclosed in the publication No. 001-8486, in a device provided with a speed control device for speed control, the exciting current id is changed when the load torque is constant, and the previous exciting current id and current information are changed. From the above, an example is disclosed in which the exciting current id is changed in the direction in which the current decreases and the phase approaches the phase in which the current becomes the minimum.

An example of a conventional synchronous motor drive device is disclosed in Japanese Patent Laid-Open No. 2001-86782. In the synchronous motor drive device disclosed in Japanese Unexamined Patent Publication No. 2001-86782, in which speed control is performed by a speed control device, the d-axis inductance, the q-axis inductance, the torque constant, and the number of pole pairs of the brushless DC motor are set as motor constants. Current phase angle β that maximizes generated torque
Is calculated to obtain the exciting current command value id * and the torque current command value iq * which is the output of the speed controller so that the current phase angle β that maximizes the generated torque is obtained, and the actual current becomes the current command value. An example of performing current control is disclosed.

[0012]

In the conventional synchronous motor drive device, the exciting current calculation means 50 uses a data table corresponding to the rotation frequency command value or a function expressing the relationship with the rotation frequency command value. Is required. This excitation current command value is set to a value that can achieve maximum efficiency at the rated load torque according to the rotation frequency command value (a value for realizing the current phase angle that minimizes the current). Under the conditions, maximum efficiency can be achieved only under the assumed rated load torque condition.
When there is a difference between the rated load torque condition and the actual load torque condition, the current phase angle deviates from the maximum efficiency point, and thus the drive is performed in a poor efficiency state.

In addition, when the load target of the synchronous motor is the compressor of the air conditioner, the load torque varies under the same rotation frequency as in the cooling operation and the heating operation. Therefore, under the same rotation frequency condition, since there is only one condition for the exciting current command value, it is difficult to realize high-efficiency operation with the minimum current in accordance with the load torque. In addition, if an exciting current command value condition is provided for each operating condition, the amount of memory such as a microprocessor used increases, which causes a cost increase of the control device. Also in this case, if the actual load torque condition deviates from the expected rated load torque condition,
Maximum efficiency operation could not be realized.

Further, in the devices described in JP-A-2000-209886 and JP-A-2001-8486, the energization phase that minimizes the current is not obtained in the control process, and the current phase angle that minimizes the current is determined. It took time to reach. In addition, in this method, the convergence to the energization phase that minimizes the current is not so good, so the energization phase slightly changes near the energization phase where the current minimum phase angle occurs, and as a result, the motor current fluctuates and stable operation is achieved. There were times when I couldn't. Further, in JP 2001-8486 A, since the control is performed only when the load torque is constant, it cannot be used for a load in which the load torque constantly fluctuates.

Further, in the synchronous motor drive device disclosed in JP 2001-86782 A, brushless DC
The d-axis inductance of the motor, the q-axis inductance, the torque constant, and the motor constants such as the number of pole pairs are used to calculate the current phase angle β that maximizes the generated torque, and speed control is performed so that the current phase angle β that maximizes the generated torque is obtained. The exciting current command value id * and the torque current command value iq *, which are the output of the device, are obtained, and the current is controlled so that the actual current becomes the current command value, thereby aiming for high efficiency. Therefore, id * and iq
A speed controller for obtaining * was required, and this method could not be used without a speed controller. Further, since the filter means is not provided, the variation in the exciting current command value for each calculation cycle becomes large, the operation of the synchronous motor becomes unstable, and the controllability is poor.

The present invention has been made to solve the above problems, and an object of the present invention is to obtain an inverter device or a synchronous motor drive device with high output. Another object is to obtain an efficient inverter device or synchronous motor drive device. Another object of the present invention is to obtain an inverter device or a synchronous motor drive device with good controllability. Another object of the present invention is to obtain an inverter device or a synchronous motor drive device that can be controlled in an efficient state in a short time. Another object is to obtain a low-cost inverter device or synchronous motor drive device. Also,
An object of the present invention is to obtain a highly efficient inverter device or a synchronous motor drive device that can be driven under the condition that the current becomes small even if the rotation frequency or the load torque changes. Another object of the present invention is to obtain an efficient method for controlling a synchronous motor, which is simple control.

[0017]

According to a first aspect of the present invention, there is provided a synchronous motor driving device including a current detecting means for detecting a current flowing through the synchronous motor, a current signal obtained by the current detecting means, and a synchronous motor. Current phase control means for outputting an exciting current command value for controlling the current phase angle of the current flowing through the synchronous motor so that the output torque of the synchronous motor becomes a predetermined value or more based on the voltage signal of the applied voltage, Filtering means for filtering the exciting current command value output by the current phase control means, and a compensation amount for the frequency command value is obtained from the amount of fluctuation of the current detected by the current detecting means, and the frequency command value compensated by the compensation amount is obtained. Output frequency calculation means, frequency command value compensated by the frequency calculation means, and exciting current finger filtered by the filter means An output voltage command value calculating means for calculating an output voltage command value for driving the synchronous motor based on the value, in which with a.

According to a second aspect of the present invention, there is provided a synchronous motor drive device in which a current phase for obtaining a current phase angle at which the output torque of the synchronous motor becomes a predetermined value or more based on the current signal obtained by the current detecting means. The calculating means, the voltage command value calculating means for calculating the voltage command value based on the current phase angle and the frequency command value obtained by the current phase calculating means, and the voltage larger than the voltage signal of the voltage applied to the synchronous motor. Exciting current for obtaining an exciting current command value for driving the synchronous motor based on the voltage calculating means for determining the height, and the voltage command value obtained by the voltage command value calculating means and the magnitude of the voltage obtained by the voltage calculating means. The current value control means is constituted by the command value calculation means.

According to a third aspect of the present invention, there is provided a synchronous motor driving device in which the output torque of the synchronous motor is equal to or more than a predetermined value based on the current signal detected by the current detecting means and the voltage signal of the voltage applied to the synchronous motor. Voltage command value calculating means for obtaining a voltage command value for driving at a current phase angle such that, voltage calculating means for calculating the voltage magnitude from the voltage signal of the voltage applied to the synchronous motor, and voltage command value calculation The output torque of the synchronous motor is calculated from the voltage error calculation means for calculating the voltage error by comparing the voltage command value obtained by the means with the magnitude of the voltage obtained by the voltage calculation means, and the voltage error obtained by the voltage error calculation means. A current phase control means is constituted by an exciting current command value calculating means for calculating an exciting current command value for driving at a current phase angle of a predetermined value or more.

According to a fourth aspect of the present invention, the synchronous motor drive device uses the output voltage command value obtained from the output voltage command value calculating means for the voltage signal.

According to a fifth aspect of the present invention, there is provided a synchronous motor drive device in which a current detecting means for detecting a current flowing through the synchronous motor, a current signal obtained by the current detecting means and a voltage of a voltage applied to the synchronous motor. The current phase control means outputs an exciting current command value for controlling the current phase angle of the current flowing through the synchronous motor so that the output torque of the synchronous motor becomes a predetermined value or more based on the signal, and is obtained by the current phase control means. Filter means for filtering the generated excitation current command value, speed control means for calculating the torque current command value from the estimated speed obtained from the current detected by the current detection means, and torque current command obtained by the speed control means Based on the value and the excitation current command value filtered by the filter means, the output torque of the synchronous motor becomes equal to or higher than a predetermined value. A current control means for calculating an output voltage command value for driving at a current phase angle, in which with a.

In the synchronous motor drive device according to a sixth aspect of the present invention, the exciting current control period controlled by the current phase control means is different from the output voltage control period for calculating the output voltage command value for driving the synchronous motor. It was done like this.

According to a seventh aspect of the present invention, the synchronous motor drive device is configured such that the excitation current control by the current phase control means is not performed during low speed rotation from startup to a predetermined rotation speed.

According to the eighth aspect of the present invention, in the synchronous electric drive device, the excitation current control by the current phase control means is not performed in the region where the PWM modulation is overmodulation.

According to a ninth aspect of the present invention, there is provided a synchronous motor drive device which drives a synchronous motor using a predetermined exciting current command value without using current phase control means at the time of start-up or acceleration / deceleration operation. After the predetermined operating condition is reached, the excitation current command value obtained by the current phase control means is switched to use.

A synchronous motor drive device according to a tenth aspect of the present invention controls a synchronous motor based on the electric motor constant measuring means for measuring the electric motor constant of the synchronous motor and the electric motor constant measured by the electric motor constant measuring means. A control constant changing means for changing the control constant is provided, and the synchronous motor is drive-controlled according to the change of the surrounding environment of the synchronous motor.

According to the eleventh aspect of the present invention, there is provided a synchronous motor driving device which is driven without using a position sensor for detecting the rotor position of the synchronous motor.

An inverter device according to a twelfth aspect of the present invention is provided with the synchronous motor drive device according to any one of the first to eleventh aspects.

An inverter device according to a thirteenth aspect of the present invention is an inverter main circuit for converting a direct current from a direct current power source into an alternating current by turning on and off a switching element, and an on-state of a switching element in the inverter main circuit. An inverter control unit that outputs a voltage command value for controlling the off operation, and a drive load that is controlled by applying a voltage output by an inverter main circuit controlled by the inverter control unit, And an inverter control means, which is a current phase control means for calculating an exciting current command value for driving the drive load at a current phase angle such that the output torque of the drive load becomes a predetermined value or more based on the current flowing through the drive load. A filter means for filtering the exciting current command value obtained from the current phase control means, and a current flowing through the drive load. Frequency calculation means for compensating the frequency command value from the amount of fluctuation and outputting it as the primary frequency, and output voltage command for calculating the voltage command value for driving the drive load based on the filtered excitation current command value and the primary frequency. And a value calculation means.

According to a fourteenth aspect of the present invention, in the inverter device, a current phase calculating means for obtaining a current phase angle at which the output torque becomes a predetermined value or more by the current flowing through the driving load, and the current obtained by the current phase calculating means. Based on a voltage command value calculation means for calculating a voltage command value such that a phase angle is obtained, and an output voltage command value calculated by the output voltage command value calculation means and a voltage command value calculated by the voltage command value calculation means. The current phase control means is constituted by an exciting current command value calculating means for obtaining an exciting current command value for driving the driving load.

According to a fifteenth aspect of the present invention, in the method for controlling a synchronous motor drive device, a current for obtaining a current phase angle based on a current value flowing through the synchronous motor so that the output torque of the synchronous motor becomes a predetermined value or more. Excitation current command value calculation step for obtaining an excitation current command value for driving the synchronous motor based on the phase calculation step and the current phase angle obtained in the current phase calculation step and the voltage value applied to the synchronous motor And a filter step for removing high-frequency components of the excitation current command value obtained in the excitation current command value calculation step, and a synchronous motor is driven based on the excitation current command value filtered in the filter step. Output voltage command value calculating step for obtaining an output voltage command value for

According to a sixteenth aspect of the present invention, there is provided a method of controlling a synchronous motor drive device, wherein the synchronous motor is driven at a current phase angle such that an output torque of the synchronous motor becomes a predetermined value or more based on a current value flowing through the synchronous motor. Voltage command value calculation step for obtaining the voltage command value for driving the motor, voltage calculation step for calculating the magnitude of the voltage applied to the synchronous motor, voltage magnitude and voltage command value obtained in the voltage calculation step Based on the voltage error obtained in the voltage error calculation step and the voltage error obtained in the voltage error calculation step, the output torque of the synchronous motor is determined to be a predetermined value or more. Excitation current command calculation step for calculating the excitation current command value for driving the synchronous motor at the current phase angle such that A filter step for performing filtering for removing high frequency components of the command value, and an output voltage command value calculation step for obtaining an output voltage command value for driving the synchronous motor based on the excitation current command value filtered in the filter step. , Are provided.

According to a seventeenth aspect of the present invention, there is provided a method of controlling a synchronous motor driving device, wherein a frequency command value compensated by a frequency compensation amount obtained based on a current flowing through the synchronous motor and a filtering obtained by a filter step. An output voltage command value calculation step for obtaining an output voltage command value for driving the synchronous motor based on the generated excitation current command value is provided.

[0034]

BEST MODE FOR CARRYING OUT THE INVENTION Embodiment 1. The inverter device and the synchronous motor drive device according to the first embodiment of the present invention will be described below. 1 is a diagram showing a configuration of an inverter device and a synchronous motor drive device according to a first embodiment of the present invention. In the figure, reference numeral 1 is a synchronous motor drive device, which is composed of a plurality of switching elements, and is an inverter main circuit 3 and an inverter main circuit 3 for converting direct current into alternating current.
The inverter control means 2 outputs a PWM signal for controlling ON / OFF of the plurality of switching elements.

Further, 4 is a DC power source connected to the inverter main circuit 3, and 5 is a synchronous motor which is a drive load driven by the inverter main circuit 3. The synchronous motor drive device 1, the DC power source 4, and the drive load 5 are provided. The inverter device is configured by. The switching element in the inverter main circuit 3 is turned on / off based on the PWM signal output from the inverter control means 2, and the DC voltage of the DC power supply 4 is converted into three-phase AC and applied to the synchronous motor 5 as a drive load. As a result, the inverter device operates.

Reference numeral 6a denotes one-phase current (for example, U-phase current iu) of the current flowing into the synchronous motor 5 which is a driving load.
And 6b is a current detecting means for detecting a current (for example, V-phase current iv) of a phase different from the current (U-phase current iu) detected by the current detecting means 6a. The current detecting means 6a and 6b convert the detected current into a current value or a current signal and output it.

Here, the inverter control means 2 includes the following three-phase / two-phase conversion means 7, frequency compensator 8, primary frequency calculation means 9, output voltage command value calculation means 10, two-phase / three-phase conversion means 11, It is composed of a PWM signal generating means 12, an electrical angle phase calculating means 13, a current phase controlling means 14, and a filter means 19.

The three-phase / two-phase converting means 7 is a current detecting means 6a,
The current value (for example, iu, iv) or current signal detected by 6b is converted into an exciting current component (γ-axis current iγ) and a torque current component (δ-axis current i) by using the electrical angle phase θ.
Convert to the current value or current signal on the γ-δ axis represented by δ). The frequency compensator 8 calculates the compensation amount ωd of the rotation frequency from the δ-axis current obtained by the three-phase / two-phase conversion means 7. The primary frequency calculation means 9 uses the frequency compensation amount ωd obtained by the frequency compensator 8 to compensate the frequency command value ωm * which is the target frequency given from the outside (the primary frequency ω1 which is the compensated frequency command value). Ask.).
The electrical angle phase calculation means 13 integrates the primary frequency ω1 that is the compensated frequency command value to obtain the electrical angle phase θ.

The current phase control means 14 is obtained by the output voltage calculation means 15 for determining the magnitude Va of the output voltage based on the voltage signal of the voltage applied to the synchronous motor 5 and the two-phase / three-phase coordinate conversion means 7. Based on the γ-axis current iγ and the δ-axis current iδ, the current value Ia, which is the magnitude thereof, is calculated based on the current calculation means 21, iγ, iδ, Ia. Current phase calculation means 22 for calculating the current phase angle βm, and the voltage command value V based on the current phase angle βm at which the current value is below a predetermined value (or minimum).
Voltage command value calculation means 16 for obtaining a *, reference excitation current command value calculation means 18 for obtaining a reference excitation current command value iγ0, excitation current command for driving a drive load based on Va *, Va, iγ0 The exciting current command value calculation means 17 for obtaining the value iγ * is provided, and the filter means 19 is an output of the current phase control means 14 and the exciting current command value iγ.
* Removes high frequency components with a low-pass filter,
It is a means for performing filtering for suppressing variations in each calculation.

In the present embodiment, the exciting current command value iγ * output by the exciting current command value calculating means 17 is filtered by the filter means 19 such as a low pass filter to remove high frequency components. The stability and calculation accuracy of the output voltage command values Vγ * and Vδ * can be improved, and the synchronous motor 5 can be stably controlled. That is, the current phase control means 1
The excitation current command value iγ * obtained by 4 may vary greatly in each operation cycle depending on the operating state, and the output voltage calculation means 10 uses this iγ * to output voltage command value Vγ *, Since Vδ * is calculated, the fluctuation of the exciting current command value iγ * directly changes the output voltage command values Vγ *, Vγ
Reflected in δ *, the operation of the synchronous motor 5 becomes unstable.

As in the present embodiment, the filter means 19
Is provided to eliminate fluctuations in each calculation cycle due to high frequency components generated in the exciting current command value iγ *,
By suppressing the fluctuation of * for each operation cycle, it is possible to avoid the instability of the operation of the synchronous motor 5 due to the fluctuation of the exciting current command value iγ *, and it is possible to obtain a device with good controllability and high reliability. it can. Here, in a steady operation state in which the rotation frequency and the load torque of the synchronous motor do not substantially change, the exciting current command value iγ * hardly changes. When a low-pass filter is used as the filter means 19 in such a case, the time constant of the low-pass filter may be set to a value sufficiently larger than the exciting current control cycle (calculation cycle in the current phase control means 14). For example, the time constant of the low-pass filter is the output voltage means 10 excluding the exciting current control cycle (calculation cycle in the current phase control means 14) and the calculation cycle other than the exciting current calculation (for example, the calculation cycle in the current phase control means 14). It should be designed considering the calculation cycle of
When the exciting current control period is about 10 msec, the time constant of the low pass filter may be set to about 1 sec.

Here, unless the filter means 19 such as a low-pass filter is provided, the exciting current command value iγ * cannot be averaged, so that there is a large variation in each operation cycle and accurate and stable operation cannot be performed. The calculation time becomes long, the convergence is poor, and stable control cannot be performed.

The output voltage command value calculation means 10 is obtained from the γ-δ axis currents i γ and i δ obtained from the three-phase / two-phase conversion means 7, the primary frequency ω1 obtained from the primary frequency calculation means 9 and the filter means 19. The voltage command values Vγ * and Vδ * for driving the synchronous motor 5, which is the drive load, are calculated from the excitation current command value passed through the filter iγ *. The electrical angle phase calculation means 13 calculates the electrical angle phase θ based on the primary frequency ω1 obtained by the primary frequency calculation means 9. The two-phase / three-phase conversion means 11 uses the electrical angle phase θ obtained by the electrical angle phase computation means 13 to calculate the γ-δ axis voltage instruction values Vγ *, Vδ * obtained by the output voltage command value computation means 10
The phase voltage command values Vu *, Vv *, and Vw * are converted. PW
The M signal generating means 12 is a PWM for on / off controlling the switching elements in the inverter main circuit 3 based on the three-phase voltage command values Vu *, Vv *, Vw * obtained from the two-phase / three-phase converting means 11. It is a means for generating a signal.

Next, the operation will be described with reference to FIG.
In the figure, the synchronous motor drive device 1 detects currents of two phases (for example, U-phase current iu and V-phase current iv) out of phase currents flowing into the synchronous motor 5 which is a drive load of an inverter, as current detection means 6a, 6b. To detect more. The inverter control means 2 uses a current for two phases (for example, a U-phase current iu and a V-phase current iv) detected by the current detection means 6a and 6b or a current signal for two phases to perform a frequency compensator 8 from the outside. A compensation amount ωd for compensating the given frequency command value ωm * is calculated, and this frequency command value ωm * is compensated by the frequency calculation means 9 by the compensation amount ωd to obtain the primary frequency ω.
Output as 1. Then, the output voltage command value calculation means 1
At 0, the output voltage command value (Vγ *, Vδ *) for driving the synchronous motor 5 that is the drive load is calculated by using the primary frequency ω1 that is the compensated frequency command value. Thereafter, the 2-phase / 3-phase conversion means 11 converts the output voltage command values (Vγ *, Vδ *) of the 2-phase coordinate system into the output voltage command values (Vu *, Vv *, Vw *) of the 3-phase coordinate system. Then
Output as a voltage value or voltage signal. Based on the output voltage command values (Vu *, Vv *, Vw *) of this three-phase coordinate system, the PWM signal generating means 12 outputs a PWM signal for controlling on / off of the switching element in the inverter main circuit 3. .

That is, the inverter control means 2 outputs the PWM signal by the following operation. The γ-δ axis currents iγ, iδ are obtained by the three-phase / two-phase conversion means 7 using the phase currents (for example, iu, iv) for the two phases detected by the current detection means 6a, 6b and the electrical angle phase θ. The frequency compensator 8 obtains the frequency compensation amount ωd from the δ-axis current iδ obtained by the three-phase / two-phase conversion means 7. In the primary frequency calculating means 9, the frequency command value ωm * is compensated by the difference between the rotation frequency command value ωm * which is a target frequency given from the outside and the frequency compensation amount ωd obtained by the frequency compensator 8, and the primary frequency ω1. Output as.

In the output voltage command value calculation means 10, the primary frequency ω1 obtained by the primary frequency calculation means 9 and the excitation current command value iγ * obtained by the current phase control means 14 are filtered by the filter means 19 to generate an exciting current. Command value iγ *, output voltage command values Vγ *, Vδ * for driving the synchronous motor 5, which is the drive load of the inverter, from the γ-δ axis currents iγ, iδ obtained by the three-phase / two-phase conversion means 7.
Ask for.

The current phase control means 14 calculates the exciting current command value iγ * in the following manner. 2 phase 3
Based on the γ-axis current iγ and the δ-axis current iδ obtained by the phase coordinate conversion means 7, the current calculation means 21 calculates the magnitude Ia of the current vector. Based on Ia obtained by the current calculation means 21, the output torque is equal to or higher than a predetermined value (or maximum), that is, the current value is equal to or lower than a predetermined value (or minimum).
The current phase angle βm is calculated by the current phase calculation means 22. Then, the voltage command value calculating means 16 is based on the current phase angle βm obtained by the current phase calculating means 22 and the primary frequency ω1 which is the compensated frequency command value which is the previous output value of the frequency calculating means 9. Then, the voltage command value Va * is calculated.

In the voltage calculation means 15, the magnitude Va of the output voltage command value is based on the output voltage command values Vγ * and Vδ * which are the previous output values of the output voltage command value calculation means 10.
Is calculated. Further, the reference exciting current command value calculating means 18 obtains the reference exciting current command value iγ0 based on the compensated frequency command value (primary frequency) ω1 which is the previous output value of the frequency calculating means 9. Then, in the exciting current command value calculating means 17, the voltage command value Va * obtained by the voltage command value calculating means 16, the magnitude Va of the output voltage command value obtained by the voltage calculating means 15, and the reference exciting current command value. Based on the reference excitation current command value iγ0 obtained from the calculating means 18, the current of the synchronous motor 5 is below a predetermined value (or minimum).
The exciting current command value iγ * for driving at the current phase angle βm is calculated. Then, this exciting current command value iγ
The * is passed through a filter means 19 such as a low-pass filter to remove high-frequency components and averaged to suppress variations in the exciting current command value for each operation cycle, and the exciting current command filtered by the output voltage command value calculating means 10 is suppressed. Output as the value iγ *.

The electrical angle phase computing means 13 computes the electrical angle phase θ based on the compensated frequency command value (primary frequency) ω1 which is the output value of the frequency computing means 9. And
In the 2-phase / 3-phase conversion means 11, the output voltage command value calculation means 1
The γ-δ axis voltage command values Vγ * and Vδ * obtained by 0 are used as the voltage command values Vu *, Vv *, Vw * of the three-phase coordinate system by using the electrical angle phase θ obtained by the electrical angle phase calculation means 13.
Convert to. The PWM signal generating means 12 has the three-phase voltage command values Vu *, Vv obtained by the two-phase / three-phase converting means 11.
Based on *, Vw *, a PWM signal for controlling ON / OFF of the switching element in the inverter main circuit 3 is generated. Based on this PWM signal, the switching element in the inverter main circuit 3 is turned on / off, the DC voltage of the DC power supply 4 is converted into a three-phase AC, which is applied to the synchronous motor 5 which is a drive load to drive the drive load. .

Here, the relationship between the current phase angle β and the output torque τ in a predetermined operating state when a synchronous motor, particularly an embedded magnet type motor / motor (IPMSM) is used as a drive load will be described with reference to FIG. . Figure 2 shows IPMSM
FIG. 3 is a diagram showing an example of the relationship between the current phase angle β and the output torque τ in the constant current operation state of FIG. In the figure, the horizontal axis represents the current advance angle (current phase angle) β, and the vertical axis represents the torque. In the figure, the magnet torque by the rotor magnet of the IPMSM, the d-axis inductance Ld and the q-axis inductance Lq are shown. The relationship between the reluctance torque caused by the difference between and the output torque, which is the combined torque, is shown with respect to the current phase angle β.

As shown in the figure, in the IPMSM, the output torque changes depending on the current phase angle β even under the same current condition. Therefore, even under the same current condition, the output can be achieved by controlling the current phase angle β to a predetermined phase. It is possible to output the torque at a predetermined value or higher (or maximum). Therefore, under the condition that the load torque is constant, by appropriately controlling the current phase angle β, the current value required to obtain the output torque that is the same as the load torque can be made a predetermined value or less (or minimum). . Also, if the output torque is controlled to a value higher than or equal to a predetermined value (or maximum) under constant current conditions, the current value will be lower than or equal to the predetermined value (or minimum) under constant load torque. ) And a highly efficient device can be obtained.

Further, FIG. 3 shows the IP under the condition that the rotation frequency is constant.
FIG. 6 is a diagram showing a relationship between a current phase angle β and a current value when the load torque of the MSM is changed and driven. In the figure, the horizontal axis represents the current advance angle (current phase angle) β, and the vertical axis represents the current value. Τn (n = 0, 1, 2,
3, 4) show the output torque of the IPMSM, and the magnitude relation of the magnitudes is τ4>τ3>τ2>τ1> τ0.

As shown in FIG. 3, even when the IPMSM is driven under the same rotation frequency condition, when the load torque changes (τ0 → τ1 → ... → τ4), the current value becomes the minimum. The phase angle β changes. However, even when the rotation frequency and the load torque change, if the current phase angle β can be controlled to the current phase angle βm that minimizes the current value as shown in FIG. It is possible to drive the IPMSM in the minimum state, the copper loss of the IPMSM can be minimized, and the conduction loss of the switching element of the inverter main circuit can also be minimized, so that highly efficient operation can be realized.

Here, in the present embodiment, the synchronous motor 5, which is the drive load, is driven so that the current becomes equal to or less than the predetermined value (or the current becomes the minimum). Therefore, the method will be described below. To do.

The voltage equation in the dq axis coordinates of the synchronous motor, particularly the embedded magnet type synchronous motor (IPMSM) is generally given by the mathematical expression 1. Here, the d-axis and the q-axis are defined as the q-axis, which is the magnetic pole direction of the rotor of the electric motor as the d-axis and the phase which leads the rotational direction by a predetermined angle (90 degrees in the case of two poles) from the d-axis. is doing.

[0056]

[Equation 1]

In Expression 1, Vd is the d-axis voltage of the embedded magnet type synchronous motor (IPMSM), Vq is the q axis voltage of the embedded magnet type synchronous motor (IPMSM), and R is the embedded magnet type synchronous motor (IPMSM). Per phase, Ld is the d-axis inductance of the embedded magnet synchronous motor (IPMSM), Lq is the q-axis inductance of the embedded magnet synchronous motor (IPMSM), and Φf is the induction of the embedded magnet synchronous motor. Voltage constant, id is the d-axis current of the embedded magnet type synchronous motor, iq
Is the q-axis current of the embedded magnet type synchronous motor, ω is the rotation frequency of the embedded magnet type synchronous motor, p is the differential operator with respect to time t,
Is represented. Further, when the number of pole pairs of the embedded magnet type synchronous motor is Pm, the output torque τm of the embedded magnet type synchronous motor (IPMSM) is generally given by Equation 2.

[0058]

[Equation 2]

The d-axis current id, the q-axis current iq, and the magnitude Ia of the current are defined by the equation (3).

[0060]

[Equation 3]

Substituting Equation 3 into Equation 2, the torque τm is expressed as Equation 4.

[0062]

[Equation 4]

The procedure for obtaining βm, where βm is the current phase angle at which the output torque is maximum (or the current is minimum), will be described below. By partially differentiating the equation 4 by the current phase angle β and setting it as = 0, the maximum torque is obtained (or the current is the minimum).
Is obtained as in Expression 5.

[0064]

[Equation 5]

However, in the calculation of βm, even if the calculation is not performed so that the output torque becomes maximum as in the formula 5,
Formula 6 shown below so that the output torque becomes a predetermined value or more
You may use an approximate expression like this.

[0066]

[Equation 6]

Here, the expression (6) is obtained by expanding the expression (5) into the Maclaurin series to obtain a quadratic term. When this approximate expression 6 is used, the output torque does not become maximum (current does not become minimum), but the output torque can be made higher than a predetermined value (current value is lower than a predetermined value), so that high output and low current can be obtained. The device can be obtained. Further, since the calculation of βm is an approximate expression and is simplified, the calculation load on the microcomputer can be reduced, so that the calculation time can be reduced, and further, the low-cost microcomputer can be used. , The degree of freedom in selecting a microcomputer is expanded. Further, since the calculation of βm is an approximate expression and is simplified, the convergence is also improved, so that a device and a control method having good followability in control can be obtained. However, if it is necessary to improve the control accuracy, the accuracy of the approximation formula may be improved by using up to the third term of the Maclaurin series, or the formula 5 may be used as it is.

In this embodiment, the γ-δ axes are used as the coordinate system instead of the dq axes.
The relationship between the dq axis coordinate system and the γ-δ coordinate system will be described using. In the present embodiment, when the synchronous motor is driven without using the position sensor for detecting the rotor position, the embedded magnet (IPM: Interior Perm) is actually used.
Since it is difficult to detect the d-q axes of the rotor, the γ-δ axes shown in FIG. 4 are used as the axes for control.

FIG. 4 is a vector diagram of a synchronous motor (in particular, an embedded magnet type synchronous motor (IPMSM)) under certain operating conditions. In FIG. 4, the horizontal axis represents the γ axis and the vertical axis represents the δ axis. Further, the d axis and the q axis are located at angular positions rotated by Δθ with respect to the γ axis and the δ axis, respectively. Where iγ is the IPMSM γ-axis current and iδ is IP
The δ-axis current of the MSM, Ia, is the magnitude of the current flowing through the IPMSM, and is the combined vector of iγ and iδ. Also,
Ia is deviated from the q axis by β (current phase angle). As described above, the current phase angle β is an angle formed by the q axis and the current vector Ia.

ΦF is a magnetic flux vector generated by the rotor magnet, and its magnitude is the induced voltage constant Φf, which is located on the d-axis. Further, Lq · iδ is a δ-axis component of the stator magnetic flux, Ld · iγ is a γ-axis component of the stator magnetic flux, Φ0 is a combined magnetic flux of the stator magnetic flux and the rotor magnetic flux, and the rotor magnetic flux (Φ
F) and the stator magnetic flux (Lq · iδ, Ld · iγ). Further, ω · Φ0 is a voltage vector generated by the combined magnetic flux, R · Ia is a voltage vector that compensates for the voltage effect generated by the resistance of the motor, and Va is IPMS.
It is a voltage applied to M and is a composite vector of ω · Φ0 and R · Ia. Here, as the control axis, the dq axes may be used as they are without using the γ-δ axes, and either of them can be used for control, but in the present embodiment, the current component is easily detected. The case where the γ-δ axes are used will be described.

Here, in the present embodiment, the γ-δ axes are used as the control axes, and the phase difference Δθ between the dq axes and the γ-δ axes is approximated to Δθ≈0 for simplification of control. Then, the inductance of each axis is treated as Lγ≈Ld and Lδ≈Lq.

In the present embodiment, the exciting current is controlled as a control amount for driving at the current phase angle where the current is the minimum, so this exciting current will be described below. As shown in FIG. 4, in the γ-δ axis defined as the control axis (or dq axis based on the rotor position of IPMSM), the γ axis (or d axis)
The current iγ (or id) corresponding to is generally called an exciting current, and the magnetic flux (Ld · iγ + Lq · iδ) on the stator side and the magnetic flux (Φ) of the rotor magnet generated by this exciting current.
F) produces a combined magnetic flux vector (Φ0).

On the other hand, the current corresponding to the δ-axis (or q-axis) is generally called a torque component of the current component (torque current) and has a magnitude corresponding to the output torque of the motor.

Here, the current vector Ia is the exciting current iγ
(Or id) and a torque current iδ (or iq) are combined vectors. When the output voltage vector Va is changed according to the operating state (predetermined rotation speed, predetermined load torque) of the IPMSM, the exciting current iγ changes (the δ-axis current iδ hardly changes because of the torque component), and this exciting current iγ Changes in the current phase angle β. As shown in FIG. 4, the exciting current iγ (or id) and the torque current iδ
Since the combined vector of (or iq) is the current vector Ia, the exciting current iγ (or id) and the torque current iδ
By controlling (or iq) and controlling the output voltage vector Va, the current phase angle β of the current vector Ia can be changed.

In this embodiment, no speed controller is provided,
In other words, even in the control without the torque current command value (iδ *), the output voltage vector is controlled by controlling only the exciting current iγ so that the current phase angle at which the current becomes the predetermined value or less (or the minimum) is obtained. The Va is controlled so that the current becomes equal to or lower than a predetermined value (current minimum). The current phase angle βm at which the current is minimum (or the current is equal to or less than a predetermined value) is an angle formed by the q-axis and the current vector Ia in the dq axis coordinate system. However, in a control system without a speed controller or a control system without a position sensor, the actual position of the dq axes cannot be known. Therefore, in this embodiment, the control axis (γ-δ axes) and the motor Axis (dq
The magnitude Va of the voltage vector, the voltage command value Va *, and the magnitude Ia of the current vector are used as variables used for control so that the influence of the phase difference Δθ with the axis) can be ignored.

In the present embodiment, the current minimum control can be realized even in the control system in which the drive means of the synchronous motor does not have the speed control loop or the current control loop. Therefore, the current phase control means 14 outputs the current. Excitation current command value iγ *
Is calculated as follows.

The output voltage calculation means 15 uses the output voltage signal (for example, Vγ * and Vδ * which are the previous output values of the output voltage command value calculation means 10) to output the voltage of the inverter according to the following equation (equation 7). The magnitude Va of is calculated.

[0078]

[Equation 7]

Here, the output voltage (Vu, Vv, Vw) output by the inverter main circuit 3 may be used as the output voltage signal. In that case, a separate voltage detecting means (not shown) is used for the inverter main circuit. 3 may be provided on the output side. In this embodiment, since the previous output values Vγ * and Vδ * of the output voltage command value calculation means 10 are used in order to eliminate the need for the voltage detection means and the like, the voltage detection means and the like are unnecessary. It is possible to obtain a highly efficient device with low current while having a simple structure and low cost.

Further, the current calculation means 21 uses the γ-axis current iγ and the δ-axis current iδ obtained by performing the three-phase / two-phase coordinate conversion of the current flowing through the synchronous motor 5 which is the driving load, and determines the magnitude of the current. Ia is calculated by Expression 8.

[0081]

[Equation 8]

Here, in Equation 8, the control axis (γ-δ axis)
Although the magnitude Ia of the current vector is calculated above, the magnitude of the current vector is different only by the phase difference Δθ between the control axis (γ-δ axis) and the motor axis (dq axis). Ia
Are the same. Therefore, when the magnitude Ia of the current vector is used for control as in this embodiment, the control axis (γ−
Since it is not necessary to consider the phase difference Δθ between the δ axis) and the motor axes (dq axes), control is possible even in a control method that does not have a speed control loop or a current control loop. The current phase calculating means 22 calculates the current phase angle βm at which the current becomes equal to or less than a predetermined value (or minimum) according to Equation 6 (or Equation 5) using Equation 8. Voltage command value calculation means 1
In 6, the voltage command value Va * for driving the drive load 5 at the current phase angle βm at which the current is below a predetermined value (or minimum) is set by using the arithmetic expression of the dq axis coordinate system. Ask for. Since Vd and Vq are obtained by substituting Equation 3 into Equation 1, the magnitude Va of the resultant force is represented by the square root of the square sum of Vd and Vq (Equation 9).

[0083]

[Equation 9]

However, the control axis (γ-δ axis) and the motor axis (d-q axis) are deviated by the phase difference of Δθ, but the magnitude of the voltage vector on the control axis (γ-δ axis). Since Va has the same magnitude Va as the voltage vector on the motor axes (dq axes), the coordinate axes can be replaced. That is, in terms of control, the coordinate system actually obtained is the γ-δ axis coordinate system, so
It can be considered by replacing Equation 9 representing the magnitude of the voltage vector on the motor shaft (d-q axes) with Equation 7 representing the magnitude Va of the voltage vector on the control axis (γ-δ axes).

Then, Ia obtained by the equation 8 in the equation 7
And the current phase angle β obtained by Equation 6 (or Equation 5)
Substituting m, the voltage command value V when the current phase angle β is βm
a * is obtained as in Expression 10. Again, the control axis (γ
The command value is set to the magnitude V of the voltage vector so that the phase difference Δθ between the −δ axis) and the motor axes (dq axes) need not be considered.
I am instructed by a *.

[0086]

[Equation 10]

Here, as described above, Ia is the magnitude of the current flowing into the synchronous motor 5 on the γ-δ axes, βm is the current phase angle at which the output torque becomes a predetermined value or more (or maximum),
That is, it is a current phase angle at which the current is below a predetermined value (minimum).

Here, when calculating the voltage command value Va * in the equation 10, if the equation (5) or (6) is used to obtain the current phase angle βm at which the current becomes the minimum (the current is less than a predetermined value), the computation load is obtained. In some cases, the calculation load increases, the calculation load increases, and the calculation time increases. In this case, it is not necessary to calculate up to βm, and it is sufficient if sin βm can be calculated. To do this, transform Equation 5 (or Equation 6) into sin
βm is calculated, and Ia obtained by Equation 8 and the primary frequency ω1 obtained by the primary frequency calculating means 9 are set as ω, Equation 10
, The voltage command value Va * can be calculated, and since only sinβm is obtained compared to the case where βm is obtained, the number of calculations can be reduced and the calculation time can be shortened. Where sinβm and cosβm
Can be obtained from, for example, Equation 6 as Equation 11.

[0089]

[Equation 11]

Here, by utilizing the fact that Va * is a function of ω and Ia, a numerical value table having a relationship between ω and Ia or ω and Ia ² is previously stored in the voltage command value calculation means 16 or separately. You may make it memorize | store in a means etc.

With such a table, it is possible to reduce the number of operations such as the square root, which takes a long time, and the number of operations, which increases the degree of freedom in selecting a microcomputer and reduces the cost. Can be achieved. Further, the determination as to whether to use the arithmetic expression or the table may be made according to the performance of the microcomputer to be used.

In the reference exciting current calculating means 18, the exciting current command value iγ is calculated based on a numerical table, a function with the frequency ω, a constant of a predetermined value, or the like.
Iγ0 which is the reference value of * is obtained. Exciting current calculation means 1
7, the reference excitation current command value iγ0 obtained by the reference excitation current command value calculation unit 18, the output voltage magnitude Va obtained by the output voltage calculation unit 15, and the voltage command obtained by the voltage command value calculation unit 16 Using the value Va *, the exciting current command value iγ * is calculated by the following formula 12. Here, Khe in Expression 12 is a control gain of the exciting current given in advance by experiments or the like.

[0093]

[Equation 12]

The calculation of the exciting current command value iγ * in the mathematical expression 12 is an example of a proportional control method in which the difference between the voltage command value Va * and the output voltage magnitude Va is multiplied by the proportional gain Khe. In the proportional control method, since the difference between the voltage command value Va * and the magnitude Va of the output voltage is simply multiplied by the control gain given in advance, a steady error occurs with respect to the voltage command value Va *, and If it is desired to accurately control the current phase angle β, integral control may be used. Formula 13 shows an example of a control method using integral control.

[0095]

[Equation 13]

Here, iγ * _prv is the output value of the exciting current command value in the previous control cycle in the current phase control means 14, and the output value of the exciting current command value in this previous control cycle is the reference excitation. By using the current command value instead of iγ0, the difference between the voltage command value Va * and the output voltage magnitude Va can be made as close to 0 as possible, so that the error with respect to the voltage command value Va * can be reduced. , The calculation accuracy is improved. Here, although it is difficult to make the current phase angle β accurately follow the current minimum phase angle βm with the proportional control method as shown in Formula 12, a control method or a control method that does not have a speed control loop or a current control loop is used. When performing control in the modulation region, more stable controllability can be obtained by the proportional control shown in Formula 12 than in the integral control shown in Formula 13.

In the current phase control means 14, the above equations 1-
By using Expression 13, the exciting current command value iγ
Calculate *. Then, the exciting current command value iγ * calculated by the current phase control means 14 is filtered by the filter means 19 such as a low-pass filter in order to obtain stability of control of the synchronous motor, and the filtered exciting current command value iγ *. * Means output voltage command value calculation means 10
Is used for calculation in. Here, the filter means 19
Need not be provided, but it is better to provide the filter means 19 in order to obtain control stability.

As described above, as a method of controlling the current phase angle in order to drive the synchronous motor 5 at the current phase angle at which the current becomes equal to or less than the predetermined value (or the minimum value), the synchronous motor in the present embodiment is used. The output torque of the synchronous motor 5 is a predetermined value or more based on the current detection means 6a, 6b for detecting the flowing current, and the current signal obtained by the current detection means 6a, 6b and the voltage signal of the voltage applied to the synchronous motor 5. The current phase control means 14 for outputting the exciting current command value iγ * for controlling the current phase angle β of the current flowing through the synchronous motor 5 so as to obtain the exciting current command value iγ * output by the current phase means 14 And a compensation amount ωd for the frequency command value ωm * from the variation amount of the current detected by the current detection units 6a and 6b.
And the frequency command value ω1 compensated by the compensation amount ωd
, Frequency compensation value ω1 compensated, and filtered excitation current instruction value i
Since the output voltage command value calculating means 10 for obtaining the output voltage command values Vγ * and Vδ * for driving the synchronous motor 5 based on γ * is provided, the excitation of the current phase angle at which the current becomes equal to or less than the predetermined value. Since the output voltage command value is calculated based on the current command value and the compensated frequency command value, even if the operating conditions of the synchronous motor change due to frequency fluctuations or load fluctuations, the current will be below the specified value depending on the conditions. It is possible to control the current to the current phase angle that is (or the minimum), and it is possible to accurately keep the current value below the predetermined value (or the minimum) by following the changes in the operating conditions. A drive can be obtained. Further, since the filter means 19 is provided, the exciting current command value iγ for each calculation
The variation of * becomes small and stable control can be performed.

Further, in this embodiment, the current detecting means 6
The current phase angle β such that the output torque of the synchronous motor 5 becomes a predetermined value or more based on the current signals obtained by a and 6b.
A voltage command based on the current phase calculation means 22 for obtaining m and the current phase angle βm obtained by the current phase calculation means 22 and the primary frequency ω1 which is the previous frequency command value obtained by the frequency command value calculation means 9. The voltage command value calculation means 16 for calculating the value Va *, the voltage calculation means 15 for calculating the magnitude of the voltage based on the voltage signal of the voltage applied to the synchronous motor 5, and the voltage command value calculation means 16 are obtained. An exciting current command value calculating means 17 for obtaining an exciting current command value iγ * for driving the synchronous motor 5 based on the obtained voltage command value Va * and the voltage magnitude Va obtained from the voltage calculating means 15. , The current phase control means 14 is configured to control the current phase angle β to the current phase angle βm such that the current becomes a predetermined value or less (or a minimum). Although it is a simple configuration to obtain the output voltage at the (or minimum) current phase angle βm and the actual output voltage, it is not necessary to have a numerical table or function of the excitation current command value for each operating condition, so inverter control An inexpensive microcomputer with a small memory capacity can be used without increasing the memory usage of the microcomputer used for the means, and a simple control circuit can be used to obtain a reliable and inexpensive synchronous motor drive device. be able to.

Further, in the present embodiment, the DC power supply 4 for supplying the DC, the inverter main circuit 3 for converting the DC from the DC power supply 4 into the AC by turning the switching element on and off, and the inverter main circuit By applying the voltage output from the inverter control means 2 that outputs a voltage command value for controlling the on / off operation of the switching element in 3 and the inverter main circuit 3 controlled by the inverter control means 2. Drive load controlled 5
And an exciting current command value for driving the drive load 5 at the current phase angle βm such that the output torque of the drive load becomes a predetermined value or more based on the current flowing through the drive load. current phase control means 22 for calculating iγ *, filter means 19 for filtering the excitation current command value iγ * obtained by the current phase control means 22,
The frequency command value ωm based on the fluctuation amount of the current flowing through the driving load 5.
Based on the frequency calculation means 9 for compensating * and outputting as the primary frequency ω1, the excitation current command value iγ * filtered by the filter means 19 and the primary frequency ω1 obtained by the frequency calculation means 9, the drive load 5 is determined. Since the inverter device is constituted by the output voltage command value calculation means 10 for calculating the voltage command values Vγ * and Vδ * for driving, the current phase control means and the frequency calculation means are simply provided. However, even if the driving condition of the driving load changes, the output torque under that condition can be controlled to be a predetermined value or more. Further, the current can be controlled at the current phase angle at which the current value becomes equal to or less than the predetermined value according to the output torque, and the synchronous motor drive device with low current and high efficiency can be realized. Moreover, since the current value can be reduced, the rated capacity of the switching element can be reduced, and a low-cost inverter device can be provided. Further, since the filter means 19 is provided, variations in the exciting current command value iγ * are reduced, and a highly reliable inverter device capable of stably controlling the drive load can be obtained.

Further, the current phase calculating means 22 for obtaining the current phase angle βm at which the output torque becomes a predetermined value or more by the current flowing through the driving load 5, and the current phase angle βm obtained by the current phase calculating means 22 are obtained. Voltage command value V
The voltage command value calculation means 16 for calculating a * and the voltage calculation means for calculating the voltage magnitude Va based on the previous output voltage command values Vγ *, Vδ * obtained by the output voltage command value calculation means 10. 15 and the voltage magnitude Va obtained by the voltage calculation means 15 and the voltage command value Va * obtained by the voltage command value calculation means 16 based on the driving load 5
The current phase control means 1 by the excitation current command value calculation means 17 for obtaining the excitation current command value iγ * for driving
Since No. 4 is configured, it is possible to control the current phase angle such that the value of the current flowing in the drive load is equal to or less than a predetermined value, and to control in an efficient state in a short time. Further, since the convergence and stability are good in the vicinity of the current phase angle where the current value is equal to or less than the predetermined value, it is possible to realize an inverter device with good controllability and low current. Further, since only the exciting current command value is controlled, the magnitude Va of the voltage vector is used as a value used for control.
By using Va * and the magnitude Ia of the current vector, it is not necessary to consider the phase difference between the control axis (γ-δ axis) and the motor axis (dq axis), and further, the speed controller and the current controller. Even if the inverter is not provided, a low current and highly efficient inverter device can be realized.

Further, in the present embodiment, even when the operating condition of the synchronous motor changes due to frequency fluctuations or load fluctuations, the current phase angle becomes such that the output torque becomes a predetermined value or more (or maximum) according to the conditions. Since the current can be controlled, a high output inverter device and a synchronous motor drive device can be obtained.

Further, since the current phase angle is controlled so that the current is always a predetermined value or less (or minimum), the followability to the current minimum phase angle is good, and the convergence and near the current minimum phase angle are good. Since the stability is good, an inverter device and a synchronous motor drive device with good controllability can be obtained. Further, since only the excitation current command value is controlled, the magnitudes Va and Va * of the voltage vector and the magnitude Ia of the current vector are used as the values used for the control so that the control axis (γ-δ axis) and the motor axis (d -Q-axis), there is no need to consider the phase difference, and even if the speed controller and the current controller are not provided, the inverter device that can realize the current phase control in which the current becomes the predetermined value or less (or the minimum), A synchronous motor drive device is obtained.

Further, in the present embodiment, since the electric motor is driven without the position sensor for detecting the rotor position, the position sensor is unnecessary, the cost is low, the reliability is high, and Since current phase control in which the current is below a predetermined value (or minimum) can be realized, a highly efficient inverter device and synchronous motor drive device can be obtained.
Further, since the position sensor is unnecessary, it is possible to realize an inverter device and a synchronous motor driving device that can perform high-efficiency, high-efficiency and high-efficiency control even in a synchronous motor used in a compressor or the like to which the position sensor cannot be attached.

Further, in this embodiment, the voltage signal used in the output voltage calculation means 15 is the output voltage command value calculation means 1
Previous output voltage command values Vγ * and Vδ obtained from 0
Although it is not limited to *, it has been explained that the voltage value actually output from the inverter main circuit 3 may be detected by providing voltage detection means (not shown). (Not shown) is provided to detect the voltage value output from the inverter main circuit 3, and then the three-phase / two-phase conversion means is used to
The voltage values Vγ and Vδ on the −δ axis may be converted, and the converted voltage values Vγ and Vδ may be used. By detecting and using the actual voltage value, the synchronous motor 5, which is the drive load, can be controlled more accurately, and a highly efficient and highly reliable device can be obtained.

Further, in FIG. 1, the calculation cycle of the current phase control means 14 in the inverter control means 2 and the calculation processing cycle other than the current phase control means 14 may be different cycles. Current phase control means 1 in the inverter control means 2
The calculation within 4 does not require a quick response compared with the calculation processing other than the current phase control means 14, so the calculation cycle (excitation current control cycle) by the current phase control means 14 is different from that of the current phase control means 14. It may be longer than the control cycle (output voltage control cycle). That is, the current phase control means 1
The calculation processing cycle (excitation current control cycle) in 4 may be set to about once every several to several hundreds of calculation processing cycles (output voltage control cycle) other than the current phase control means 14. By doing so, when the arithmetic processing of the inverter control means 2 is realized by a microcomputer or the like, the processing load on the microcomputer can be reduced, a low-cost inverter control means can be obtained, and a low-cost motor drive device can be obtained. Can be obtained.

Next, referring to FIG. 5, the first embodiment of the present invention will be described.
An example of the method of controlling the synchronous motor in FIG. Figure 5
FIG. 6 is a flowchart showing a control method for highly efficiently driving the synchronous motor according to the present embodiment with a current of a predetermined value or less. In the figure, ST1 is a current detection / calculation step for detecting the current flowing in the synchronous motor and calculating current values iγ, iδ of the two-phase coordinate system, and ST2 is a current detection / calculation step based on the current signal obtained in the current detection / calculation step ST1. A current phase calculation step for obtaining the current phase angle βm at which the current is equal to or smaller than a predetermined value (or minimum) by the current phase calculation means 22, ST3
Is the voltage command value Va * based on the primary frequency ω1 which is the previously compensated frequency command value obtained by the frequency calculation means 9 and the current phase angle βm obtained in the current phase calculation step ST2. Voltage command value calculation step, ST4 is the magnitude Va of voltage applied to the synchronous motor.
Is a voltage calculation step, and ST5 is a reference excitation current calculation step for obtaining the reference excitation current command value iγ0.

In ST6, the actual current phase angle is calculated based on the voltage command value Va * obtained in the voltage command value calculation step ST3 and the voltage magnitude Va obtained in the voltage calculation step ST2. Is a predetermined value or less (or is the minimum), the exciting current command value calculation step for obtaining the exciting current command value iγ * for driving the motor motor 5 at the current phase angle βm, ST7 is the exciting current command value obtained in ST6. iγ *
Is filtered by passing it through the filter means 19, ST8 is output voltage command values Vγ *, Vδ * for driving the synchronous motor based on the filtered excitation current command value iγ * obtained by the filter step ST7. Is an output voltage command value calculation step for obtaining.

First, at ST1, at least two-phase currents of the synchronous motor 5 which is a drive load are detected and converted into current values of a two-phase coordinate system. That is, currents for two phases (for example, U-phase current iu, V-phase current iv) flowing in the synchronous motor 5 are detected by the current detecting means 6a, 6b, and currents of the two-phase coordinate system (for the three-phase / two-phase converting means 7). For example, γ-axis current iγ, δ
It is converted into the axial current iδ), and Ia is calculated by Expression 8.

At ST2, the γ-axis current i obtained at ST1
Using the γ and δ axis currents iδ, the current phase angle βm at which the current value is equal to or smaller than the predetermined value (or the minimum value) is calculated as the current phase calculation means 22.
Is calculated using Equation 6 (or Equation 5). (Here, sin βm and cos βm can be calculated as in Expression 11 even if the current phase angle βm is not calculated as in Expressions 6 and 5, so sin βm and cos β in Expression 11 can be obtained.
The calculation load may be reduced by obtaining up to m. )

At ST3, the primary frequency ω which is the previously compensated frequency command value obtained by the frequency calculation means 9
1 and the current phase angle βm obtained in ST2, the voltage command value calculation means 16 calculates the voltage command value Va * using the mathematical formula 10.

In ST4, the voltage calculation means 15 calculates the voltage value applied to the drive load from the voltage command value obtained by the output voltage command value calculation means 10 shown in FIG.
That is, in ST4, the voltage calculating means 15 determines the magnitude Va of the voltage for driving the synchronous motor 5 using the previous voltage command values Vγ * and Vδ * calculated by the output voltage command value calculating means 10. Calculation is performed using Equation 7. here,
As the voltage command values Vγ * and Vδ *, a voltage detection means (not shown) or the like may be separately provided to detect the voltage applied to the drive load 5, and the voltage signal may be used. Then,
Since the latest voltage information can be used at all times, it is possible to obtain a control method for a synchronous motor that enables fine and reliable control.

In ST5, based on the frequency command value (for example, the primary frequency ω1 which is the previously compensated frequency command value obtained by the frequency calculation means 9), the reference excitation current calculation means 18 outputs the reference excitation current command. Determine the value iγ0.

At ST6, the voltage command value Va * obtained at ST3, the voltage magnitude Va obtained at ST4,
And the reference excitation current command value iγ * obtained in ST5
The excitation current command value iγ * for driving the drive load 5 at the current phase angle βm based on
7 is calculated using Equation 12 (or Equation 13). Then, in ST7, the exciting current command value iγ * obtained in ST6 is used as a filter means 1 such as a low-pass filter.
9 to remove the high-frequency component of the excitation current command value iγ * to reduce the variation in each operation cycle and average them.
The calculation stability and control stability are improved. Here, unless the filter means 19 such as a low-pass filter is provided, the exciting current command value iγ * cannot be averaged, so that the variation becomes large and stable control cannot be performed accurately.

At ST8, output voltage command values Vγ * and Vδ * for driving the drive load 5 are calculated based on the excitation current command value iγ * filtered by the filter means 19 obtained at ST7. Obtained by means 10. Here, the output voltage command value calculation step ST8
Since the arithmetic expression used in 1 differs depending on the control method of the synchronous motor, any arithmetic expression may be used as long as Vγ * and Vδ * can be calculated.

For example, Equation 14 shows an example of a mathematical expression for calculating the γ-axis voltage command value Vγ * and the δ-axis voltage command value Vδ * which are the outputs of the voltage command value calculating means 10.

[0117]

[Equation 14]

Expression 14 represents an arithmetic expression in the case of using the position sensorless control system of the synchronous motor by the primary magnetic flux control, for example. Here, in Expression 14, Φγ *
Is a primary magnetic flux command value in primary magnetic flux control, Φγerr is a primary magnetic flux error, Kγ is a γ-axis control gain, and Kδ is a δ-axis control gain.

As described above, in the present embodiment, the current phase calculation step ST3 for obtaining the current phase angle such that the output torque of the synchronous motor becomes a predetermined value or more based on the value of the current flowing through the synchronous motor, and the synchronization step ST3. Voltage calculation step ST4 for calculating the magnitude of the voltage applied to the electric motor 5, and reference excitation current calculation step ST5 for obtaining the reference excitation current command value.
And the voltage command value Va * obtained in the voltage command value calculation calculation step ST3, the voltage magnitude Va obtained in the voltage calculation step ST4, and the reference excitation current command obtained in the reference excitation current command value calculation step. Excitation current command value i for driving the synchronous motor 5 based on the value iγ0
Excitation current command value calculation step ST6 for obtaining γ * and high frequency components of the excitation current command value iγ * obtained in the excitation current command value calculation step ST6 are removed by the filter means 19 to suppress variation in each calculation cycle. And a filter step for performing the filtering for controlling the synchronous motor. Therefore, it is possible to obtain a control method for a synchronous motor, which enables simple and efficient control with high output and low current, and efficiently and stably.

Further, even when the operating condition of the synchronous motor changes due to the rotation frequency or the load change, the current can be controlled at the low current phase angle so that the output torque according to the condition becomes a predetermined value or more, and the low current can be controlled. In addition, the synchronous motor can be driven with high efficiency. Also, good followability to low current phase,
Good convergence and stability near low current phase,
Controllability can be improved. In addition, since it is a simple control that controls only the excitation current command value without considering the coordinate axes, it is possible to accurately drive the synchronous motor with low current and high efficiency even in control means that does not have a speed controller or current controller. A possible control method is obtained.

Here, in FIG. 5, the voltage calculation step ST4 may be performed anywhere as long as it is performed in a step prior to the exciting current command value calculation step ST6.
It may be performed anywhere from T1 to ST5. The output voltage command value calculation step ST8 is a step of obtaining not only the control of the exciting current but also the output voltage command value for driving the synchronous motor according to the rotation frequency command value and the load torque. In FIG. 5, current detection / calculation step ST1
To the exciting current command value calculation step ST6 is defined as a processing cycle A, and the processing cycle after the output voltage command value calculation step ST7 in the inverter control means 2 is a processing cycle B.
Then, the processing cycle A and the processing cycle B may be set to different cycles in order to reduce the processing load of the microcomputer or the like that performs the processing. In this case, since the processing cycle A does not require responsiveness as compared with the processing cycle B, it is possible to reduce the load on the microcomputer by making the processing cycle A longer than the processing cycle B, so that a low-cost microcomputer can be used. Thus, the degree of freedom in selecting a microcomputer is expanded, and a low-cost and highly-reliable motor control method can be obtained with simple control.

Further, the method of performing the high efficiency control of the minimum current without obtaining the current phase angle βm at which the current becomes the predetermined value or less (or the minimum) described in the first embodiment will be described with reference to FIG. . FIG. 6 is a diagram showing the configuration of another inverter device and a synchronous motor drive device representing the first embodiment of the present invention. In the figure, the same parts as those in FIG. 1 are designated by the same reference numerals and the description thereof will be omitted.

In FIG. 6, 140 is a current phase control means, which is the output voltage calculation means 15 for obtaining the magnitude Va of the output voltage, and the γ-axis current iγ, δ-axis current obtained by the two-phase / three-phase coordinate conversion means 7. The current calculation means 21 for calculating the magnitude Ia of the current based on iδ, the current phase angle βm at which the current value is less than or equal to a predetermined value (or the minimum) is calculated as sin βm, and the driving load is calculated at this current phase angle βm. Voltage command value calculation means 16 for obtaining a voltage command value Va * for driving
0, voltage error calculation means 225 for calculating an error between voltage command value Va * and output voltage magnitude Va, reference excitation current command value calculation means 1 for obtaining a reference excitation current command value iγ0
8. Exciting current command value calculating means 170 for obtaining an exciting current command value iγ * for driving a driving load.

Therefore, in the apparatus having the configuration shown in FIG.
The current value or current signal detected by the current detecting means 6a, 6b, and the voltage signal applied to the synchronous motor 5 (for example, the output voltage signals Vγ *, Vδ * and its magnitude V).
Based on a), the voltage command value calculating means 160 for obtaining the voltage command value Va * for driving at the current phase angle βm such that the output torque of the synchronous motor 5 becomes a predetermined value or more, and the voltage command value Va * and the voltage. A voltage error calculating means 225 for calculating a voltage error by comparing with the signal Va, and a voltage error calculating means 225.
The excitation current command value calculation means 1 for calculating the excitation current command value iγ * for driving at the current phase angle βm at which the output torque of the synchronous motor 5 becomes a predetermined value or more from the voltage error obtained from
The current phase control means 140 is constituted by 70 and, and the voltage error calculation means 225 for calculating the voltage error is provided. Therefore, the calculation formula can be easily calculated by a simplified calculation formula without complication. The time can be shortened. Further, even when the operating condition of the synchronous motor changes due to the rotation frequency or the load change, the current can be controlled with a low current phase angle at which the output torque according to the condition becomes equal to or higher than a predetermined value, resulting in high efficiency at low current. It is possible to realize a simple synchronous motor drive device.

Further, a follower to the current phase angle βm at which the current value is equal to or less than a predetermined value is excellent, and a synchronous motor drive device with a low current and a high efficiency, which has good convergence and stability near this current phase angle, is realized. it can. Further, since it is not necessary to consider the coordinate axes and only the exciting current command value is controlled, it is possible to realize a highly efficient synchronous motor drive device with a good low current even in control means that does not have a speed controller or a current controller.

Next, the operation will be described with reference to the flowchart. FIG. 7 is a flowchart showing another control method of the synchronous motor of this embodiment. In the figure, ST11
Is a current detection / calculation step for detecting the current flowing through the synchronous motor to determine the magnitude of the current, and ST12 is a current signal obtained in the current detection / calculation step ST11 and the current is equal to or less than a predetermined value from the rotation frequency of the synchronous motor (or minimum)
Is a voltage command value calculation step for obtaining an output voltage command value Va * for driving the drive load at a current phase angle of

ST13 is a voltage calculation step for calculating the magnitude Va of the voltage applied to the synchronous motor, and ST14 is the voltage command value Va * obtained in the voltage command value calculation step ST12 and the voltage obtained in the voltage calculation step ST13. Voltage error calculation step for obtaining the difference (Va * -Va) from the magnitude Va, ST15 is a reference excitation current command value calculation step for calculating the reference excitation current command value iγ0, and ST16 is a reference excitation current command value calculation step ST15. Excitation current command value calculation step for obtaining the excitation current command value iγ * based on the obtained reference excitation current command value iγ0 and the voltage error (Va * -Va) obtained in the voltage error calculation step ST14 The exciting current command value iγ * obtained in the exciting current command value calculation step ST16 is increased by the filter means 19. A filter step for performing filtering for removing wave components, ST18 is an output voltage for obtaining output voltage command values Vγ *, Vδ * for driving the synchronous motor 5 based on the excitation current command value iγ * filtered by the filter step ST17. This is a command value calculation step.

First, at ST11, at least two-phase currents of the synchronous motor 5 as a drive load are detected and converted into a current value of a two-phase coordinate system, and the magnitude thereof is obtained. That is, currents for two phases (for example, U-phase current iu, V-phase current iv) flowing in the synchronous motor 5 are detected by the current detecting means 6a, 6b, and currents of the two-phase coordinate system (for the three-phase / two-phase converting means 7). For example, γ-axis current iγ, δ-axis current iδ) is converted to the magnitude I
a is calculated by Expression 8.

In ST12, γ-δ obtained in ST11
A current phase angle β at which the current value is equal to or smaller than a predetermined value (or minimum) using the axis currents iγ, iδ and the magnitude Ia of the current.
m is calculated by equation 11 as sin βm and cos βm, and the voltage command value computing means 160 computes the voltage command value Va * using formula 10.

Then, in ST13, the voltage calculation means 15 calculates the voltage value applied to the drive load from the voltage command value obtained by the output voltage command value calculation means 10 or the like. That is, in ST13, the voltage calculation unit 15 calculates the magnitude Va of the voltage for driving the synchronous motor 5 using the previous voltage command values Vγ * and Vδ * calculated by the output voltage command value calculation unit 10. Calculation is performed using Equation 7. here,
In ST13, the voltage applied to the driving load 5 as the voltage command value may be detected by separately providing a voltage detecting means and the voltage signal may be used. Then, since the latest voltage information can be used at all times, it is possible to obtain a device capable of fine and reliable control.

At ST14, the voltage command value Va * obtained at ST12 and the voltage magnitude Va obtained at ST13.
From the voltage error calculation means 225, the voltage error (Va * -V
Find a). Then, in ST15, the reference excitation current command value iγ0 is calculated using the compensated frequency command value (primary frequency) ω1 that is the previous output value of the frequency calculation means 9 as a book. In ST16, the reference excitation current command value iγ0 obtained in ST15 and the voltage error (V
a * -Va), the excitation current command value iγ * for driving the drive load 5 with a current value at which the current phase angle is βm.
Is calculated by using the exciting current command value calculating means 170 using Expression 12 (or Expression 13). Here, in the present embodiment, since the voltage error (Va * -Va) is directly calculated in ST14, the calculation of the exciting current command value iγ * in ST16 is simplified and the calculation speed is increased. You can do it.

In ST17, the exciting current command value iγ * obtained in ST16 is filtered by the filter means 19 to remove high frequency components and suppress variations in each operation cycle. In ST18, output voltage command values Vγ *, Vδ for driving the drive load 5 based on the excitation current command value iγ * filtered in ST17.
The output voltage command value calculation means 10 obtains *. here,
Since the calculation formula used in the output voltage command value calculation step ST18 differs depending on the control system of the synchronous motor, Vγ
Any arithmetic expression can be used as long as * and Vδ * can be calculated.
For example, Equation 14 is used.

As described above, in the present embodiment, the voltage command for driving the synchronous motor at the current phase angle such that the output torque of the synchronous motor becomes a predetermined value or more based on the value of the current flowing in the synchronous motor. A voltage command value calculation step ST12 for obtaining a value, a voltage command value Va * obtained in the voltage command value calculation step ST12, and a voltage calculation step ST13 for calculating the magnitude Va of the voltage value applied to the synchronous motor.
And the voltage command value Va * obtained in the voltage command value calculation step ST12 and the voltage magnitude VA obtained in the voltage calculation step ST13 are compared to obtain a voltage error (Va * −
VA) voltage error calculation step ST14, reference excitation current command value iγ0 reference excitation current command value calculation step, voltage error calculation step ST14 voltage error (Va * -VA), and reference excitation Reference excitation current command value i obtained in the current calculation step ST15
an exciting current command calculation step ST16 for calculating an exciting current command value iγ * for driving the synchronous motor 5 at a current phase angle βm such that the output torque of the synchronous motor becomes a predetermined value or more based on γ0; Current command value calculation step S
A filter step ST17 for filtering the exciting current command value iγ * obtained by T16, and an output voltage command value Vγ * for driving the synchronous motor based on the exciting current command value iγ * filtered by the filter step ST17, Since the output voltage command value calculating step ST18 for calculating Vδ * is provided, the control of the synchronous motor is simple, even though it is a simple control for calculating the voltage error, and the controllability is good, and the synchronous motor can be efficiently driven with high output and low current. You can get the way. Further, since it has a filter step, it is possible to obtain a highly reliable method for controlling a synchronous motor that can control the synchronous motor in a stable state.

Further, the frequency command value (primary frequency) ω1 compensated by the frequency compensation amount obtained based on the current flowing through the synchronous motor, and the exciting current command value calculation step S
Excitation current command value iγ * obtained in T6 and ST16
And an output voltage command value calculation step ST18 for calculating output voltage command values Vγ * and Vδ * for driving the synchronous motor on the basis of the above, the exciting current at the current phase angle βm at which the current becomes a predetermined value or less. Based on the command value iγ * and the compensated frequency command value ω1, the output voltage command values Vγ *, V
Since δ * is calculated, even if the operating condition of the synchronous motor changes due to the rotation frequency or the load change, it is possible to obtain a control method for the synchronous motor that can accurately follow and perform stable control.

Further, even when the operating condition of the synchronous motor changes due to the rotation frequency or the load variation, the current can be controlled at a low current phase angle whose current value becomes a predetermined value or less according to the condition, and the low current can be controlled. In addition, the synchronous motor can be driven with high efficiency. Further, the followability to the phase of the low current is good, and the convergence and stability near the phase of the low current are good, so that the controllability can be improved. Further, since it is not necessary to consider the coordinate axes and only the exciting current command value is controlled, it is possible to obtain a control method capable of driving the synchronous motor with low current and high efficiency even in the control means having no speed controller or current controller.

Here, in FIG. 7, the voltage calculation step ST13 may be performed at any step before the voltage error calculation step ST14. The output voltage command value calculation step ST18 is a step of obtaining not only the control of the exciting current but also the output voltage command value for driving the synchronous motor according to the rotation frequency command value and the load torque. In FIG. 7, assuming that the processing cycle from the current detection / calculation step ST11 to the exciting current command value calculation step ST16 is the processing cycle X, and the processing cycle after the filter step ST17 in the inverter control means 2 is the processing cycle Y, the processing cycle. X and the processing cycle Y may be set to different cycles in order to reduce the processing load of a microcomputer or the like that performs processing. In this case, the processing cycle X does not require responsiveness as compared with the processing cycle Y. Therefore, if the processing cycle X is longer than the processing cycle Y, the load on the microcomputer can be reduced, so that a low-cost microcomputer can be used. , The degree of freedom in selecting a microcomputer has expanded,
It is possible to obtain a low-cost and highly reliable electric motor control method that is simple control.

In the inverter device, the synchronous motor drive device, and the synchronous motor control method shown in the first embodiment of the present invention, the synchronous motor drive device (or the inverter device) and the synchronous motor that is a drive load are connected at a ratio of 1: 1. The relationship in the above case is shown as an example, and in such a case, by incorporating the synchronous motor drive device and the control method of the present embodiment into the synchronous motor 5 which is a drive load, the synchronous motor can be integrated. It is possible to obtain a highly efficient and highly reliable synchronous motor equipped with a drive device and a control method.

Embodiment 2. A synchronous motor drive device according to Embodiment 2 of the present invention will be described below. Figure 8
2 is a diagram showing configurations of an inverter device and a synchronous motor driving device according to a second embodiment of the present invention. FIG. In the figure,
The same parts as those in FIGS. 1 to 7 of the first embodiment are designated by the same reference numerals and the description thereof will be omitted. In the figure, reference numeral 1 is a synchronous motor driving device, which is composed of a plurality of switching elements, and an inverter main circuit 3 for converting direct current into alternating current, and a PWM signal for controlling on / off of a plurality of switching elements of the inverter main circuit 3. It is comprised by the inverter control means 2 which outputs.

Further, 4 is a DC power source connected to the inverter main circuit 3, and 5 is a synchronous motor which is a driving load driven by the inverter main circuit 3. The synchronous motor driving device 1, the DC power source 4, and the driving load 5 are provided. The inverter device is configured by. The switching element in the inverter main circuit 3 is turned on / off based on the PWM signal output from the inverter control means 2, and the DC voltage of the DC power supply 4 is converted into three-phase AC and applied to the synchronous motor 5 as a drive load. As a result, the inverter device operates.

Reference numeral 6a denotes one-phase current (for example, U-phase current iu) of the current flowing into the synchronous motor 5 which is a driving load.
And 6b is a current detecting means for detecting a current (for example, V-phase current iv) of a phase different from the current (U-phase current iu) detected by the current detecting means 6a.
The current detecting means 6a and 6b convert the detected current into a current value or a current signal and output it.

Here, the inverter control means 2 comprises three-phase / two-phase conversion means 7 and rotor position / speed estimation means 2 shown below.
0, current phase control means 145, frequency comparison means 210,
Speed control means 220, γ-axis current comparison means 230, δ-axis current comparison means 240, current control means 250, two-phase / three-phase conversion means 11, PWM signal generation means 12, filter means 19
It is composed of and.

The three-phase / two-phase converting means 7 is a current detecting means 6a,
The current value for two phases (for example, U-phase current iu, V-phase current iv) or current signal detected by 6b is calculated by using the electrical angle phase θ for the excitation current component (γ-axis current iγ) and torque current component (δ-axis). It is converted into a current value or current signal on the γ-δ axis represented by the current iδ).

The rotor position / speed estimating means 20 uses the γ-δ axis currents iγ, iδ obtained by the three-phase / two-phase converting means 7 and the previous output voltage command values Vγ *, Vδ * to synchronize the drive load. The rotor position value θ and the rotation speed (rotation frequency) ωm of the electric motor 5 are estimated. The frequency comparison means 210 compares the frequency command value ωm * given from the outside with the rotor rotation frequency ωm obtained by the rotor position / speed estimation means 20, and calculates a frequency difference (ωm * −ωm). The speed control means 220 outputs the output (ωm * −) of the frequency comparison means 210.
The δ-axis current command value iδ * is obtained based on ωm).

The γ-axis current comparing means 230 has an exciting current command value iγ * obtained by filtering the exciting current command value (also called γ-axis current command value) iγ * output from the current phase control means 145 by the filter means 19. The γ-axis current iγ obtained by the coordinate conversion means 7 is compared, and the γ-axis current difference (i
γ * −iγ) is calculated. The δ-axis current comparison means 240
Δ-axis current command value iδ obtained by the speed control means 220
* Is compared with the δ-axis current iδ obtained by the coordinate conversion means 7, and the δ-axis current difference (iδ * -iδ) is calculated. The current control unit 250 outputs the output value of the γ-axis current comparison unit 230 (i
γ * -iγ) and the output value (iδ * -iδ) of the δ-axis current comparison means 240 based on the output voltage command value Vγ *,
Calculate Vδ *.

The current phase control means 145 includes an output voltage calculation means 15 for obtaining the magnitude of the output voltage, a voltage command value calculation means 165 for obtaining a voltage command value at which the current is below a predetermined value (or a minimum), and a current is a predetermined value. An excitation current command value calculation means 175 for obtaining an excitation current command value for driving a drive load so as to be the following (or a minimum), and a reference excitation current command value calculation means 18 for obtaining a reference excitation current command value. Then, the exciting current command value calculator 175 outputs the exciting current command value iγ *.

Further, reference numeral 19 is equivalent to the filter means described in the first embodiment, and the exciting current command value iγ * which is the output of the current phase control means 145 is subjected to removal of high frequency components by a low-pass filter or the like for every calculation cycle. It is a filter means for performing filtering for suppressing the variation of. In the present embodiment, as described in the first embodiment, since the filtering means 19 is provided to perform the filtering for removing the high frequency component, the stability of the calculation of the output voltage command values Vγ * and Vδ * is performed. Further, the calculation accuracy can be improved, the instability of the operation of the synchronous motor 5 due to the fluctuation of the exciting current command value iγ * can be avoided, and the controllability and the reliability of the device can be obtained.

Numeral 11 is a three-phase voltage command value using the γ-δ axis voltage command values Vγ * and Vδ * obtained by the current control means 250 using the electrical angle phase θ obtained by the rotor position / speed estimation means 20. Two-phase / three-phase conversion means for converting into Vu *, Vv *, Vw *, 12 is obtained by the two-phase / three-phase conversion means 11
It is a PWM signal generating means for generating a PWM signal for ON / OFF controlling the switching elements in the inverter main circuit 3 with the phase voltage command values Vu *, Vv *, Vw *.

Next, the operation will be described with reference to FIG.
In the figure, the synchronous motor drive device 1 detects currents of two phases (for example, U-phase current iu and V-phase current iv) out of phase currents flowing into the synchronous motor 5 which is a drive load of an inverter, as current detection means 6a, 6b. To detect more. Current detection means 6
The inverter control means 2 uses the currents of the two phases (for example, the U-phase current iu and the V-phase current iv) or the two-phase current signals detected by a and 6b to drive the synchronous motor 5, and the inverter main unit 2 drives the inverter main unit. PWM for calculating the voltage (or voltage signal) output from the circuit 3 and performing on / off control of the switching element in the inverter main circuit 3
Output a signal.

That is, the inverter control means 2 outputs the PWM signal by the following operation. Using the electrical angle phase θ from the phase currents iu and iv detected by the current detecting means 6a and 6b, the three-phase / two-phase converting means 7 uses γ-δ.
The axial currents iγ and iδ are obtained. The rotor position / speed estimation means 20 has the γ-δ axis currents iγ, iδ obtained by the three-phase / two-phase conversion means 7 and the γ-δ axis voltage command value Vγ *, which is the previous output value of the current control means 250. The current rotor position (positional electrical angle phase) θ and speed (rotation frequency) ωm of the synchronous motor 5 are estimated from Vδ *.

The frequency comparison means 210 compares the rotation frequency command value ωm * given from the outside with the rotation frequency ωm which is the output of the rotor position / speed estimation means 20, and obtains the difference Δωm. The speed control unit 220 obtains a torque current command value (δ-axis current command value) iδ * by proportional-plus-integral control (speed control) based on the rotation frequency difference Δωm. The δ-axis current comparison means 240 compares the torque current command value iδ * and the δ-axis current iδ to obtain the difference (δ-axis current difference) Δiδ.

In the current phase control means 145, the γ-axis current iγ and the δ-axis current i obtained by the two-phase / three-phase coordinate conversion means 7 are obtained.
δ and Vγ which is the previous output of the current control means 250
Based on *, Vδ *, the exciting current command value iγ * for driving the synchronous motor at the current phase angle βm at which the current is below a predetermined value (or minimum) is output. The current phase control means 145 constituted by the voltage command value calculation means 165, the voltage calculation means 15, the reference excitation current command value calculation means 18, and the excitation current command value calculation means 175 will be described below.

The voltage command value calculating means 165 has the γ-axis current iγ and the δ-axis current i obtained by the two-phase / three-phase coordinate converting means 7.
Based on δ and the estimated speed ωm obtained by the rotor position / speed estimation means 20, a current phase angle βm at which the current becomes a predetermined value or less (or a minimum) is calculated, and this current phase angle β
The voltage command value Va * for driving the synchronous motor is calculated at m. The voltage calculation means 15 calculates the magnitude Va based on the voltage command values Vγ * and Vδ * which are the previous output values of the current control means 250. Further, in the reference exciting current command value calculating means 18, the reference exciting current command value i is based on the estimated speed ωm obtained from the rotor position / speed estimating means 20.
Calculate γ0. Then, the exciting current command value calculating means 17
5, the voltage command value Va *, the voltage magnitude Va (or the voltage error between the voltage command value Va * and the output voltage command value (Va
* -Va)) and the reference excitation current command value iγ0 obtained from the reference excitation current command value calculating means 18, and an excitation current command value for driving the synchronous motor at a current below a predetermined value (or minimum). Output iγ *. In this way, the current phase control means 145 outputs the exciting current command value iγ * by the exciting current command value calculating means 175.

The filter means 19 is the current phase control means 1
The high-frequency component of the exciting current command value iγ * is removed by a low-pass filter or the like in order to remove the high-frequency component of the exciting current command value iγ *, which is the output of 45, and suppress the variation in each operation cycle to perform stable control. Perform filtering. Here, in a steady operation state in which the rotation frequency and the load torque of the synchronous motor do not substantially change, the exciting current command value iγ * hardly changes. When a low-pass filter is used as the filter means 19 in such a case, the time constant of the low-pass filter may be set to a value sufficiently larger than the exciting current control cycle (calculation cycle in the current phase control means 145). For example, the time constant of the low-pass filter is the excitation current control cycle (calculation cycle in the current phase control means 145) or the calculation cycle other than the excitation current calculation (for example, the current control means 250 excluding the calculation cycle in the current phase control means 145). It should be designed considering the calculation cycle of
When the exciting current control period is about 10 msec, the time constant of the low pass filter may be set to about 1 sec.

In the γ-axis current comparison means 230, the γ-axis current command value (excitation current command value) iγ * filtered by the filter means 19 and the γ-axis current value iγ obtained by the three-phase / two-phase conversion means 7 are obtained. Compare and compare (γ-axis current difference) Δ
Find iγ. In the current control means 250, the γ-axis current difference Δ
Output voltage command values Vγ * and Vδ * are obtained from the iγ and δ-axis current difference Δiδ by proportional-plus-integral control (current control).

The 2-phase / 3-phase conversion means 11 converts the γ-δ axis voltage command values Vγ *, Vδ * into the voltage command values Vu *, Vv *, Vw * of the three-phase coordinate system using the electrical angle phase θ. . PW
The M signal generating means 12 uses three-phase voltage command values Vu *, Vv *,
A PWM signal for controlling ON / OFF of the switching element in the inverter main circuit 3 is generated from Vw *. The switching element in the inverter main circuit 3 is turned on / off based on this PWM signal, the DC voltage of the DC power supply 4 is converted into a three-phase AC and applied to the synchronous motor 5 as a drive load, and the synchronous motor 5 is driven. To be done.

The inverter device and the synchronous motor drive device according to the present embodiment are another example of the device that is driven without using the position sensor for detecting the rotor position, and the inverter device and the synchronous motor drive device according to the first embodiment are used. It is different from the device and shows an example of a control block diagram in the case of vector-controlling the synchronous motor 5 which is a drive load. The difference from the first embodiment is that the rotor position / speed estimation means 20 and the speed control means 22 for estimating the rotor position.
0 and the current control means 250 are provided. Even in the drive device including the speed controller and the current controller as in the present embodiment, the synchronous motor 5 is in a state where the torque is equal to or higher than a predetermined value (or maximum), that is, the current is equal to or lower than a predetermined value (or minimum). ), It is possible to drive with high efficiency. In the high-efficiency driving method (current phase control method) in which the current is equal to or lower than the predetermined value (or the minimum value), if the control method similar to the method described in Embodiment 1 with reference to FIGS. It is possible to obtain a highly reliable motor control method that is simple in cost, low in cost, high in efficiency, high in output.

Further, in this embodiment, the current detection means 6a, 6b for detecting the current flowing through the synchronous motor, the current signal obtained by the current detection means 6a, 6b and the voltage of the voltage applied to the synchronous motor 5 are detected. A current phase control means for outputting an exciting current command value iγ * for controlling the current phase angle of the current flowing through the synchronous motor 5 so that the output torque of the synchronous motor 5 becomes a predetermined value or more (or maximum) based on the signal. 145, a filter means 19 for filtering the excitation current command value iγ * obtained by the current phase control means 145, and the torque detected by the current detected by the current detection means 6a, 6b or the estimated speed ωm obtained from the current signal. Speed control means 220 for calculating the command value iδ *
And the torque current command value iδ * obtained by the speed control means 220 and the excitation current command value iγ * filtered by the filter means 19 so that the output torque of the synchronous motor becomes equal to or greater than a predetermined value (or maximum). Output voltage command value Vγ for driving at a different current phase angle βm
Since the current control means 250 for obtaining * and Vδ * is provided, the torque current is also controlled, so that speed controllability is good, and since torque control is possible, noise and vibration can be suppressed.
Further, since the filter means 19 is provided, it is possible to suppress the variation of the exciting current command value iγ * for each calculation cycle, and to obtain a motive motor drive device and an inverter device that can perform stable control.

Further, in the present embodiment and the first embodiment, when the synchronous motor 5 which is the drive load is driven, the current phase control means is provided from the start up until the predetermined rotation speed (low speed rotation) is reached. When you execute 14, 140, 145,
There is a case where the synchronous operation cannot be maintained due to fluctuations in the output torque of the synchronous motor 5 during the execution of control, and a step out occurs. For this reason,
If the current phase control means 14, 140, 145 are not executed until the synchronous motor 5 reaches a predetermined number of revolutions or more, step-out can be avoided and a highly reliable device can be obtained. .

Further, in the present embodiment and the first embodiment, in the region where the PWM signal is overmodulated, it is difficult to control the output voltage due to overmodulation, and therefore the current phase control means 14, 140, 145. There is almost no effect of executing. Therefore, in the overmodulation region, the current phase control means 14, 140, 145 may not be executed to simplify the control. In particular, when integral control is used to control the excitation current, the operation of the integrator becomes abnormal during overmodulation, so it is better not to perform current phase control in the overmodulation region because stable control can be performed and high efficiency can be achieved. It is possible to obtain a highly reliable device.

Further, in the present embodiment and the first embodiment, if the current phase control means 14, 140, 145 are executed in an unsteady operation state such as startup or acceleration / deceleration operation, the control is unsuccessful. If it becomes stable or loses step, it is in an unsteady operating state (such as during startup or acceleration / deceleration).
Sets the reference excitation current command value iγ0 obtained from the reference excitation current command value calculation means 18 to a strong level, and sets the strengthened iγ0 to the excitation current command value iγ * (iγ * = i
When used as (γ0), unstable control or step-out does not occur, and a highly reliable device capable of stable control can be obtained (strong excitation control). (Here, "strong excitation" generally refers to a case where an exciting current is passed so that the combined magnetic flux vector Φ0 strengthens the magnetic flux vector ΦF due to the rotor magnet (when id> 0). Conversely, the combined magnetic flux vector rotates. When exciting current is passed so as to weaken the magnetic flux vector ΦF due to the child magnet (id <0
In the case of), weakening is called excitation. )

Further, the inverter device and the synchronous motor drive device described in the present embodiment and the first embodiment are examples of the sensorless drive system which does not use the position sensor for detecting the rotor position of the synchronous motor 5 which is the drive load. Since a position sensor is unnecessary, a low cost and highly reliable device can be obtained. Further, the control method described in the present embodiment can be implemented in a drive method (with a position sensor) that detects the rotor position, and the same effect can be obtained.

Further, in the inverter device and the synchronous motor drive device described in the present embodiment and the first embodiment, the control amount (exciting current) for performing the high-efficiency control in which the current is the predetermined value or less (or the minimum). In the calculation of the command value, R (resistance value per one phase of IPMSM), Ld
(D-axis inductance of IPMSM), Lq (IPMS
The calculation is performed using a motor constant indicating the characteristics of the motor such as M q-axis inductance) and Φf (IPMSM induced voltage constant). For this reason, when the temperature of the electric motor changes due to heat generation of the electric motor or changes in the surrounding environment, the electric motor constant changes, so that an error may occur between the actual electric motor constant and the electric motor constant used in the calculation. When this error occurs, when performing high-efficiency control with the minimum current, the operating point with the minimum current deviates from the actual minimum current point, making control at the minimum current point difficult.

Even for such a change in the motor constant,
In order to realize high-efficiency driving with the minimum current, for example, the value of the current flowing through the driving load 5 in the output voltage command value calculating means 10 described in the first embodiment or the current control means 250 described in the second embodiment. By providing a motor constant measuring means for measuring the motor constant from the voltage value and a control constant changing means for changing the control constant such as control gain and frequency compensation gain, etc., online or offline, using the motor constant measuring means. It suffices to measure the electric motor constant, correct the electric motor constant with respect to changes in the ambient environment such as temperature, and change the control constant by the control constant changing means according to the corrected electric motor constant. By doing this, the motor constant can be used to change the characteristics of the motor (temperature change, etc.).
Since the control constant can also be changed by following the above, even if the motor constant fluctuates due to changes in the ambient environment such as the temperature change of the motor, the output torque is maximum and the current is minimum, so high efficiency and high output. Control becomes feasible, and a highly efficient inverter device and electric motor drive device can be obtained.

In the present embodiment and the first embodiment, the embedded magnet type synchronous motor (IP
Although the MSM) is shown as an example, the same effect as the effect described in the first and second embodiments can be obtained with any motor as long as it is a synchronous motor. For example, a synchronous motor having a permanent magnet such as a permanent magnet type synchronous motor (PMSM) or a brushless DC motor (BLDCM) can be realized by using the mathematical formulas equivalent to those in the first and second embodiments. Especially in the permanent magnet type synchronous motor (PMSM), the surface magnet type synchronous motor (SPM)
In SM), the d-axis inductance Ld and the q-axis inductance Lq in Equations 1 to 13 may be set to Ld = Lq, and the reluctance motor (RM), the synchronous reluctance motor (SyRM), and the switched reluctance motor ( In the case of a synchronous motor having no permanent magnet such as SRM), the induced voltage constant Φf in Expressions 1 to 13 may be calculated with Φf = 0.

[0165]

According to the first aspect of the present invention, there is provided a synchronous motor driving device, which includes a current detecting means for detecting a current flowing through the synchronous motor, a current signal obtained by the current detecting means, and a voltage of a voltage applied to the synchronous motor. A current phase control means for outputting an exciting current command value for controlling the current phase angle of the current flowing through the synchronous motor so that the output torque of the synchronous motor becomes a predetermined value or more based on the signal, and output by the current phase control means. Filter means for filtering the excited magnetic current command value, and frequency calculating means for obtaining a compensation amount for the frequency command value from the amount of fluctuation of the current detected by the current detection means and outputting the frequency command value compensated by the compensation amount. , Synchronization based on the frequency command value compensated by the frequency calculation means and the excitation current command value filtered by the filter means The output voltage command value calculating means for obtaining the output voltage command value for driving the motive is provided, and therefore, based on the excitation current command value and the compensated frequency command value at the current phase angle at which the current becomes a predetermined value or less. Since the output voltage command value is calculated based on the output voltage command value, even if the operating condition of the synchronous motor changes due to the rotation frequency or load fluctuation, the current value is less than the predetermined value according to the output torque under that condition. It is possible to realize a synchronous motor drive device that can control the electric current and has a low current and high efficiency. Further, since the filter means 19 is provided, the variation of the exciting current command value iγ * for each calculation becomes small, and stable control can be performed.

The synchronous motor drive device according to claim 2 of the present invention is a current phase calculating means for obtaining a current phase angle such that the output torque of the synchronous motor becomes a predetermined value or more based on the current signal obtained by the current detecting means. And a voltage command value calculation means for calculating a voltage command value based on the current phase angle and the frequency command value obtained by the current phase calculation means, and a voltage magnitude from the voltage signal of the voltage applied to the synchronous motor. Excitation current command value for obtaining the excitation current command value for driving the synchronous motor based on the voltage calculation means to be obtained and the voltage command value obtained by the voltage command value calculation means and the magnitude of the voltage obtained by the voltage calculation means Since the current phase control means is constituted by the computing means, the difference between the output voltage and the actual output voltage at the current phase angle at which the current is below a predetermined value (or minimum) is obtained. Despite its simple configuration, it is not necessary to have a numerical table or function of the excitation current command value for each operating condition, so the memory capacity of the microcomputer used for the inverter control means is not increased and the memory capacity is small. Since an inexpensive microcomputer can be used and a simple control circuit can be used, a highly reliable and inexpensive synchronous motor drive device can be obtained.

In the synchronous motor drive device according to a third aspect of the present invention, the output torque of the synchronous motor becomes a predetermined value or more based on the current signal detected by the current detecting means and the voltage signal of the voltage applied to the synchronous motor. With a voltage command value calculating means for obtaining a voltage command value for driving at such a current phase angle, a voltage calculating means for obtaining a voltage magnitude from a voltage signal of a voltage applied to the synchronous motor, and a voltage command value calculating means. The output torque of the synchronous motor is a predetermined value based on the voltage error calculation means for obtaining the voltage error by comparing the obtained voltage command value with the magnitude of the voltage obtained by the voltage calculation means, and the voltage error obtained by the voltage error calculation means. Since the current phase control means is configured by the exciting current command value calculating means for calculating the exciting current command value for driving at the current phase angle as described above, the rotation frequency Even if the operating condition of the synchronous motor changes due to load fluctuations or load fluctuations, the current can be controlled with a low current phase angle at which the output torque exceeds the specified value according to the condition, and the synchronous motor drive with low current and high efficiency can be achieved. The device can be realized. Further, it is possible to realize a synchronous motor drive device that has a good followability to a current phase angle at which the current value is equal to or less than a predetermined value, has good convergence and stability near this current phase angle, and has a low current and high efficiency. Also, since only the excitation current command value is controlled,
Even in the control means having no speed controller or current controller, it is possible to realize a synchronous motor drive device having a good low current and high efficiency.

In the synchronous motor drive device according to the fourth aspect of the present invention, the output voltage command value obtained from the output voltage command value calculating means is used for the voltage signal. An efficient synchronous motor drive device can be realized.

According to a fifth aspect of the present invention, there is provided a synchronous motor driving device, wherein a current detecting means for detecting a current flowing through the synchronous motor, a current signal obtained by the current detecting means and a voltage of a voltage applied to the synchronous motor. The current phase control means outputs an exciting current command value for controlling the current phase angle of the current flowing through the synchronous motor so that the output torque of the synchronous motor becomes a predetermined value or more based on the signal, and is obtained by the current phase control means. Filter means for filtering the generated excitation current command value, speed control means for calculating the torque current command value from the estimated speed obtained from the current detected by the current detection means, and torque current command obtained by the speed control means Based on the value and the excitation current command value filtered by the filter means, the output torque of the synchronous motor becomes equal to or higher than a predetermined value. Since the current control means for determining the output voltage command value for driving at the current phase angle is provided, the torque current is also controlled, so that the speed controllability is good, and since the torque control is possible, noise and vibration are suppressed. it can.

In the synchronous motor drive device according to claim 6 of the present invention, the exciting current control period controlled by the current phase control means is made different from the output voltage control period for calculating the output voltage command value for driving the synchronous motor. Since the processing load on the microcomputer can be reduced, an inexpensive microcomputer can be used and the degree of freedom in selecting a microcomputer can be expanded. Further, it is possible to realize a highly efficient synchronous motor drive device with a low current while having a simple structure.

In the synchronous motor drive device according to the seventh aspect of the present invention, the exciting current control by the current phase control means is not performed during low speed rotation from the start up to the predetermined number of rotations. It is possible to realize a highly efficient and highly efficient synchronous motor drive device.

In the synchronous motor drive device according to the eighth aspect of the present invention, the excitation current control by the current phase control means is not performed in the region where the PWM modulation is overmodulation, so that the control becomes unstable during overmodulation. Can be suppressed, and a highly efficient and highly efficient synchronous motor drive device can be realized.

According to the ninth aspect of the present invention, in the synchronous electric drive device, at the time of start-up or acceleration / deceleration operation, the synchronous motor is driven using a predetermined exciting current command value without using the current phase control means, and a predetermined value is set. After the operating conditions are reached, the excitation current command value obtained by the current phase control means is switched to use, so that the stability of the control can be secured at the time of start-up or transition of the synchronous motor, and the reliability can be improved. A highly efficient synchronous motor drive device can be realized.

According to a tenth aspect of the present invention, there is provided a synchronous motor driving device, wherein a motor constant measuring means for measuring the motor constant of the synchronous motor and a control for controlling the synchronous motor based on the motor constant measured by the motor constant measuring means. Since the control constant changing means for changing the constant is provided and the synchronous motor is driven and controlled according to the change in the surrounding environment of the synchronous motor, the change in the characteristic due to the temperature change of the electric motor or the environment in which the electric motor is used is changed. Even when the motor characteristics such as the motor constant change due to the change, it is possible to automatically adapt to the change of the motor constant, and it is possible to realize a synchronous motor drive device with low current and high efficiency.

According to the eleventh aspect of the present invention, there is provided a synchronous motor drive device which is driven without using a position sensor for detecting the rotor position of the synchronous motor.
It is possible to realize a synchronous motor drive device capable of performing high-output and highly-efficient control even in a synchronous motor to which a position sensor cannot be attached.

Since the inverter device according to claim 12 of the present invention includes the synchronous motor drive device according to any one of claims 1 to 10, a high output and high efficiency inverter device can be obtained.

An inverter device according to a thirteenth aspect of the present invention is an inverter main circuit for converting a direct current from a direct current power source into an alternating current by turning on and off a switching element, and an on-state of a switching element in the inverter main circuit.
An inverter control unit that outputs a voltage command value for controlling the off operation, and a drive load that is controlled by applying a voltage output by an inverter main circuit controlled by the inverter control unit, are provided. A current phase control means for calculating an exciting current command value for driving the drive load at a current phase angle such that the output torque of the drive load becomes a predetermined value or more based on the current flowing through the drive load; Filtering means for filtering the exciting current command value obtained from the current phase control means, frequency calculating means for compensating the frequency command value from the fluctuation amount of the current flowing through the driving load and outputting as a primary frequency, filtered exciting current An output voltage command value calculating means for calculating a voltage command value for driving the drive load based on the command value and the primary frequency; Since it is configured, even if the operating conditions of the driving load changes can be controlled so that the output torque at the conditions to be a predetermined value or more. Further, the current can be controlled at the current phase angle at which the current value becomes equal to or less than the predetermined value according to the output torque, and the synchronous motor drive device with low current and high efficiency can be realized. Further, since the filter means 19 is provided, variations in the exciting current command value iγ * are reduced, and a highly reliable inverter device capable of stably controlling the drive load can be obtained.

According to the fourteenth aspect of the present invention, in the inverter device, the current phase calculating means for obtaining the current phase angle at which the output torque becomes a predetermined value or more due to the current flowing through the drive load, and the current phase obtained by the current phase calculating means. Based on the voltage command value calculating means for calculating the voltage command value for obtaining the angle, and the output voltage command value calculated by the output voltage command value calculating means and the voltage command value calculated by the voltage command value calculating means Since the current phase control means is constituted by the exciting current command value calculating means for obtaining the exciting current command value for driving the driving load, the current phase angle is set to the current phase angle such that the current value flowing in the driving load becomes the predetermined value or less. Since the followability is good, and the convergence and stability are good near the current phase angle where the current value is less than a predetermined value, the controllability is good, and the inverter device is low current and highly efficient. It can be realized. Further, since only the exciting current command value is controlled, it is possible to realize an inverter device having a low current and a high efficiency even when the speed controller and the current controller are not provided.

According to a fifteenth aspect of the present invention, there is provided a control method for a synchronous motor drive device, wherein a current phase angle for obtaining a current phase angle such that an output torque of the synchronous motor becomes a predetermined value or more based on a current value flowing through the synchronous motor. An excitation current command value calculation step for obtaining an excitation current command value for driving the synchronous motor based on the calculation step and the current phase angle obtained in the current phase calculation step and the voltage value applied to the synchronous motor; , For driving the synchronous motor based on the excitation current command value filtered in the filter step and the filtering step for removing the high frequency component of the excitation current command value obtained in the excitation current command value calculation step Since the output voltage command value calculation step for obtaining the output voltage command value of is provided, high output and low current can be achieved with simple control. The method of rate and stably performed for controlling the synchronous motor can be obtained. Further, since the simple control is performed to control only the exciting current command value, it is possible to obtain a control method capable of accurately driving the synchronous motor with low current and high efficiency even in the control means having no speed controller or current controller.

According to a sixteenth aspect of the present invention, there is provided a control method for a synchronous motor drive device, wherein the synchronous motor is driven at a current phase angle such that the output torque of the synchronous motor becomes a predetermined value or more based on the current value flowing through the synchronous motor. Voltage command value calculation step for obtaining the voltage command value for driving, voltage calculation step for calculating the magnitude of the voltage applied to the synchronous motor, voltage magnitude and voltage command value calculation obtained in the voltage calculation step The output torque of the synchronous motor becomes equal to or greater than a predetermined value based on the voltage error calculation step for obtaining the voltage error by comparing the voltage command value obtained in the step and the voltage error obtained in the voltage error calculation step. Excitation current command calculation step for calculating the excitation current command value for driving the synchronous motor with the current phase angle as described above, and the excitation current obtained in the excitation current command value calculation step A filtering step for removing high frequency components of the command value, and an output voltage command value calculating step for obtaining an output voltage command value for driving the synchronous motor based on the excitation current command value filtered in the filtering step. Since the simple control is provided, the controllability is good and the synchronous motor can be driven with a low current. Further, even when the operating condition of the synchronous motor changes due to the rotation frequency or the load change, the current can be controlled with the low current phase angle that makes the current equal to or less than the predetermined value according to the condition, and the current is low and the efficiency is high. The synchronous motor can be driven with. Further, the followability to the phase of the low current is good, and the convergence and stability near the phase of the low current are good, so that the controllability can be improved. Further, since it has a filter step, it is possible to obtain a highly reliable method for controlling a synchronous motor that can control the synchronous motor in a stable state.

According to a seventeenth aspect of the present invention, there is provided a method of controlling a synchronous motor drive device, wherein a frequency command value compensated by a frequency compensation amount obtained based on a current flowing through the synchronous motor and an exciting current command value calculation step are used. Since the output voltage command value calculation step for obtaining the output voltage command value for driving the synchronous motor based on the generated excitation current command value is provided, the excitation current command at the current phase angle at which the current becomes the predetermined value or less Since the output voltage command value is calculated based on the value and the compensated frequency command value, even if the operating condition of the synchronous motor changes due to the rotation frequency or load fluctuation,
It can follow accurately and can perform stable control.

[Brief description of drawings]

FIG. 1 is a diagram showing an example of a configuration of a synchronous motor drive device representing a first embodiment of the present invention.

FIG. 2 is a diagram illustrating a relationship between a current phase angle and a generated torque of the synchronous motor according to the first embodiment of the present invention.

FIG. 3 is a diagram illustrating a relationship between a current phase angle and a current of the synchronous motor according to the first embodiment of the present invention with respect to a change in output torque.

FIG. 4 is a diagram showing a vector diagram of a general synchronous motor in the first embodiment of the present invention.

FIG. 5 is a flowchart diagram illustrating a method of controlling the synchronous motor drive device according to the first embodiment of the present invention.

FIG. 6 is a diagram showing an example of another configuration of the synchronous motor drive device representing the first embodiment of the present invention.

FIG. 7 is a flowchart illustrating another control method of the synchronous motor drive device according to the first embodiment of the present invention.

FIG. 8 is a diagram showing an example of a configuration of a synchronous motor drive device that represents Embodiment 2 of the present invention.

FIG. 9 is a diagram showing an example of a configuration of a conventional synchronous motor drive device.

[Explanation of symbols]

1 Synchronous motor drive device, 2 Inverter control means, 3
Inverter main circuit, 4 DC power supply, 5 synchronous motor,
6a, 6b Current detection means, 7 3 phase 2 phase conversion means, 8
Frequency compensator, 9 primary frequency calculating means, 10 output voltage command value calculating means, 11 2 phase 3 phase coordinate converting means, 1
2 PWM signal generating means, 13 electrical angle phase calculating means, 1
4 current phase control means, 15 output voltage calculation means, 16
Voltage command value calculating means, 17 exciting current command calculating means,
18 reference excitation current command calculation means, 19 filter means, 20 rotor position / speed estimation means, 21 current calculation means, 22 current phase calculation means, 140, 145 current phase control means, 160, 165 voltage command value calculation means,
170, 175 excitation current command value calculation means, 210 frequency comparison means, 220 speed control means, 225 voltage error calculation means, 230 γ-axis current comparison means, 240 δ-axis current comparison means, 250 current control means.

   ─────────────────────────────────────────────────── ─── Continued front page    (72) Inventor Yosuke Shinomoto             2-3 2-3 Marunouchi, Chiyoda-ku, Tokyo             Inside Ryo Electric Co., Ltd. (72) Inventor Koichi Arisawa             2-3 2-3 Marunouchi, Chiyoda-ku, Tokyo             Inside Ryo Electric Co., Ltd. (72) Inventor Namihei Suzuki             2-3 2-3 Marunouchi, Chiyoda-ku, Tokyo             Inside Ryo Electric Co., Ltd. (72) Inventor Isao Kawasaki             2-3 2-3 Marunouchi, Chiyoda-ku, Tokyo             Inside Ryo Electric Co., Ltd. (72) Inventor Kazuyuki Mitsushima             2-3 2-3 Marunouchi, Chiyoda-ku, Tokyo             Inside Ryo Electric Co., Ltd. (72) Inventor Tsutomu Makino             5, Kamigome, Higashioosone Town, Kita Ward, Nagoya City, Aichi Prefecture             Address 1071 Mitsubishi Electric Mechatronics Software             Wear Co., Ltd. F term (reference) 5H560 BB04 BB12 DA14 DB20 DC12                       EB01 GG04 RR04 SS01 XA02                       XA04 XA12 XA13                 5H576 BB02 CC01 DD07 EE01 EE02                       EE11 GG02 GG04 HB02 JJ03                       LL14 LL22

Claims (17)

[Claims] 1. An output torque of the synchronous motor is determined based on a current detecting means for detecting a current flowing through the synchronous motor, and a current signal obtained by the current detecting means and a voltage signal of a voltage applied to the synchronous motor. Current phase control means for outputting an exciting current command value for controlling the current phase angle of the current flowing through the synchronous motor so as to be a predetermined value or more, and filtering the exciting current command value output by the current phase control means. Filter means, a frequency calculation means for obtaining a compensation amount for a frequency command value from a variation amount of the current detected by the current detection means, and outputting a frequency command value compensated by the compensation amount, and the frequency calculation means. The synchronous motor drive is based on the compensated frequency command value and the exciting current command value filtered by the filter means. And an output voltage command value calculating means for obtaining an output voltage command value for driving the machine. 2. A current phase calculation means for obtaining a current phase angle such that the output torque of the synchronous motor becomes a predetermined value or more based on the current signal obtained by the current detection means,
A voltage command value calculation means for calculating a voltage command value based on the current phase angle and the frequency command value obtained by the current phase calculation means, and a voltage magnitude from the voltage signal of the voltage applied to the synchronous motor. An exciting current command value for driving the synchronous motor based on the voltage calculating means to be obtained and the voltage command value obtained by the voltage command value calculating means and the magnitude of the voltage obtained by the voltage calculating means. The exciting current command value calculating means to be obtained constitutes the current phase control means.
The synchronous motor drive device according to. 3. A drive is performed at a current phase angle such that the output torque of the synchronous motor becomes a predetermined value or more based on the current signal detected by the current detecting means and the voltage signal of the voltage applied to the synchronous motor. Voltage command value calculating means for obtaining a voltage command value for calculating, a voltage calculating means for calculating a voltage magnitude from a voltage signal of a voltage applied to the synchronous motor, and the voltage command obtained by the voltage command value calculating means. The output torque of the synchronous motor is equal to or more than a predetermined value based on the voltage error calculation means for obtaining a voltage error by comparing the value with the magnitude of the voltage obtained by the voltage calculation means, and the voltage error obtained by the voltage error calculation means. And the exciting current command value calculating means for calculating an exciting current command value for driving at a current phase angle The synchronous motor drive device according to claim 1. 4. The synchronous motor drive according to claim 1, wherein an output voltage command value obtained by the output voltage command value calculation means is used for the voltage signal. apparatus. 5. An output torque of the synchronous motor is determined based on a current detecting means for detecting a current flowing through the synchronous motor, and a current signal obtained by the current detecting means and a voltage signal of a voltage applied to the synchronous motor. Current phase control means for outputting an exciting current command value for controlling the current phase angle of the current flowing through the synchronous motor so as to be a predetermined value or more, and the exciting current command value obtained by the current phase control means. Filtering means for filtering, speed control means for calculating a torque current command value by an estimated speed obtained from the current detected by the current detection means, torque current command value obtained by the speed control means, and the filter means Such that the output torque of the synchronous motor becomes a predetermined value or more based on the excitation current command value filtered by A synchronous motor drive device, comprising: a current control unit that obtains an output voltage command value for driving at a current phase angle. 6. The exciting current control cycle controlled by the current phase control means is different from the output voltage control cycle for calculating an output voltage command value for driving the synchronous motor. The synchronous motor drive device according to claim 5. 7. The synchronous motor according to any one of claims 1 to 6, wherein the exciting current control by the current phase control means is not performed during low-speed rotation from startup to a predetermined rotation speed. Drive. 8. The synchronous motor according to claim 1, wherein the excitation current control by the current phase control means is not performed in a region where the PWM modulation is overmodulation. Drive. 9. At the time of start-up or acceleration / deceleration operation, the synchronous motor is driven by using a predetermined exciting current command value without using the current phase control means, and after the predetermined operating condition is reached, 9. The synchronous motor drive device according to claim 1, wherein the excitation current command value obtained by the current phase control means is switched to use. 10. Motor constant measuring means for measuring the motor constant of the synchronous motor, and control constant changing means for changing the control constant for controlling the synchronous motor based on the motor constant measured by the motor constant measuring means. 10. The synchronous motor drive device according to claim 1, wherein the synchronous motor drive control is performed according to a change in the surrounding environment of the synchronous motor. 11. The synchronous motor drive device according to claim 1, wherein the synchronous motor is driven without using a position sensor for detecting the rotor position of the synchronous motor. 12. An inverter device comprising the synchronous motor drive device according to any one of claims 1 to 11. 13. A direct current power supply for supplying direct current, an inverter main circuit for converting direct current from the direct current power supply into alternating current by turning on and off a switching element, and turning on a switching element in the inverter main circuit. An inverter control unit that outputs a voltage command value for controlling the OFF operation, and a drive load that is controlled by applying a voltage output by the inverter main circuit controlled by the inverter control unit are provided. , The inverter control means calculates an excitation current command value for driving the drive load at a current phase angle such that the output torque of the drive load becomes a predetermined value or more based on the current flowing through the drive load. Current phase control means and filter means for filtering the exciting current command value obtained from the current phase control means A frequency calculation means for compensating a frequency command value based on a variation amount of a current flowing through the drive load and outputting it as a primary frequency; and driving the drive load based on the filtered exciting current command value and the primary frequency. And an output voltage command value calculating means for calculating a voltage command value for performing the inverter device. 14. A current phase calculating means for obtaining a current phase angle at which an output torque becomes a predetermined value or more based on a current flowing through the drive load, and a current phase angle obtained by the current phase calculating means is obtained. A voltage command value calculating means for calculating a voltage command value, and the drive load based on the output voltage command value calculated by the output voltage command value calculating means and the voltage command value calculated by the voltage command value calculating means. 14. The inverter device according to claim 13, wherein the current phase control means is constituted by an exciting current command value calculating means for obtaining an exciting current command value for driving. 15. A current phase calculation step for obtaining a current phase angle such that an output torque of the synchronous motor becomes a predetermined value or more on the basis of a current value flowing in the synchronous motor, and the current phase calculation step. Obtained in an exciting current command value calculating step for obtaining an exciting current command value for driving the synchronous motor based on the current phase angle and the voltage value applied to the synchronous motor, and the exciting current command value calculating step. A filtering step for removing high frequency components of the excitation current command value, and an output for obtaining an output voltage command value for driving the synchronous motor based on the excitation current command value filtered in the filtering step. A method of controlling a synchronous motor, comprising: a voltage command value calculating step. 16. A voltage command value calculation for obtaining a voltage command value for driving the synchronous motor at a current phase angle such that an output torque of the synchronous motor becomes a predetermined value or more based on a current value flowing in the synchronous motor. A voltage calculation step for calculating the magnitude of the voltage applied to the synchronous motor, a voltage magnitude obtained in the voltage computation step, and a voltage command value obtained in the voltage command value computation step. And a voltage error calculation step of calculating a voltage error by comparing
Excitation that calculates an excitation current command value for driving the synchronous motor at a current phase angle such that the output torque of the synchronous motor becomes a predetermined value or more based on the voltage error obtained in the voltage error calculation step A current command calculation step, a filter step for performing filtering for removing high frequency components of the excitation current command value obtained in the excitation current command value calculation step, and an excitation current command value filtered in the filter step based on An output voltage command value calculation step of obtaining an output voltage command value for driving the synchronous motor, the control method of the synchronous motor. 17. The synchronization is performed based on a frequency command value compensated by a frequency compensation amount obtained based on a current flowing through the synchronous motor and the filtered excitation current command value obtained by the filtering step. An output voltage command value calculating step for obtaining an output voltage command value for driving an electric motor is provided.
The method of controlling the synchronous motor according to claim 5 or claim 16.