JP2009124812A - Motor driving device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To carry out sensorless sine wave driving by V/f control, according to motor load torque. <P>SOLUTION: The direct-current power from a direct-current power source 1 is converted into alternating-current power through an inverter circuit 2 and a motor 3 is thereby driven to rotationally drive a motor load 4. A peak current corresponding to the peak value of motor current is detected by a current detecting means 5, and a peak current target value at which a predetermined current phase is obtained is set by a motor load current estimated by computation. The output voltage orthe output frequency of the inverter circuit 2 is controlled by a controlling means 6 so that a detected motor peak current becomes equal to the set value. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention relates to a motor drive device, and more particularly to a motor control means by V / f control of a permanent magnet motor.

Conventionally, this type of motor drive device detects the inverter circuit DC current using a shunt resistor, estimates the motor effective current Iδ from the DC current, and performs V / f control so that the motor current becomes a predetermined value. (For example, see Patent Document 1).
JP 2005-218273 A

  However, the conventional motor drive device uses a filter circuit and an integration circuit (average circuit) to detect the DC average current from the shunt resistance voltage that changes according to the switching state of the inverter circuit, so that the circuit becomes complicated and the current detection accuracy When the value is increased, there is a problem that current detection responsiveness deteriorates. Furthermore, because the V / f control of the permanent magnet motor is a sensorless control method that does not calculate and estimate the rotor position, turbulence is likely to occur, and if control responsiveness is poor, turbulence is likely to occur, so detection accuracy and control responsiveness are traded. There was an issue that turned off.

  Further, since the conventional method is a method in which the motor effective current Iδ is estimated and calculated to change the drive frequency, there is a problem that when the load torque increases, the rotational speed decreases and the target rotational speed cannot be controlled.

  The present invention solves the above-described conventional problems, and is a sensorless sine wave drive by detecting a motor peak current and controlling it to a predetermined value. The load on the q axis and the motor applied voltage axis (δ axis) Sensorless control with excellent stability is possible by constant control of the angle, current detection accuracy, detection response, and control response are good, and motor load current is estimated and calculated, and motor peak current is calculated according to motor load current. Is set, and the objective is to perform the maximum efficiency operation by the advance angle control that advances the current from the q-axis.

  In order to solve the above-described conventional problems, the motor driving device of the present invention drives a permanent magnet motor with a sensorless sine wave by an inverter circuit that converts DC power into AC power, and generates a peak value of the motor current and a motor load current. A motor peak current set value set in accordance with the corresponding torque current or motor effective current is compared, and the inverter frequency or the inverter output voltage is controlled so that the target peak current is obtained by the deviation signal.

  The motor drive device of the present invention is a permanent magnet by controlling the inverter output voltage or the frequency so that the peak value of the motor current is substantially equal to or larger than the torque current corresponding to the load torque or the motor effective current. The motor is driven sensorlessly, coordinate conversion is not required, stable control can be performed without turbulence regardless of the saliency or non-saliency motor, and lead angle control and maximum efficiency operation are possible. Moreover, since control is possible without high-speed A / D conversion means and high-speed coordinate conversion means, the program capacity is reduced to 1/10 to 1/20 compared with an inexpensive processor and a conventional position estimation type sensorless drive, and simple control is possible. A motor drive device capable of sensorless sine wave drive can be realized by a program. In addition, since a simple and inexpensive current sensor can be used and the motor parameters and control parameters are very small, it is possible to realize a motor drive device that is excellent in robustness, inexpensive and highly reliable. Furthermore, since the program capacity is small and high-speed computation such as coordinate transformation is not required, the burden on the processor is reduced, and a single processor can control a plurality of motors simultaneously, and a one-processor multi-motor simultaneous drive system can be easily configured.

  The first invention is a DC power source, an inverter circuit that converts DC power of the DC power source into AC power, a permanent magnet motor driven by the inverter circuit, a load driven by the motor, and the motor current Current detecting means for detecting the current and a control means for controlling the inverter circuit according to an output signal of the current detecting means to drive the motor in a sine wave. The control means detects a peak current of the motor. Current detection means; load current detection means for detecting current according to torque current or effective current of the motor; peak current setting means for setting motor peak current based on an output signal of the load current detection means; Comparison means for comparing the output signal of the current detection means and the output signal of the peak current setting means, and the output of the inverter circuit The frequency control means for controlling the wave number, the voltage control means for controlling the inverter circuit output voltage by the output signal of the frequency control means, and the correction means for correcting the inverter output voltage or output frequency, the output of the comparison means A signal is added to the correction means to control the motor peak current, and maximum efficiency operation control or lead angle control is possible without coordinate conversion, and low cost is achieved by an inexpensive processor and simple current detection means. A highly reliable and highly efficient sensorless sine wave drive system can be realized.

  In the second invention, the set value of the peak current setting means is changed by the output signal of the load current detecting means in the first invention, and the motor current phase is controlled, and the torque current or the motor effective current is controlled. It is possible to control the current phase β from the q-axis by adjusting the ratio of the peak current setting value corresponding to, and the surface magnet motor (SPMSM) is controlled to be substantially in phase with the q-axis to provide an embedded magnet motor (IPMSM). ) Enables maximum efficiency operation by current advance control.

  According to a third aspect of the invention, the motor current phase is controlled by changing the voltage coefficient of the voltage control means for controlling the inverter circuit output voltage according to the output signal of the frequency control means in the first invention. It is possible to control the current phase β from the q axis by adjusting the ratio of the motor applied voltage to the motor induced voltage. In the surface magnet motor (SPMSM), the current phase β is controlled almost in phase with the q axis, and the embedded magnet motor (IPMSM). Then, maximum efficiency operation becomes possible by current advance angle control.

  According to a fourth aspect of the present invention, the correction means for correcting the inverter output voltage or the inverter output frequency according to the first aspect of the invention corrects the output voltage when starting the motor and corrects the output frequency after starting the motor. At startup, the torque is controlled by controlling to a predetermined motor current by voltage control, and after startup, the inverter output voltage adds a voltage at a predetermined ratio to the motor induced voltage, and the current phase from the q-axis is set to a predetermined value by frequency control. By controlling so that stable rotation can be controlled, it is possible to start even with a load having a large starting torque.

  According to a fifth aspect of the invention, the correction means for correcting the inverter output voltage or the inverter output frequency in the first aspect of the invention corrects the output voltage and the output frequency when starting the motor, and corrects the output frequency after starting the motor. Yes, at the time of motor start-up, torque is secured by controlling to a predetermined motor current by voltage control and frequency control, and after starting, the inverter output voltage adds a predetermined ratio of voltage to the motor induced voltage, and from the q-axis by frequency control Since stable rotation control is possible by controlling the current phase to be a predetermined value, it is possible to reduce turbulence even with a load having a large starting torque, and stable starting is possible.

  The sixth invention is such that the torque current or effective current of the motor is detected from the inverter circuit DC voltage and DC current in the first invention, and the inverter input power is obtained by the DC voltage detecting means and one shunt resistor. Since it can be easily detected and the motor peak current can also be detected from the shunt resistor, an optimum current phase can be realized by an inexpensive and simple current detecting means, and an inexpensive and highly reliable motor driving apparatus can be realized.

  In the seventh invention, the output signal of the load current detecting means in the first invention is applied to the peak current setting means via the delay means, and not only can the vibration of the load current feedback control be suppressed by the delay means. Since the transition from the constant peak current control at the start to the load current feedback control after the start can be smoothly controlled, the optimum load torque tracking control can be performed.

  The eighth invention is such that the voltage coefficient of the voltage control means for controlling the inverter circuit output voltage in the first invention is reduced in accordance with the inverter output frequency so as to reduce the motor applied voltage. This can be increased to reduce the effect of voltage saturation.

(Embodiment 1)
FIG. 1 shows a block diagram of a motor drive apparatus according to Embodiment 1 of the present invention.

  In FIG. 1, DC power is supplied from a DC power source 1 to a three-phase full bridge inverter circuit 2, converted into three-phase AC power by the inverter circuit 2, and the permanent magnet motor 3 is driven. The motor 3 drives a motor load 4 such as a fan, a pump, and a compressor. A current detecting means 5 comprising a shunt resistor is connected to the negative voltage side of the inverter circuit 2 and a direct current corresponding to the output current of the inverter circuit 2, that is, the peak current of the motor 3 is detected by detecting the direct current flowing through the inverter circuit 2. The peak current Ip and the inverter output, that is, the DC average current Idc corresponding to the motor output are detected.

  The control means 6 detects the current corresponding to the motor current peak value and the load torque from the output signal of the current detection means 5, and outputs the inverter output voltage and the output so that the motor current peak value Ip becomes the optimum value corresponding to the load torque. The frequency is controlled to drive the motor 3 with a sensorless sine wave. The peak current detection means 60 detects the motor peak current, that is, the shunt resistance current peak value corresponding to the maximum positive and negative current of the motor current, and is a simple peak current detection circuit comprising an operational amplifier, a diode, a resistor, and a capacitor. Can be configured. The load current detecting means 61 estimates and calculates the motor torque current Iq or the motor effective current component Iδ, and details of the estimation calculation will be described later. The output signal of the load current detecting means 61 is applied to the peak current setting means 62, and a current value substantially equal to the torque current Iq or the motor effective current component Iδ or a peak current setting signal ip * having a predetermined magnification is set. Then, the current peak signal ip and the peak current setting signal ip * are added to the comparison means 63 to output a deviation signal Δip.

  FIG. 2 shows a vector diagram of an embedded magnet motor (IPMSM).

  FIG. 2 shows the relationship between the dq coordinate seen from the rotor magnet and the γ-δ coordinate seen from the inverter output voltage, and the phase difference δ between the dq coordinate and the γ-δ coordinate is the load angle δ (δ = φ + β). Called. Iq and Id are current vectors obtained by vector decomposition of the motor current I into dq coordinates, and Iγ and Iδ are current vectors obtained by vector decomposition of the motor current I into γ-δ coordinates. The phase β is the current phase from the q axis, and the phase φ indicates the power factor angle with respect to the motor voltage Va. Since IPMSM uses reluctance torque, it is necessary to control the advance angle. Generally, when the current phase is advanced by 30 degrees with respect to the q axis, the maximum efficiency operation is assumed. In order to increase the advance angle β, it is necessary to increase the load angle δ. In the output current constant method according to the present invention, the inverter output voltage is set to be substantially equal to or smaller than the induced voltage, and the motor current I is torqued. The load angle δ can be increased by setting it larger than the current Iq (= Icos β) or the effective current Iδ (= Icos φ). That is, if the torque current Iq is known, it can be understood that the motor current I may be set to a value Ip = Iq / cosβ obtained by gradually reducing the torque current Iq by cosβ in order to set the predetermined current phase β. Further, if the motor applied voltage Va is set to a value substantially equal to the motor induced voltage Em or lower than the motor induced voltage Em, the advance angle β can be increased.

  FIG. 3 shows a PWM signal and a shunt resistance voltage waveform during two-phase modulation.

  In FIG. 3, vc is a triangular wave carrier signal, vu and vv are u-phase and v-phase modulation signals, up, vp and wp are upper arm control signals for each UVW phase, and Vsh is a shunt resistance voltage waveform. Since the w-phase lower arm transistor is forced to conduct, the w-phase modulation signal is not shown.

  The pattern in which the motor peak current appears in the two-phase modulation is a section (t0 to t2, t4 to t5) where only the upper arm of one phase is turned on as shown in FIG. 3, or the upper arm of the two phases is turned on. It appears in the section (t5 to t7). Unlike the three-phase modulation, the two-phase modulation is PWM-controlled only for the two phases, so that the section where the peak current appears is widened, so that the peak current can be easily detected.

  FIG. 4 shows a phase in which the two modulation signal waveforms of each phase of UVW and each phase current appear in the shunt resistor. The interval from 0 to 1 / 3π is the W phase current Iw and the V phase current Iv, the interval from 1 / 3π to 2 / 3π is the U phase current Iu and the V phase current Iv, and the interval from 2 / 3π to π is the U phase. A phase current Iu, a W-phase current Iw, and each phase current appear sequentially. As shown by the arrows in the figure, the section where the current peak value appears is delayed by 30 degrees from the phase at which the voltage from the neutral point of each phase reaches its peak. The peak value of each phase current appears. That is, the interval 0 to 1 / 3π is the peak value of Iw, the interval 1 / 3π to 2 / 3π is the peak value of Iv, the interval 2 / 3π to π is the peak value of Iu, and the peak value is 6 times in one period. Appears. When the current phase is delayed by 30 degrees from the voltage phase, it is easy to detect the peak current. However, when the current phase is delayed by 60 degrees, the pulse width is narrowed, indicating that current detection becomes difficult. However, in the case of IPMSM, since the power factor angle φ between the voltage phase and the current phase is small, the detection of the current peak value is easy, and in the case of SPMSM, the advance angle β is set to be slightly or almost zero. The case where the power factor angle φ becomes large is very rare, and practically no problem occurs.

  The one-shunt current detection method and the method in which the peak current detection means is constituted by the voltage amplifier and the peak hold circuit have the feature that not only the hardware configuration is simple but also the processor software is less burdensome and simple. Further, the A / D conversion timing for detecting the current may be the valley or peak (t0, t3, t6 in FIG. 3) of the carrier signal in which all the switching transistors of the inverter circuit are on or off, and the current detection is simple. In addition, it has a strong feature against noise.

  In the conventional method of reproducing a sine wave current by the single shunt current detection method, when the carrier frequency is high or the modulation degree becomes large, a current undetectable region appears, so an instantaneous current corresponding to each phase. The three-shunt current detection method is superior when detecting. However, since the present invention does not require coordinate conversion, it is not necessary to reproduce the sine wave current, and it is only necessary to detect the peak value of the motor current. Current detection is easy. Further, when the inverter circuit PWM control is two-phase modulation, peak current detection is facilitated. Of course, there is no problem with the three-shunt current detection method, but it is necessary to increase the A / D conversion functions of the current circuit and the processor, which increases the price.

  The frequency control means 64 sets the output frequency of the inverter circuit 2, and adds a frequency signal ω to the voltage control means 65 to generate a voltage Vf proportional to the frequency. Since the voltage Vf generates a voltage corresponding to the motor induced voltage Em, Vf = ω × Ke × kr, and Ke is an induced voltage constant. kr is set to a value of 0.8 to 1.2. Normally, when it is smaller than 1, the current advance angle β is increased, and when it is larger than 1, the current phase β approaches zero. The correction means 66 is composed of a voltage correction means 66a and a frequency correction means 66b, and each is controlled by a current deviation signal Δip. The output signal Vδ of the voltage correction means 66 is added to the sine wave PWM control means 67, and the frequency correction means. The output signal ω1 of 66b is applied to the phase signal generating means 68. The phase signal generating means 68 integrates the angular frequency ω1, generates an electrical angle signal θ, and adds the electrical angle signal θ to the sine wave PWM control means 67.

  The sine wave PWM control means 67 performs PWM control of the inverter circuit 2 in accordance with the voltage signal Vδ and the DC power supply voltage Edc of the inverter circuit 2 to drive the motor 3 in a sine wave, and normally uses two-phase modulation. When the two-phase modulation is used, the motor interphase voltage can be made higher than that of the three-phase modulation, the switching loss of the inverter circuit 2 can be reduced, and the peak current detection accuracy of the motor 3 can be improved. A sinusoidal PWM control means 65 so that a voltage corresponding to a control voltage Vδ obtained by adding a correction voltage ΔVδ and a starting voltage Vs to a voltage Vf obtained by multiplying the output signal ω of the frequency setting means 64 by an induced voltage constant Ke is applied to the motor 3. To control. Vδ is obtained from Equation 1.

  Here, kr is a voltage coefficient for lead angle control as described above, and is adjusted to a value of 0.8 to 1.2. Further, when the motor rotates at high speed, the voltage coefficient kr is decreased as the inverter frequency is increased, so that the motor applied voltage can be reduced and the lead angle can be controlled, and the influence of voltage saturation can be reduced.

  The motor phase control signal is obtained from Equation 2 from the voltage control signal Vδ and the electrical angle θ.

  The voltage correction means 66a performs correction control by changing and correcting the inverter output voltage signal Vδ in proportion to the current deviation signal Δip, and obtains the correction voltage ΔVδ according to Equation 3 and performs PI control. When the peak current ip increases from the set value ip * (because Δip becomes negative), the output voltage is decreased. When the peak current ip decreases from the set value ip * (since Δip becomes positive), the output voltage is reversed. Increase.

  In Equation 3, Kp is a proportionality constant and Ki is an integration constant.

  When the motor is SPMSM, the motor torque current Iq is a value obtained by dividing the inverter circuit input power (Pin = Edc × Idc) by the motor induced voltage (Em = Ke × ω).

  In Equation 4, kη is a constant corresponding to efficiency. The control signal Vf is a signal corresponding to the induced voltage Em.

  Further, the effective current Iδ can be obtained from Equation 5.

  In Equation 5, Va is a motor applied voltage, and kd is a correction coefficient corresponding to efficiency. The motor applied voltage Va can be obtained from the modulation degree obtained from the inverter output voltage control signal Vδ and the inverter circuit DC power supply voltage Edc. Therefore, it can be seen from Equation 5 that Iδ is obtained when the inverter circuit DC average current Idc is gradually decreased by the modulation degree. As shown in the vector diagram of FIG. 2, the torque current Iq and the effective current Iδ are very close to each other. However, since Iq is affected by variations in the induced voltage constant Ke, the detection accuracy is Iδ. However, the peak current set value ip * becomes more stable when set larger than the actual torque current. Therefore, it is more stable to use Iq obtained from Equation 4.

  The frequency correction means 66b corrects the inverter output frequency ω1 by the deviation signal Δip between the peak current signal ip and the peak current setting signal ip *, and is corrected by the relational expression expressed by Equation 6.

  That is, the frequency correction signal Δω is proportionally controlled by the deviation signal Δip, and the proportionality constant is Kf. When the peak current signal ip increases with respect to the peak current setting signal ip *, the deviation signal Δip becomes negative and the inverter output frequency becomes lower. Conversely, when the peak current signal ip is lower than the peak current setting signal ip *, the deviation signal Δip becomes positive and the inverter output frequency becomes high. When the inverter frequency is increased, the γ-δ axis is more advanced than the dq axis and the current increases. Therefore, Equation 6 can control the current so that the inverter output frequency ω1 is substantially equal to the frequency target value ω. Recognize. In other words, Equation 6 is considered to control the phase difference δ (load angle) between the γ-δ axis and the dq axis to be constant. Unlike the voltage control, the frequency control method can be stabilized by proportional control. Therefore, there is no response time delay, and the frequency control method can be stabilized by the proportionality constant Kf.

  Here, P is the number of magnet poles, J is the moment of inertia of the motor 3 and the motor load 4, and L is the inductance of the motor coil. As shown in Equation 7, the method of the present invention for controlling the frequency of the peak current is superior to the conventional method of controlling the frequency of the effective current.

  As described above, the first embodiment describes the basic method of the present invention. The inverter circuit DC current peak value Ip is detected and the inverter circuit is set so that the peak value Ip becomes the target set value Ip *. The output voltage or the output frequency is corrected. The inverter circuit DC current average value and the motor induced voltage, or the torque current Iq obtained from the motor applied voltage, or a value larger than Iq is set as the target set value Ip *. The inverter output voltage or output frequency is corrected by the error signal Δip between the peak current Ip and the peak current set value Ip *, and the peak current Ip is controlled to be equal to the current set value Ip *. When controlled in this way, the motor current is controlled according to the load torque, and the advance angle β from the q axis can be controlled, so that the maximum efficiency operation control is always performed regardless of the load fluctuation.

  Although the voltage / frequency control is shown in the embodiment, when the moment of inertia is small, the motor current I is controlled by directly controlling the load angle δ by correcting the inverter frequency with the error signal Δip with a constant inverter output voltage. can do. With a rotational speed reduction load such as a fan or a pump load, the torque changes with the square of the rotational speed, so the braking coefficient is large, and stabilization control is possible only with voltage control. In the case of a constant torque load or the like, turbulence occurs in voltage control. However, if frequency control is performed, the braking coefficient increases, so that control stability is excellent and turbulence hardly occurs.

  As shown in Equations 4 and 5, in order to obtain the torque current Iq or the effective current Iδ, it is necessary to detect not only the inverter circuit DC average current Idc but also the DC power supply voltage Edc. However, in the case of a small motor, the DC power supply voltage Edc is constant and is known in advance, so there is no need to detect it, and it is omitted because only the torque estimation calculation coefficient of the processor needs to be changed.

  Needless to say, a DC power supply voltage detecting means is required when a DC power supply is constructed by rectifying a home AC power supply or when the fluctuation of the DC power supply voltage is large. Further, in motor control by V / f control, when the DC power supply voltage fluctuation is large, it is necessary to detect the DC power supply voltage Edc to control the modulation degree, and the DC power supply voltage detecting means is indispensable. Omitted.

  It is obvious that the method for estimating the torque current Iq or the effective current Iδ can be obtained by calculation from the motor constant, the motor peak current Ip, and the motor applied voltage Va, other than those shown in the first embodiment. The method shown in the first embodiment is simple in calculation, can be executed by an 8-bit or 16-bit processor, and can realize the lowest price sensorless sine wave drive.

(Embodiment 2)
FIG. 5 shows a detailed block diagram of the control means of the motor drive apparatus in Embodiment 2 of the present invention.

  Hereinafter, a detailed block diagram of the control means 6 will be described.

  The load current detecting means 61 is composed of a DC current detecting means 61a for detecting an average value of inverter DC current and a load current estimating means 61b, and a delay means 62a and a current setting means for constituting a peak current setting means 62 for the load current estimation signal. Add to 62b. The direct current detection means 61a detects a direct current component of the inverter circuit bus current of the shunt resistor 5 and is composed of a low-pass filter. The load current estimation unit 61b includes a DC voltage detection unit 611 that detects the DC voltage Edc of the inverter circuit 2, and a motor power calculation unit 612 that calculates the motor input Pin from the DC current average value signal idc detected by the DC current detection unit 61a. Torque calculating means 613 for gradually calculating the motor power signal pin by a predetermined value and the output signal Vf of the voltage control means 65 proportional to the motor induced voltage multiplied by a predetermined calculation coefficient to output the torque calculating means A motor induced voltage estimation unit 614 that outputs to 613 is configured. The load current estimating means 61 b performs the calculation shown in Equation 4 to estimate the torque current Iq, and adds a signal to the delay means 62 a of the peak current setting means 62. The delay means 62a is composed of a first-order lag filter element or a first-order integration means, prevents abrupt fluctuation of the peak setting signal ip *, and suppresses vibration of the current control system. In order to increase the response to a sudden change in load torque, the response of the current setting value ip * is increased when the current is increased, and the response of the current setting value ip * is decreased when the current is decreased. There is an inhibitory effect. The output signal ip * of the current setting means 62b is applied to the comparison means 63 and compared with the output signal ip of the peak current detection means 61. Further, the advance signal β is controlled by adding the output signal ip * of the current setting means 62b to the voltage control means 65 or the motor induced voltage estimation unit 614 and controlling the applied voltage Va or the voltage coefficient according to the current value. be able to. That is, the voltage saturation occurs as the output current increases, so that control for increasing the advance angle is possible in order to prevent voltage saturation.

  When the motor is an IPMSM, the target set value ip * set by the load current estimating means 61b has a small power factor angle φ as can be seen from the vector diagram of FIG. It is better to set a larger value. However, in the case of SPMSM, since there is almost no need for the lead angle control, the torque current Iq is estimated, and the torque current Iq is made approximately equal to the peak current set value ip *, thereby enabling the maximum efficiency operation. In high-speed rotation, the effect of voltage saturation can be prevented by controlling the lead angle even in SPMSM. Therefore, by changing the voltage coefficient kr so that the motor applied voltage Va decreases according to the frequency setting value ω, Can be controlled.

  The voltage correction unit 66a includes a PI control unit 660a that performs feedback control of the error signal Δip, a voltage limiter 661a, and a voltage addition unit 662a. The voltage adding unit 662 a adds the correction voltage ΔVδ to the voltage control signal Vf corresponding to the motor induced voltage, and adds the control signal Vδ to the sine wave PWM control means 67. The voltage limiter 661a limits the voltage control range. When the peak current Ip is controlled by the voltage priority control, the voltage limiter 661a is increased, and when the control is performed by the frequency priority control, the control range is decreased.

  The output signal ω and the error signal Δip of the frequency setting unit 64 are applied to the frequency correction unit 66b. The frequency correction unit 66b includes a proportional unit 660b that calculates a signal proportional to the error signal Δip, an addition unit 661b for the frequency setting signal ω, a frequency proportional calculation unit 662b for the frequency setting signal ω, and output signals (Kf · The signal Δω0 from the multiplication unit 663b that calculates the product of Δip) and the output signal (K · ω) of the frequency proportional calculation unit 662b is added to the addition unit 661b via the frequency limiter 664b. The proportional constant Kf of the proportional unit 660b is set to about 1 to 50, and the proportional constant K of the frequency proportional calculation unit 662b is 1 in which the output signal Δω0 from the multiplication unit 663b is almost zero at start-up, and becomes Kf · Δip in a steady state. Choose a smaller value. The output signal ω1 of the frequency correction unit 66b is applied to the phase signal generation unit 68 and integrated to generate a phase signal θ. The phase signal θ is added to the sine wave generation unit 67a of the sine wave PWM control unit 67 and is a three-phase sine wave signal. vu, vv, vw are generated, and three-phase PWM signals up, un, vp, vn, wp, wn are generated via the PWM control means 67b. As shown in FIG. 3, the PWM control unit 67b includes a carrier signal generation unit, a signal comparison unit, a dead time insertion unit (all not shown), and the like, but the details are omitted.

  The frequency correction means 66b not only prevents the turbulence when the load fluctuates, but also sets the control voltage substantially constant in the case of the lead angle control, and sets the inverter output voltage substantially equal to the motor induced voltage. By controlling the peak current Ip by frequency control, a frequency priority control system is achieved.

  FIG. 6 shows a method for controlling the inverter control voltage, frequency, and frequency gain when the motor is started.

  A so-called V / f control is performed in which the inverter frequency ω and the applied voltage corresponding to the frequency are linearly increased from the start of operation to the target rotational speed, and the frequency gain Kf is increased in proportion to the inverter frequency ω. If the motor peak current set value ip * is kept constant and the start control is performed by voltage control or voltage / frequency control, the start becomes easy.

  In the case of a fan or a pump load, high-efficiency operation control can be performed by changing the peak current set value ip * according to the drive frequency. In addition, since the reluctance torque increases when the current of the saliency motor is advanced from the q-axis, the set current ip * is preferably increased in proportion to the frequency in order to control the advance angle. Further, as described above, the advance angle β can be increased by decreasing the voltage coefficient kr.

  Further, the startup time ts until the target rotational speed is reached can be reduced by changing the start time ts according to the moment of inertia of the load. That is, as the moment of inertia increases, the turbulence can be reduced by increasing the startup time ts.

  By changing the frequency correction gain Kf in accordance with the drive frequency, it is possible to reduce fluctuations in the motor speed at the time of low start-up. Therefore, the frequency control gain is set low at start-up, and constant current control is performed by the voltage correction means 66a. Do. When the frequency gain Kf is increased in proportion to the inverter frequency ω (Kfa), voltage / frequency control is performed. When the frequency gain Kf is set to zero until the start time t1 (Kfb), voltage control is performed. The current phase β can be controlled by fixing the control signal ΔVδ of the voltage correction means 66a at the starting time t1 and performing constant current control by the frequency correction means 66b.

  FIG. 7 is a control flowchart of the motor driving apparatus according to the second embodiment.

  The program starts from step 100, and after various initial settings are made in step 101, the process proceeds to step 102 to set the inverter frequency ω, and then proceeds to step 103 to detect the motor peak current Ip and the load current Iq. Alternatively, Iδ is estimated and calculated. Next, the routine proceeds to step 104 where the peak current target value ip * is set, and then the routine proceeds to step 105 where the frequency gain Kf is set according to the start-up time as shown in FIG. Next, the routine proceeds to step 106, where the deviation signal Δip between the peak current set value ip * and the detection signal ip is calculated, then the routine proceeds to step 107, where the inverter output frequency ω1 is controlled, and the motor current peak value is controlled to a predetermined value. . At the time of start-up, the peak current for start-up is set at step 101, and after start-up, control is performed so that the peak current according to the load current is obtained. Next, the routine proceeds to step 108, where it is determined whether the startup time has passed the predetermined time t1, and if it is within the predetermined time t1, the routine proceeds to step 109, step 110 to perform peak current control by voltage control, and then to step 111. Go ahead and execute sine wave PWM control. When the predetermined time t1 has elapsed, the routine jumps to step 112, the control voltage ΔVδ is fixed, and the routine proceeds to step 113, where the motor applied voltage is set to a value corresponding to the motor drive frequency, voltage control is not performed, and the frequency is determined by the deviation signal Δip. Control the current. After step 111, the routine proceeds to step 114, where it is determined whether or not the operation has been completed. If the operation has been completed, the routine proceeds to step 115, where the motor drive control is terminated.

  As described above, the current lead angle β can be controlled by the ratio (1 / cos β) of the peak current set value ip * to the load current (Iq, Iδ). Furthermore, since the control can also be performed from the ratio of the motor induced voltage Ke · ω and the motor applied voltage Va (that is, the voltage coefficient kr), by controlling the peak current set value ip * or the motor applied voltage Va according to the inverter frequency ω. The advance angle β can be controlled to a predetermined value, voltage saturation in high speed operation is reduced, and maximum efficiency operation control is possible.

  As described above, according to the present invention, the inverter output voltage or the output frequency is controlled so that the motor peak current becomes a predetermined value, and the peak current set value is set according to the motor load current Iq or Iδ. By changing the ratio of the peak current set value to the load current, the current lead angle can be controlled and maximum efficiency operation becomes possible. In particular, the load angle control that makes the phase relationship between the γ-δ axis and the dq axis constant by feeding back the deviation signal of the peak current to the frequency enables stable control regardless of position sensorlessness.

  Further, even if the motor load fluctuates greatly from no load to the rated load, current control according to the load torque is possible. Therefore, high efficiency control is possible because the output current is reduced even at low output. In addition, since coordinate conversion is not required and peak hold type 1 shunt method can be realized, sensorless sine wave drive can be easily realized by a low-cost processor without requiring high-speed calculation and high-speed A / D converter. Since it can be realized with a simple and low-cost current sensor, the number of parts is small, and low-cost, high-reliability, high-performance sensorless sine wave drive becomes possible.

  In addition, the conventional sensorless sine wave drive system has many motor parameters and control parameters, and there are problems such as motor variation, motor temperature characteristics, parameter tuning, etc. However, the present invention is robust because there are very few motor parameters and control parameters. It has excellent characteristics and simplifies gain tuning on the motor control program, facilitating processor software development.

  In addition, since the rotational speed control is based on the open loop method, the rotational speed control is easy and the rotational speed fluctuation is small, so there is a feature that the rotational speed fluctuation is small against load torque fluctuations and power supply voltage fluctuations. Control can be realized.

  As described above, the motor drive device of the present invention is a sensorless sine wave drive of a permanent magnet motor by an inverter circuit that converts DC power to AC power, and feeds back a deviation signal of motor peak current to the frequency. Since the phase relationship between the −δ axis and the dq axis can be made constant and position sensorless control can be easily performed, it can be applied to most motor drive devices that drive a permanent magnet motor, and the rotational speed fluctuation Spindle motors, fan motors, pump motors, dishwasher washing pump drive devices, washing machine washing / dehydration motor drive devices, vacuum cleaner motor drive devices, fan motor drive devices such as ventilation fans and combustors, air conditioners It can be applied to compressor motor drive devices for refrigerators and refrigerators. Furthermore, since the control program is simple and the burden on the processor is small, it can be applied to a multi-motor simultaneous drive system such as a heat pump washer / dryer or an air conditioner. In addition, since it can be realized without a processor, a sensorless sine wave drive 1-chip motor control integrated circuit, an intelligent power module (IPM) in which a power semiconductor and a drive circuit are integrated, or a sensorless sine wave drive 1-chip inverter is easy. Can be realized.

1 is a block diagram of a motor drive device according to Embodiment 1 of the present invention. Control vector diagram of the motor drive device Figure showing the same shunt resistance voltage waveform and current detection timing Diagram showing current detection timing during two-phase modulation The block diagram of the control means of the motor drive unit in Embodiment 2 of this invention The figure which shows the starting control method of the motor drive device Control flowchart of the motor drive device

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 DC power supply 2 Inverter circuit 3 Motor 4 Motor load 5 Current detection means 6 Control means 60 Peak current detection means 61 Load current detection means 62 Peak current setting means 63 Comparison means 64 Frequency control means 65 Voltage control means 66 Correction means

Claims (8)

DC power source, inverter circuit for converting DC power of the DC power source into AC power, a permanent magnet motor driven by the inverter circuit, a load driven by the motor, and current detection means for detecting the motor current And a control means for controlling the inverter circuit by an output signal of the current detection means to drive the motor in a sine wave, the control means including a peak current detection means for detecting a current peak value of the motor, Load current detecting means for detecting a current corresponding to a torque current of the motor or an effective current, a peak current setting means for setting a motor peak current by an output signal of the load current detecting means, and an output signal of the peak current detecting means Comparing means for comparing the output signal of the peak current setting means and the output frequency of the inverter circuit A frequency control unit; a voltage control unit that controls the inverter circuit output voltage based on an output signal of the frequency control unit; and a correction unit that corrects the inverter output voltage or output frequency, and the output signal of the comparison unit is corrected. A motor driving apparatus for controlling the motor peak current in addition to the means. 2. The motor driving apparatus according to claim 1, wherein the setting value of the peak current setting means is changed by the output signal of the load current detection means to control the motor current phase. 2. The motor driving apparatus according to claim 1, wherein the motor current phase is controlled by changing a voltage coefficient of the voltage control means for controlling the inverter circuit output voltage by an output signal of the frequency control means. 2. The motor driving apparatus according to claim 1, wherein the correcting means for correcting the inverter output voltage or the inverter output frequency corrects the output voltage when the motor is started and corrects the output frequency after starting the motor. 2. The motor driving apparatus according to claim 1, wherein the correcting means for correcting the inverter output voltage or the inverter output frequency corrects the output voltage and the output frequency when starting the motor and corrects the output frequency after starting the motor. 2. The motor drive device according to claim 1, wherein the torque current or effective current of the motor is detected from the inverter circuit DC voltage and DC current. 2. A motor driving apparatus according to claim 1, wherein the output signal of the load current detecting means is applied to the peak current setting means via the delay means. 2. The motor drive apparatus according to claim 1, wherein the voltage coefficient of the voltage control means for controlling the inverter circuit output voltage is reduced in accordance with the inverter output frequency to reduce the motor applied voltage.