JP4983393B2 - Motor drive device - Google Patents

Description

  The present invention relates to a motor drive device, and more particularly to a motor control means by V / f control of a permanent magnet motor.

Conventionally, this type of motor drive device detects an inverter circuit output current, performs coordinate conversion of the motor current to a motor applied voltage coordinate axis, and performs V / f control so that the motor current after the coordinate conversion becomes a predetermined value. (For example, see Patent Document 1).
JP 2000-236694 A

  However, since the conventional motor drive device instantaneously detects the motor current corresponding to the motor applied voltage phase, and converts the coordinate to the motor applied voltage axis to convert it into a direct current component, the high speed A / D conversion means or the high speed current Detection means and high-speed coordinate conversion means are required, and there is a problem that a processor such as a microcomputer for controlling the inverter circuit is high-speed and high-priced. Further, since the control is performed by the effective current vector after the coordinate conversion, there is a problem that the effective current setting is complicated in the control of a fan or a pump in which no load or the load changes according to the rotation speed.

  Further, when the lead angle control is required at a high speed rotation such as a saliency motor (IPMSM), there is a problem that the lead angle control becomes difficult.

  The present invention solves the above-described conventional problems, and is basically capable of sensorless sine wave drive with an inexpensive current sensor without coordinate conversion, and with a torque that varies depending on the load with large torque fluctuations and the rotational speed. Sine wave drive by V / f control is possible even for saliency motors that require variable lead and lead angle control, and a motor drive device that can drive sensorless sine waves with an inexpensive, low-speed processor and simple control program It is intended to do.

In order to solve the above-described conventional problems, a motor driving device of the present invention includes a DC power supply, an inverter circuit that converts DC power of the DC power supply into AC power, a permanent magnet synchronous motor driven by the inverter circuit, A fan or pump load driven by the motor, current detection means for detecting a peak value of the inverter circuit DC current, and controlling the inverter circuit by an output signal of the current detection means to drive the motor in a sine wave The control means comprises: frequency setting means for setting an output frequency of the inverter circuit; voltage control means for controlling the inverter circuit output voltage according to an output signal of the frequency setting means; and current of the motor Current setting means for setting a peak value; output signal of the current detection means; and output of the current setting means A current comparing means for comparing the item, and the frequency correcting means for correcting the output frequency by the output signal of the current comparator means, those made of V / f control means for controlling the motor voltage in accordance with the frequency.

  The motor driving device of the present invention detects the motor current corresponding to the peak value of the motor current or the rotating magnetic field, and controls the inverter circuit output frequency and voltage so as to be a set value. Regardless of the motor, stable control can be performed without turbulence, and also stable control can be easily performed in advance angle control, and control can be performed without high-speed A / D conversion means and high-speed coordinate conversion means. A motor drive device capable of sensorless sine wave drive can be realized by a control program. In addition, a simple and inexpensive current sensor can be used, and the control program can be simplified, so that an inexpensive and highly reliable motor drive device can be realized. Furthermore, since the processor is lightly loaded, a single processor can control multiple motors simultaneously. In addition, a single processor multiple motor simultaneous drive system can be easily configured.

A first invention includes a DC power source, an inverter circuit that converts DC power of the DC power source into AC power, a permanent magnet synchronous motor driven by the inverter circuit, and a fan or pump load driven by the motor. The inverter circuit DC current is detected by a current detection means, and the inverter circuit is controlled by an output signal of the current detection means to control the inverter circuit to drive the motor in a sine wave. Frequency setting means for setting an output frequency of the inverter circuit, voltage control means for controlling the inverter circuit output voltage by an output signal of the frequency setting means, current setting means for setting a current peak value of the motor, and the current Current comparison means for comparing the output signal of the detection means and the output signal of the current setting means; and the current comparison means And frequency correction means for correcting the output frequency by the output signal of, which consists of V / f control means for controlling the motor voltage in accordance with the frequency, can be prevented hunting in V / f control, no load from maximum load The sensorless sine wave motor driving apparatus can be realized with a simple and simple current detection means and motor control program, and with a low cost and high reliability .

(Embodiment 1)
FIG. 1 is a block diagram of a motor drive device according to Embodiment 1 of the present invention.

  In FIG. 1, a DC power source 2 is constructed by applying AC power from an AC power source 1 to a DC power source circuit composed of a rectifier circuit, and DC power is converted into three-phase AC power by a three-phase full-bridge inverter circuit 3. 4 is driven. In the DC power supply 2, capacitors 21a and 21b are connected in series to the DC output terminal of the full-wave rectifier circuit 20, and the connection point of the capacitors 21a and 21b is connected to one terminal of the AC power supply input to constitute a voltage doubler rectifier circuit. The voltage applied to the inverter circuit 3 is increased to reduce the current and reduce the inverter circuit and motor loss. The motor 4 drives a motor load 5 such as a compressor of an air conditioner, a dewatering drum of a washing machine, or a fan / pump. By connecting the current detection means 6 to the negative voltage side of the inverter circuit 3 and detecting the current flowing through the inverter circuit 3, it corresponds to the output current of the inverter circuit 3, that is, the peak current Ip of the motor 4 or the rotating magnetic field. Drive current is detected.

  The current detection means 6 is a so-called one-shunt current detection method, and is a shunt resistor 60 commonly connected to the emitter terminal side of the lower arm transistor of the inverter circuit 3 and a current detection circuit that detects a current flowing through the shunt resistor 60. 61. The current detection circuit 61 can be composed of a signal level amplification conversion circuit and a peak current detection circuit for detecting current by an A / D conversion circuit built in a processor such as a microcomputer. There are a case where the peak current detection circuit is constituted by hardware such as a diode, a capacitor, and a resistor, and a case where the magnitude of the A / D converted input signal is determined by software, and the details are omitted.

  In the 1-shunt current detection method, when the carrier frequency is high or the modulation degree becomes large, a current undetectable region appears. Therefore, when detecting an instantaneous current corresponding to each phase, 3-shunt current detection Although the method is superior, in the present invention, since the peak value of the motor current is detected, the circuit configuration is simpler and the current detection is easier in the one-shunt current detection method. Further, when the inverter circuit PWM control is two-phase modulation, peak current detection is facilitated. Of course, there is no problem with the three-shunt current detection method.

  The control means 7 controls the output frequency and output voltage of the inverter circuit 3 so that the peak current Ip of the motor 4 or the drive current corresponding to the rotating magnetic field becomes a set value, and sets the inverter circuit output frequency. Frequency setting means 70, voltage control means 71 for controlling the inverter output voltage ratio according to the output signal ω of frequency setting means 70, current setting means 72 for setting the motor current peak value Ip, and output of the current setting means 72 The current comparison means 73 that compares the setting signal ips with the output signal ip of the current detection means 6, and the output signal ω of the frequency setting means 70 is corrected according to the output signal Δip of the current comparison means 73, or the inverter circuit output voltage phase is changed. The inverter circuit 3 is subjected to PWM control in a sine wave form in accordance with the frequency correction means 74 for correction and the output signal Vδ of the voltage control means 71. A converter control unit 75, and the phase signal generation means 76 for applying the inverter control unit 75 generates a phase signal theta integrates the output angular frequency signal ω1 frequency correction means 74 the phase signal theta.

  Further, the current error signal Δip is applied to the voltage correction means 77, and after proportional integration, the voltage correction signal ΔVδ is added to the voltage control means 71, and the voltage is controlled and the constant current is controlled when the motor is started. The inverter control means 75 applies PWM control to the inverter circuit 3 in accordance with the voltage signal Vδ and the DC bus voltage of the inverter circuit 3 and applies a sinusoidal voltage to the motor 4, and usually uses two-phase modulation. When the two-phase modulation is used, the motor interphase voltage can be made higher than the three-phase modulation, the switching loss of the inverter circuit 3 can be reduced, and the peak current detection accuracy of the motor 4 can be improved. The inverter control means 75 is controlled so that a voltage corresponding to the control voltage Vδ obtained by adding the correction voltage ΔVδ and the starting voltage Vs to the voltage obtained by multiplying the inverter circuit drive frequency ω by the induced voltage constant Ke is applied to the motor 4. Vδ is obtained from Equation 1.

  The motor phase control signal is obtained from Equation 2 from the voltage control signal Vδ and the electrical angle θ.

  The frequency correction means 74 corrects and changes the inverter frequency in proportion to the current error signal Δip to perform stabilization control, and performs turbulence prevention and advance angle control. That is, according to Equation 3, when the motor current ip increases from the set value ips, Δip becomes negative. Therefore, when the drive frequency is decreased and when the motor current ip decreases below the set value ips, Δip becomes positive. Increase the drive frequency.

  The frequency correcting unit 74 includes a proportional unit 74a and an adding unit 74b, which will be described in detail later. The proportional unit 74a calculates a product of the current error signal Δip and the proportionality constant Kf, and the adding unit 74b outputs the output of the frequency setting unit 70. The inverter ω1 is obtained by adding the output signal of the proportional section 74a to the signal ω. Kf shown in Equation 3 is obtained from the product (Kf = ω · k) of the output signal ω of the frequency setting means 70 and the proportionality constant k. That is, it means that the frequency control gain Kf increases in proportion to the set frequency ω. At a low frequency, the proportionality constant Kf is lowered to reduce frequency change or phase change, and at a high frequency, the proportionality constant Kf is increased. Control is performed so that the motor peak current becomes the set value Ips.

  FIG. 2 is a control vector diagram of the non-saliency motor (SPMSM) showing the present invention, and shows the relationship between the rotor magnet axis coordinates (dq coordinates) of the motor and the motor applied voltage axis coordinates (γ-δ coordinates). ing. Since the motor applied voltage axis coordinate (γ-δ coordinate) is advanced by a load angle δ than the dq coordinate, the motor applied voltage (= inverter circuit output voltage) Va is equal to the δ axis voltage, and only the δ axis is controlled. Since Va = Vδ and Vγ = 0, the reverse coordinate transformation is unnecessary and the calculation can be performed from Equation 2. The motor induced voltage Em is on the q axis, and the vector of the motor current I is set to be approximately equal to or larger than the q axis current Iq at the rated load. The motor current vector I is displayed with a phase β advance from the q axis, and the motor applied voltage Va is set to be substantially equal to the motor induced voltage Vm. The phase φ indicates the phase (power factor angle) between the motor applied voltage Va and the current I. If the load torque is constant, Iq is constant, so the motor coil voltage vector ωLI is on the one-dot chain line, and the vector relationship (I = Iq) parallel to the q axis is the maximum efficiency operation (β = 0). Assuming that the motor applied voltage is Va0, a lead angle is obtained when the motor applied voltage Va is smaller than Va0, and a delay angle is produced when the motor applied voltage Va is larger than Va0.

In the present invention, the frequency is controlled so that the peak value Ip of the motor current I becomes a predetermined value, the angular velocity ω1 of the motor applied voltage coordinate (γ-δ coordinate) is controlled, and the current advance angle β from the q axis is controlled. As a result, it means that the load angle δ is controlled. Further, since the magnetic flux Ψ of the rotating magnetic field is a product of the inductance L and the current I, that is, Ψ = L · I, controlling the current I means controlling the rotating magnetic flux Ψ constant. Further, not only the method of controlling the motor phase currents Iu, Iv, the peak value of Iw, is evident from the this to be also constant control the effective value same.

  In the case of Iδ constant control in which the δ-axis current (effective current) Iδ is controlled to a predetermined value, the δ-axis current Iδ varies due to load fluctuations. Therefore, it is necessary to change the Iδ setting value according to the load state. When the motor current (or motor peak current Ip) is controlled to be constant, the load angle δ and the phase φ automatically change according to the load, so that there is a feature that it is not necessary to change the current value.

  In the conventional constant control of the effective current Iδ or the constant control of the reactive current Iγ, the lead angle control is difficult. However, in the present invention for controlling the current vector I or the current peak value Ip, the torque fluctuation and the load angle fluctuation are controlled. Since the influence directly appears as a fluctuation of the current vector I or the peak current Ip, it is excellent in detecting a load fluctuation, and has an advantage of being excellent in stabilization against the load fluctuation. In particular, in the advance angle control in which the current vector I is advanced from the q axis, the I control becomes more significant than the Iδ control and the Iγ control, and the torque fluctuation becomes more significant, and the coordinate conversion is unnecessary, which is advantageous as the current control target.

  If the current setting value ips is optimally set according to the load, high-efficiency operation can be performed. Therefore, the current setting value ips may be changed according to the load torque and the rotational speed. Further, according to the present invention, if the set value of the current setting means 72 is made constant from the time of start-up, the start-up torque can be increased, and further stable operation can be achieved even if the motor current is controlled constant up to the advance angle control region. .

  To further explain the frequency correction operation, if the motor current ip increases from the set value ips, the error signal Δip becomes a negative value, and the frequency correction signal Δω (Δω = Kf × Δi) also becomes a negative value, so ω1 = The corrected frequency ω1 is lowered by the control of ω + Δω, the γ-δ axis approaches the dq axis, the load angle δ decreases, and the motor current I decreases, so that the constant current control operation is performed. V / f control causes irregularity when there is torque fluctuation, but by adding frequency control, the irregularity is suppressed and rotation speed fluctuation or load angle fluctuation is reduced. The phenomenon is also suppressed. Further, since the frequency correction gain increases in proportion to the set frequency ω, the higher the frequency, the larger the constant current action and the lower the turbulence. Since the motor coil voltage vector ωLI in the vector diagram of FIG. 2 is proportional to the frequency, it is considered that the present invention corrects the motor driving frequency in proportion to the motor coil voltage vector. In other words, the motor coil voltage vector ωLI increases as the drive frequency increases, and the motor coil voltage vector ωLI and the load angle δ are correlated, so that the load angle fluctuation, that is, the turbulence phenomenon is also proportional to the motor coil voltage vector ωLI. . Therefore, it can be seen that the turbulence can be reduced by correcting the frequency in proportion to the motor coil voltage vector ωLI.

  FIG. 3 is a graph showing a method for controlling the inverter control voltage, frequency, and frequency correction gain when the motor is started.

  A so-called V / f control is performed in which the drive frequency and the applied voltage corresponding to the frequency are linearly increased from the start to the target rotational speed, and the frequency correction gain Kf is increased in proportion to the drive frequency. If the motor current set value is kept constant and the starting control is performed by voltage control, the starting becomes easy. In the case of a fan or pump load, high-efficiency operation control can be performed by changing the current set value according to the drive frequency. Further, since the reluctance torque increases when the saliency motor advances the current more than the q axis, the set current ips may be increased in proportion to the frequency in order to control the advance angle.

  Further, the startup time ts until the target rotational speed is reached can be reduced by changing the start time ts according to the moment of inertia of the load. That is, as the moment of inertia increases, the turbulence can be reduced by increasing the startup time ts.

  By changing the frequency correction gain Kf in accordance with the drive frequency, it is possible to reduce the motor rotational speed fluctuation at the low start-up speed. Therefore, at the start-up, the frequency control gain is set to almost zero, and the voltage correction means 77 makes the constant current constant. By performing control and reducing the control signal ΔVδ of the voltage correction means 77 after startup and performing constant current control by the frequency correction means 74, the motor current can be made close to a sine wave to facilitate startup. As shown in Kfa of FIG. 3, the frequency control gain is not increased linearly in proportion to the time, the frequency control gain at the start of startup is set to almost zero, and the control gain is increased during the startup as shown in Kfb. May be. It should be noted that if the frequency proportional control gain is increased, oscillation tends to occur and noise is weakened. Therefore, a low-pass filter and a limiter may be appropriately provided in each of the voltage correction unit 77 and the frequency correction unit 74. If a delay element or an integral element is added to the frequency correction means 74, the turbulence cannot be suppressed, and conversely, the turbulence increases. Therefore, a combination of proportional control, a filter and a limiter is used.

  FIG. 4 is a vector diagram of a saliency motor (IPMSM).

  Since the saliency motor uses reluctance torque, it is necessary to control the advance angle. Generally, when the current phase is advanced by 30 degrees with respect to the q axis, the maximum efficiency operation is assumed. In order to increase the lead angle β, it is necessary to increase the load angle δ (δ = φ + β). In the constant output current method according to the present invention, the inverter output voltage is set substantially equal to or less than the induced voltage. The load angle δ can be increased by setting the current I large.

  In the case of a non-saliency motor (SPMSM), when the inverter output voltage Va and the motor induced voltage Em are substantially equal, the relationship between the motor current I and the load angle δ is given by Equation 4 from the cosine theorem.

  Here, kL is a ratio of the induced voltage constant Ke and the inductance L, and is given by kL = Ke / L. The motor current I is independent of the drive frequency ω1 and is determined by the load angle δ. By controlling the motor current I, the load angle δ can be controlled conversely, and the current advance angle β can be controlled. In the case of IPMSM, the output voltage ratio with respect to the frequency of the voltage control means 71 is set so that the motor induced voltage Em and the inverter output voltage Va are substantially equal, and the motor peak current is set by the current setting means 72. The load angle δ or the current advance angle β can be controlled.

  Since the induced voltage Em increases as the driving speed increases, voltage saturation may occur where the induced voltage Em becomes higher than the DC bus voltage Vp. Therefore, when the motor applied voltage Va is saturated, it is necessary to further advance the current. For this purpose, the current set value ips is preferably increased. The set current value ips may be increased as the frequency increases, but the motor efficiency can be improved by changing the current set value to be increased after detecting voltage saturation. In the detection of voltage saturation (Em> Va), the motor induced voltage Em is proportional to the drive frequency ω (Em = Ke × ω), and the motor applied voltage peak value Vap is obtained from the DC bus voltage Vp and the modulation factor m (Vap = m × Vp), voltage saturation determination can be made only by comparing the drive frequency, the DC bus voltage Vp, and the modulation degree.

  FIG. 5 is a diagram illustrating a PWM signal, a shunt resistance voltage waveform, and a current detection timing during two-phase modulation.

  In FIG. 5, vc is a triangular wave carrier signal, vu and vv are u-phase and v-phase modulation signals, up, vp and wp are upper arm control signals for each phase of UVW, and Vsh is a shunt resistance voltage waveform. Since the w-phase lower arm transistor is forced to conduct, the w-phase modulation signal is not shown.

  As shown in FIG. 5, the pattern in which the motor peak current appears in the two-phase modulation is a section in which only the upper arm of one phase is turned on (t0 to t2, t4 to t5), or the upper arm of the two phases is turned on. Appear in a certain section (t5 to t7). Unlike the three-phase modulation, the two-phase modulation is PWM-controlled only for the two phases, so that the section where the peak current appears is widened, so that the peak current can be easily detected.

  FIG. 6 is a current detection timing chart during two-phase modulation, and shows the two modulation signal waveforms of each phase of UVW and the phase at which each phase current appears in the shunt resistor. The interval from 0 to 1 / 3π is the W phase current Iw and the V phase current Iv, the interval from 1 / 3π to 2 / 3π is the U phase current Iu and the V phase current Iv, and the interval from 2 / 3π to π is the U phase. A phase current Iu, a W-phase current Iw, and each phase current appear sequentially. As shown by the arrows in the figure, the section where the current peak value appears is delayed by 30 degrees from the phase at which the voltage from the neutral point of each phase reaches its peak. The peak value of each phase current appears. That is, the interval 0 to 1 / 3π is the peak value of Iw, the interval 1 / 3π to 2 / 3π is the peak value of Iv, the interval 2 / 3π to π is the peak value of Iu, and the peak value is 6 times in one period. Appears. When the current phase is delayed by 30 degrees from the voltage phase, it is easy to detect the peak current. However, when the current phase is delayed by 60 degrees, the pulse width is narrowed, indicating that current detection becomes difficult. However, in the case of IPMSM, the power factor angle φ of the voltage phase and the current phase becomes small, so that the detection of the current peak value is easy, and in the case of SPMSM, the degree of advance is set slightly, so the power factor angle When φ becomes large, it is very rare and practically no problem occurs.

  The one-shunt current detection method and the method constituted by the voltage amplifier and the peak hold circuit have the feature that not only the hardware configuration is simple, but also the processor software is light and simple. Further, the A / D conversion timing for detecting the current may be the valley or peak (t0, t3, t6 in FIG. 5) of the carrier signal in which all the switching transistors of the inverter circuit are turned on or off, and the current detection is simple. In addition, it has a strong feature against noise.

  As described above, the present invention controls the motor peak current or the motor current corresponding to the rotating magnetic flux to a predetermined value in order to drive the permanent magnet motor by sensorless sine wave by V / f control. Constant output current control is performed by controlling the motor drive frequency based on an error signal from the set value, and the phase relationship between the rotor magnet axis coordinates (dq coordinates) and the applied voltage axis coordinates (γ-δ coordinates) is a constant load angle. Since it is controlled to δ, the turbulence in the V / f control can be almost eliminated, and the stable operation can be achieved even with the advance control. According to the present invention, stabilization control is possible regardless of a saliency motor or a non-saliency motor, and in particular, the advance angle control can be easily performed by making the inverter output voltage and the motor induced voltage substantially constant and constant output current. There are features. Furthermore, since the pulse width in which the peak current appears is widened when two-phase modulation is performed using the one-shunt method, the peak current can be easily detected, and a simple peak current detection means can be configured by the voltage amplifier and the peak hold circuit. Wear can be simplified.

  Although the motor voltage is V / f controlled by the signal ω before frequency correction in the first embodiment, the effect is the same even if the V / f control is performed by the signal ω1 after correction.

(Embodiment 2)
Hereinafter, a second embodiment of the present invention will be described with reference to FIG.

  FIG. 7 is a block diagram of the control means of the motor drive device according to Embodiment 2 of the present invention.

  The block diagram of the control means shown in FIG. 7 is obtained by partially changing the control means 7 shown in FIG. 1 or adding a detailed block. Only the change addition unit will be described below.

  The voltage control means 71 comprises a V / f control section 71a and a voltage addition section 71b. The V / f control section 71a multiplies the voltage proportional to the output signal ω of the frequency setting means 70, that is, ω by an induced voltage constant Ke. The voltage Vf is output to make the inverter circuit output voltage ratio to the drive frequency constant. The voltage adder 71 b adds the correction voltage ΔVδ and the starting voltage Vs (not shown) to the voltage Vf to output the voltage Vδ, and adds the voltage signal Vδ to the inverter control means 75. The output signals ip and ips of the current detection means 6 and the current setting means 72 are compared by the current comparison means 73, and the error signal Δip is added to the voltage correction means 77. The voltage correction unit 77 proportionally integrates the error signal Δip by the proportional integration control unit 77 a, limits the control voltage of the proportionally integrated signal via the voltage limiting unit 77 b, and adds the addition signal ΔVδ to the voltage control unit 71. Since the addition signal ΔVδ is limited by the voltage limiting unit 77b, the addition signal ΔVδ is effective only at the time of low-frequency startup, and a voltage approximately equal to the motor induced voltage Vm is applied to the motor during steady rotation.

  The output signal ω of the frequency setting means 70 and the output signal Δip of the current comparison means 73 are applied to the frequency correction means 74. The frequency correction means 74 includes a proportional section 74a for calculating a signal proportional to the error signal Δip, an adder section 74b for the frequency setting signal ω, a frequency proportional calculation section 74c for the frequency setting signal ω, and an output signal (Kf · The signal Δω0 from the multiplying unit 74d that calculates the product of Δip) and the output signal (K · ω) of the frequency proportional calculating unit 74c is added to the adding unit 74b via the frequency limiting unit 74e. The proportional constant Kf of the proportional unit 74a is set to about 1 to 10, and the proportional constant K of the frequency proportional calculation unit 74c is 1 in which the output signal Δω0 from the multiplication unit 74d is almost zero at start-up and becomes Kf · Δip at the steady state. Choose a smaller value. The output signal ω1 of the frequency correction unit 74 is applied to the phase signal generation unit 76, and the phase signal θ generates the three-phase sine wave signals vu, vv, vw in addition to the sine wave generation unit 75a of the inverter control unit 75, and PWM control. Three-phase PWM signals up, un, vp, vn, wp, wn are generated via means 75b. As shown in FIG. 5, the PWM control means 75b includes a carrier signal generation unit, a signal comparison unit, a dead time insertion unit (none of which are shown), and the details are omitted.

  As described above in the second embodiment, the voltage control unit 71 is provided with the voltage limiting unit, and the frequency correction unit 74 is provided with the frequency proportional calculation unit 74c. Since the current Ip is controlled and the motor current Ip is controlled by frequency control during high-speed rotation, the inverter circuit output voltage and the motor-induced voltage are almost equal during high-speed rotation, leading to lead angle control and field-weakening control during high-speed rotation is possible. It becomes.

  As described above, according to the present invention, the inverter circuit drive frequency is controlled so that the motor peak current Ip or the motor drive current I corresponding to the motor rotating magnetic field becomes a set value. The phase relationship between the axis and the dq axis becomes a set value corresponding to the load, and the phase relationship between the rotating magnetic field and the rotor magnetic flux can be kept constant, so that sensorless sine wave driving is possible. In particular, even if the motor load fluctuates greatly from no load to the rated load, it is possible to operate with constant current control, so the control is very simple. In addition, by changing the gain of the frequency correction means that corrects the drive frequency by the current error signal according to the drive frequency, the high frequency range of turbulence can be suppressed, and the inverter output voltage and the motor induced voltage can be made substantially equal. The field-weakening control can be performed by controlling the voltage according to the above. Further, since it is not necessary to perform coordinate conversion or vector decomposition of the motor current, the control program is simplified with little computation, and the motor can be easily controlled by an 8-bit or 16-bit microcomputer.

  Since the V / f control does not estimate the rotor position, the present invention uses almost no motor parameters. Further, since the rotation speed is open loop control, the fluctuation in the rotation speed is very small, the control method is simple, and the current detection is also easy. Low noise, low price, high reliability motor drive device can be realized. In particular, control is possible regardless of saliency motors and non-saliency motors, and lead angle control is easy, so the field control field can be expanded to a high speed range by field weakening, and the motor control program and current detection are simplified. The burden on the processor is lightened, and it can be applied to the compressor motor, washing motor, drying fan motor simultaneous sine wave drive system such as heat pump washer / dryer, and low-cost and highly reliable multi-motor simultaneous drive device can be realized .

  As described above, the motor drive device of the present invention detects the motor current corresponding to the peak value of the motor current or the rotating magnetic field by driving the permanent magnet motor with the inverter circuit that converts the DC power into the AC power. The inverter circuit output voltage and the motor drive frequency are controlled so as to be set to the set value, so that it can be applied to most motor drive devices that drive permanent magnet motors. The present invention can be applied to a motor driving device for a vacuum cleaner, a motor driving device for a vacuum cleaner, a fan motor driving device such as a ventilation fan or a combustor, and a compressor motor driving device for an air conditioner or a refrigerator. Furthermore, the present invention can also be applied to a multiple motor simultaneous drive system such as a heat pump washer / dryer or an air conditioner.

1 is a block diagram of a motor drive device according to Embodiment 1 of the present invention. Motor control vector diagram of motor drive device in Embodiment 1 of the present invention The graph which shows the starting control method of the motor drive unit in Embodiment 1 of this invention Control vector diagram of saliency motor in Embodiment 1 of the present invention Shunt resistance voltage waveform and current detection timing chart in Embodiment 1 of the present invention Current detection timing chart during two-phase modulation in Embodiment 1 of the present invention Block diagram of control means in embodiment 2 of the present invention

2 DC power supply 3 Inverter circuit 4 Motor 5 Motor load 6 Current detection means 7 Control means 70 Frequency setting means 71 Voltage control means 72 Current setting means 73 Current comparison means 74 Frequency correction means

Claims (1)

DC power supply, inverter circuit for converting DC power of the DC power supply to AC power, a permanent magnet synchronous motor driven by the inverter circuit , a fan or pump load driven by the motor, and the inverter circuit DC current Current detecting means for detecting the peak value of the current and control means for controlling the inverter circuit by the output signal of the current detecting means to drive the motor in a sine wave, the control means for controlling the output frequency of the inverter circuit. A frequency setting means for setting, a voltage control means for controlling the output voltage of the inverter circuit according to an output signal of the frequency setting means, a current setting means for setting a current peak value of the motor, and an output signal of the current detection means; The current comparison means for comparing the output signal of the current setting means and the output signal of the current comparison means Ri and frequency correction means for correcting the output frequency, the motor drive device having the V / f control means for controlling the motor voltage in accordance with the frequency.

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