JP3163655B2 - Inverter device - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an inverter device for converting a DC voltage obtained by rectifying and smoothing an AC power supply to a high frequency and supplying it to a load.

[0002]

FIG. 14 is a circuit diagram of a conventional inverter device (Japanese Patent Application No. 2-405558). Hereinafter, the circuit configuration will be described. An AC power supply Vs is connected to an AC input terminal of the full-wave rectifier DB via a filter circuit including a transformer L3 and capacitors C5 and C6.
A smoothing capacitor C1 is connected to a DC output terminal of the full-wave rectifier DB via a diode D3. A series circuit of transistors Q1 and Q2 is connected to the smoothing capacitor C1. Diodes D1 and D2 are connected in anti-parallel to the transistors Q1 and Q2, respectively. At the connection point between the diode D3 and the rectifier DB, a capacitor C
3, one end of the inductor L1 is connected to the connection point of the transistors Q1 and Q2.
The power supply side terminal of the filament of the discharge lamp La is connected between the other end of the capacitor C3 and the other end of the inductor L1. A capacitor C2 is connected in parallel between the non-power supply terminals of the filament of the discharge lamp La. Further, an inductor L2 and a capacitor C are provided at both ends of the diode D3.
4 series circuits are connected in parallel.

The operation of the above circuit will be described below.
First, the operation of the inverter will be described. The inverter comprises transistors Q1, Q2 and diodes D1, D2,
Inductor L1, capacitors C2 and C3 and discharge lamp La
It is composed of The transistors Q1 and Q2 are alternately turned on and off at a high speed to convert the DC voltage of the smoothing capacitor C1 to a high frequency, thereby lighting the discharge lamp La at a high frequency. The capacitor C2 constitutes a path for supplying a preheating current to the filament of the discharge lamp La, and also functions as a capacitor for resonance with the inductor L1. The capacitor C3 is a coupling capacitor for cutting a DC component.

In the above circuit, when the transistor Q2 is turned on, a current flows from the capacitor C1 through a path returning to the capacitor C1 via the capacitor C4, the inductor L2, the capacitor C3, the discharge lamp La and the capacitor C2, the inductor L1, and the transistor Q2. . At this time, the voltage appearing in each element includes V1 ≒ V4 + V5 + V3 + V2 +
There is a relationship of V6. Connected to the output terminal of the full-wave rectifier DB is a series circuit of the capacitor C3, the discharge lamp La, the capacitor C2, and the inductor L1.
When Vin |> V3 + V2 + V6 ≒ V1-V4-V5, a current flows from the full-wave rectifier DB through a path returning to the full-wave rectifier DB via the capacitor C3, the discharge lamp La and the capacitor C2, the inductor L1, and the transistor Q2. Will be.

That is, the inductor L2 and the capacitor C2 inserted between the output terminal of the full-wave rectifier DB and the capacitor C1.
4 shares the voltage of the difference between the rectified output voltage of the full-wave rectifier DB and the voltage V1 of the capacitor C1. Even if the input voltage | Vin | is lower than the voltage V1 of the capacitor C1, The current Iin flows. Therefore,
The input power factor increases. The input current waveform from which high-frequency components have been removed by the filter circuit including the capacitors C5 and C6 and the transformer L3 can be a waveform close to a sine wave with less harmonic components. Further, in this circuit, when the transistor Q2 is turned on, a current flows directly from the rectifier DB to the load, so that the overall efficiency of the circuit is high and a circuit method suitable for a relatively small and small capacity inverter device. Met.

[0006]

In the above conventional example, when the input voltage Vin is considerably higher than the load voltage, when the resistance value of the load is extremely small (at a light load), or when the resonance state of the inverter is low. Under certain conditions, such as when the voltage is weak, the voltage generated in the series circuit of the inductor L2 and the capacitor C4 becomes small, and the voltage sharing the difference between the rectified output voltage of the full-wave rectifier DB and the voltage V1 of the capacitor C1 is shared. And the pause time T occurs in the waveform of the input current as shown in FIG. For this reason, although the circuit configuration is a simple and high-efficiency inverter, there is a limit to the improvement of the input power factor and the reduction of the input current harmonic under certain conditions as described above, and there is still room for improvement. was there. From this, when the output control is performed, the above condition may be satisfied, and it is difficult to perform the output control while maintaining a high input power factor and a low harmonic component of the input current. .

SUMMARY OF THE INVENTION The present invention has been made in view of the above circumstances, and has as its object to improve the input power factor by supplying an input current from an AC power supply via an inverter load and a switching element. The present invention is to reduce the idle period of the input current and reduce the harmonic component of the input current regardless of the controlled condition in the inverter device provided with.

[0008]

According to the present invention, in order to solve the above-mentioned problems, as shown in FIG. 1, a full-wave rectifier DB for full-wave rectifying an AC power supply Vs, a full-wave rectifier DB
A smoothing capacitor C1 connected via a diode D3 to a DC output terminal of the first and second transistors Q1 and Q2 connected in series at both ends of the smoothing capacitor C1 and alternately turned on and off; Anti-parallel diodes D1 and D2 of the first and second transistors Q1 and Q2,
An inverter load (capacitors C2, C3 and load F) is connected between a connection point between the DC output terminal of the full-wave rectifier DB and the diode D3 and a connection point between the first and second transistors Q1 and Q2.
And the inductor L1), an impedance element (a series circuit of the inductor L2 and the capacitor C4) is connected to both ends of the diode D3 to control the operating frequency of the first and second transistors Q1 and Q2. And a control unit for shortening the idle period of the input current Iin from the AC power supply Vs.

[0009]

The operation of the present invention will be described below with reference to FIG. The circuit of FIG. 1 is substantially the same as the circuit of the conventional example shown in FIG. 14, in which a frequency control circuit K2 for controlling the switching frequency of the transistors Q1 and Q2 is added, and the load F is released. The only difference is that it is not limited to the electric lamp La. As shown, the smoothing capacitor C1
Is V1, the voltage of the resonance capacitor C2 connected to both ends of the load F is V2, and the DC component cutting capacitor C is
3, the voltage of the capacitor C4 is V4, the voltage of the inductor L2 is V5, and the voltage of the inductor L1 is V6. The input voltage from the AC power supply Vs is Vin, the input current is Iin, and the output current of the full-wave rectifier DB is Id.
And

As described in the conventional example, Vin ≒ 0 V depends on the voltage generated in the capacitor C4 and the inductor L2.
Even in the vicinity, the input current Iin flows when the transistor Q2 is turned on. Voltage V generated at the inverter load composed of capacitors C2 and C3, load F and inductor L1
3 + V2 + V6 is substantially equal to V1-V4-V5, and when the input voltage Vin becomes higher, from the rectifier DB,
The current Id flows through a path returning to the rectifier DB via the capacitor C3, the load F, the capacitor C2, the inductor L1, and the transistor Q2. Further, when the input voltage Vin is near the peak, a current flows from the rectifier DB to a path that returns to the rectifier DB via the diode D3 and the capacitor C1, in addition to the path described above.
1 is charged.

As is apparent from the above analysis, to reduce the idle period of the input current Iin, V = V3 + V2 + V6
It is necessary to control the value of ≒ V1-V4-V5 to be low when the transistor Q2 is turned on. Now, since the smoothing capacitor C1 is used as a power supply for the inverter, it is a large-capacity electrolytic capacitor. Therefore, the voltage V1 of the capacitor C1 becomes substantially constant. Also, the collector-emitter voltage Vc of the transistor Q2
The relationship between e and the voltage V4 and the voltage V (= V3 + V2 + V6) is as shown in FIG. In order to shorten the idle period of the input current Iin, it is necessary to make the lower envelope of the voltage V (= V3 + V2 + V6) as low as possible. This corresponds to increasing the amplitudes of the voltages V4 and V5, which can be realized by setting the switching frequency near the resonance frequency of the resonance system including the capacitor C4 and the inductor L2.

[0012]

FIG. 1 is a circuit diagram of a first embodiment of the present invention. This circuit is different from the conventional circuit shown in FIG.
A frequency control circuit K2 for controlling the switching frequency of the transistors Q1 and Q2 is added. Under the conditions described in the conventional example, if the amplitude of the voltage of the capacitor C4 and the voltage of the inductor L2 is small, | Vin | <V in a period T near Vin ≒ 0V as shown in FIG. Pauses occur in the current. Here, when the switching frequency is changed by the frequency control circuit K2 to approach the resonance frequency, the capacitor C4 and the inductor L2
Has a large amplitude, and the voltage V (= V3 + V2 + V
6) The lower envelope shifts down. For this reason, FIG.
As shown in FIG. 7, there is a period where | Vin |> V exists even near Vin ≒ 0 V. Therefore, the pause of the input current is eliminated, and the input current waveform becomes a waveform close to a sine wave. Therefore, the input power factor is higher than in the conventional example, and the harmonic component of the input current is reduced.

In order to actually perform the above control, | Vi
It is necessary to detect n | and V in advance. For example, as shown in the circuit diagram of FIG. 5, V (= V2 + V3 + V6) is detected by a detection circuit J1 connected to a terminal d, and | Vin | is detected by a detection circuit J2 connected to a terminal e. In comparison, the frequency control circuit K2 must determine whether it is necessary to change the switching frequency. However, the present invention does not specify the means for detecting | Vin | or V, and the gist of the present invention is to control the voltage amplitude of the capacitor C4 and the inductor L2 to eliminate the pause of the input current.

FIG. 6 is a circuit diagram of a second embodiment of the present invention. In this embodiment, MO is used as the switching element.
SFET is used. The diodes D1 and D2 are M
It can be omitted because it can be replaced by a parasitic diode in the OSFET. The switching element is not limited to a bipolar transistor or MOSFET, but may be an electrostatic induction thyristor or another semiconductor element.

In this circuit, only the capacitor C4 is connected instead of the series circuit of the capacitor C4 and the inductor L2 in the embodiment of FIG. The cause of the pause in the input current Iin is the same, and the input current Iin is controlled by controlling the amplitude of the voltage of the capacitor C4 to be large.
Can be eliminated. Therefore, by adding a frequency control circuit K2 and performing the same control as described in the embodiment of FIG. 1, the pause of the input current Iin is eliminated, the input power factor is increased, and the harmonic component of the input current is increased. Can be reduced.

FIG. 7 is a circuit diagram of a third embodiment of the present invention. In this circuit, in the embodiment of FIG.
4 and an inductor L2 instead of a series circuit.
Only 2 is connected. In this case, since the impedance element is the inductor L2, the manner of generating the voltage is different, but when the transistor Q2 is turned on, the inductor L2 determines the difference between the output voltage of the full-wave rectifier DB and the voltage V1 of the capacitor C1. Even if the input voltage Vin is lower than the voltage V1 of the capacitor C1, the input current Iin flows, and the input power factor and the input current harmonic can be improved. From this, although there is a difference between the inductor L2 and the capacitor C4, the operation is the same as in each of the above embodiments. When the impedance element is only the inductor L2,
The diode D3 is not essential and can be omitted.

Also in this embodiment, by adding the frequency control circuit K2 and controlling the switching frequency to approach the resonance frequency, the amplitude of the voltage V5 of the inductor L2 is controlled to be large, and the input current Iin is stopped. Can be eliminated. Further, in the circuit configuration of FIG. 7, if the inductance values of the inductors L1 and L2 are designed to satisfy L2> L1, the voltage generated in the inductor L2 increases, so that the output voltage of the rectifier DB and the capacitor C
The function of sharing the difference of 1 voltage V1 is increased. Therefore, the idle period of the input current Iin is shortened. An embodiment circuit utilizing this will be described below.

FIG. 8 is a circuit diagram of a fourth embodiment of the present invention. In this circuit, the inductance value of the inductor L2 in the circuit of FIG. 7 is made variable. Thus, when the pause period of the input current becomes longer, the pause period of the input current can be shortened by controlling the inductance value of the inductor L2 to be increased. In this circuit, the pause of the input current is reduced by making the inductance value of the inductor L2 variable, so that the width of frequency change by the frequency control circuit K2 can be reduced.

In general, when the frequency is changed in the inverter device, the output changes. However, if the inductance value of the inductor L2 is properly adjusted, the output can be kept constant. For example, when the inverter is operated in a region where the switching frequency is higher than the resonance frequency of the circuit, the output can be reduced by increasing the switching frequency. However, when the switching frequency is increased, the frequency becomes farther from the resonance frequency of the circuit, so that the voltage generated in the inductor L2 decreases. Then, as described in the conventional example, a pause occurs in the input current. In order to shorten the pause of the input current, the switching frequency must be close to the resonance frequency of the circuit, so that the switching frequency is reduced. In this case, the output cannot be reduced.
In such a case, in this circuit, when the inductance value of the inductor L2 is increased, the idle period of the input current is shortened. At the same time, since the inductor L2 also functions as a current limiting element of the inverter, the output is reduced.
As described above, if the inductance value of the inductor L2 is variably controlled, the idle period of the input current can be kept short while maintaining a short period.
Output control can be performed. Even if it is necessary to change the switching frequency, the change width is small, so that the control is easy. The effect of improving the input power factor and the input current harmonic is obtained in the same manner as in each of the above embodiments.

FIGS. 9 to 12 show circuit examples in which the inductance value of the inductor L2 is variable. In the circuit example of FIG. 9, a saturable reactor is used, and its inductance value is determined by the control voltage Vc applied to the control winding Tc. In the circuit example of FIG.
21 and an inductor L22 constitute an inductor L2, and a bidirectional switch S is provided at both ends of the inductor L22.
W are connected in parallel, and the inductance value is small when the switch SW is on and large when the switch SW is off. FIG.
In the circuit example of (2), the inductor L2 is configured by a parallel circuit of the inductor L21 and the inductor L22, and the inductor L22
, A bidirectional switch SW is connected in series, and its inductance value is small when the switch SW is on and large when it is off. In the circuit example of FIG.
Is provided with an intermediate tap, and the winding up to the intermediate tap can be short-circuited by the bidirectional switch SW. The inductance value is small when the switch SW is on and large when the switch SW is off. In each of the circuit examples shown in FIGS. 10 to 12, the inductance value of the inductor L2 can be variably controlled only in two stages. However, if more stages of variable control are required, the number of inductors connected in series or in parallel can be adjusted. Or just increase the number of taps.

FIG. 13 is a circuit diagram of a fifth embodiment of the present invention. In this circuit, the inductor L in the circuit of FIG.
2 is arranged between the series circuit of the transistors Q1 and Q2 and the smoothing capacitor C1. In this circuit, when the transistor Q2 is turned on, the capacitor C1, the inductor L2, the capacitor C3, the load F and the capacitor C
2, inductor L1, transistor Q2, capacitor C
1 flows, and the voltage relationship of each element is V1 ≒ V
5 + V3 + V2 + V6. At this time, the voltage between the DC output terminals of the full-wave rectifier DB is V = V3 + V2 +
V6 ≒ V1−V5, and the input current flows even if the input voltage | Vin | is lower than the voltage V1 of the capacitor C1 by the voltage V5 of the inductor L2. This full-wave rectifier DB
The function of the inductor L2 sharing the difference between the output voltage of the capacitor C1 and the voltage V1 of the capacitor C1 is the same as that described in the embodiment of FIG. Therefore, the control method is the same. To reduce the pause of the input current, the switching frequency is set close to the resonance frequency of the circuit, and a large voltage V5 is generated in the inductor L2. Then, the voltage difference between the output voltage of full-wave rectifier DB shared by inductor L2 and voltage V1 of capacitor C1 also increases, and the input current flows even during the period when input voltage | Vin | is low, so that the input power factor Can be increased, and the harmonic component of the input current can be reduced. Note that also in this embodiment, the inductance value of the inductor L2 is
1 or the inductor L2
Is preferably variable as in the above-described embodiment.

In the example described above, attention is paid to the fact that if the input voltage | Vin | is higher than the voltage of the element connected to the output terminal of the rectifier DB, the input current Iin will flow. It is to control the voltage amplitude of C4. However, in practice, as long as the voltage relationship as described in the description of the embodiment is satisfied, the pause of the input current is always eliminated, and the means for realizing the voltage relationship is not limited to frequency control.
For example, the following method can be adopted.

(A) A method in which the oscillation state of the resonance system is changed by changing the impedance values of the inverter element and the impedance element, and as a result, control is performed so as to approach the resonance state. Thus, by changing the impedance value of each element, it is possible to satisfy the voltage relationship as described in the embodiment.

(B) A method of increasing the load. That is, by changing the impedance value of the load, the oscillation state of the resonance system is changed, and as a result, control is performed so as to approach the resonance state. By changing the impedance value of the load in this way, it is possible to satisfy the voltage relationship as described in the embodiment.

Although a specific example of a circuit for controlling the above (a) and (b) is not specifically shown, for example, if means for turning on and off the impedance element by a bidirectional switch is used, Obviously, it can be easily realized.

[0026]

According to the present invention, a full-wave rectifier for full-wave rectification of an AC power supply, a smoothing capacitor connected to a DC output terminal of the full-wave rectifier via a diode, and a DC terminal at both ends of the smoothing capacitor or a full-wave rectifier. A DC output terminal of a full-wave rectifier, comprising first and second switching elements connected in series between output terminals and alternately turned on and off, and antiparallel diodes of the first and second switching elements. And an inverter load having an inverter load connected between a connection point of the diode and a connection point of the first and second switching elements, wherein an impedance element is connected to both ends of the diode, and the first and second switching elements are connected. Control means for controlling the operating frequency of the rectifier to reduce the idle period of the input current from the AC power supply. By performing the frequency control so as to increase the voltage amplitude of the impedance element even in a low period, the voltage difference between the charging voltage of the smoothing capacitor and the output voltage of the full-wave rectifier can be shared by the impedance element. There is an effect that the rest period of the input current can be reduced, the input power factor can be increased, and the harmonic component of the input current can be reduced.

Further, if the inductance value of the impedance element is set to be larger than the inductance value of the inverter load, the voltage amplitude of the impedance element increases, so that the difference between the output voltage of the full-wave rectifier and the voltage of the smoothing capacitor is calculated. Therefore, there is an advantage that the effect of sharing is increased.

Further, if the inductance value of the impedance element is made variable, the voltage amplitude of the impedance element can be controlled so that the input current does not pause even if the output of the inverter is changed by frequency control.
There is an effect that it is possible to prevent the input power factor from decreasing and the harmonic component of the input current from increasing.

[Brief description of the drawings]

FIG. 1 is a circuit diagram of a first embodiment of the present invention.

FIG. 2 is a waveform chart for explaining the operation of the first embodiment of the present invention.

FIG. 3 is a waveform chart showing an operation before frequency control according to the first embodiment of the present invention.

FIG. 4 is a waveform chart showing an operation after frequency control according to the first embodiment of the present invention.

FIG. 5 is a circuit diagram in which a detection circuit is added to the first embodiment of the present invention.

FIG. 6 is a circuit diagram of a second embodiment of the present invention.

FIG. 7 is a circuit diagram of a third embodiment of the present invention.

FIG. 8 is a circuit diagram of a fourth embodiment of the present invention.

FIG. 9 is a circuit diagram illustrating an inductor used in a fourth embodiment of the present invention.

FIG. 10 is a circuit diagram illustrating an inductor used in a fourth embodiment of the present invention.

FIG. 11 is a circuit diagram illustrating an inductor used in a fourth embodiment of the present invention.

FIG. 12 is a circuit diagram illustrating an inductor used in a fourth embodiment of the present invention.

FIG. 13 is a circuit diagram of a fifth embodiment of the present invention.

FIG. 14 is a circuit diagram of a conventional example.

FIG. 15 is a waveform chart showing the operation of the conventional example.

[Explanation of symbols]

 D1, D2, D3 Diode C1, C2, C3 Capacitor C4, C5, C6 Capacitor Q1, Q2 Transistor L1, L2 Inductor L3 Transformer Vs AC power supply DB Full-wave rectifier F Load

Claims (5)

(57) [Claims] 1. A full-wave rectifier for full-wave rectification of an AC power supply, a smoothing capacitor connected to a DC output terminal of the full-wave rectifier via a diode, and both ends of the smoothing capacitor connected in series and turned on alternately. First and second turned off
A switching element, and an anti-parallel diode of the first and second switching elements, wherein an inverter is provided between a connection point between the DC output terminal of the full-wave rectifier and the diode and a connection point between the first and second switching elements. In an inverter device connected to a load, a control for connecting an impedance element to both ends of the diode and controlling an operation frequency of the first and second switching elements to reduce a pause of an input current from an AC power supply. An inverter device comprising means. 2. A full-wave rectifier for full-wave rectifying an AC power supply, a smoothing capacitor connected to a DC output terminal of the full-wave rectifier via a diode, and a series connection between the DC output terminal of the full-wave rectifier. First and second switching elements, which are alternately turned on and off, and an anti-parallel diode of the first and second switching elements. A connection point between the DC output terminal of the full-wave rectifier and the diode, the first and second switching elements, In an inverter device having an inverter load connected to a connection point of a second switching element, an impedance element is connected to both ends of the diode, and an operating frequency of the first and second switching elements is controlled to obtain an AC power supply. An inverter device comprising control means for shortening the idle period of the input current from the inverter. 3. The inverter device according to claim 1, wherein the impedance element is a series circuit of a capacitor and an inductor. 4. The inverter device according to claim 3, wherein the inductance value of the impedance element is set to be larger than the inductance value of the inverter load. 5. The inverter device according to claim 1, further comprising a control unit that changes the impedance value of the impedance element or the inverter load so as to enhance the resonance effect.