JP3261706B2 - Inverter device - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an inverter device for converting a DC voltage obtained by rectifying and smoothing an AC power supply to a high frequency and supplying it to a load.

[0002]

2. Description of the Related Art FIG. 23 is a circuit diagram of a conventional inverter device (Japanese Patent Application No. 2-327321). Hereinafter, the circuit configuration will be described. An AC power supply Vs is connected to an AC input terminal of the full-wave rectifier DB via a filter circuit including an inductor L3 and a capacitor C5. The DC output terminals of the full-wave rectifier DB include inductors L2 and L1.
Is connected to a smoothing capacitor C1. A series circuit of transistors Q1 and Q2 is connected to the smoothing capacitor C1. Diodes D1 and D2 are connected in anti-parallel to the transistors Q1 and Q2, respectively.
The connection point between the transistors Q1 and Q2 and the inductors L1 and L
2 between the discharge lamps L through the capacitor C3.
a is connected. A capacitor C2 is connected in parallel between the non-power supply terminals of the filament of the discharge lamp La.

The operation of the above circuit will be described below.
First, the operation of the inverter will be described. The inverter comprises transistors Q1, Q2 and diodes D1, D2,
Inductor L1, capacitors C2 and C3 and discharge lamp La
It is composed of The transistors Q1 and Q2 are alternately turned on and off at a high speed to convert the DC voltage of the smoothing capacitor C1 to a high frequency, thereby lighting the discharge lamp La at a high frequency. The capacitor C2 constitutes a path for supplying a preheating current to the filament of the discharge lamp La, and also functions as a capacitor for resonance with the inductor L1. The capacitor C3 is a coupling capacitor for cutting a DC component.

In the above circuit, when the transistor Q2 is turned on, current flows from the capacitor C1 through the path of the inductor L1, the capacitor C3, the discharge lamp La, and the transistor Q2, but the current flowing to the load is partially passed from the rectifier DB to the inductor L2, The current also flows from the path of the capacitor C3, the discharge lamp La, and the transistor Q2. When transistor Q2 turns off, the oscillating circuit of the inverter turns on diode D1 to form a closed loop. At this time, the energy stored in the inductor L2 is discharged to the capacitor C1 via the capacitor C3, the discharge lamp La, and the diode D1, and the capacitor C1 is charged. Next, when the transistor Q1 is turned on, a current flows through the load from the capacitor C3 to the load through the path of the inductor L1, the transistor Q1, and the discharge lamp La in the opposite direction. At this time,
The energy remaining in the inductor L2 is the inductor L1
Is discharged to the capacitor C1 via the capacitor C1, and further charges the capacitor C1.

[0005] The above process is repeated over the entire section of the commercial cycle of the AC power supply Vs, so that the input current always flows. Therefore, the input power factor increases. Also, an appropriate filter circuit is added to the input side, and the input current waveform from which the high-frequency component has been removed can be a waveform close to a sine wave with few harmonic components. Further, in this circuit, since current flows directly from the rectifier DB to the load via the inductor L2, the overall efficiency of the circuit increases, and a circuit method suitable for a relatively small and small-capacity inverter device is used. there were.

[0006]

In the above conventional example, when the input voltage Vin is considerably higher than the load voltage, when the resistance value of the load is extremely small (at a light load), or when the resonance state of the inverter is low. Under a certain condition such as a weak case, a pause T occurs in the waveform of the input current Iin as shown in FIG. For this reason, although the circuit configuration is a simple and high-efficiency inverter, there is a limit to the improvement of the input power factor and the reduction of the input current harmonic under certain conditions as described above, and there is still room for improvement. was there. From this, when the output control is performed, the above condition may be satisfied, and it is difficult to perform the output control while maintaining a high input power factor and a low harmonic component of the input current. .

SUMMARY OF THE INVENTION The present invention has been made in view of the foregoing, and an object of the present invention is to improve the input power factor by supplying an input current from an AC power supply through a vibration element and a switching element of an inverter. It is an object of the present invention to reduce an idle period of an input current and reduce a harmonic component of an input current regardless of a controlled condition in an inverter device provided with such a circuit.

[0008]

According to the present invention, in order to solve the above-mentioned problems, as shown in FIG. 1, a full-wave rectifier DB for full-wave rectifying an AC power supply Vs, a full-wave rectifier DB
And a first and second transistor Q1, Q2 connected in series to both ends of the smoothing capacitor C1 and alternately turned on and off at a high frequency. , The antiparallel diodes D1 and D2 of the first and second transistors Q1 and Q2,
And a connection between the diode D3 and the smoothing capacitor C1.
Connection between the connection point and the first and second switching elements Q1 and Q2.
The load F and the capacitor C2 for resonance are connected in parallel between
Circuit, DC component cutting capacitor C3, and resonance
Is connected to the series circuit of the inductor L1.
A first element including at least one of a plurality of circuit elements constituting
Circuit elements (load F and capacitors C2 and C3) and remaining 1
Connection with a second circuit element (inductor L1) including
Connect one end of impedance element (inductor L2) to the continuation point
Connected, the DC output terminal of the full-wave rectifier DB and the diode D
Connect the other end of the impedance element to the connection point 3
Of the AC power supply voltage
Voltage around the switching circuit
During the period, there is a period when the voltage becomes lower than the rectified output voltage of the full-wave rectifier.
The duty of the first and second switching elements
And control means for controlling the switching frequency .

[0009]

The operation of the present invention will be described below with reference to FIG. The circuit of FIG. 1 is substantially the same as the circuit of the related art shown in FIG. 23. The configuration of the filter circuit between the AC power supply Vs and the full-wave rectifier DB is changed, and the load F is connected to a discharge lamp. The only difference is that it is not limited to La. As shown in the figure, the voltage of the smoothing capacitor C1 is V1, the voltage of the resonance capacitor C2 connected to both ends of the load F is V2, the voltage of the DC component cutting capacitor C3 is V3, and the voltage of the inductor L2 is V4. . The input voltage from the AC power supply Vs is Vin, the input current is Iin, and the output current of the full-wave rectifier DB is Id.

According to the study of the present inventor, the cause of the pause of the input current Iin from the AC power supply Vs is caused by the voltage Vin of the AC power supply Vs and the voltage V (= V3 + V) of the circuit element of the inverter.
It was found that there was a magnitude relationship with 2). Input current Ii
The waveforms of the respective parts when n was stopped were as shown in FIG. | Vin |> V
When (= V3 + V2), it was found that the input current Iin flows. Strictly speaking, the current Id flows when the relationship of the above expression is satisfied and the transistor Q2 is turned on. However, when the voltage V becomes minimum in the switching cycle and the transistor Q2 is turned on. The phase is almost the same as the period, and if the condition of the above equation is satisfied,
Input current flows. In the case of the relationship shown in FIG.
In the period T of ≒ 0 V, | Vin | is small and the input current Iin does not flow. Here, it can be seen that in order for the input current Iin to always flow, it is sufficient that a period where | Vin |> V exists even in the period T where Vin ≒ 0 V.

It suffices if the voltage relationship is as shown in FIG. In FIG. 3, even near Vin ≒ 0 V, it is always | Vin |> V within one cycle of inverter switching.
, The idle period of the input current Iin is eliminated, the input power factor is high, and the harmonic components of the input current are reduced. As a means for achieving or approaching such a voltage relationship, it is conceivable to lower the voltage V3 of the capacitor C3 or increase the amplitude of the voltage V2 of the capacitor C2. It is effective to increase the duty ratio of the lower transistor Q2 as a means for realizing the above, and it is effective to make the switching frequency close to the resonance frequency of the circuit as a means for realizing the above. Therefore, by taking these measures, the problems of the conventional example can be solved.

[0012]

FIG. 4 is a circuit diagram of a first embodiment of the present invention. This circuit is obtained by adding a duty control circuit K1 having a duty control function for the transistors Q1 and Q2 to the basic configuration of the present invention shown in FIG. Each output terminal a, b, c of the duty control circuit K1 is connected to a transistor Q
1 and Q2 are connected to control terminals a, b and c, respectively, to turn on and off the transistors Q1 and Q2 alternately.

FIGS. 5 and 6 are operation waveform diagrams of the present embodiment. Before the duty control, as shown in FIG.
Since | Vin | <V in the period T near V, the input current pauses. Therefore, the duty ratio of the transistors Q1 and Q2 is changed by the duty control circuit K1,
If the ON time of the lower transistor Q2 is shortened ,
The voltage V3 of the DC component cutting capacitor C3, which is a circuit element of the inverter, decreases. For this reason, as shown in FIG. 6, V (= V2 + V3) shifts downward compared to before the duty control in FIG. 5, and there is a period where | Vin |> V even when Vin ≒ 0V. Therefore, the pause of the input current is eliminated, and the input current waveform becomes a waveform close to a sine wave. Therefore, the input power factor is higher than in the conventional example, and the harmonic component of the input current is smaller.

In order to actually perform the above-described control, | Vi
It is necessary to detect n | and V in advance. For example, as shown in the circuit diagram of FIG. 7, V (= V2 + V3) is detected by a detection circuit J1 connected to a terminal d, and | Vin | is detected by a detection circuit J2 connected to a terminal e. In comparison, the duty control circuit K1 must determine whether the duty ratio needs to be changed. However, the present invention does not specify the means for detecting | Vin | or V, and the gist of the present invention is to control the voltage V3 of the DC component cutting capacitor C3 to eliminate the pause of the input current. is there.

FIG. 8 is a circuit diagram of a second embodiment of the present invention. This circuit is obtained by adding a frequency control circuit K2 instead of the duty control circuit K1 of the embodiment of FIG.
Now, assuming that the switching frequency is considerably higher than the resonance frequency of the circuit, the amplitude of the voltage V2 of the resonance capacitor C2 is small, and | Vin | <V (V) in the period T near Vin ≒ 0V as shown in FIG. = V2 + V3),
Pauses occur in the input current. Therefore, the frequency control circuit K2
When the switching frequency is lowered to approach the resonance frequency of the circuit, the oscillation increases as shown in FIG.
The amplitude of the voltage V2 of the resonance capacitor C2 increases. As a result, there is a period where | Vin |> V exists even when Vin ≒ 0 V. Therefore, the pause of the input current is eliminated, and the input current waveform becomes a waveform close to a sine wave. Therefore, the input power factor is higher and the harmonic component of the input current is smaller than in the conventional example.

What has been described above is a control method in the case where the switching frequency is higher than the resonance frequency and the input current pauses. Conversely, if the switching frequency is lower than the resonance frequency and the input current pauses, the switching frequency may be increased to approach the resonance frequency.

FIG. 11 is a circuit diagram of a third embodiment of the present invention. In this embodiment, M is used as the switching element.
OSFET is used. The diodes D1 and D2 are
It can be omitted because a parasitic diode in the MOSFET can be used instead. The switching element is not limited to a bipolar transistor or MOSFET,
It may be an electrostatic induction thyristor or another semiconductor element.

In this circuit, the arrangement of the inductor L1 and the parallel circuit of the load F and the capacitor C2 in the embodiment of FIG. 4 are interchanged, and a capacitor C4 is connected in place of the inductor L2. As shown in the figure,
The voltage of the capacitor C4 is Vc, the voltage of the inductor L1 is V5, and the voltage of the diode D2 is Vd. In this circuit configuration, the condition under which the input current flows is | Vin |> V
(= Vc + V3 + V5), and the transistor Q2 is on. The waveforms of each part are as shown in FIG. The relationship between the waveform of the voltage V5 of the inductor L1 and the on / off state of the transistor Q2 is as shown in FIG. 13, and the transistor Q2 is turned on during the period T1 when the value of the voltage V5 of the inductor L1 is large, and becomes small. In a certain period T2, the transistor Q2 is turned off. Period T
The change in the voltage V3 of the capacitor C3 and the change in the voltage Vc of the capacitor C4 are smaller than the change in the voltage V5 of the inductor L1 at 1 and T2, and the voltage V (= Vc + V3 + V5)
Drop in the period T1 and rise in the period T2 are almost equal to the voltage V5.
Is the same as Therefore, as the degree of drop of voltage V5 in period T1 increases, the above condition | Vin |> V (=
Vc + V3 + V5), and the input current easily flows. Here, the change in the voltage V5 of the inductor L1 is closely related to the oscillation state of the resonance system, and the closer the switching frequency is to the resonance frequency, the more the inductor L1 changes.
Of the voltage V5 is large. This is because the circuit current i increases near the resonance frequency because the inductor L1 and the capacitor C2 form a series resonance system. The voltage V5 of the inductor L1 has a rate of change di / d
Since it is proportional to t, when the circuit current i increases, the change in the voltage V5 of the inductor L1 also increases.

From the above, in this circuit, a frequency control circuit K2 having a frequency control function is added, and when a pause occurs in the input current, the frequency control described in the second embodiment is performed. By controlling the switching frequency to approach the resonance frequency and increasing the drop of the voltage V5 of the inductor L1, the above condition | Vi
n |> V (= Vc + V3 + V5). Thereby, the pause of the input current Iin is eliminated, the input power factor is increased, and the harmonics of the input current can be reduced.

FIG. 14 is a circuit diagram of a fourth embodiment of the present invention. This circuit has a configuration in which an inductor L2 is connected in series to a capacitor C4 in the embodiment shown in FIG. In this circuit configuration, the condition for the input current to flow is | Vin |> V (= Vc + V4
+ V3 + V5), and the transistor Q2 is on. The difference from the embodiment shown in FIG. 11 is that the voltage V4 of the inductor L2 is added to the conditional expression.
However, the one having the largest change among the above conditional expressions is the voltage V5 of the inductor L1. Therefore, as in the embodiment shown in FIG. 11, if the switching frequency is controlled to be close to the resonance frequency of the resonance system, the voltage V (= Vc + V4 + V3 + V) when the transistor Q2 is turned on.
The drop of 5) becomes large, and the above conditional expression is satisfied. As a result, the pause of the input current Iin is eliminated, the input power factor is increased, and harmonics of the input current can be reduced.

FIG. 15 shows a modification of the circuit shown in FIG. This circuit has a configuration in which the capacitor C4 in series with the inductor L2 is removed from the circuit of FIG. Even in this circuit configuration, the cause of the pause in the input current Iin is the same. However, since the impedance element is only the inductor L2, the current does not reverse, so that the diode D3 may be omitted. Even in this circuit configuration, if the switching frequency is controlled so as to approach the resonance frequency of the resonance system, the pause of the input current Iin is eliminated, the input power factor is increased, and the harmonics of the input current can be reduced.

FIG. 16 is a circuit diagram of a fifth embodiment of the present invention. In this circuit, the parallel circuit of the load F and the capacitor C2 in the first embodiment shown in FIG.
3 is connected in series with the inductor L1. In this circuit, the condition for the input current to flow is | Vin |> V3 + V4 and the transistor Q2
Is on. Here, the voltage V4 of the inductor L2 is relatively small and has little effect. Therefore,
By lowering the voltage V3 of the capacitor C3, the above conditional expression can be satisfied. As a means for achieving this, it is effective to reduce the on-duty of the transistor Q2. Therefore, in this embodiment, a duty control circuit K1 having a duty control function is added, and when a pause occurs in the input current, the transistor Q2 is turned on and off.
Reduce the duty. Then, the voltage V3 of the capacitor C3 decreases, and the conditional expression | Vin |> V3 + V4
And therefore the pause in the input current is reduced, and the input current waveform approaches a sine wave. This makes it possible to increase the input power factor and reduce the harmonic components of the input current.

FIG. 17 is a circuit diagram of a sixth embodiment of the present invention. In this circuit, the connection between the capacitor C3 and the inductor L1 in the first embodiment shown in FIG. 4 is replaced. In this circuit, the condition for the input current to flow is | Vin |> V4 + V5 + V2 and the transistor Q2 is on. Here, too, the effect of the voltage V4 of the inductor L2 is small, and the magnitude of V5 + V2 may be substantially examined. When the transistor Q2 is on, V5 + V2 ≒ V1-
Since V1 and V3 maintain a substantially constant voltage, V5 + V2 is also a substantially constant voltage. Therefore, in order to lower V5 + V2, control may be performed so as to increase the voltage V3 of the capacitor C3. As a means for achieving this, it is effective to increase the on-duty of the transistor Q2. Therefore, in this embodiment, a duty control circuit K1 having a duty control function is added, and when a pause occurs in the input current, the on-duty of the transistor Q2 is increased. Then, the voltage V3 of the capacitor C3 increases, and the above conditional expression | Vi
n |> V4 + V5 + V2, thus reducing the rest of the input current and bringing the input current waveform closer to a sine wave. This makes it possible to increase the input power factor and reduce the harmonic components of the input current.

FIG. 18 is a circuit diagram of a seventh embodiment of the present invention. In this circuit, in the circuit of FIG. 11, the capacitor C3 is replaced with the load F and the capacitor C2 instead of the inductor L1.
Are connected in series to the parallel circuit. In this circuit, the condition for the input current to flow is | Vin |
> Vc + V5 and the transistor Q2 is on. This is because, in the circuit of FIG. 11, the conditional expression | Vin |> V (= Vc + V3 +
The voltage V3 of the capacitor C3 is removed from the right side of V5). Since the voltage V3 of the capacitor C3 is a substantially constant voltage, in order to satisfy the conditional expression of the present embodiment from which the voltage V3 is removed, the change in the voltage V5 of the inductor L1 during the ON period of the transistor Q2 is increased. You can do it. Therefore, in this embodiment, a frequency control circuit K2 having a frequency control function is added.
When a pause occurs in the input current, by bringing the switching frequency close to the resonance frequency, the change in the voltage V5 of the inductor L1 increases, and the above-described conditional expression | Vin |> V
c + V5 is satisfied. Therefore, the pause of the input current is reduced, and the input current waveform approaches a sine wave. This makes it possible to increase the input power factor and reduce the harmonic components of the input current.

FIGS. 19 and 20 show modifications of the circuit shown in FIG. The circuit of FIG. 19 has a configuration in which an inductor L2 is inserted in series with the capacitor C4 in the circuit of FIG. Also, in the circuit of FIG. 20, in the circuit of FIG. 18, an inductor L2 is used instead of the capacitor C4.
Are connected. In these circuit configurations, the frequency control circuits K2 each having a frequency control function are added, and when a pause occurs in the input current, the switching frequency is brought close to the resonance frequency so that the voltage V5 of the inductor L1 is reduced.
Changes, the pause of the input current decreases, and the input current waveform approaches a sine wave. This makes it possible to increase the input power factor and reduce the harmonic components of the input current.

FIG. 21 is a circuit diagram of an eighth embodiment of the present invention. In this circuit, the load F is the discharge lamp FL in the first embodiment shown in FIG. Here, assuming that the input voltage Vin is 200V AC, the voltage V1 of the capacitor C1 becomes approximately 282V. Assuming that the duty of the transistors Q1 and Q2 is 50%, the voltage V3 of the capacitor C3
Becomes about 141V. Now, assuming that the load is a low-pressure discharge lamp FL having a low wattage, the lamp voltage at the time of lighting is low, the amplitude of the load voltage V2 is small, and the condition for the input current to flow | Vin |> V3 + V2 Is not satisfied, and a pause occurs in the input current. Therefore, when the control circuit K3 controls the on-time of the transistor Q2 to be shortened, the voltage V3 of the capacitor C3 decreases, and accordingly, the idle period of the input current is shortened. By appropriately controlling the on-duty of the transistor Q2, it is possible to completely eliminate the pause of the input current. Further, the discharge lamp FL generally has a negative resistance characteristic,
When the duty ratios of the transistors Q1 and Q2 are unbalanced, the lamp current decreases and the lamp voltage increases. This corresponds to an increase in the amplitude of the load voltage V2. In this regard, the pause of the input current can be eliminated. As described above, the input power factor of the inverter-type discharge lamp lighting device can be increased, and the harmonic component of the input current can be reduced.

FIG. 22 is a circuit diagram of a ninth embodiment of the present invention. In this circuit, the load F is the resistor RL in the first embodiment shown in FIG. The resistor RL corresponds to, for example, an incandescent lamp, and in that case, a high-frequency lighting device of the incandescent lamp is configured. In this lighting device, a control circuit K3 having a duty control function and a frequency control function is provided, and the light output can be controlled by changing the switching frequency. When reducing the light output,
Keep the switching frequency away from the resonance frequency. Then, the oscillation is weakened, and the amplitude of the load voltage V2 is reduced.
At this time, a pause occurs in the input current Iin. Therefore, when the duty ratio is unbalanced and the on-duty ratio of the transistor Q2 is controlled to increase, the voltage V3 of the capacitor C3 decreases, and the pause of the input current Iin is eliminated. Therefore, it is possible to realize a lighting device having a high input power factor and a low harmonic component of the input current. In the lighting device of FIG. 22, the switching frequency is set higher than the resonance frequency of the circuit, and when performing output control, the switching frequency is increased to reduce the light output. At this time, the transistor Q2 is turned on at the same time. By controlling the duty ratio to be large, it is possible to control the optical output while always keeping the input power factor high and keeping the harmonic components of the input current small.

In the example described above, if the input voltage | Vin | is higher than the voltage of the element connected to the output terminal of the rectifier DB, the input current Iin flows. V3 and the voltage V2 of the load are controlled. However, in practice, as long as the voltage relationship as described in the description of the embodiment is satisfied, the pause of the input current is always eliminated, and the means for realizing the voltage relationship is limited to duty control and frequency control. Instead, for example, the following method can be adopted.

(A) By changing the impedance values of the circuit elements and impedance elements of the inverter ,
A method in which the oscillation state of a resonance system is changed, and as a result, control is performed so as to approach the resonance state. Thus, by changing the impedance value of each element, it is possible to satisfy the voltage relationship as described in the embodiment.

(B) A method of increasing the load. That is, by changing the impedance value of the load, the oscillation state of the resonance system is changed, and as a result, control is performed so as to approach the resonance state. By changing the impedance value of the load in this way, it is possible to satisfy the voltage relationship as described in the embodiment.

Although a specific circuit example for controlling the above (a) and (b) is not specifically shown, for example, if means for turning on and off the impedance element by a bidirectional switch is used, It is clear that this can be achieved.

[0032]

According to the present invention, a full-wave rectifier for an AC power supply is provided.
And a diode at the DC output terminal of the full-wave rectifier.
Of the smoothing capacitor
It is connected in series at both ends and alternately turned on and off at high frequency.
First and second switching elements, and first and second switching elements.
And an anti-parallel diode of a switching element,
The connection point between the diode and the smoothing capacitor and the first and second
A load and a resonance coil are connected between the connection points of the switching elements.
Capacitors and capacitors for cutting DC components
And a series circuit of a resonance inductor,
Includes one or more of a plurality of circuit elements constituting a column circuit
A second circuit element including the first circuit element and the remaining one or more
Connect one end of the impedance element to the connection point with
At the connection point between the DC output terminal of the rectifier and the diode,
Inverter device that connects the other end of the impedance element
In addition, the first circuit is also provided near the zero crossing of the AC power supply voltage.
If the voltage of the element is within
And the first and second
2 switching element duty or switching cycle
Controlling the wave number, or the first or second circuit element,
Or impedance element or load impedance value
Control means for changing the
There is an effect that the rest period of the input current from the source can be shortened, the input power factor can be increased, and the harmonic component of the input current can be reduced.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a basic configuration of the present invention.

FIG. 2 is a waveform chart showing an operation before duty control according to the present invention.

FIG. 3 is a waveform chart showing an operation after duty control of the present invention.

FIG. 4 is a circuit diagram of a first embodiment of the present invention.

FIG. 5 is a waveform chart showing an operation before duty control according to the first embodiment of the present invention.

FIG. 6 is a waveform chart showing an operation after duty control according to the first embodiment of the present invention.

FIG. 7 is a circuit diagram in which a detection circuit is added to the first embodiment of the present invention.

FIG. 8 is a circuit diagram of a second embodiment of the present invention.

FIG. 9 is a waveform chart showing an operation before frequency control according to the second embodiment of the present invention.

FIG. 10 is a waveform chart showing an operation after frequency control according to the second embodiment of the present invention.

FIG. 11 is a circuit diagram of a third embodiment of the present invention.

FIG. 12 is a waveform chart showing the overall operation of the third embodiment of the present invention.

FIG. 13 is a waveform chart showing the operation of the main part of the third embodiment of the present invention.

FIG. 14 is a circuit diagram of a fourth embodiment of the present invention.

FIG. 15 is a circuit diagram of a modification of the fourth embodiment of the present invention.

FIG. 16 is a circuit diagram of a fifth embodiment of the present invention.

FIG. 17 is a circuit diagram of a sixth embodiment of the present invention.

FIG. 18 is a circuit diagram of a seventh embodiment of the present invention.

FIG. 19 is a circuit diagram of a modification of the seventh embodiment of the present invention.

FIG. 20 is a circuit diagram of another modification of the seventh embodiment of the present invention.

FIG. 21 is a circuit diagram of an eighth embodiment of the present invention.

FIG. 22 is a circuit diagram of a ninth embodiment of the present invention.

FIG. 23 is a circuit diagram of a conventional example.

FIG. 24 is a waveform chart showing the operation of the conventional example.

[Explanation of symbols]

 D1, D2, D3 Diode C1, C2, C3 Capacitor C4, C5, C6 Capacitor Q1, Q2 Transistor L1, L2 Inductor L3 Transformer Vs AC power supply DB Full-wave rectifier F Load

──────────────────────────────────────────────────続 き Continued on the front page (58) Field surveyed (Int.Cl. ⁷ , DB name) H02M 7/48 H02M 1/12 H02M 7/5387 H04B 41/24

Claims (10)

(57) [Claims] 1. A full-wave rectifier for full-wave rectification of an AC power supply, a smoothing capacitor connected via a diode to a DC output terminal of the full-wave rectifier, and alternately connected at both ends of the smoothing capacitor in series at a high frequency. First turned on and off
And it includes a second switching element and a reverse parallel diode of the first and second switching elements, the diode
Connection point between the capacitor and the smoothing capacitor and the first and second switches
Between the load and the resonance capacitor
Parallel circuit, DC component cut capacitor and resonance
Connected to a series circuit of inductors for
A first round including at least one of a plurality of circuit elements to be configured
Connection between a circuit element and a second circuit element including one or more of the remaining ones
Connect one end of the impedance element to the point
Connect the impedance at the connection point between the DC output terminal and the diode.
In the inverter device connected to the other end of the
Even when the AC power supply voltage is near zero crossing,
Output voltage of the full-wave rectifier within one cycle of switching.
The first and second switches are connected so that a period during which the pressure is lower than the pressure occurs.
Control the duty or switching frequency of the switching element.
Inverter, characterized in that it comprises a control means for control. 2. A first circuit element for cutting a DC component.
Including a capacitor, wherein the control means is for cutting a DC component.
And the second switch so as to lower the voltage of the
Controlling the duty of the switching element
The inverter device according to claim 1. 3. The control circuit according to claim 1, wherein the control means is a first or second circuit.
The first is to increase the voltage amplitude of the load contained in the element.
And the switching frequency of the second switching element.
The inverter device according to claim 1, wherein
Place. 4. A full-wave rectifier for full-wave rectifying an AC power supply.
To the DC output terminal of the full-wave rectifier via a diode.
Connected to both ends of the smoothing capacitor
First connected in a row and alternately turned on and off at high frequency
And a second switching element, and a first and a second switch.
And an anti-parallel diode of the switching element.
Connection point between the capacitor and the smoothing capacitor and the first and second switches
Between the load and the resonance capacitor
Parallel circuit, DC component cut capacitor and resonance
Connected to a series circuit of inductors for
Structure First circuit including one or more of a plurality of circuit elements to be formed
Connection between a circuit element and a second circuit element including one or more of the remaining ones
Connect one end of the impedance element to the point
Connect the impedance at the connection point between the DC output terminal and the diode.
In the inverter device connected to the other end of the
A first or second circuit element, or an impedance element, or
Changes the impedance value of the load to
The voltage of the first circuit element is switched even near zero crossing of
Output voltage of the full-wave rectifier within one cycle of
Control means for controlling the time period to occur.
Features inverter device. 5. The first circuit element comprises a load and a resonance capacitor.
Connect a DC component cut capacitor directly to the parallel circuit of the capacitor.
The second circuit element is connected in a column,
And the impedance element is an inductor or inductor.
5. A circuit comprising a series circuit of an inductor and a capacitor.
The inverter device according to any one of the above. 6. A first circuit element includes a load and a resonance capacitor.
Connect a resonance inductor in series with the parallel circuit of the capacitor.
The second circuit element is a capacitor for cutting a DC component.
The impedance element consists of an inductor.
The inverter device according to claim 1 or 4, wherein: 7. A first circuit element for cutting a DC component.
The second circuit element consists of a capacitor for load and resonance.
A resonance inductor is connected in series with a parallel circuit of capacitors
The impedance element is composed of an inductor.
An inverter device according to claim 1, 2 or 4,
Place. 8. A first circuit element for cutting a DC component
Consists of a series circuit of a capacitor and a resonance inductor,
The second circuit element is a parallel circuit of a load and a capacitor for resonance.
And the impedance element is an inductor or capacitor.
Or a series of inductors and capacitors in series.
The inverter device according to any one of claims 1 to 4. 9. The first circuit element is a resonance inductor.
The second circuit element is a load and a capacitor for resonance.
DC component cutting capacitor connected in series to the parallel circuit of
The impedance element is an inductor or core.
Composed of a capacitor or a series circuit of an inductor and a capacitor.
In any one of claims 1, 3 and 4. Mounted inverter
Place. 10. The impedance element is an inductor.
The DC output terminal of the full-wave rectifier and the smoothing capacitor
Characterized by omitting the diode connected between them
The inverter device according to claim 1.