JPH10271846A - Power device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a power device capable of decreasing input distortion by a simple circuit configuration, and having higher circuit efficiency as well. SOLUTION: Concerning this device, the midpoint of the series circuit of diodes D1, D2 out of those D1-D4 constituting a full-wave rectifier DB is connected to one end of an AC power source Vs, and a capacitor C1 which produces a filter effect is connected in parallel with the series circuit of the diodes D1, D2. In parallel with the capacitor C1, the series circuit of a pair of a first and second switching element Q1, Q2 to be turned on and off alternately by a high frequency is connected. In reverse/parallel with the switching elements Q1, Q2, the diode D3, D4 are connected respectively. In parallel with the diode D2, a capacitor C2 is connected. And a load circuit 1 is connected between one end on the side no being connected to the series circuit of the diodes D1, D2, out of both ends of the AC power source Vs, and the midpoint of the series circuit of the switching elements Q1, Q2.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power supply device for converting a DC voltage obtained by rectifying and smoothing an AC power supply to a high frequency and supplying it to a load.

[0002]

2. Description of the Related Art FIG. 53 shows a conventional circuit (Japanese Patent Laid-Open No. 22110/1990).
In this prior art circuit, a series circuit of an inductor L2 and an AC power supply Vs is connected between the AC terminals of a full-wave rectifier composed of bridge-connected diodes D1 to D4, and the DC terminal of the full-wave rectifier is connected. , A switching element Q1 and Q2 are connected in parallel with each of the diodes D1 and D2 of one arm of the full-wave rectifier, and a DC cut capacitor is connected to both ends of the switching element Q2. It is configured by connecting a series circuit of C30 and the load circuit 1, and has a configuration in which a step-up chopper for improving input distortion and switching elements Q1 and Q2 of an inverter for supplying high frequency to the load circuit 1 are also used.

The conventional circuit shown in FIG. 53 has an AC power supply Vs
Polarity in the direction of the arrow V _in, the current flows in the AC power source Vs → diode D3 → the switching element Q1 → inductor L2 → the AC power source Vs when the switching element Q1 is turned on. When the polarity of the AC power supply Vs is _in the arrow direction of Vin and the switching element Q1 is off, the inductor L ₂
→ charging the AC power source Vs → diode D3 → the capacitor 1 → the diode D2 → the capacitor C1 in the induced electromotive voltage of the inductor L2 current flows through a path of the inductor L _2.

On the other hand, the switching elements Q1 and Q2 are alternately turned on and off, so that a high-frequency voltage is applied to the load circuit 1. As described above, the switching element Q1 also serves as a chopper element and an inverter element. When the polarity of the AC power supply Vs is opposite to the direction of the arrow, the roles of the switching elements Q1 and Q2 change, and the switching element Q2
Serves also as a chopper element and an inverter element.

[0005] As described above, a chopper circuit for improving distortion,
Since the switching element is also used in the high frequency generation inverter, the number of circuit components is reduced, and the circuit configuration is simple and inexpensive.

[0006]

In the circuit of the prior art, the switching elements Q1 and Q2 are shared by the inverter and the chopper circuit, thereby simplifying the circuit configuration. Another. Further, there are two conversion processes of AC power supply Vs → chopper circuit and DC power supply (capacitor C1) → AC (inverter), and the circuit efficiency is limited.

The present invention has been made in view of the above problems, and has as its object to provide a power supply device with a simple circuit configuration, improved input distortion, and high circuit efficiency. .

[0008]

According to a first aspect of the present invention, there is provided a full-wave rectifier formed by connecting first to fourth diodes in a bridge, and an AC terminal of the full-wave rectifier. A series circuit of an AC power supply and a load circuit, a first smoothing capacitor connected between the DC output terminals of the full-wave rectifier, and first and second diodes of one arm of the full-wave rectifier. First and second switching elements connected in anti-parallel, and at least one of the DC output terminals of the full-wave rectifier and the midpoint of the arm of the full-wave rectifier to which the first and second switching elements are not connected. Characterized in that the first and second switching elements are alternately turned on and off alternately, and the input current can flow over substantially the entire power supply cycle,
Therefore, the input distortion is small, and the charging current to the smoothing capacitor flows only through the load circuit when the power is turned on, so that the inrush current can be suppressed without adding a special circuit, and furthermore, it can be realized with a relatively small number of parts. .

According to a second aspect of the present invention, in the first aspect of the present invention, both ends of the load circuit are formed by both ends of a primary winding of a transformer forming a resonance system with the second capacitor, and the secondary winding of the transformer is formed. Power is supplied to a load connected to the line, and a substantially constant load current can flow to the secondary side. In the invention of claim 3, in the invention of claim 1,
Both ends of the load circuit are constituted by both ends of the series circuit of the first inductor and the discharge lamp, and the non-power supply side terminal of the discharge lamp is connected to the first inductor and the third capacitor constituting the resonance circuit. Feature and does not use a transformer,
The loss due to the transformer can be reduced, and an efficient power supply device can be obtained. In addition, when there is no load, there is no path for charging the first capacitor, so that the inrush current can be made substantially zero.

According to a fourth aspect of the present invention, in the first aspect of the present invention, the second inductor is connected to both ends of the load circuit, and a transformer is not used. Therefore, loss due to the transformer can be reduced, and Since the inductor serves as a low-pass filter, the load current can be made substantially constant. According to a fifth aspect of the present invention, in the second, third or fourth aspect of the present invention, a control means capable of arbitrarily changing the frequencies of the first and second switching elements is provided. Even if it is almost constant, the input current can be arbitrarily varied and the power of the load can be varied. For example, if the load is a discharge lamp, dimming can be performed, and switching between preheating, starting, and lighting can be performed. In addition, when the power of the load suddenly changes and the element is stressed,
This can be avoided by changing the frequency.

According to a sixth aspect of the present invention, in the second, third, or fourth aspect of the present invention, there is provided control means capable of arbitrarily changing the on-periods of the first and second switching elements. Even if the frequency is substantially constant, the input current can be arbitrarily varied, so that the power of the load can be varied. For example, if the load is a discharge lamp, dimming can be performed. Lighting can be switched, and when the power of the load suddenly changes and stress is applied to the switching element, the ON period can be changed to avoid this.

According to a seventh aspect of the present invention, in the fifth or sixth aspect of the present invention, means for detecting the voltage of the first capacitor is provided, and the first and second capacitors are detected in accordance with the voltage of the first capacitor.
Characterized in that at least one of the driving frequency and the ON period of the switching element is made variable, and when the smoothed voltage is abnormally boosted, the oscillation is stopped or the dimming is performed to apply the voltage to the switching element. If the frequency or the ON period is controlled so that the applied stress can be avoided or the smoothed voltage becomes substantially constant, a stable output can be obtained. For example, in the case of a discharge lamp, light with less flicker An output can be obtained, and furthermore, by detecting a smoothed voltage and shifting the power-on and control start times, it is possible to prevent the destruction of the circuit due to an inrush current.

According to an eighth aspect of the present invention, in the second aspect of the present invention, there is provided means for detecting a voltage generated in the transformer,
At least one of the frequency and the ON period of the first and second switching elements is made variable in accordance with the voltage of the transformer, and the oscillation is stopped when the voltage across the load is abnormally increased. By dimming, the stress applied to the switching element can be avoided, and when the load is short-circuited, the oscillation is stopped to avoid the stress applied to the switching element. Can be.

According to a ninth aspect of the present invention, in the fifth or sixth aspect of the present invention, a means for detecting the polarity of the AC power supply is provided, and the frequency of the first and second switching elements or the ON state of the first and second switching elements are determined according to the polarity of the AC power supply. It is characterized in that at least one of the periods is variable, and the input current can be adjusted every half cycle of the power supply, so that the input current can be made closer to a sine wave, the input distortion can be further reduced, and a constant switching waveform can be obtained. Since the output of the load can be made smaller than when controlling,
Since deeper dimming can be performed and dimming can be performed while keeping the smoothed voltage substantially constant, it is possible to prevent abnormal stress from being applied to the switching element.

According to a tenth aspect of the present invention, in the fifth or sixth aspect of the present invention, there is provided means for detecting voltages generated in the first and second diodes, respectively.
And at least one of the drive frequency or the ON period of the second switching element is made variable,
By improving the asymmetry of the input current every half cycle of the power supply and making the input current closer to the sine, the peak value of the input current can be suppressed, preventing abnormal stress from being applied to the input filter. it can.

According to an eleventh aspect of the present invention, in the fifth or sixth aspect, means for detecting the voltage of the second capacitor is provided, and the voltage of the second capacitor is reduced according to the voltage of the second capacitor. By the way, the drive frequency is lowered or the on-period is changed in a direction to increase the load output. Where the voltage of the second capacitor is high, the drive frequency is increased or the on-period is changed in a direction to decrease the load output. Since the output of the load can be kept substantially constant, for example, when the load is a discharge lamp, an optical output with less flicker can be obtained.

According to a twelfth aspect of the present invention, in the invention of the fifth or sixth aspect, means for detecting a current flowing through the transformer or the load is provided so that the current flowing through the load becomes substantially constant according to the detected current. It is characterized in that at least one of the drive frequency or the ON period of the first and second switching elements is made variable, and even when the power supply fluctuates, the load current can be made substantially constant. If the load is a discharge lamp, the pulsating flow of the lamp current can be reduced, and a light output with less flicker can be obtained.

According to a thirteenth aspect of the present invention, in the second, third or fourth aspect of the present invention, means for varying the capacity of the second capacitor is provided, and the capacity of the second capacitor is made variable. This makes it possible to adjust the input current according to the load. For example, if the load is a discharge lamp, the input current can be reduced even when dimming or when the output of the load is switched. .

According to a fourteenth aspect of the present invention, in the second, third or fourth aspect, power is supplied to a plurality of loads. For example, if the load is a discharge lamp, the plurality of discharge lamps are turned on simultaneously. It is possible. According to a fifteenth aspect, in the second, third or fourth aspect, the second load circuit is connected in parallel to one of the first and second switching elements or in parallel with the first capacitor. Characterized by
Since the output of the added load circuit can be made substantially constant without special control, for example, if the load is a discharge lamp, an optical output with less flickering can be obtained as a whole, and even when one of the loads is disconnected, Since the remaining discharge lamps can be kept lit and the respective loads can be output at an arbitrary ratio, for example, even when the power supply fluctuates, the overall load current can be made substantially constant, and By reducing the pulsating current, an optical output with less flicker can be obtained.

According to a sixteenth aspect of the present invention, in the second, third or fourth aspect of the invention, at least one of the first and second switching elements is driven by another winding provided on an inductor of a transformer or a load. It is characterized in that the driving circuit of the switching element can be omitted,
By driving from the secondary-side inductor, which can realize a circuit with a smaller number of components, the circuit is automatically stopped at the time of abnormality such as no load, for example. If the driving of the other switching element is controlled by an external signal from the middle inductor, control such as dimming and stopping can be easily performed.

According to a seventeenth aspect of the present invention, in the second, third or fourth aspect of the present invention, the fifth capacitor is connected in parallel with the first capacitor.
A sixth diode series circuit is connected, a middle point of the series circuit is connected to a terminal of the AC power supply to which the first and second diodes are not connected, and a connection point between the AC power supply and the second capacitor is connected. And a load circuit connected between the power supply and the load circuit, so that even if the power supply is almost doubled, a common substrate can be used with the addition of a small number of components, and the voltage applied to the load circuit can be increased. Voltage can be made almost constant, so that almost the same output can be obtained even if the power supply is different.
Further, if the components are modularized, it is possible to cope with different types of power supplies only by combining the modules.

According to an eighteenth aspect of the present invention, in the second, third, or fourth aspect, the third and fourth switching elements which are alternately turned on and off at a high frequency are connected in parallel with the first and second diodes. The surplus power can be regenerated to the power supply when the load suddenly becomes lighter.

[0023]

BEST MODE FOR CARRYING OUT THE INVENTION

(Embodiment 1) FIG. 1 shows a circuit configuration of the present embodiment. In this embodiment, an AC power supply Vs and diodes D1 to D4 forming a full-wave rectifier DB at one end of the AC power supply Vs are shown.
Of the series circuits of the diodes D1 and D2 are connected,
A capacitor C1 that performs a smoothing action is connected in parallel with the series circuit of the diodes D1 and D2, and a series circuit of a pair of first and second switching elements Q1 and Q2 that are alternately turned on and off at a high frequency is connected in parallel with the capacitor C1. And a diode D3 in anti-parallel with the switching elements Q1 and Q2, respectively.
D4 and a capacitor C2 in parallel with diode D2.
And the diode D is connected between both ends of the AC power supply Vs.
The load circuit 1 is connected between one end of the series circuit of the switching elements Q1 and D2 that is not connected to the series circuit of the switching elements Q1 and Q2.

In this embodiment, the switching elements Q1 and Q2 are alternately turned on at a high frequency by the drive signal of the drive circuit 2. By turning it off, high frequency power can be supplied to the load circuit 1. Next, FIGS. 2 to 6 show operation explanatory diagrams over one cycle of the switching operation of the switching elements Q1 and Q2. It is assumed that the AC power supply Vs has a polarity (positive time) in a direction indicated by an arrow in FIG.

First, in an operation mode in which the switching element Q1 shown in FIG. 2 is turned on and the switching element Q2 is turned off,
When the switching element Q1 is turned on and the switching element Q2 is turned off, as shown in FIG.
3 → Capacitor C1 → Capacitor C2 → AC power supply Vs →
A current flows through the path of the load circuit 1.

When all the energy stored in the load circuit 1, which is an inductive load circuit, is released, the capacitor C1 is used as a power source, and the capacitor C1 → the switching element Q1 → the load circuit 1 → the capacitor C2 → the capacitor C1 as shown in FIG. A current flows through the path, and power is supplied to the load circuit 1. At this time, the voltage across the capacitor C2 gradually increases.

Further, in the operation mode in which the switching element Q1 shown in FIG. 4 is turned on and the switching element Q2 is turned off,
Eventually, when the voltage of the capacitor C2 becomes equal to the voltage of the AC power supply Vs, no more current flows into the capacitor C2. However, since the current still tends to flow in the load circuit 1 in the same direction, the AC power supply Vs → the diode D1 → the switching element Q1 →
A current flows through a path from the load circuit 1 to the AC power supply Vs.

As shown in FIG. 5, when the switching element Q1 is turned off and the switching element Q2 is turned on in the operation mode, the switching element Q1 is turned off and the switching element Q2 is turned on. In order to release the energy, as shown in FIG. 5, the AC power supply Vs → the diode D1 → the capacitor C1 →
A current flows through a path of the diode D4 → the load circuit 1 → the AC power supply Vs.

When all the energy of the load circuit 1 is released, a current flows through the path of the capacitor C2 → the load circuit 1 → the switching element Q2 → the capacitor C2 as shown in FIG. The voltage between both ends gradually decreases. Eventually, the switching element Q1 turns on again and the switching element Q
2 is turned off, and the operation mode shifts to the operation mode shown in FIG. 2 in order to release the energy stored in the load circuit 1 until then.

When the AC power supply Vs has the opposite polarity (negative polarity) to the direction indicated by the arrow in FIG. 1, the circuit operates almost in the same manner. The point that the direction of the AC power supply Vs is different from the positive
When the switching element Q1 is on, power is supplied directly from the AC power supply Vs to the load circuit 1 via the diode D1 when the switching element Q1 is on. On the other hand, when the direction of the AC power supply Vs is negative, the switching element Q2 is on. In this case, power is supplied directly to the load circuit 1 via the path of the AC power supply Vs → the load circuit 1 → the switching element Q2 → the diode D2 → the AC power supply Vs via the diode D2.

When the AC power supply Vs has a polarity opposite to the direction indicated by the arrow in FIG. 1, that is, when the voltage is negative, the voltage across the capacitor C2 does not rise more than the voltage across the capacitor C1 because of the presence of the diode D5. Therefore, the peak voltage of the voltage between both ends of the capacitor C2 is cut, and the stress on the capacitor C2 is reduced. It is desirable that the capacitor C2 has such a small capacity that charge and discharge can be performed within one cycle of switching.

With the above operation, the input current from the AC power supply Vs can flow over substantially the entire power supply cycle.
A circuit that is a more specific example of FIG. 1 is the circuit shown in FIG.
This circuit has a capacitor C21 and an inductor LF at the input.
1, a filter circuit FL composed of LF2, and as a load circuit 1, one end of an AC power supply Vs, a switching element Q
1, a primary winding of a transformer T1 composed of a leakage transformer is connected between the two windings, a preheating type discharge lamp La1 is connected to the secondary winding, and a filament of the discharge lamp La1 is further connected. A capacitor C3 that resonates with the leakage inductance of the transformer T1 is connected between the non-power supply terminals. The switching element Q1 and the diode D in FIG.
3 is replaced by a switching element Q1 'formed by a MOS-FET having a parasitic diode, and similarly, the switching element Q2 and the diode D4 in FIG. 1 are also replaced by a switching element Q2' formed by a MOS-FET having a parasitic diode. By doing so, the diode D
3, D4 is omitted to further reduce the number of parts. In addition, by providing the filter circuit FL, the input current becomes a substantially sine wave with suppressed harmonics, and the input power factor can be increased. Further, since the load circuit 1 has a transformer configuration, even when a voltage having a relatively low frequency such as a power supply cycle is applied to the primary side, only the AC component is transmitted to the secondary side. Therefore, the high-frequency current flowing through the discharge lamp La1 as a load is substantially constant.

[0033] Figure 8 shows an operation waveform diagram over one period of the AC power source Vs, Fig. (A) is a waveform of the input current I _in and the voltage his type of AC power source Vs. FIG. 3B shows the capacitor C
2 (c) shows the voltage waveform appearing at both ends of the discharge lamp L.
3 shows a lamp current waveform flowing through a1. Also, FIG.
Is the current flowing through the primary winding of the transformer T1, and the currents of the switching elements in (a), (b) and (c) are shown in FIGS. 9 (a), (b) and (c), respectively.

When the polarity of the AC power supply Vs is positive, the current waveform in the switching element Q1 'near the peak of the voltage of the AC power supply Vs (A)
The current waveform at 2 'is as shown in (c). Conversely, when the polarity of the AC power supply Vs is negative, the switching element Q1' is at (c) near the peak of the voltage of the AC power supply Vs.
The switching element Q2 'is as shown in FIG. Further, the currents of the switching elements Q ₁ ′ and Q 2 ′ when the AC power supply Vs is near 0 (V) are as shown in FIG. This is because the voltage of the AC power supply Vs is superimposed on the primary winding of the transformer T because the AC power supply Vs is connected to the load circuit 1.

As described above, by using the circuit of this embodiment, it is possible to suppress the harmonic component of the input current and increase the input power factor with a relatively small number of parts, thereby realizing the miniaturization and cost reduction of the circuit. can do. In the present embodiment, a leakage transformer utilizing a leakage inductance has been described as the transformer T1, but a normal inductor L is used instead of the leakage inductance as shown in FIG.
1 is connected in series to the primary side of the transformer T1. Alternatively, the same operation is performed even when a normal inductor L1 is connected in series to the secondary side of the transformer T1 as shown in FIG. Further, as shown in FIG. 12, a configuration without using the transformer T1 may be used. However, FIG.
In the configuration of FIG. 8, the current flowing through the load circuit 1 is as shown in FIG.
The waveform becomes a waveform in which the pulsating flow of the power supply cycle as shown by.

As shown in FIG. 13, an inductor L2 may be further connected in parallel to the load circuit 1 in FIG. In this case, since the inductor L2 serves as a low-pass filter, a substantially constant lamp current as described in the above embodiment can be obtained. Furthermore, the capacitor C
Although the connection point of No. 2 has been described with a configuration parallel to the diode D2, the circuit operation is the same even in parallel to the diode D1 as shown in FIG.
Among the above-described operations, the switching element Q1 ′
And the current flowing through the switching element Q2 'is reversed, and only the mode in which the input current flows is changed. Or Figure 1
Even if the positions of the capacitor C2 and the diode D5 are reversed as shown in FIG. 5, the effect is substantially the same. That is, when the diode D5 and the diode D2 are connected in parallel as shown in FIG. 15, the only difference is that the voltage across the capacitor C2 is clamped at 0 V as shown in FIG. 16 and no negative voltage is generated. .

Alternatively, as shown in FIG. 17, the circuits of FIGS. 1 and 15 are combined to form a system, and the switching elements Q1 'and Q2' are turned on. By synchronizing the OFF states, the input current can flow even when any of the switching elements Q1 'and Q2' is ON. Therefore, for example, in a case where the input filter units are integrated in the system, the peak value of the current flowing through the input filter unit can be reduced, and the filter unit can be reduced in size. The number of systems referred to here does not need to be two, but may be plural. In this case, the number of circuits in FIG. 1 and the number of circuits in FIG. 16 may be set to be substantially the same.

Alternatively, capacitors C2 'and C2' each having half the capacity of the capacitor C2 may be connected in parallel to the diodes D1 and D2 as shown in FIG. 18. In this case, the current flowing into the capacitor C2 in FIG. In FIG. 18, since the current can be shunted to the upper and lower capacitors C2 'and C2' and halved, the stress on the element can be reduced.

The AC power supply Vs is not limited to a single phase, but may be, for example, three phases. At that time, the circuit is configured as shown in FIG. 19, and four diodes D5 to D8 are added to the single-phase circuit shown in FIG. 1 to provide a three-phase full-wave rectifier with switching elements Q1 ', Q
A three-phase circuit can be easily realized simply by configuring with a 2 'parasitic diode.

The case where the discharge lamp La1 is turned on by a high-frequency alternating current as a load has been described. However, the load is not limited to the discharge lamp La1. It can be easily realized by the configuration of the load circuit 1 for applying the DC obtained by rectifying the secondary output of the transformer T1 with the full-wave rectifier 1 to the load R as shown in FIG.

(Embodiment 2) A circuit diagram of this embodiment is shown in FIG.
It is shown in FIG. In the present embodiment, a control circuit 3 that can arbitrarily control the drive frequency, the on-time, the duty ratio, and the like of the switching elements Q1 ′ and Q2 ′ is provided as a drive circuit in the circuit of FIG. Note that the load circuit 1 is connected to a transformer T1 of a leakage transformer 2 as in FIG.
The discharge lamp La1 and the capacitor C3 for resonance are connected to the next side, and the diodes D3 and D4 of the full-wave rectifier are omitted using the parasitic diodes of the switching elements Q1 'and Q2' of the MOS-FET. The filter circuit FL is omitted.

In the circuit of the present embodiment, when the driving frequency of the switching elements Q1 'and Q2' is shifted from the state shown in FIG. 22A to the state shown in FIG. 22B, the resonance between the leakage inductance of the transformer T1 and the capacitor C3 is increased. The power of the load (discharge lamp La1) can be varied by the action, and the dimming of the discharge lamp La1 can be performed. However, if only the number of driving cycles changes, the voltage across the capacitor C1 increases. As for the dimming, the duty ratio may be adjusted while the frequency is kept constant as shown in FIGS. In this case, although the voltage is hardly boosted, the input current is distorted.
This is because switching element Q1 'and switching element Q
If the ON period of 2 ′ is different, the input current I _in (AC power supply Vs → diode D1 → switching element Q1 ′) _{in which the} AC power supply Vs flows in a positive period (the direction of the arrow shown in FIG. 22).
→ Load circuit 1 → AC power supply Vs) and input current flowing in the negative period of AC power supply Vs (AC power supply Vs → Load circuit 1 → Switching element Q2 ′ → Diode D2 → AC power supply V
This is because the time during which the current flows in both loops of s) differs, and the input current becomes vertically asymmetric.

Alternatively, both the driving frequency and the duty ratio may be changed as shown in FIG. 24 (a) to FIG. 24 (b). FIG. 25 shows an example of a circuit configuration for eliminating the distortion of the input current described above. In the illustrated circuit, a power polarity discriminating circuit 4 is provided which divides voltages at both ends of the AC power source Vs by R34 and R35 and R36 and R37, and determines the polarity of the AC power source Vs by a comparator COMP1 based on these voltage values. ing. FIG. 26B shows the output (point A) of the converter COMP1. FIG.
When the voltage polarity of the AC power supply Vs shown in (a) is illustrated (positive), an “H” signal is output, and when the voltage polarity of the AC power supply Vs is negative, an “L” signal is output. The exclusive OR of this signal and the control signals of the switching elements Q1 'and Q2' is determined by using exclusive OR circuits EXOR1 and EXOR.
It is taken by OR2 and used as a new control signal.

By adding the power supply polarity discriminating circuit 4 and the exclusive OR circuits EXOR1 and EXOR2 in this way, duty control is performed and the switching element Q
Even when the switching waveforms of 1 'and Q2' are unbalanced, the ON period is switched every half cycle, so that the same input current of positive and negative AC power supply Vs flows.

The configuration of this embodiment is shown in FIGS.
5, the same operation can be realized by using the circuit configuration shown in FIGS. (Embodiment 3) This embodiment controls the switching elements Q1 'and Q2' as shown in FIGS. 27 to 29 in preheating, starting and lighting of the discharge lamp La1 as a load. 27A to 29A show the preheating state, FIG. 27B shows the starting state, and FIG. 27C shows the lighting state. In the case of FIG. The drive frequency is shifted so as to be gradually lowered.

FIG. 28 shows a case where the duty ratio is changed while the drive frequency is kept constant. FIG. 29 shows a case where both the drive frequency and the duty ratio may be changed at the same time. In the present embodiment, by using the configuration of the embodiment of FIG. 25 and switching the on-period every half cycle as in the embodiment of FIG. Lighting is facilitated.

It is needless to say that a similar effect can be obtained in this embodiment even when the operation is adopted in the circuit configurations of FIGS. 10 to 15 and FIGS. 17 to 21. (Embodiment 4) The configuration of this embodiment is shown in FIG. FIG.
In the circuit configuration shown by 0, a circuit for connecting the voltage between both ends of the capacitor C1 to the capacitor C31 which is the power supply of the control circuit 3 via the resistor R1 is added to the circuit of FIG. If the polarity of the AC power supply Vs is positive, the inrush current is the AC power supply Vs → the diode D1 → the capacitor C1 → the parasitic diode of the switching element Q2 ′ → the transformer T of the load circuit 1.
1 → switch SW → AC power supply Vs, and in a negative period, AC power supply Vs → switch SW → load circuit 1
T1 → parasitic diode of switching element Q1 ′ → capacitor C1-diode D2 → AC power supply Vs
Flows along the path. The switch SW is a power switch.

In any case, since the transformer T1 passes through the path, if the impedance of the transformer T1 is made sufficiently large, the rush current can be suppressed without adding any circuit. For example, the number of primary windings of the transformer T1 may be increased, the core gap may be reduced as much as possible, and the inductance value may be increased.

Next, the circuit operation of FIG. 30 will be described.
At the moment when the switch SW1 is closed and the AC power supply Vs is turned on, the voltage across the capacitor C1 is rapidly charged from 0 since no charge is stored in the capacitor C1. At the moment when the power is turned on, the power supply voltage V
Since C31 is also 0, the control circuit 3 does not operate immediately after the power is turned on. Control starts only when the capacitor C1 is charged and exceeds a certain reference value. FIG. 31 shows the waveform of each part at this time. FIG. 3A shows the input current, and FIG. 3B shows the shape of the voltage VC31 across the capacitor C31. In FIG. 3B, the voltage Vth indicated by a dashed line indicates a threshold level at which the control circuit 3 starts. FIG. 3C shows the voltage waveform between the gate and the source of the switching element Q1 ', and FIG. 4D shows the voltage waveform between the gate and the source of the switching element Q2'.

In the present embodiment, when the AC power supply Vs is turned on at time t0, the switching elements Q1 'and Q
2 'is started. If the switching elements Q1 'and Q2' operate almost simultaneously with turning on the power, for example, when a current is flowing through the parasitic diode of the switching element Q2 ', the switching element Q1'
When 1 'is turned on, the capacitor C1 is short-circuited. In the present embodiment, the destruction of the circuit due to the inrush current is prevented by shifting the start time of the control from the time when the power is turned on.

The present embodiment can be implemented in any of the configurations shown in the first to third embodiments. (Embodiment 5) The circuit configuration of this embodiment is shown in FIG.
This embodiment is different from the configuration of FIG.
1 and C2 are respectively referred to as a voltage detection circuit 5 and a voltage detection circuit 6, and a circuit for detecting a voltage generated at both ends of C2.
A circuit for detecting a voltage generated at both ends of the series circuit of the capacitor C2 and the capacitor C2 is referred to as a voltage detection circuit 7. The output of the voltage detection circuit 6 and the output of the voltage detection circuit 7 are used to determine the polarity of the power supply polarity 4
And the detection voltage of the voltage detection circuit 5 or 6 is input to the control circuit 3 depending on the polarity of the AC power supply Vs, and the drive frequency and the duty ratio of the switching elements Q1 ′ and Q2 ′ are changed according to those voltages. Things.

FIG. 33 shows the waveform of each part. 5A shows the voltage of the AC power supply Vs, FIG. 5B shows the detection voltage V32 selected through the changeover switch element S controlled by the power supply polarity discriminating circuit 4, and FIG.
Is a voltage waveform VC32 obtained by filtering the higher harmonics of the voltage waveform by the capacitor C32. In FIG.
The period Ta indicates the detection voltage of the voltage detection circuit 5, and the period Tb indicates the detection voltage of the voltage detection circuit 6.

According to the present embodiment, for example, the voltage waveform VC
By using 32, the frequency is set low near the peak of the AC power supply Vs, and set high near 0 V of the AC power supply Vs, so that the crest factor of the lamp current can be improved. Alternatively, the duty ratio of the switching elements Q1 'and Q2' may be controlled using the waveform of the VC32. In this case, the duty ratio is made closer to 50% near the peak of the AC power supply Vs, and the duty ratio is further unbalanced near 0 V of the AC power supply Vs, thereby improving the crest factor and eliminating flickering of the discharge lamp La1. be able to.

Needless to say, the number of driving cycles and the duty ratio may be changed simultaneously. Further, for example, V
When the voltage value of C32 becomes abnormally high, by stopping the oscillation of the switching elements Q1 'and Q2', it is possible to avoid destruction of the elements due to overvoltage.

Another example of the control regarding the capacitor C2 is shown in FIGS. The circuit of FIG. 34 includes a capacitor C2 ″ and a full-wave rectifier DB in parallel with the capacitor C2.
2. A series circuit of an AC switch composed of a transistor r2 is connected. When the transistor Tr2 is turned on, the capacitor C2 "is connected in parallel to the capacitor C2, and the capacitance at both ends of the capacitor C2 is increased by the combined capacitance. This is a circuit that can be used.

The circuit of FIG. 35 connects a capacitor C2 "in series with a capacitor C2, a full-wave rectifier DB2, and a parallel circuit of an AC switch including a transistor Tr2.
2 are connected to a capacitor C2 "to reduce the capacity. These circuits are, for example, discharge lamps L
When the dimming of a1 is performed, when the load becomes lighter, it can be used in a case where a pause section occurs in the input current. In general, in this embodiment, as the dimming is made deeper, the rest period of the input current increases. Therefore, by using the circuit of FIG.
And the series circuit of C2 ″ and the combined capacitance of the series circuit is reduced, the voltage across the series circuit is reduced, and the input current can be approximated to a substantially sinusoidal wave again. In this case, the waveform jumps near the zero cross of the input current.
Capacitor C2 in parallel with capacitor C2
By connecting 2 "to increase the combined capacitance, the voltage generated at both ends of the capacitor C2 can be increased to make the input current approximate to a substantially sine wave again. The circuit configurations shown in FIGS. For example, it is effective as a switching means when it is desired to provide two types of output characteristics with one discharge lamp La 1. Alternatively, the switching elements Q1 'and Q2' are not limited to the capacitor C2 but may be changed according to the load current. 36 can improve the pulsating current of the lamp current by controlling the number of driving cycles and the duty ratio of the lamp.The example shown in FIG. The current detection circuit 8 controls the driving frequency and the duty ratio of the switching elements Q1 'and Q2' according to the magnitude of the lamp current. During a period in which the current is relatively large, the lamp current can be suppressed by increasing the frequency, and the pulsating current can be reduced.

By controlling the switching elements Q1 'and Q2' according to the voltage generated in the capacitor C2, or by changing the impedance of the capacitor C2, flicker due to the pulsating flow of the lamp current can be reduced, and the input current can be reduced. And distortion can be improved. Or,
Even when the current flowing to the load is detected, the lamp current can also reduce the flicker due to the pulsating flow.

It should be noted that the configuration of this embodiment is the same as that of the first to fourth embodiments.
The present invention can be similarly implemented by adopting any of the configurations. (Embodiment 6) A circuit diagram of this embodiment is shown in FIG. The circuit of FIG. 37 is different from the circuit of FIG.
Voltage detection circuits 10 and 11 for detecting respective voltages of C2 '
And the magnitude of those voltage values is determined by the comparator COMP.
3, the control circuit 3 changes the drive frequency and the duty ratio based on the comparison result. Specifically, for example, when the input current _Iin is as shown _in FIG. 38A, if the voltage value of the lower capacitor C2 ′ is larger than the voltage value of the upper capacitor C2 ′, the oN period of the switching element Q2 'shortened, it is possible to input current I _in a substantially sinusoidal wave as shown in FIG. 38 (b) if Shiteyare longer oN period of the capacitor C1.

The locations to be detected are the capacitors C2 'and C2'.
Not only the voltage of 2 'but also a circuit for detecting a voltage generated in the diodes D1 and D2 may be provided. When the voltage value across the diode D1 is greater than the voltage across the diode D2, the ON period of the switching element Q2 'is shortened and the ON period of the switching element Q1' is increased, as shown in FIG. Substantially sinusoidal input current I _in
Can be

[0060] or may be directly detected by the current detection circuit 9 which inserts the input current I _in to the path of the AC power source Vs as shown in the circuit of Figure 39. The control method is the same as described above and will not be described. In FIG. 39, a capacitor C3 that resonates with the leakage inductance of the transformer T1 is connected in parallel to the primary winding of the transformer T1. The configuration of the present embodiment can be similarly implemented by adopting any of the configurations of the first to fifth embodiments.

(Embodiment 7) A circuit diagram of this embodiment is shown in FIG.
0 to FIG. FIG. 40 shows a case where a series circuit of resistors R10 and R11 is connected to both ends of the capacitor C1 in the circuit of FIG. 21, and the voltage at the midpoint of the series circuit and an arbitrary reference value V
dc is input to a comparator COMP4 and compared therewith, and the switching elements Q1 ′ and Q2 ′ are stopped or dimmed according to the output signal of the comparator COMP4.

Alternatively, instead of detecting the voltage between both ends of the capacitor C1, as shown in FIG. 41, the signal level obtained from the tertiary winding provided in the transformer T1 via the diode D32 and the reference value Vla are compared with the comparator COMP5. And when the reference value Vla is exceeded under the control of the control circuit 3, the control may be stopped or the light control state may be set.

As shown in FIG. 42, in the circuit of FIG. 21, a resistor R12 is inserted in series with the source terminal of the switching element Q2 ', and a voltage value generated at both ends of the resistor R12 is detected. The detected voltage value and the reference value Vsw are compared by a comparator COMP6, and the output of the comparator COMP6 is used to stop the switching elements Q1 'and Q2' under the control of the control circuit 3 or to perform a dimming state. Is also good.

When the circuits shown in FIGS. 40 to 42 are used, for example, when the discharge lamp La1 comes off and becomes unloaded, or when the discharge lamp La1 is almost in a half-wave state due to exhaustion of the filament emitter of the discharge lamp La1. The device can be prevented from being destroyed when it becomes impossible to maintain the lighting only by the above. The configuration of the present embodiment can be similarly implemented by adopting any of the configurations of the first to sixth embodiments.

(Embodiment 8) A circuit diagram of this embodiment is shown in FIG.
3 to FIG. 45. The circuit of FIG. 43 is a circuit in which the transformer T1 of the load circuit 1 is used as a drive circuit of the switching elements Q1 and Q2 in the circuit of FIG. That is, the switching elements Q1 and Q2 are alternately turned on depending on the direction of the current flowing through the primary winding of the transformer T1. The winding direction of the driving winding is determined so as to turn off. However,
Only at the time of activation, the switching element Q2 is turned on by the activation means 12 provided separately.

Alternatively, as shown in FIG. 43, when a normal transformer T2 is used for the load circuit 1 instead of the leakage transformer and an inductor L1 is inserted on the secondary side, the driving circuit for the switching elements Q1 and Q2 is Instead of the transformer T2, a drive winding may be wound around the inductor L1, and a drive signal may be obtained from the output of the drive winding. Also in this case, the switching elements Q1 and Q2 are alternately turned on depending on the direction of the current of the inductor L1. The winding direction of the drive winding is determined so that it turns off. With this circuit configuration, for example, when the discharge lamp La1 comes off,
Since no current flows through the inductor L1, the oscillation stops automatically. However, in the case of this circuit configuration, as in the circuit of FIG.

Using the circuits shown in FIGS. 43 and 44, a circuit that oscillates by itself without applying a signal from the outside can be realized, so that the number of components can be reduced. Alternatively, as shown in FIG. 45, the switching element Q2 'composed of a MOS-FET is driven by a signal from the control circuit 3, and only the switching element Q1 composed of a transistor is driven from the driving winding provided in the transformer T1. A signal may be obtained. According to this circuit, the switching element Q2 ′
The control such as dimming and stopping as described in the above embodiments can be performed only by the control on the side, so that a circuit that can be controlled with a small number of components can be realized. In FIG. 45, FIG.
As shown in FIG. 44, the diode D4 constituting the full-wave rectifier with the diodes D1 to D3 is eliminated by using the parasitic diode of the switching element Q2 composed of a MOS-FET.

The present embodiment can be applied to any of the configurations of the first to seventh embodiments. (Embodiment 9) A circuit diagram of this embodiment is shown in FIG. This embodiment adopts a load circuit 1 in which a plurality of discharge lamps are connected as loads in the configuration of the first to seventh embodiments.
FIG. 46 shows a configuration example of the load circuit 1.

The circuit shown in FIG. 46A has two discharge lamps La1 and La2 connected in series to the secondary side of a transformer T1. The circuit of FIG. 46B connects two primary windings in parallel using two transformers T1 and T1 'each composed of a leakage transformer, and a discharge lamp La is connected to each secondary winding.
1 and La2 are connected in parallel, and the capacitor C3 'is a resonance capacitor similar to the capacitor C3.

The circuit shown in FIG. 46 (c) further includes current limiting inductors L1 and L1 'and a discharge lamp La in the secondary winding of the transformer T1.
1 and La2 and a circuit composed of capacitors C3 and C3 'for resonance. Further, in the circuit of FIG. 46D, a balancer L2 is provided on the secondary winding of the transformer T1,
A circuit composed of discharge lamps La1, La2 and resonance capacitors C3, C3 'is connected to each terminal of the balancer L2.

The circuit shown in FIG. 46 (e) has two discharge lamps L connected in series on the secondary side of the transformer T1 as in FIG. 46 (b).
a1 and La2 are connected to perform preheating of each filament from the secondary winding of the inductor L1 provided on the primary side. With these configurations, it is possible to light a plurality of discharge lamps as loads. FIG. 46
(A) to (e) show the configuration diagram in the case where two discharge lamps La1 and La2 are connected, but it goes without saying that even when there are a plurality of discharge lamps, they can be lit by connecting them in a similar manner.

FIG. 47 shows another circuit configuration of the present embodiment. The circuit shown in FIG. 47 is different from the circuit of FIG. 1 in that a load circuit 1 'comprising a transformer T1' comprising a leakage transformer, a discharge lamp La1 'and a resonance capacitor C3' is connected in parallel with a switching element Q2 '. They are connected via C20. Since the circuit operation of the discharge lamp La1 has been described in the first embodiment, only the operation of the discharge lamp La1 'will be briefly described here.

At the moment when the switching element Q1 'is turned on and the switching element Q2' is turned off, the transformer T1 'is turned on in the same direction as the current flowing through the load circuit 1'.
A current flows for a while in the path of switching element Q1 '→ capacitor C1 → capacitor C20 → transformer T1 ′. Then, using capacitor C1 as a power supply, capacitor C1 → switching element Q1 ′ → transformer T1 ′ → capacitor C20.
→ Current flows through the path of the capacitor C1.

Next, when the switching element Q1 'is turned off and the switching element Q2' is turned on, the transformer T1 '→
After a current flows for a while on the path of the capacitor C20 → the switching element Q2 ′, a current flows on the path of the capacitor C20 → the transformer T1 ′ → the switching element Q2 ′ → the capacitor C20 using the capacitor C20 as a power supply. Q1 'is turned on, and switching element Q2' is turned off.

That is, the above operation is the operation of the half-bridge inverter itself using the capacitor C1 as a power supply, and this circuit itself does not have a function of improving the distortion of the input current. Since the lamp current of the discharge lamp La1 'becomes substantially constant if the capacity of the capacitor C1 is sufficiently large, the combined light output of the discharge lamps La1 and La1' is, for example, as shown in FIG.
The flicker is reduced as compared with the circuit configuration shown in FIG. In addition, even if either of the discharge lamps La1 and La1 'comes off, the remaining discharge lamps can maintain lighting. For example, by adjusting the capacity of the inductor and the capacitor C20 on the primary side of the transformer T1 'when the discharge lamp La1' comes off slightly to the leading side with respect to the driving frequency, the remaining discharge lamp is obtained. When La1 is turned on, the current flowing through the switching elements Q1 'and Q2' is a combined current of a leading phase and a lagging phase, and the peak value of the current flowing through the element is suppressed more than the lagging current for lighting only the discharge lamp La1. Can be.

Alternatively, by designing the discharge lamps La1 and La1 'to have slightly shifted frequency characteristics, it is possible to obtain a substantially constant output with respect to power supply fluctuations and load fluctuations.
Alternatively, as in the circuit shown in FIG. 48, a lamp La3 for direct lighting in the circuit of FIG. 1 may be connected between both ends of the capacitor C1 via a switching element Tr3, and the discharge lamp La1 and the lamp La3 may be turned on simultaneously. . Also in this case, since the lamp current of the lamp La3 is substantially constant if the capacity of the capacitor C1 is sufficiently large, the combined light of the discharge lamp La1 and the lamp La3 has, for example, the circuit configuration shown in FIG. Less flickering. Alternatively, the switching element Tr3 is turned on. By turning off, the lighting ratio of the discharge lamp La1 and the lamp La3 can be arbitrarily controlled.

Note that this embodiment can be implemented in the same manner regardless of the configuration of any of the first to eighth embodiments. (Embodiment 10) This embodiment is composed of the circuit shown in FIG. 49, and corresponds to the case where the power supply voltage is different in the configurations of Embodiments 1 to 8.

The circuit shown in FIG. 49 is different from the circuit of FIG.
A series circuit of the diodes D5 and D6 is connected in parallel to the capacitor C1, and a middle point of the series circuit is connected to one end of both ends of the AC power supply Vs which is connected to the load circuit 1,
Further, a DC cut capacitor C4 is connected between the connection point between the switching elements Q1 and Q2 and the load circuit 1.

FIG. 50 is a circuit specifically showing the circuit of FIG. 49. A filter circuit FL including a capacitor C21 and inductors LF1 and LF2 is provided at the input part of FIG.
1, D2, D5, and D6 are connected to a low-speed diode bridge DB.
10 is a circuit in which a high-speed diode D7 is added in addition to the circuit of FIG. Further, a small-capacity capacitor C5 is connected to the output terminal of the diode bridge DB10 in order to absorb high-frequency current flowing through the diode D7 and reduce stress applied to the diode bridge DB10.
Further, as in the first embodiment, the load circuit 1 uses a transformer T1 composed of a leakage transformer, and replaces the switching element Q1 and the diode D1 with a switching element Q1 ′ composed of a MOS-FET and its parasitic diode. Switching element Q2, diode D
2 is replaced with a switching element Q1 ′ composed of a MOS-FET and its parasitic diode. The operation of the circuit is simply performed with reference to FIG. The basic operation is the same as the operation shown in the first embodiment, but the difference from the first embodiment is that the capacitor C is connected in series with the load circuit 1.
4 is connected, the voltage generated across capacitor C2 is about half that in the first embodiment. This is because when current flows from the capacitor C1 or the AC power supply Vs to the capacitor C2,
→ Switching element Q1 '→ Load circuit 1 → Capacitor C
4 → capacitor C2 → capacitor C1 and AC power supply Vs → diode D1 → switching element Q1 ′ → load circuit 1 → capacitor C4 → capacitor C2 → diode D6 → AC power supply Vs. Is divided by the capacitors C4 and C2.

Further, by adding the diodes D5 and D6, the AC power supply Vs-diode D1 → capacitor C1 → diode D6 → AC power supply Vs or AC power supply Vs → diode D5 → capacitor C1 → diode D
2 → Since the path of the AC power supply Vs exists, the capacitor C
1 is almost equal to the peak value of the AC power supply Vs.

Therefore, for example, when the power supply voltage is 100 V, the circuit of the first embodiment is used, and when the power supply voltage is 200 V, if the diodes D5, D6 and the capacitor C4 are added as shown in FIG. With this circuit, it is possible to realize a circuit having substantially the same characteristics. As a result, the circuit board is shared, and if the power supply is 200 V, the configuration shown in FIG. 43 is adopted. If the power supply is 100 V, the diodes D5 and D6 and the capacitor C4 are removed.
If the terminal from which the capacitor C4 is removed is short-circuited with a jumper wire or the like, the circuit shown in FIG. 1 can be easily realized.

Alternatively, as shown in FIGS. 51A to 51D, by modularizing the circuit, if the voltage of the AC power supply Vs is 100 V, for example, the configurations of FIGS. Combination, AC power supply Vs voltage is 200V
Then, the configurations of FIGS. 51 (a), (c) and (d) are used in combination. The effect is obtained that the circuit can be shared between 100V and 200V.

The power supply voltage is 100 V and 20 V as described above.
Not limited to 0V, for example, 120 and 240V
Needless to say, any combination such as 120 V and 200 V may be used. It should be noted that the configuration of this embodiment and any of the configurations of Embodiments 1 to 9 can be adopted.

(Embodiment 11) This embodiment is composed of the circuit shown in FIG. 52.
In this configuration, switching elements Q3 and Q4 are connected in parallel with the diodes D1 and D2, respectively, to enable regeneration to the AC power supply Vs. The switching elements Q3 and Q4 are alternately turned on and off by the drive circuit. The switching element Q4 is turned on while the switching element Q1 is on, and the switching element Q3 is turned on while the switching element Q2 is on. . For example, when the switching element Q3 is turned off and the switching element Q4 is turned on in the state of the third operation mode of the first embodiment, the capacitor C1 → the switching element Q1 → the load circuit 1 → the AC power supply Vs → the switching element Q
4 → Current flows through the path of the capacitor C1, so that power can be regenerated to the AC power supply Vs. Conversely, when the AC power supply Vs is in the negative direction, if the switching element Q3 is turned on and the switching element Q4 is turned off, the capacitor C1 → the switching element Q3 → the AC power supply Vs → the load circuit 1 → the switching element Q2 → the capacitor Since current flows through the path of C1, power can be regenerated to the AC power supply Vs.

When the switching elements Q3 and Q4 are turned off at the same time, it goes without saying that the operation is exactly the same as that of the first embodiment. Note that this embodiment can be adopted in any of the configurations of the first to tenth embodiments.

[0086]

According to a first aspect of the present invention, there is provided a full-wave rectifier configured by connecting a first to a fourth diode in a bridge, and an AC power supply and a load circuit connected between AC terminals of the full-wave rectifier. Circuit, a first smoothing capacitor connected between the DC output terminals of the full-wave rectifier, and first and second anti-parallel connected to each of the first and second diodes of one arm of the full-wave rectifier.
A second switching element, at least one of the DC output terminals of the full-wave rectifier, and first and second arms of the arm of the full-wave rectifier;
Having a second capacitor connected between the switching element and the midpoint of the arm to which the switching element is not connected.
Since the switching elements are alternately turned on and off, the input current can flow over substantially the entire power cycle.
Therefore, the input distortion is small, and the charging current to the smoothing capacitor flows only through the load circuit when the power is turned on, so that the inrush current can be suppressed without adding a special circuit, and furthermore, it can be realized with a relatively small number of parts. This has the effect.

According to a second aspect of the present invention, in the first aspect, both ends of the load circuit are formed by both ends of a primary winding of a transformer constituting a resonance system with a second capacitor, and the secondary winding of the transformer is formed. Supply power to the load connected to the
There is an effect that a substantially constant load current can flow on the secondary side. According to a third aspect of the present invention, in the first aspect of the present invention, both ends of the load circuit are constituted by both ends of a series circuit of the first inductor and the discharge lamp, and the first terminal is connected to the non-power supply side terminal of the discharge lamp.
Since the inductor and the third capacitor forming the resonance circuit are connected, the transformer is not used. Therefore, the loss due to the transformer can be reduced, and an efficient power supply device can be obtained. Since there is no path for charging the capacitor, there is an effect that the rush current can be made substantially zero.

According to a fourth aspect of the present invention, since the second inductor is connected to both ends of the load circuit in the first aspect of the present invention, a transformer is not used. Therefore, the loss due to the transformer can be reduced, and the inductor has a low-pass characteristic. Since the filter functions as a filter, there is an effect that the load current can be made substantially constant. The invention of claim 5 is the invention of claim 2 or 3
Alternatively, in the invention according to the fourth aspect, since the control means capable of arbitrarily changing the frequencies of the first and second switching elements is provided, the input current can be arbitrarily varied even when the ON period is substantially constant, The power of the load can be varied,
For example, if the load is a discharge lamp, dimming can be performed, and preheating, starting, and switching of lighting can be performed. In addition, when the load power suddenly changes and the element is stressed, the frequency is changed. This has the effect of avoiding this.

The invention of claim 6 is the invention of claim 2 or 3 or 4
In the invention, since the control means capable of arbitrarily changing the on-period of the first and second switching elements is provided, the input current can be arbitrarily varied even when the frequency is substantially constant. The power can be varied,
For example, if the load is a discharge lamp, dimming can be performed,
Furthermore, there is an effect that switching between preheating, starting, and lighting can be performed, and when the power of the load suddenly changes and stress is applied to the switching element, the ON period can be changed to avoid this.

According to a seventh aspect of the present invention, in the fifth or sixth aspect of the present invention, a means for detecting a voltage of the first capacitor is provided, and the first and second switching elements are provided according to the voltage of the first capacitor. At least one of the drive frequency or the ON period is variable, so that when the smoothed voltage is abnormally boosted, the oscillation applied to the switching element is avoided by stopping oscillation or dimming. Or so that the smoothing voltage is substantially constant,
If the frequency or the ON period is controlled, a stable output can be obtained. For example, in the case of a discharge lamp, an optical output with less flicker can be obtained, and a smooth voltage is detected.
By shifting the power-on and control start times, it is possible to prevent the destruction of the circuit due to the rush current.

According to an eighth aspect of the present invention, in the second aspect of the present invention, a means for detecting a voltage generated in the transformer is provided, and the frequency of the first and second switching elements or the on-period of the first and second switching elements is determined according to the voltage of the transformer. Since at least one of them is variable, if the voltage across the load is abnormally boosted,
By stopping the oscillation or dimming, the stress applied to the switching element can be avoided, and when the load is short-circuited, the oscillation is stopped and applied to the switching element. There is an effect that stress can be avoided.

According to a ninth aspect of the present invention, in the fifth or sixth aspect of the present invention, a means for detecting the polarity of the AC power supply is provided, and the frequency of the first and second switching elements or the ON state of the first and second switching elements are determined according to the polarity of the AC power supply. Since at least one of the periods is variable, the input current can be adjusted every half cycle of the power supply, so that the input current can be made closer to a sine wave, input distortion can be further reduced, and control is performed with a constant switching waveform Since the output of the load can be made smaller than this, deeper dimming can be performed, and dimming can be performed while the smoothed voltage is kept substantially constant, so that an effect that abnormal stress is applied to the switching element can be prevented.

According to a tenth aspect of the present invention, in the fifth or sixth aspect of the present invention, there is provided means for respectively detecting voltages generated in the first and second diodes, and the first and second diodes are provided in accordance with the voltages.
And at least one of the drive frequency and the ON period of the second switching element is made variable, so that the asymmetry of the input current in each power supply half cycle can be improved, and the input current can be made closer to the sine. Has the effect of suppressing the peak value of, and preventing abnormal stress from being applied to the filter of the input section.

According to an eleventh aspect of the present invention, in the invention of the fifth or sixth aspect, means for detecting the voltage of the second capacitor is provided, and the voltage of the second capacitor is lowered according to the voltage of the second capacitor. By the way, lower the driving frequency or change the on-period in the direction to increase the load output,
Where the voltage of the second capacitor is high, the drive frequency is increased or the ON period is changed in a direction in which the load output becomes smaller, so that the output of the load can be kept substantially constant. In this case, there is an effect that an optical output with less flicker can be obtained.

According to a twelfth aspect of the present invention, in the fifth or sixth aspect, means for detecting a current flowing through a transformer or a load is provided so that the current flowing through the load becomes substantially constant in accordance with the detected current. Since at least one of the drive frequency and the ON period of the first and second switching elements is made variable, the load current can be made substantially constant even when the power supply fluctuates. In the case of an electric lamp, there is an effect that the pulsating flow of the lamp current can be reduced and an optical output with less flicker can be obtained.

According to a thirteenth aspect of the present invention, in the second, third or fourth aspect of the present invention, a means for varying the capacity of the second capacitor is provided. The input current according to the load can be adjusted. For example, if the load is a discharge lamp, the input current can be reduced even when dimming or when the output of the load is switched. is there.

According to a fourteenth aspect of the present invention, in the second, third, or fourth aspect of the present invention, power is supplied to a plurality of loads.
For example, if the load is a discharge lamp, there is an effect that a plurality of discharge lamps can be turned on simultaneously. The invention of claim 15 is
In the invention according to claim 2, the second load circuit is connected in parallel to one of the first and second switching elements or in parallel with the first capacitor. Can be made substantially constant without special control. For example, if the load is a discharge lamp, light output with less flickering can be obtained as a whole, and even if one of the lamps comes off, the remaining discharge lamps are turned on. Since each load can be output at an arbitrary ratio, for example,
Even when the power supply fluctuates, the load current as a whole can be made substantially constant, and the pulsating current of the lamp current can be reduced, and an optical output with less flicker can be obtained.

According to a sixteenth aspect, in the second or third or fourth aspect, at least one of the first and second switching elements is driven by another winding provided in a transformer or an inductor of a load. Since the drive circuit for the switching element can be omitted, a circuit with a smaller number of components can be realized. By driving from the secondary side inductor, for example, the circuit is automatically stopped in the event of an abnormality such as no load. Further, if the driving of one switching element is controlled by an inductor in the circuit and the driving of the other switching element is controlled by an external signal, there is an effect that control such as dimming and stopping can be easily performed.

A seventeenth aspect of the present invention is the invention according to the second, third or fourth aspect of the present invention, wherein a series circuit of fifth and sixth diodes is connected in parallel to the first capacitor, and the midpoint of the series circuit is connected to an alternating current. Since the fourth capacitor is connected between the load circuit and the connection point between the AC power supply and the second capacitor, the power supply is connected to the terminal of the power supply to which the first and second diodes are not connected. When the power supply differs, it is possible to use a common board with the addition of a small number of components and to keep the voltage applied to the load circuit almost constant, so that almost the same output can be obtained even if the power supply differs. If the components are further modularized, it is possible to cope with different types of power supplies only by combining the modules.

According to an eighteenth aspect of the present invention, in the second, third or fourth aspect of the present invention, the third and fourth switching elements which are alternately turned on and off at a high frequency are connected in parallel with the first and second diodes. In addition, there is an effect that surplus power can be regenerated to the power supply when the load suddenly decreases.

[Brief description of the drawings]

FIG. 1 is a circuit diagram according to a first embodiment of the present invention.

FIG. 2 is an operation explanatory view of the above.

FIG. 3 is an operation explanatory diagram of the above.

FIG. 4 is an operation explanatory view of the above.

FIG. 5 is an operation explanatory view of the above.

FIG. 6 is an operation explanatory view of the above.

FIG. 7 is a circuit diagram showing a specific example of the above.

FIG. 8 is a waveform chart for explaining the circuit operation of FIG. 7;

FIG. 9 is a waveform diagram for explaining the operation of the switching element of FIG. 7;

FIG. 10 is a circuit diagram showing another specific example of the above.

FIG. 11 is a circuit diagram showing another specific example of the above.

FIG. 12 is a circuit diagram showing another circuit example of the above.

FIG. 13 is a circuit diagram showing still another example of the circuit.

FIG. 14 is a circuit diagram showing still another example of the circuit.

FIG. 15 is a circuit diagram showing still another example of the above circuit.

FIG. 16 is a circuit diagram showing still another circuit example of the above.

FIG. 17 is a circuit diagram showing still another circuit example of the above.

FIG. 18 is a circuit diagram showing still another circuit example of the above.

FIG. 19 is a circuit diagram showing still another example of the circuit.

FIG. 20 is a circuit diagram showing another example of the above circuit.

FIG. 21 is a circuit diagram according to a second embodiment of the present invention.

FIG. 22 is an explanatory diagram of the above operation.

FIG. 23 is another operation explanatory view of the above embodiment.

FIG. 24 is another operation explanatory view of the above.

FIG. 25 is a circuit diagram of another example of the above.

FIG. 26 is a waveform chart for explaining the operation of the above.

FIG. 27 is an operation explanatory diagram of the third embodiment of the present invention.

FIG. 28 is another explanatory diagram of the above operation.

FIG. 29 is another operation explanatory view of the above.

FIG. 30 is a circuit diagram according to a fourth embodiment of the present invention.

FIG. 31 is a waveform chart for explaining the operation of the above.

FIG. 32 is a circuit diagram according to a fifth embodiment of the present invention.

FIG. 33 is a waveform diagram for explaining the above operation.

FIG. 34 is a circuit diagram showing another example of the above circuit.

FIG. 35 is a circuit diagram showing another example of the above circuit.

FIG. 36 is a circuit diagram showing another example of the above circuit.

FIG. 37 is a circuit diagram according to a sixth embodiment of the present invention.

FIG. 38 is a waveform diagram for explaining the above operation.

FIG. 39 is a circuit diagram showing another example of the circuit.

FIG. 40 is a circuit diagram according to a seventh embodiment of the present invention.

FIG. 41 is a circuit diagram showing another example of the above circuit.

FIG. 42 is a circuit diagram showing another example of the above circuit.

FIG. 43 is a circuit diagram according to an eighth embodiment of the present invention.

FIG. 44 is a circuit diagram showing another example of the above circuit.

FIG. 45 is a circuit diagram showing another example of the above circuit.

FIG. 46 is a circuit diagram of a load circuit example according to a ninth embodiment of the present invention.

FIG. 47 is a circuit diagram showing another example of the above circuit.

FIG. 48 is a circuit diagram showing another example of the above circuit.

FIG. 49 is a circuit diagram according to a tenth embodiment of the present invention.

FIG. 50 is a circuit diagram of a specific example of the above.

FIG. 51 is an explanatory diagram of another example of the above.

FIG. 52 is a circuit diagram of an eleventh embodiment.

FIG. 53 is a circuit diagram of a conventional example.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 Load circuit 2 Drive circuit Vs AC power supply D3 to D4 Diode Q1, Q2 Switching element C1 Capacitor C2 Capacitor

Claims (18)

[Claims] 1. A full-wave rectifier configured by bridge-connecting first to fourth diodes, a series circuit of an AC power supply and a load circuit connected between AC terminals of the full-wave rectifier, and a full-wave rectifier. A first smoothing capacitor connected between the DC output terminals, first and second switching elements connected in anti-parallel to each of the first and second diodes of one arm of the full-wave rectifier; A second terminal connected between one of the DC output terminals of the rectifier and a middle point of the arm of the full-wave rectifier to which the first and second switching elements are not connected.
Characterized in that the first and second switching elements are turned on and off alternately. 2. Both ends of a load circuit are formed by both ends of a primary winding of a transformer constituting a resonance system with a second capacitor;
The power supply device according to claim 1, wherein power is supplied to a load connected to a secondary winding of the transformer. 3. Both ends of a load circuit are constituted by both ends of a series circuit of a first inductor and a discharge lamp, and a non-power supply terminal of the discharge lamp is provided with a first inductor and a third capacitor constituting a resonance circuit. The power supply device according to claim 1, wherein the power supply device is connected. 4. The power supply device according to claim 1, wherein a second inductor is connected to both ends of the load circuit. 5. The power supply device according to claim 2, further comprising control means capable of arbitrarily changing the frequencies of the first and second switching elements. 6. The power supply device according to claim 2, further comprising a control unit capable of arbitrarily changing an on-period of the first and second switching elements. 7. A device for detecting a voltage of a first capacitor, wherein first and second voltages are detected in accordance with the voltage of the first capacitor.
6. The method according to claim 5, wherein at least one of the driving frequency and the ON period of the switching element is variable.
Or the power supply device according to 6. 8. A method for detecting a voltage generated in a transformer, wherein at least one of a frequency and an ON period of the first and second switching elements is variable according to the voltage of the transformer. The power supply device according to claim 2. 9. A system for detecting a polarity of an AC power supply, wherein at least one of a frequency and an ON period of the first and second switching elements is made variable in accordance with the polarity of the AC power supply. The power supply device according to claim 5. 10. A means for detecting voltages generated in the third and fourth diodes, respectively, wherein at least one of the drive frequency or the ON period of the first and second switching elements is variable according to the voltage. The power supply device according to claim 5, wherein 11. A means for detecting the voltage of the second capacitor is provided, and the drive frequency is reduced or the load output is turned on during the ON period when the voltage of the second capacitor is low in accordance with the voltage of the second capacitor. 7. The power supply according to claim 5, wherein the drive frequency is increased or the on-period is changed in a direction where the load output becomes smaller when the voltage of the second capacitor is high. apparatus. 12. A means for detecting a current flowing in a transformer or a load, wherein the drive frequency or the ON period of the first and second switching elements is adjusted so that the current flowing in the load becomes substantially constant in accordance with the detected current. 7. The power supply device according to claim 5, wherein at least one of them is variable. 13. The power supply device according to claim 2, further comprising means for varying the capacitance of the second capacitor. 14. The power supply according to claim 2, wherein power is supplied to a plurality of loads. 15. A second load circuit connected in parallel to one of the first and second switching elements or in parallel with the first capacitor.
The power supply as described. 16. The power supply device according to claim 2, wherein at least one of the first and second switching elements is driven by another winding provided on an inductor of a transformer or a load. . 17. A series circuit of fifth and sixth diodes is connected in parallel to a first capacitor, and a middle point of the series circuit is connected to a side of the AC power supply to which the first and second diodes are not connected. 5. The power supply device according to claim 2, wherein a fourth capacitor is connected between the terminal and a connection point between the AC power supply and the second capacitor and the load circuit. 18. The power supply device according to claim 2, wherein third and fourth switching elements which are alternately turned on and off at a high frequency are connected in parallel with the first and second diodes.