JP2885618B2 - Adaptive receiver - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an adaptive receiver, and more particularly to an adaptive receiver for performing diversity combining by a plurality of adaptive matched filters and removing intersymbol interference by a decision feedback equalizer.

[0002]

2. Description of the Related Art Conventionally, there has been known an adaptive receiver for performing diversity combining using a plurality of adaptive matched filters and removing intersymbol interference using a decision feedback equalizer (Kojiro Watanabe: "Adaptation in a multipath transmission path"). Receiving method ", IEICE, Communication System Study Group, February 1979 (CS78
-203)). The conventional adaptive receiver, as shown in FIG. 2, N number of adaptive matched filter (AMF) 40 ₁ ~40 _N
When, these N adaptive matched filter 40 ₁ to 40 and combiner 20 for combining the outputs signals of the _N, a decision feedback equalizer output composite signal of the combiner 20 is input (DFE)
Becomes more and 30 are configured to control the tap coefficients of the adaptive matched filter 40 ₁ to 40 _N by the output signal of the decision feedback equalizer 30.

[0003] The adaptive receiver is for performing optimal reception at multipath channel power limit based, the adaptive matched filter 40 ₁ to 40 _N for each diversity branch, the signal-to-noise power ratio of the baseband input signal ( After maximizing the signal-to-noise ratio (SN ratio), the combiner performs maximum ratio combining to enhance the signal, and the decision feedback equalizer 30 can remove residual intersymbol interference after diversity combining by the combiner 20. Digital transmission over severe multipath fading lines. This adaptive receiver has already been put to practical use in tropospheric scattering communication, which is a typical multipath line.

The adaptive matching filters 40 ₁ to 40 _N forming the main parts of the adaptive receiver have the same configuration, and are cascaded as shown in FIG. 2, each having a delay time of T / 2 (T is a symbol period). a delay element 41 ₁ to 41 _4, the complex multiplier complex multiplier 42 ₁ to 42 ₄ and the delay element 41 ₁ of the output signal the input signal of the delay element 41 ₁ to 41 ₄ are inputted to the branch is input 42 and _5, these complex multiplier 42
Combiner 43 for combining _{1-42 5} output signals, respectively
And a tap coefficient correction circuit 44.

Next, the adaptive matched filters 40 ₁ to 40
The operation of _N will be described with reference to FIG. In FIG.
The same components as those described above are denoted by the same reference numerals. 3, 41 delay elements corresponding to the delay elements 41 ₁ to 41 _4, 42 is a complex multiplier that corresponds to the complex multiplier 42 ₁ to 42 _4. 3A shows an example of a transmission path impulse response, FIG. 3B shows an example of a tap coefficient distribution of an adaptive matched filter, and FIG. 3C shows an example of a transmission impulse response at the output of the adaptive matched filter. FIG. 4D shows an example of a tap coefficient distribution of the adaptive matched filter when the reference tap of the adaptive matched filter is shifted, and FIG. 4E shows the transmission impulse response of the transmission matched filter output at the output of the adaptive matched filter when the reference tap is shifted. One example is shown below.

Now, the impulse response of the transmission line is shown in FIG.
Assuming that it is as shown in (A), the transversal filter composed of the delay element 41, the complex multiplier 42, and the combiner 43 in this case, that is, the adaptive matching filter, has the time of the impulse response of FIG. The tap coefficient distribution shown in FIG. 11B, which is an inverted complex conjugate response, is estimated. The response of complex conjugate time reversal, 3 a sampling value of T / 2 interval of the impulse response of (A) h _-2,
Similarly, assuming that h ₋₁ , h ₀ , h ₁ , and h ₂ , sampling values of the T / 2 interval of the tap coefficient distribution of the adaptive matched filter are W ₋₂ , W ₋ , as shown in FIG. _{1, W 0, W 1,} W
_{If it is 2} , it is expressed by the following equation.

W _-2 = h ₂ ^* (1) W _-1 = h ₁ ^* (2) W ₀ = h ₀ ^* (3) W ₁ = h _-1 ^* (4) W ₂ = h _-2 ^* ( 5) In the above formula, * represents a complex conjugate. The main response of the transmission path impulse response is h ₀ at time t = ₀ , and the energy is maximum as shown in FIG.
The corresponding adaptive matched filter response is W ₀ in FIG. 3B, which is the central complex multiplier (42 _{3 in} FIG. 2) of the plurality of complex multipliers 42 as the tap coefficient of the adaptive matched filter. ). This center tap is referred to as a reference tap because it serves as a reference timing of the impulse response. Similarly, other sampling values W _i (i = −
2, -1, 1) is also multiplied by a tap coefficient in the corresponding order of the complex multiplier of the plurality of complex multiplier 42 (42 _i in FIG. 2).

Thus, a time-inverted complex conjugate response (FIG. 3A) is added to the impulse response (FIG. 3A) of the input transmission line.
The convolution of (B)) is called matched filtering in communication theory, and the impulse response at the output of the adaptive matched filter has a symmetric shape as shown in FIG. That is, the time-dispersed impulse response (FIG. 3A) converges to the reference time (t = 0), and the signal is strengthened. This is the operation of maximizing the SN ratio by the adaptive matched filter.

[0009] The first point in matched filtering is to estimate its tap coefficient, and the usual method is as follows.

First, a decision data signal is input to a replica filter, and a received input signal is estimated. Next, an error between the estimated signal and the actual input signal is obtained, and the tap coefficient of the matched filter is estimated by an algorithm that minimizes the root mean square. However, each of the adaptive matched filters 40 ₁ to 40 _{N in} the adaptive receiver shown in FIG. 2 uses a correlation method based on a decision data signal instead of using a replica filter in order to more easily estimate a matched filter tap coefficient. Used.

That is, to explain the correlation method, the transmitting side converts a transmission symbol sequence {S _n } to _.
When transmitted in the order of S ₋₁ , S ₀ , S ₁ , S ₂ ,... Every T period, the received signal is a convolution operation of this transmission symbol sequence and the transmission path impulse response (FIG. 3A). Given by Received signal r _-2 on each tap of the adaptive matched filter 40 ₁ in FIG. 2 at this _{time, r -1, r 0, r} 1, r 2 is expressed as follows.

R ₋₂ =. . . _{+ H 2 · S 0 + h} 0 · S 1 + h -2 · S 2 +. . . (6) r _-1 =. . . + H ₃ · S _-1 + h ₁ · S ₀ + h _-1 · S ₁ +. . . (7) r ₀ =. . . + H ₂ · S _-1 + h ₀ · S ₀ + h _-2 · S ₁ +. . . (8) r ₁ =. . . + H ₁ · S _-1 + h _-1 · S ₀ + h _-3 · S ₁ +. . . (9) r ₂ =. . . + H ₂ · S _-2 + h ₀ · S _-1 + h _-2 · S ₀ +. . . (10) The tap coefficient correction circuit 44 of FIG.
The received signal r _i on the tap (i = −2, −1, 0, 1, 1)
2) and a correlation operation is performed on the decision data signal S from the decision feedback equalizer 30, and the obtained correlation value is used as a tap coefficient of the corresponding tap. That is, the i-th tap coefficient Wi is represented by the following equation.

W _i = E [r _i ^* · S] (11) Here, E [] indicates a time average (expected value) process, and *
Represents a complex conjugate. Further, when the decision data signal S from the decision feedback equalizer 30 is S ₀ at time t = 0, the tap coefficients W _i The above is expressed by the following equation.

[0014] W _i = substituting ^{_{E [r i * · S 0}} ] (12) wherein in the above (12) (6) to (10), and "1" autocorrelation power for S, When the calculation is advanced with the autocorrelation at a distance of one symbol or more set to “0”, the following equation is obtained.

W−2 = E [(... + H2 S0 + h0 S1 + h−2S2 +...) * · S0] = h2 * (13) W−1 = E [(... + H3 S−1 + h1 S0) + H-1S1 + ...) *. S0] = h1 * (14) W0 = E [(... + h2 S-1 + h0 S0 + h-2S1 + ...) *. S0] = h0 * (15) W1 = E [(... + h1S-1 + h-1S0 + h-3S1 + ...) *. S0] = h-1 * (16) W2 = E [(... + h2 S-2 + h0 S-1 + h-2S0) + ..) *. S0] = h-2 * (17) The result of the correlation operation of the above equations (13) to (17) indicates the tap coefficient for operating as a matched filter from the equation (1). It can be seen that it matches the equation (5). Therefore,
Even if a replica filter is not used, the tap correction circuit 44
The tap coefficient of the adaptive matching filter is obtained by the correlation operation equation (Equation (11)).

The second point in matched filtering is that it has a timing control function. It is well known that this timing function is realized by setting the transversal filter at fractional intervals. Here, indicates that the fractionally spaced sets the delay time of the delay element 41 ₁ to 41 ₄ of FIG. 2 in half the symbol period T.

The timing control function refers to adjusting the timing of a received signal that fluctuates on the propagation path to the clock timing phase of the receiver. Specifically, the transmitted symbols propagate through independent diversity branches. The propagation path of each branch fluctuates with time,
The propagation delay time fluctuates every moment. Therefore, the reception timings of the received symbols of each diversity branch undergo independent fluctuations and do not match.

Therefore, if the diversity combining is performed as it is, the main responses of the branches do not match, and the combined impulse response adds not only the delay dispersion due to multipath but also the dispersion due to the timing deviation between the branches. . This means that the combined impulse response deviates further from the Nyquist distortion-free condition. Therefore, it is necessary to absorb the timing deviation between the branches. Accordingly, the adaptive matched filter 40 ₁ to 40 _N of FIG. 2 is a fractionally spaced transversal filter absorbs the timing phase shift by the following operation.

[0019] On each tap that is composed of delay elements 41 ₁ to 41 _4, the received signal r _-2, r _-1, are distributed in the order of _{r 0, r 1, r 2} . These are received signals sampled at T / 2 intervals from each other. Therefore, when focusing on the received signal r ₀ of the center tap, the symbol S _{0 in the} equation (8) is used.
Is also distributed to the received signals r ₋₁ and r ₁ on the front and rear taps as shown in the equations (7) and (9).

Here, the received signal r _{-1 represented} by the equation (7)
S ₀ in is only a time delay T / 2 than S ₀ in the received signal r ₀ of the center tap. On the other hand, S ₀ in the received signal r _{1 represented} by the equation (9) is the received signal r of the center tap.
It has advanced only in time T / 2 than the S ₀ in _0. In this case, if the tap coefficient W ₁ is larger than the tap coefficient W ₀ , the output of the combiner 43 can advance the timing of S ₀ equivalently. Conversely, tap coefficient W _-1
Is larger than W ₀ , S ₀ that is temporally delayed from the center tap is output from the combiner 43, so that the timing of S ₀ can be equivalently delayed.

Therefore, the adaptive matched filter controls the timing by utilizing this property, matches the reception timing of the received symbol of each diversity branch, and enables effective diversity combining.

By the way, the tap coefficients W _i for the timing control is obtained by correlation method of the first point. In the correlation method described above, the decision feedback equalizer 30 is used.
Is used. This determination data S
Is based on the timing of the clock recovery circuit built in the decision feedback equalizer 30. Therefore, the correlation processing of the adaptive matching filter is governed by the reception clock extracted from the decision data S, that is, the timing of the clock phase of the clock recovery circuit in the decision feedback equalizer 30.

This will be described with reference to FIG. 3. The rising timing of the reception clock is represented by the time t = t in FIG.
Corresponds to 0. In addition, as shown in FIG. 3C, the main response of the matched filter output during the normal operation is equal to the reference timing (t) as shown in FIG. =
0). This operation is exactly the maximization of the S / N ratio by the adaptive matched filter, and at the same time, it is a timing function for adjusting the timing to the reference timing (t = 0).

[0024]

However, the conventional adaptive receiver described above feeds back the decision data retimed by the reception clock as described above, and
₁ to 40 _N, and is controlled by the clock phase of the clock recovery circuit in the decision feedback equalizer 30, so that a feedback system using a clock as a medium is configured. When the clock phase of the circuit is slightly shifted due to fading or other causes,
Clock phase determination data S is fed back to the adaptive matched filter 40 ₁ to 40 _N be slightly shifted, thereby shifting the adaptive matched filter 40 ₁ to 40 _N respectively of the timing control.

This deviation of the timing control causes the reference tap coefficient W _0, which is the main response, to be slightly smaller, and _one of the preceding and succeeding tap coefficients W ₋₁ and W ₁ is changed from the original value. Is also slightly larger. Then, a time t = 0 the main response of the transmission system the impulse response at the output of the adaptive matched filter 40 ₁ to 40 _N are shown in FIG. 3 (C)
Is slightly shifted in either direction. Then, this shift is input to the clock recovery circuit inside the decision feedback equalizer 30, and the received clock phase is shifted.

As described above, in the conventional adaptive receiver, since the feedback loop system using the clock as a medium is configured, the deviation of the clock phase of the clock recovery circuit propagates through the loop and is returned to the clock recovery circuit again. An increase in clock phase shift occurs. Adaptive matched filter 4
Since each of 0 ₁ to 40 _N has a tap formed at an interval of T / 2, the timing control can perform an endless operation. Therefore, the phenomenon of increasing the clock phase shift is as follows.
The main response may continue until it settles at one end of the matched filter.

FIG. 3D shows an example of the transmission impulse response at the output of the adaptive matched filter when the clock phase is shifted by T due to the clock phase shift described above.
As shown in FIG. 3D, a case where the reference tap is located not at the center of the transversal filter but at W- ₂ of the first tap is shown.

In this case, the impulse response in FIG. 3D is h ^* (-t + T), which is shifted by T from the original time-reversed complex conjugate h ^* (-t) for the adaptive matched filter. Therefore, the trailing edge of the channel response (Postcurso
Information of h _1, h ₂ is r) is not included in the impulse response of FIG. 3 (D), the inability multiplying by the complex multiplier. For this reason, correct matched filtering is not performed, and as shown in FIG. 3E, symmetry of the impulse response and convergence of the time dispersion energy are not performed in the output of the adaptive matched filter. This operation creates more adverse effects than the effects of multipath distortion due to propagation.

As described above, in the conventional adaptive receiver, a feedback system using the reception clock phase as a medium is configured.
Adaptive matched filter 40 ₁ to 40 _N reference tap position of the adaptive matched filter 40 ₁ to 40 _N by the interaction of the timing functions may become unstable, and the shift from the center tap,
There is a problem that a normal matched filter function may be impaired.

The present invention has been made in view of the above points,
An object of the present invention is to provide an adaptive receiver which solves the above-mentioned problem by providing a complex multiplier at a reference tap position at a final output stage of an adaptive matched filter.

[0031]

SUMMARY OF THE INVENTION In order to achieve the above object, the present invention provides a method of synthesizing a signal having a maximum signal-to-noise ratio by a plurality of adaptive matched filters supplied with separately received input signals. Input to a decision feedback equalizer through a filter, extracts decision data from which intersymbol interference has been removed by the decision feedback equalizer, and corrects tap coefficients of the plurality of adaptive matched filters using the decision data. In the receiver, each of the plurality of adaptive matched filters includes a delay unit, a first multiplication unit, a synthesis unit, a second multiplication unit, and a tap coefficient correction circuit.

Here, the delay means converts the input signal to T /
2 (where T is the symbol period of the input signal) and outputs a plurality of delayed signals having different delay times. In addition, the first multiplying means includes a first delay signal corresponding to the input signal and a delay signal other than the delay signal of the reference tap having a median delay time among the input signal and the plurality of delay signals. Are multiplied separately. Also,
The combining means combines the output signal of the first multiplying means and the delay signal of the reference tap. The second multiplying means multiplies the output signal of the combining means by a second tap coefficient to be multiplied by the delay signal of the reference tap, and outputs the result to the combiner. Further, the tap coefficient correction circuit performs the first operation by performing a correlation operation between the determination data and the plurality of delay signals.
And the second tap coefficient are sequentially calculated.

[0033]

In the present invention, the second tap coefficient to be multiplied by the delay signal of the reference tap among the plurality of delay signals from the delay means and the input signal is determined by the output of the synthesis means by the second multiplication means. By multiplying the combined signal, the maximum ratio combining of each diversity branch is realized. Further, a first tap coefficient obtained by normalizing a conventional tap coefficient with the second tap coefficient with respect to a delayed signal at a tap position other than the reference tap among the plurality of delayed signals by the first multiplying means is used. By multiplying, matched filtering in each diversity branch is realized.

Accordingly, in the present invention, the first tap coefficient multiplied by the first multiplying means can be smaller than the second tap coefficient multiplied by the second multiplying means. The reference tap position can always be held at the center tap position of the adaptive matched filter.

[0035]

Next, an embodiment of the present invention will be described.
FIG. 1 shows a configuration diagram of an embodiment of the present invention. In FIG.
The same components as those described above are denoted by the same reference numerals, and description thereof will be omitted. In FIG. 1, N adaptive matched filters 10 ₁ to 10 _N (N is a natural number of 2 or more, usually a power of 2) are N
The input signals are provided in one-to-one correspondence. This input signal is, for example, a baseband signal obtained by separately frequency-converting signals received by N reception antennas in the microwave band.

The adaptive receiver maximizes the signal-to-noise power ratio (SN ratio) of the baseband input signal by the adaptive matching filters 10 ₁ to 10 _N of the diversity branches, and then performs maximum ratio combining by the combiner. , The residual inter-symbol interference after the diversity combining by the combiner 20 is removed by the decision feedback equalizer 30, and the data (decision data) that is diversity combined and demodulated from the decision feedback equalizer 30. the basic operation is configured to output is the same as the conventional configuration of the adaptive matched filter 10 ₁ to 10 _N is different from the conventional adaptive matched filter 40 ₁ to 40 _N.

[0037] For the adaptive matched filter 10 ₁ to 10 _N are each identical construction and will be representatively described adaptive matched filter 10 _1, as shown in FIG. 1, the adaptive matched filter 10 ₁ is connected in cascade, each Delay time T / 2
(T is the symbol period) and the delay element 11 ₁ to 11 _4, the complex multiplier 12 ₁ and 12 ₂ the input signal of the delay element 11 ₁ and 11 ₂ are inputted by branching, input and output of the delay element 11 ₄ Complex multipliers 12 ₃ and 1 to which signals are respectively input
2 ₄ and, synthesizer 13 and, synthesizer 13 complex multiplier 1 provided on the output side of synthesizing these complex multiplier 12 ₁ to 12 ₄ of the output signal and the delay element 11 _{and second} output signals and respectively
4 and a tap coefficient correction circuit 15.

That is, the adaptive matched filter 1 of this embodiment
0 ₁ , the delay elements 111 to ₁ constituting the delay means
1 wherein the _fourth input signal or output signal is input first
The complex computing unit constituting the multiplying means consist 12 ₁ to 12 _4, Izu provided in the center tap to be the reference tap, reference tap the output signal of the delay element 11 ₂ is a delayed signal of the constantly intact synthetic Therefore, the reference tap is secured.

However, if the complex multiplier is removed from the reference tap as it is, the signal corresponding to the reference tap cannot be controlled in amplitude and phase, so that not only does it not function as a matched filter, but even the maximum diversity combining is performed. Impossible.

Therefore, in this embodiment, instead of removing the complex multiplier from the reference tap, the complex multiplier 14 is provided at the final output stage of the adaptive matched filter, that is, at the output side of the synthesizer 13. This complex multiplier 14 constitutes the second multiplying means, and the second tap coefficient W _{0 to} be multiplied is obtained by the tap coefficient correction circuit 15 by the following equation.

W ₀ = E [r ₀ ^* · S] (18) (where, r ₀ is a delay signal of the reference tap) Accordingly, when focusing only on the reference tap, the conventional amplitude and phase control is performed. Becomes possible.

However, since the complex multiplier 14 multiplies the combined signal output from the combiner 13 by the tap coefficient W _{0 of the} equation (18), the above-described tap coefficient is also applied to signals from taps other than the reference tap. W ₀ is multiplied in common. In other words, the signal of each tap is multiplied by a tap coefficient obtained by multiplying the tap coefficient W _k (k = −2, −1, ₁ , 2) of the adaptive matching filter 10 ₁ by W ₀ , and a normal adaptive Matching filtering becomes impossible.

Therefore, in this embodiment, the complex multipliers 12 ₁ to 12 ₁ .
First tap coefficient W _k should multiplied by 12 ₄ is calculated based on the following equation by the tap coefficient correction circuit 15.

[0044] ^{_{W k = E [r k *}} · S] / W 0 (19) As can be seen from the above equation, the first tap coefficient W _k are normalized by the second tap coefficients W _0.

[0045] Thus, for the reference tap other signals, the complex multiplier 12 ₁ to 12 ₄ (19) the tap coefficient W _k of the formula is multiplied by the tap coefficients W further in the complex multiplier 14 _By being multiplied by ₀ , it is finally equivalent to being multiplied by a tap coefficient for normal adaptive filtering.

The above-described tap coefficient W ₀ is a correlation value between the delay signal r ₀ of the reference tap and the determination data S, as shown in the equation (18), and originally includes the main response component of the transmission path impulse response. Therefore, this correlation value is larger than the value based on the other tap coefficients. Further, the tap coefficients W _k is a value of the correlation values normalized by W ₀ of the determination data S and the signal r _k other than such, reference tap shown in equation (19). Therefore, the output signal of the complex multiplier 12 ₁ to 12 _4, the amplitude is smaller than the delay signal r ₀ of the reference tap.

[0047] This is a reference tap position is always held at the center tap position of the adaptive matched filter 10 _1, also the front edge of the impulse response other than the other tap main response (Pre
(cursor) or trailing edge. Therefore, according to the present embodiment, even if the phase of the output clock of the clock recovery circuit in the decision feedback equalizer 30 is slightly shifted, the reference tap position does not shift from the center position of the adaptive matching filter. Since the main response of the impulse response of the output of the adaptive matched filter can always be held at the reference timing position, the above-described phenomenon of increasing the clock phase shift can be prevented.

[0048]

As described above, according to the present invention,
Even if the phase of the output clock of the clock recovery circuit in the decision feedback equalizer is slightly shifted, the reference tap position can always be held at the center tap position of the adaptive matching filter, so that the phenomenon of increasing the clock phase shift is reduced. This can be prevented beforehand, so that stable adaptive matching filtering can always be performed. Therefore, according to the present invention, stable adaptive equalization and timing control can be performed even for severe multipaths such as minimum phase transition or non-minimum phase transition fading, and even in severe multipath fading lines, it is even more than before. A stable receiving operation can be performed.

[Brief description of the drawings]

FIG. 1 is a configuration diagram of an embodiment of the present invention.

FIG. 2 is a configuration diagram of a conventional example.

FIG. 3 is a diagram for explaining an operation of an adaptive matching filter and a conventional problem.

[Explanation of symbols]

10 ₁ to 10 _N Adaptive matched filter 11 _{1 to} 11 ₄ Delay element 12 _{1 to} 12 ₄ First complex multiplier 13, 20 Synthesizer 14 Second complex multiplier 15 Tap coefficient correction circuit 30 Decision feedback equalizer

Claims (2)

(57) [Claims] 1. A signal in which a signal-to-noise ratio is maximized by a plurality of adaptive matched filters to which input signals received separately are supplied, is input to a decision feedback equalizer through a synthesizer, and the decision feedback An adaptive receiver configured to take out the decision data from which the intersymbol interference has been removed by the type equalizer and correct the tap coefficients of the plurality of adaptive matched filters by the decision data, wherein each of the plurality of adaptive matched filters is Delay means for delaying the received input signal by T / 2 (where T is the symbol period of the input signal) and outputting a plurality of delay signals having different delay times from each other; Of the plurality of delay signals and input signals from
First multiplying means for separately multiplying the input signal by a corresponding first tap coefficient with respect to the other delayed signals except for the delayed signal of the reference tap having a median delay time, and the first multiplying means; Synthesizing means for synthesizing the output signal of the reference tap and the delay signal of the reference tap, respectively; multiplying the output signal of the synthesis means by a second tap coefficient to be multiplied by the delay signal of the reference tap, and outputting the result to the synthesizer Second multiplying means, and the first and second signals for maximizing a signal-to-noise ratio of an input signal received by a correlation operation between the determination data and the plurality of delay signals and the input signal . And a tap coefficient correction circuit for sequentially calculating tap coefficients. 2. The tap coefficient correction circuit according to claim 1, wherein a signal rk at a tap position k (where k is a positive or negative integer other than 0) by said delay means is Wk = E [rk * .S] / W0.
(Where E [] is a time averaging process, * is a complex conjugate, S is the determination data, and W0 is the second tap coefficient.) 2. The adaptation according to claim 1, wherein the second tap coefficient W0 is obtained based on an equation W0 = E [r0 * .S] (where r0 is a delay signal of a reference tap). Receiving machine.