JP2020028211A - Phase estimation device for synchronous reluctance motor - Google Patents

Abstract

To provide a phase estimation device for sensorless vector control of a synchronous reluctance motor, particularly, a phase estimation device for generating no phase error in principle.SOLUTION: A phase estimation device 10a is configured to: filter a differential equivalent value of a mirror phase magnetic flux having a mirror phase relationship with a stator current by a 2×2 stable filter 10a-1 showing a phase displacement of π/2 [rad] at a frequency corresponding to an electric speed of an electric motor; and extract and determine a rotor phase from a processed signal with a phase determiner 10-a2 using mirror phase characteristics.SELECTED DRAWING: Figure 4

Description

  The present invention relates to a rotor phase estimating device essential for sensorless vector control of a synchronous reluctance motor. In particular, the present invention relates to a rotor phase estimating apparatus which does not have a phase delay and a phase advance in principle.

In the present invention, a two-dimensional plane is viewed in polar coordinates, and three terms of angle, spatial position, and spatial phase are used synonymously. These units are "radians (rad)" or "degrees." In the present invention, the positive direction of the angle, the spatial position, and the spatial phase may be defined to be leftward (counterclockwise) or rightward (clockwise). However, in the present specification, in order to maintain the simplicity of the description, the positive direction of the angle, the spatial position, and the spatial phase is defined as counterclockwise (counterclockwise), and the present invention will be described. Thereby, the generality of the present invention is not lost.

As the name of the magnetic flux generated by the stator current of the synchronous reluctance motor, three terms of “stator magnetic flux”, “stator interlinkage magnetic flux”, and “stator reaction magnetic flux” are used synonymously. In principle, the stator flux, which is a simple name, is used.

In the present invention, the angular velocity and the angular frequency in the unit [rad / s] are also simply referred to as the velocity and the frequency, respectively. Further, the two-input two-output indicating the number of input terminals and the number of output terminals of the filter is simply abbreviated as “2 × 2”. Similarly, 2 rows and 2 columns indicating the size of the matrix are abbreviated as “2 × 2”, and 2 rows and 1 column indicating the size of the vector are abbreviated as “2 × 1”.

As known to those skilled in the art, there are an electric speed and a mechanical speed in the rotation speed of the rotor. There is a strict one-to-one relationship between the two velocities, allowing a unique conversion from electrical to mechanical speed and from mechanical to electrical speed. In the present invention, the rotor speed is used to mean the electric speed, in consideration of the well-known to those skilled in the art, unless the clarity of the description is lost.

In the present invention, the term "equivalent value" is used for a signal. The equivalent value is a general term for a true value, an approximate value, an estimated value, a command value, and the like of the signal.

In order for a synchronous reluctance motor to exhibit high control performance, control of stator current is indispensable, and a vector control method has conventionally been known as a control method for this purpose. The vector control method requires information on the phase of the forward salient pole (positive salient pole) or the phase of the reverse salient pole (negative salient pole) of the rotor. These phases are generally called rotor phases. Consider FIG. FIG. 3A employs a forward salient pole (positive salient pole) phase, and FIG. 2B employs a reverse salient pole (negative salient pole) phase. In both figures, three coordinate systems of a dq synchronous coordinate system, an αβ fixed coordinate system, and a γδ general coordinate system are drawn as typical two-axis orthogonal coordinate systems for the synchronous reluctance motor. The γδ general coordinate system is a highly general coordinate system that includes the dq synchronous coordinate system and the αβ fixed coordinate system as special cases. The rotor phase based on the α axis is expressed as “θα”, and the rotor phase based on the γ axis is expressed as “θγ”. Vector control of a synchronous reluctance motor basically requires a rotor phase “θα” based on the α-axis. The speed of the dq synchronous coordinate system synchronized with the forward or reverse salient pole of the rotor is the same as the electric speed ω2n of the rotor. Instead, the speed ωγ of the γδ general coordinate system is arbitrary.

In the present invention, a coordinate system aiming at convergence with a certain phase difference (including a zero phase difference) to the dq synchronous coordinate system is referred to as a “γδ rotation coordinate system”. A coordinate system aiming at convergence without a phase difference to the dq synchronous coordinate system is particularly called a “γδ quasi-synchronous coordinate system”. The γδ semi-synchronous coordinate system is also a coordinate system included in the γδ rotation coordinate system. Further, the γδ quasi-synchronous coordinate system and the γδ rotation coordinate system are also included as special cases of the γδ general coordinate system, like the dq synchronous coordinate system and the αβ fixed coordinate system.

Synchronous reluctance motors have essential features such as being suitable for high-speed rotation, robust and excellent in environmental resistance, inexpensive, and highly recyclable. In order to obtain the desired performance utilizing the characteristics of the synchronous reluctance motor, this drive is performed by the vector control accompanied with the control of the stator current. Vector control of a synchronous reluctance motor requires information on the rotor phase, and a position / speed sensor such as an encoder has conventionally been mounted on the rotor to obtain this information. However, mounting the position / speed sensor on the rotor impairs the features of the synchronous reluctance motor such as robustness and low cost, and further increases the motor volume in the axial direction. In order to take advantage of the features of the synchronous reluctance motor, driving by sensorless vector control is desirable.

The core of sensorless vector control lies in estimating the rotor phase. The method of estimating the rotor phase of a synchronous reluctance motor is roughly divided into a low-speed region and a medium-high speed region. The latter is more important as a rotor phase estimation method for a synchronous reluctance motor suitable for high-speed rotation. The medium-to-high speed phase estimation method uses a driving voltage / current, and should be called a driving voltage / current utilization method. According to the driving voltage / current utilization method of the prior invention, a part (Patent Documents 1 and 2) or all (Patent Documents 3 and 4) of a stator magnetic flux is estimated and the estimated value is processed to obtain a rotor phase estimated value. And a method of estimating the extended induced voltage and processing the estimated value to obtain a rotor phase estimated value.

The estimation method of estimating “a part of the stator magnetic flux” and processing the estimated value to obtain the rotor phase estimated value is further subdivided from the viewpoint of the “part of the stator magnetic flux” to be estimated. Patent Literature 1 selects “mirror phase magnetic flux” as a part of the stator magnetic flux to be estimated. Instead, Patent Literature 2 selects “salient magnetic flux” as a part of the stator magnetic flux to be estimated. According to the present invention, similarly to Patent Document 1, a mirror phase magnetic flux is selected as “a part of the stator magnetic flux” to be estimated, and the mirror phase magnetic flux estimated value is processed to obtain a rotor phase estimated value.

The prior invention described in Patent Literature 1, which selects “mirror phase magnetic flux” as “part of stator magnetic flux” to be estimated and processes the mirror phase magnetic flux estimate to obtain a rotor phase estimate, The rotor phase estimation can be performed on either the αβ fixed coordinate system or the γδ quasi-synchronous coordinate system. ” However, the estimation of the mirror magnetic flux requires the estimation of the stator magnetic flux, and the estimation of the stator magnetic flux requires an unstable integrator that performs a pure integration process (Equation (24) of Patent Document 1, FIG. 7). See). In practice, an unstable pure integral was not available, and this had to be replaced by a stable approximate integral (see the description under equation (24) in Patent Document 1). However, as is well known to those skilled in the art, "approximate integration causes a phase error in a low-speed region where an approximation effect is expected, in principle." As a result, in the rotor phase estimation of the synchronous reluctance according to the prior invention, there was a problem / problem that "in a low-speed region, even if ideal conditions are satisfied, a phase error occurs in principle". . In the rotor phase estimation, the estimation characteristic that "a phase error occurs in principle even in the case where ideal conditions are satisfied in a low-speed range" is a major problem or problem.

Shinji Shinnaka: "Vector Control Method for Synchronous Reluctance Motor", JP-A-2001-268998 (2000-3-7) Shinji Shinnaka: "Vector control method and device for synchronous reluctance motor", JP-A-2006-230174 (2005-2-14) Condaluho Auger-Yun Richun Foon: "Rotation Speed Control Device and Method for Synchronous Reluctance Motor", JP-A-2003-33096 (2002-7-10) Wonjung Hee Ougeyon Rikkun Horn Chondalho: "Magnetic Flux Measuring Device for Synchronous Reluctance Motor and Sensorless Control System Thereof", JP-A-2004-120993 (2003-6-26)

Shinji Shinnaka: "Vector Control Technology for Permanent Magnet Synchronous Motors, Lower Volume (the essence of sensorless drive control)", Dempa Shimbun (2008) Shinji Shinnaka: "Vector Control Technology for Permanent Magnet Synchronous Motors, Volume 1 (From Principle to Cutting Edge)", Dempa Shimbun (2008)

The present invention has been made under the above background, and an object of the present invention is to solve the above-described problems and problems in sensorless driving of a synchronous reluctance motor. More specifically, it is a phase estimating device that selects the “mirror phase magnetic flux” as the “part of the stator magnetic flux” to be estimated and processes the mirror phase magnetic flux estimated value to obtain a rotor phase estimated value, “In principle, even in a low-speed range, it is an object of the present invention to provide a phase estimating apparatus in which a phase error does not occur.

In order to achieve the above object, the invention according to claim 1 provides a stator current that contributes to torque generation on a γδ rotating coordinate system composed of an orthogonal γ axis and a δ axis indicated by a vector rotator. A phase estimating apparatus for generating a phase to be used for a vector rotator in a synchronous reluctance motor driving apparatus including a current control step of dividing and controlling an axis component and a δ axis component, wherein sgn (x) is When a sign function that extracts only the polarity of x, which is a variable, and the equivalent value of the electric speed of the synchronous reluctance motor is ω2n ＾ [rad / s], the stator current as a 2 × 1 vector amount contributing to torque generation A differential equivalent value of a mirror phase magnetic flux as a 2 × 1 vector quantity having a mirror phase relationship is subjected to filtering processing, and a phase displacement of −sgn (ω2n ＾) π / 2 [rad] at an angular frequency of ω2n ＾. And a stable two-input two-output filter, wherein at least a signal after the filtering process by the two-input two-output filter is used to generate a phase to be used in the vector rotator. .

A second aspect of the present invention is the phase estimating apparatus according to the first aspect, wherein s is a differential operator, ai is a scalar coefficient of an nth-order stable polynomial of the following equation (1),
g1 is a scalar coefficient, the speed of the two-axis orthogonal coordinate system in which the 2 × 1 mirror phase magnetic flux is defined is ωγ, and a D factor that is a 2 × 2 matrix is defined by the following equation (2):
A polynomial with the D factor is defined by the following equation (3),
Further, when G is a filter gain of a 2 × 2 matrix, the two-input two-output filter is expressed by the following equation (4).
Or the following equation (5)
The filter is characterized by the following.

In the present invention, the control of the stator current is performed on the γδ rotating coordinate system, as repeatedly described in the “claims” section and the “means” section of the present invention. Alternatively, the rotor phase estimation according to the invention can be performed on any two-axis Cartesian coordinate system encompassed by the γδ general coordinate system. The rotor phase estimation according to the present invention can be performed on any of the γδ quasi-synchronous coordinate system, the γδ rotating coordinate system, and the αβ fixed coordinate system. Considering the generality of the present invention with respect to the coordinate system, the effect of the present invention will be described using a signal defined on the γδ general coordinate system.

The mathematical model (circuit equation, torque generation equation) of the synchronous reluctance motor on the γδ general coordinate system is given by the following equation, particularly when iron loss can be ignored.

The 2 × 1 vectors v1, i1, and φ1 in the above equation are the voltage, current, and magnetic flux of the stator defined on the γδ general coordinate system. R1 is a winding resistance. Li and Lm are an in-phase inductance and a mirror-phase inductance, respectively, having the following relationship with the d-axis inductance Ld and the q-axis inductance Lq.
Generally, when the forward salient pole phase is the d-axis phase as shown in FIG. 1A, the mirror phase inductance Lm is positive, and when the reverse salient pole phase is the d-axis phase as shown in FIG. , The mirror phase inductance Lm becomes negative. The definition of a Q matrix (called a mirror matrix) that is a 2 × 2 matrix is as follows.
Further, τ is the generated torque, and Np is the number of pole pairs. As is clear from (7), in a synchronous reluctance motor in which iron loss can be ignored, the stator current itself (in other words, all components of the stator current) contributes to torque generation.

The magnetic fluxes φin-i and φmr that constitute the stator magnetic flux φ1 are referred to as “in-phase magnetic flux” and “mirror-phase magnetic flux”, respectively, according to the relationship of Expression (6c). The in-phase magnetic flux has an in-phase relationship with the stator current. On the other hand, the mirror phase magnetic flux is in a mirror phase relationship with the stator current. That is, the mirror phase magnetic flux φmr and the stator current i1 satisfy the mirror phase relationship expressed by the following equation.
In the present invention, the magnetic flux that satisfies the relationship of the expression (10) with respect to the stator current that contributes to the torque generation is defined as the mirror phase magnetic flux. In equation (10), the phase of the 2 × 1 vector is simply represented using an arctangent function. Hereinafter, this expression method is adopted. The function sgn means a sign function (signum function) that extracts only the polarity of a variable targeted by the function. The definition of the sign function is the same as that known to those skilled in the art. In the equation (6), the 2 × 2D factor acts on the stator magnetic flux, and thus on the in-phase magnetic flux and the mirror-phase magnetic flux. The definition of the 2 × 2D factor acting on these magnetic fluxes complies with equation (2a).

As is apparent from the circuit equation of the equation (6), the rotor phase θγ appears only in the mirror phase magnetic flux. Under the realization, when the circuit equation of the equation (6) is arranged by focusing on the mirror phase magnetic flux, the following equation is obtained.

In the left-hand side of equation (11), the D factor acting on the mirror phase magnetic flux exhibits a differential action as understood from the case where ωγ = 0. Eventually, the right side of the equation (11) means a differential equivalent value of the mirror phase magnetic flux. The differential equivalent value of the mirror phase magnetic flux indicates a phase displacement of sgn (ω2n) π / 2 [rad] with respect to the phase of the mirror phase magnetic flux itself when rotating at an electric speed ω2n. This is a "stable filter showing a phase displacement of -sgn ([omega] 2n) [pi] / 2 [rad] at the same frequency [omega] 2n as the electric velocity, which is equivalent to the differential of the mirror phase magnetic flux in order to obtain the phase of the mirror phase magnetic flux correctly. You just have to process the value. " The asterisk * on the right side of the equation (11) means the command value of the related signal. Hereinafter, this symbol is used in the same meaning.

According to the first aspect of the present invention, when the equivalent value of the electric speed of the synchronous reluctance is ω2n ＾ [rad / s], there is a mirror-phase relationship with the stator current as a 2 × 1 vector amount that contributes to torque generation. A two-input, two-output (2 × 2) stable filter showing a differential equivalent value of a mirror phase magnetic flux as a 2 × 1 vector quantity and a phase displacement of −sgn (ω2n∥) π / 2 [rad] at an angular frequency of ω2n ＾. Will be used at least. The phase of this signal is in principle the same as the phase of the mirror phase magnetic flux. Since the phase of the stator current can be immediately specified from the stator current itself, if both phases are applied to the equation (10), an effect is obtained that the rotor phase θγ can be correctly estimated in principle.

The above effects are realized on the γδ general coordinate system. Therefore, according to the first aspect of the present invention, the same effects can be obtained on the αβ fixed coordinate system, the dq synchronous coordinate system, the γδ rotational coordinate system, and the γδ quasi-synchronous coordinate system included in the γδ general coordinate system. Can be In other words, according to the first aspect of the present invention, the rotor phase θγ can be correctly estimated in principle by processing the differential equivalent value of the mirror phase magnetic flux defined on an arbitrary two-axis orthogonal coordinate system. The effect is obtained.

Next, the effect of the invention of claim 2 will be described. It is not always easy to design a two-input, two-output (2 × 2) stable filter that exhibits a phase shift of −sgn (ω2n ＾) π / 2 [rad] at an angular frequency of ω2n ＾. If only a phase shift of -sgn (ω2n ＾) π / 2 [rad] is pursued at an angular frequency of ω2n ＾, it is possible to use two integrators arranged in parallel. However, this is unstable and not practical. On the other hand, if the two integrators are changed to approximate integrators with emphasis on stability, -sgn (ω2n ＾) π / 2 [rad], which is the desired phase characteristic, cannot be obtained. According to the invention of claim 2, which specifically shows a 2 × 2 filter transfer function, it is possible to obtain an effect that a 2 × 2 filter capable of simultaneously achieving a desired phase characteristic and stability can be realistically configured. Can be. As a result, the effect of claim 1 can be enhanced.

In order to improve the phase characteristics and the stability and enhance the effect of the invention of claim 2, the 2 × 2 filter gain G used in the expressions (4) and (5) is expressed by the following expression. It is better to choose.
The scalar coefficient g1 used in the expressions (3), (4), (5), and (12) is preferably selected in the following range.

  Diagram showing the relationship between three coordinate systems and rotor phase (forward salient pole phase, reverse salient pole phase)   1 is a block diagram illustrating a basic configuration of a motor drive system according to one embodiment.   FIG. 1 is a block diagram showing a basic configuration of a phase velocity estimator in one embodiment.   1 is a block diagram illustrating a configuration of a phase deviation estimator or a phase estimator according to an embodiment.   FIG. 1 is a block diagram showing a configuration of a phase deviation estimator in one embodiment.   FIG. 1 is a block diagram showing a configuration of a phase deviation estimator in one embodiment.   Block diagram showing one implementation example of the inverse matrix of D factor   1 is a block diagram illustrating a basic configuration of a motor drive system according to one embodiment.   FIG. 1 is a block diagram showing a basic configuration of a phase velocity estimator in one embodiment.   FIG. 2 is a block diagram illustrating a configuration of a phase estimator according to one embodiment.   FIG. 2 is a block diagram illustrating a configuration of a phase estimator according to one embodiment.

Hereinafter, preferred embodiments of the present invention will be specifically described with reference to the drawings.

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. FIG. 2 shows one basic structure of a drive system using a drive device provided with the phase estimation device of the present invention for a synchronous reluctance motor. 1 is a synchronous reluctance motor, 2 is a power converter, 3 is a current detector, 4a and 4b are 3 phase to 2 phase converters, 2 phase to 3 phase converters, and 5a and 5b are vector rotators. , 6 indicate a cosine sine signal generator, 7 indicates a current controller, 8 indicates a command converter, 9 indicates a speed controller, and 10 indicates a phase speed estimator. In FIG. 2, various devices from 2 to 10 excluding the synchronous reluctance motor 1 constitute a drive device. This phase velocity estimator 10 corresponds to the phase estimating device according to the first and second aspects. In this figure, the 2 × 1 vector signal and the 3 × 1 vector signal are represented by one thick signal line to ensure simplicity. The following block diagram representation follows this.

The six types of devices 2, 3, 4a, 4b, 5a, 5b, 6, and 7 convert the stator current contributing to torque generation into a γδ rotating coordinate system composed of two orthogonal axes γ and δ. And a current control step of controlling each component of the γ-axis and the δ-axis to follow each axis current command value has been established. Further, the phase velocity estimator 10 (corresponding to the phase estimating device according to claims 1 and 2) is responsible for generating a phase to be used for the vector rotators 6a and 6b. That is, the phase estimation value, which is the output signal of the phase velocity estimator, is converted into a cosine / sine signal by the cosine / sine signal generator 6 and then passed to the vector rotators 6a and 6b that determine the γδ rotation coordinate system. .

The three-phase stator current detected by the current detector 3 is converted into a two-phase current (two-axis current) on an αβ fixed coordinate system by a three-phase two-phase converter 4a, and then by a vector rotator 5a. The current is converted into a two-phase current (γ-axis current, δ-axis current) in the rotating coordinate system and sent to the current controller 7. The current controller 7 controls the two-phase (two-axis) voltage command value on the γδ rotating coordinate system so that the two-phase current (two-axis current) on the γδ rotating coordinate system follows the current command value of each phase and each axis. Is generated and sent to the vector rotator 5b. At 5b, the two-phase (two-axis) voltage command value on the γδ rotating coordinate system is converted into a two-phase (two-axis) voltage command value on the αβ fixed coordinate system, and sent to the two-phase three-phase converter 4b. In 4b, the two-phase signal is converted into a three-phase voltage command value, and output as a command value to the power converter 2. The power converter 2 generates electric power according to the command value, applies the electric power to the synchronous reluctance motor 1, and drives the electric motor. In FIG. 3, in order to clearly show the coordinate system in which the voltage and the current are defined, a footnote r (γδ rotating coordinate system), s (αβ fixed coordinate system), and t (uvw coordinate system) are added to indicate these. I have.

The speed controller 9 divides the rotor speed estimated value (electrical speed estimated value) ω2n}, which is one of the output signals from the phase speed estimator 10, by the number of pole pairs Np to obtain a mechanical (angular) speed estimated value. ω2m} and then sent. In this embodiment of FIG. 2, an example in which a speed control system is configured is shown, and thus a torque command value τ * is obtained as an output of the speed controller 9. As is well known to those skilled in the art, the speed controller 9 is unnecessary when the control purpose is generated torque and a speed control system is not configured. In this case, the torque command value is directly applied from the outside.

The core of the present invention is inside a phase velocity estimator 10 (corresponding to the phase estimating device according to claims 1 and 2). In any of the speed control and the torque control, the phase speed estimator 10 does not require any change. Hereinafter, embodiments of the phase speed estimator 10 will be described without losing generality with respect to control modes such as speed control and torque control. This phase velocity estimator receives as input signals a stator voltage command value and a stator current, which are estimated values of a stator voltage evaluated on a γδ rotating coordinate system of two orthogonal axes where current control is performed. , And outputs a rotor phase estimation value and a speed estimation value (electric speed estimation value). In other words, the phase estimation by the phase velocity estimator is performed on the same coordinate system as the γδ rotating coordinate system in which the current control is performed. In this embodiment, the γδ rotating coordinate system is substantially a γδ quasi-synchronous coordinate system.

FIG. 3 shows the internal structure of the phase velocity estimator 10 on the γδ rotating coordinate system in FIG. The phase velocity estimator 10 on the γδ rotating coordinate system includes a phase deviation estimator 10a and a phase synchronizer 10b.

In the present embodiment, the true value (measured value) of the stator current and the command value of the stator voltage defined on the γδ rotating coordinate system are used as the input signals of the phase deviation estimator 10a. In response to this, the phase deviation estimator outputs the estimated rotor phase value θγ ＾ evaluated on the γδ rotating coordinate system with good consistency. The phase deviation estimator outputs, as other input signals, the speed ωγ of the γδ rotating coordinate system and the estimated value ω2n ＾ of the rotor speed (electrical speed) in order to output a rotor phase estimated value on the γδ rotating coordinate system. It has gained. The heart of the present invention resides in the phase deviation estimator 10a.

FIG. 4 schematically shows the structure of one embodiment of the phase deviation estimator 10a on the γδ rotating coordinate system (the drawing of velocity signal lines of ω2n ＾ and ωγ is omitted to avoid congestion in the figure). are doing). This phase deviation estimator mainly includes a 2 × 2 stable filter 10a-1 and a phase determiner 10a-2. The 2 × 2 stable filter 10a-1 is a stable filter according to the first aspect of the present invention. More specifically, it is configured based on the invention of claim 2. That is, FIG. 4A shows an example in which the filter described by the equation (4) is used as a 2 × 2 stable filter, and FIG. 4B is described by an equation (5) as a 2 × 2 stable filter. An example using a filter is shown. The input signal to the 2 × 2 stable filter is a differential equivalent value of the mirror phase magnetic flux based on the first aspect of the present invention. Further, in the present embodiment, the coordinate system in which the differential equivalent value of the mirror phase magnetic flux is defined is the same as the γδ rotating coordinate system in which current control is performed. In the example of FIG. 4, based on the third equation of the equation (11), the differential equivalent value of the mirror phase magnetic flux is synthesized using the stator voltage (command value) and the stator current (true value). .

The phase determiner 10a-2 in FIG. 4 is implemented by changing the equation (10), which expresses the mirror phase characteristic, into a form that reduces the amount of calculation. One example of a specific implementation is as follows.

The phase synchronizer, which is a post-processor of FIG. 3, incorporates a generalized integral PLL method developed in connection with sensorless vector control of a permanent magnet synchronous motor. The configuration of the phase synchronizer based on the generalized integral PLL method is described in detail in Non-Patent Document 1, which is well known to those skilled in the art, and further description will be omitted. The generalized integral PLL method for permanent magnet synchronous motors applies to synchronous reluctance motors without significant changes.

In the embodiment using FIGS. 2 to 4, the 2 × 2 stable filter 10a-1 is configured based on the equation (4) or the equation (5). In other words, the order n of the filter could be chosen arbitrarily. The filter order is set to 1 order for the 2 × 2 stable filters of the equations (4) and (5), the equation (12) is applied to the filter gain G, and the zero-order coefficient a0 of the stable polynomial of the equation (1) is When selecting as shown in equation (15),
FIGS. 5 and 6 are obtained as the 2 × 2 stable filter 10a-1 constituting the phase deviation estimator 10a of FIG. FIG. 5 corresponds to equation (4) in which the filter gain is on the input end side, and FIG. 4 (a). Instead, FIG. 6 corresponds to the equation (5) in which the filter gain is on the output end side, and FIG. 4B. The filter gain G in the embodiment shown in FIGS. 5 and 6 is a simple one obtained by considering the expression (13) in the expression (15).
The 2 × 2 inverse D factor “D−1” used in FIGS. 5 and 6 is realized as shown in FIG. 5 and 6, the 2 × 2D factor generally required for synthesizing the differential equivalent value of the mirror phase magnetic flux is replaced by the 2 × 2 inverse D factor “D-1” of the 2 × 2 stable filter 10a-1. And offset it. The cancellation is achieved by inputting a part of the signal of the differential equivalent value of the mirror phase magnetic flux from another part of the filter. As a result, a simpler configuration is achieved as the phase deviation estimator.

In the first and second embodiments described with reference to FIG. 2 to FIG. 7, the differential equivalent value of the mirror phase magnetic flux on the γδ rotating coordinate system, in other words, the input signal to the phase deviation estimator, The signal defined on the coordinate system was used. More specifically, the true value (actually measured value) of the stator current and the same command value as the equivalent value of the stator voltage were used as the stator current and stator voltage signals. Instead, using the same command value as the equivalent value of the defined stator current on the γδ rotating coordinate system, the true value of the stator voltage, and other equivalent values related to the stator current and the stator voltage are used. , The derivative equivalent value of the mirror phase magnetic flux on the γδ rotating coordinate system may be synthesized.

Next, as another embodiment according to the present invention, as a differential equivalent value of the mirror phase magnetic flux to be processed by the two-input two-output (2 × 2) stable filter, a coordinate different from the γδ rotating coordinate system in which current control is performed. An example using a signal defined in a system, more specifically, an example using a signal defined on an αβ fixed coordinate system will be described. FIG. 8 schematically shows a structural example of a sensorless vector control system using the present invention. The decisive difference between the structural example of FIG. 2 and the structural example of FIG. 8 is that, in the embodiment of FIG. 8, the input signal to the phase velocity estimator 10 (corresponding to the phase estimating device according to claims 1 and 2). Are the current and voltage signals of the stator defined on the αβ fixed coordinate system. The other devices in FIG. 8 are the same as those in FIG.

As shown in FIG. 8, the phase velocity estimator 10 (corresponding to the phase estimating device according to claims 1 and 2) has a true value (measured value) of the stator current defined on the αβ fixed coordinate system. ) And the same command value as the equivalent value of the stator voltage are input, and the phase estimation value (that is, the phase of the γδ rotating coordinate system) θα ＾ finally used by the vector rotator and the rotor speed (electric speed) estimation Value ω2n}. The estimated value of the electric speed of the rotor is divided by the number of pole pairs Np and converted into an estimated value of the machine speed ω2m ＾, and then sent to the speed controller 9.

FIG. 9 shows the internal structure of one embodiment of the phase velocity estimator 10 in FIG. The phase speed estimator 10 includes a phase estimator 10a and a speed estimator 10c.

In the present embodiment, the true value (measured value) of the stator current defined on the αβ fixed coordinate system and the same command value as the estimated value of the stator voltage are used as input signals to the phase estimator 10a. ing. In response to this, the phase estimator outputs a rotor phase estimated value (initial phase estimated value) “θγθ = θα ＾” evaluated on the αβ fixed coordinate system having good consistency. The general configuration of the phase estimator 10a for this embodiment is shown in FIG. The 2 × 2 stable filter 10a-1 of FIG. 4 in the present embodiment is a stable filter according to the first aspect of the present invention. A specific filter is configured based on the invention of claim 2. However, the 2 × 2 stable filters in FIGS. 4A and 4B correspond to the conditions of the αβ fixed coordinate system (that is, the velocity of the coordinate system is zero, ωγ = 0) in Expressions (4) and (5), respectively. ) Must be implemented. The input signal to the 2 × 2 stable filter is a differential equivalent value of the mirror phase magnetic flux based on the first aspect of the present invention. As a matter of course, the differential equivalent value of the mirror phase magnetic flux at this time must be defined on the αβ fixed coordinate system. In the example of FIG. 4, based on the third equation of the equation (11), the differential equivalent value of the mirror phase magnetic flux is synthesized using the stator voltage (command value) and the stator current (true value). When the condition of the αβ fixed coordinate system (that is, the velocity of the coordinate system is zero and ωγ = 0) is applied to the expression (11), the 2 × 2D factor of the expression (11) is obtained by changing the differential operator to only the diagonal element. 2 × 2 matrix. The configuration of the phase determiner 10a-2 of FIG. 4 in this embodiment is the same as that of the first and second embodiments described with reference to FIGS.

The speed estimator 10c, which is a post-processor of FIG. 9, incorporates an integral feedback speed estimation method developed in connection with sensorless vector control of a permanent magnet synchronous motor. The configuration of the speed estimator based on the integral feedback type speed estimation method is described in detail in Non-Patent Document 1, which is well known to those skilled in the art, and further description will be omitted. The integral feedback type speed estimation method for a permanent magnet synchronous motor is applied to a synchronous reluctance motor without major changes.

In the embodiment described with reference to FIGS. 8, 9, and 4, the 2 × 2 stable filter 10a-1 is configured based on the equation (4) or the equation (5). In other words, the order n of the filter could be chosen arbitrarily. The filter order is set to the first order for the 2 × 2 stable filters of the equations (4) and (5), the equation (12) is applied to the filter gain G, and the zero-order coefficient a0 of the stable polynomial of the equation (1) is expressed by ( In the case of selecting as in the expression 15), FIGS. 10 and 11 are obtained as the 2 × 2 stable filter 10a-1 constituting the phase estimator 10a in FIG. FIG. 10 corresponds to the equation (4) in which the filter gain is on the input end side, and FIG. Instead, FIG. 11 corresponds to equation (5) in which the filter gain is on the output end side, and FIG. 4 (b). The filter gain G in the embodiment shown in FIGS. 10 and 11 is a simple one according to the equation (16). In both figures, due to the addition of the condition of the αβ fixed coordinate system (that is, the speed of the coordinate system is zero and ωγ = 0), the inverse D factor is an inverse differentiator, that is, an integrator “1 / s”. In FIGS. 10 and 11, the 2 × 2 differential operator “s” required for synthesizing the differential equivalent value of the mirror phase magnetic flux on the αβ fixed coordinate system is replaced by the 2 × 2 stable filter 10a-1. X2 Integrator "1 / s" is offset. The cancellation is achieved by inputting a part of the signal of the differential equivalent value of the mirror phase magnetic flux from another part of the filter. As a result, a simpler configuration is achieved as the phase estimator.

In the fourth and fifth embodiments described with reference to FIGS. 4 and 8 to 11, as the differential equivalent value of the mirror phase magnetic flux on the αβ fixed coordinate system, in other words, as the input signal to the phase estimator, The signal defined on the αβ fixed coordinate system was used. More specifically, the true value (actually measured value) of the stator current and the same command value as the equivalent value of the stator voltage were used as the stator current and stator voltage signals. Instead, the same command value as the equivalent value of the stator current defined on the αβ fixed coordinate system, the true value of the stator voltage, etc., and other equivalent values related to the stator current and the stator voltage are used. It should be pointed out that the derivative equivalent value of the mirror phase magnetic flux on the αβ fixed coordinate system may be synthesized.

In the first to sixth embodiments, the synchronous reluctance motor to which the present invention is applied is a synchronous reluctance motor that does not require iron loss, and the “stator current contributing to torque generation” is “stator current itself”. In the case of a synchronous reluctance motor requiring iron loss consideration, the "stator current contributing to torque generation" is different from the "stator current itself". More specifically, as described in Non-Patent Document 2 (in this document, a wide range of AC motors is used), the stator current of a synchronous reluctance motor that requires consideration of iron loss has a It is divided into a stator load current that is a contributing component and a stator iron loss current that is a component that does not contribute to torque generation. In a synchronous reluctance motor requiring iron loss, “stator current contributing to torque generation” means a stator load current. It is pointed out that the present invention of claims 1 and 2 can be applied to a synchronous reluctance motor requiring consideration of iron loss by using a stator load current as a "stator current contributing to torque generation". .

As described above, the phase velocity estimator according to the present invention (corresponding to the phase estimating device according to claims 1 and 2) has been described in detail and in detail using a plurality of embodiments with reference to various drawings. Although the phase velocity estimator of the present invention can be realized in an analog manner, it is preferable that the phase velocity estimator be constructed digitally in view of recent remarkable progress in digital technology. The digital configuration includes a hardware configuration and a software configuration, but any one of the present invention can be configured as will be apparent to those skilled in the art.

INDUSTRIAL APPLICABILITY The present invention is suitable for a drive device for a synchronous reluctance motor that requires high-speed drive in a severe environment.

DESCRIPTION OF SYMBOLS 1 Synchronous reluctance motor 2 Power converter 3 Current detector 4a Three-phase two-phase converter 4b Two-phase three-phase converter 5a Vector rotator 5b Vector rotator 6 Cosine sine signal generator 7 Current controller 8 Command converter 9 Speed Controller 10 Phase speed estimator 10a Phase deviation estimator or phase estimator 10a-1 Two-input two-output (2 × 2) stable filter 10a-2 Phase determiner 10b Phase synchronizer 10c Speed estimator

Claims (2)

A current control step of dividing and controlling a stator current contributing to torque generation into a γ-axis component and a δ-axis component on a γδ rotating coordinate system composed of orthogonal γ and δ axes indicated by a vector rotator. In a synchronous reluctance motor drive device having a phase estimating device responsible for generating a phase to be used for the vector rotator,
When sgn (x) is a sign function that extracts only the polarity of x, which is a variable, and the equivalent value of the electric speed of the synchronous reluctance motor is ω2n ＾ [rad / s],
A stator current as a 2 × 1 vector quantity contributing to torque generation and a differential equivalent value of a mirror phase magnetic flux as a 2 × 1 vector quantity having a mirror-phase relationship are subjected to filtering processing, and −sgn at an angular frequency of ω2n ＾. (Ω2n ＾) π / 2 [rad], and a stable two-input two-output filter is provided.
A phase estimating apparatus for a synchronous reluctance motor driving device, wherein a phase to be used for a vector rotator is generated by using at least a signal after a filtering process by the two-input two-output filter. s is a differential operator, ai is a scalar coefficient of the following nth-order stable polynomial,
g1 is a scalar coefficient, the speed of the two-axis orthogonal coordinate system in which the 2 × 1 mirror phase magnetic flux is defined is ωγ [rad / s], and a D factor that is a 2 × 2 matrix is defined by the following equation.
A polynomial with D factor is defined by the following equation,
Further, when G is a filter gain of a 2 × 2 matrix,
The two-input two-output filter is expressed by the following equation.
Or the following equation
The phase estimating device for a synchronous reluctance motor driving device according to claim 1, wherein the filter is a filter described in (1).