JP2007305010A - Reference voltage generation circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To generate a reference voltage while suppressing the impact of noise of an external power source. <P>SOLUTION: The gate of the transistor NM31 of the current control part 30 is connected to the gates of the transistors NM13 and NM14, and the same amount of the drain current as the drain current that flows through the transistors NM13 and NM14 respectively flows through the transistor NM31. This drain current flows through the transistor PM22 of the constant current circuit part 20. The channel width ratio of the transistor PM21 and the transistor PM22 is set at 3 to 1, and the current three times as much as the current that flows through the transistor PM22 flows through the transistor PM21. The transistors PM21 and PM22 operate in the saturated region, make the current supplied from the external power source of the voltage VC constant, and supply it to a bandgap circuit part 10 that generates the reference voltage Vref. Since the current supplied to the bandgap circuit part 10 is made constant, the impact of the noise superimposed on the external power source can be avoided. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to a reference voltage generation circuit.

  An analog circuit is often provided with a reference voltage generation circuit that generates a reference voltage for comparison with an analog signal. In particular, analog integrated circuits are required to have a reference voltage generation circuit that does not depend on power supply voltage or temperature as the analog integrated circuit becomes larger and has higher functions.

  As such a reference voltage generation circuit, there are an operational amplifier type that uses a P-channel bulk as a power source, an operational amplifier type that self-biases the bulk of the P-channel input stage, and a current mirror circuit type.

Among these, the current mirror type reference voltage generation circuit is advantageous in that it has a simple configuration compared to an operational amplifier type and can reduce current consumption. This current mirror type reference voltage generation circuit generates a reference voltage independent of temperature by using a bipolar transistor to add a forward-biased pn junction potential and a voltage proportional to absolute temperature ( For example, see Patent Document 1).
JP 2005-128939 A (page 11, FIG. 1, FIG. 5)

  However, in such a conventional reference voltage generation circuit, noise generated by the influence of the inductance component between the wires of the external power supply may be superimposed on the power supply line, and the band gap voltage may be reduced.

  This current mirror type reference voltage generation circuit is more susceptible to noise than an operational amplifier type that self-biases the bulk of the P-channel input stage, and the bandgap voltage decreases when affected by noise. .

  The present invention has been made in view of such a conventional problem, and an object thereof is to provide a reference voltage generation circuit capable of preventing the influence of noise.

In order to achieve this object, a reference voltage generation circuit according to the first aspect of the present invention includes:
A reference voltage generator that generates a reference voltage based on a band gap voltage of the semiconductor by generating a potential difference proportional to the absolute temperature, and adding the generated potential difference and a potential difference of a pn junction of a forward-biased semiconductor;
A constant current supply unit including a transistor having a current path, wherein the transistor operates in a saturation region, constant current supplied to the current path of the transistor from an external power source, and supplies the current to the reference voltage generation unit; , Provided.

The reference voltage generation unit includes a first transistor and a second transistor that form a current mirror circuit, and a current path of the first transistor includes a pn junction through a resistor. A semiconductor is connected, a second semiconductor having a pn junction is connected to the current path of the second transistor, and an area ratio of the pn junction between the first semiconductor and the second semiconductor is set. And configured to generate a potential difference proportional to absolute temperature across the resistor,
The constant current supply unit includes a third transistor and a fourth transistor constituting a current mirror circuit, and a current outflow end of a current path of the third transistor is connected to the reference voltage generation unit And
Obtaining the amount of current flowing in the current path of the second transistor of the reference voltage generation unit, and based on the obtained amount of current, the constant current supply so that the third transistor operates in a saturation region A current control unit that controls the amount of current flowing through the current path of the fourth transistor of the unit may be provided.

  The ratio of the channel width or the pn junction area between the third transistor and the fourth transistor of the constant current supply unit corresponds to the ratio of the current path of the current control unit and the current path of the reference voltage generation unit. May be set.

  According to the present invention, the influence of noise can be prevented.

Hereinafter, a reference voltage generation circuit according to an embodiment of the present invention will be described with reference to the drawings.
The configuration of the reference voltage generation circuit according to this embodiment is shown in FIG.
The reference voltage generation circuit according to the present embodiment includes a band gap circuit unit 10, a constant current circuit unit 20, and a current control unit 30.

  The band gap circuit unit 10 generates a potential difference proportional to absolute temperature, and generates a reference voltage based on the semiconductor band gap voltage by adding the generated potential difference and the potential difference of the forward-biased semiconductor pn junction. It is a circuit part.

  The band gap circuit unit 10 is configured by a self-biased cascode current mirror type band gap circuit, and includes transistors PM11 to PM16, transistors NM11 to NM14, transistors Q11 to Q13, resistors R1 to R4, and a capacitor C1. .

  The transistors PM11 to PM16 are P-channel MOSFETs, are formed on an N-type substrate, and the bulk is biased to the voltage VH. The transistors PM11 to PM16 have substantially the same characteristics.

  The transistors PM11 and PM12 are a pair of transistors constituting a current mirror circuit. Both gates as control terminals are connected to each other, and the sources of the transistors PM11 and PM12 are connected to the voltage line of the voltage VH. The

  The transistor PM13 and the transistor PM14 are a pair of transistors constituting a current mirror circuit, and both gates are connected to each other. The source of the transistor PM13 is connected to the drain of the transistor PM11, and the source of the transistor PM14 is connected to the drain of the transistor PM12.

  Thus, the band gap circuit unit 10 includes the two-stage current mirror circuit in order to improve the accuracy of the reference voltage Vref.

  The transistors PM15 and PM16 are transistors for causing the same amount of drain current to flow between the source and drain of the transistors PM11 and PM13 as current paths.

  The source of the transistor PM15 is connected to the voltage line of the voltage VH, and the source of the transistor PM16 is connected to the drain of the transistor PM15.

  The gate of the transistor PM15 is connected to the gates of the transistors PM11 and PM12, and the gate of the transistor PM16 is connected to the gates of the transistors PM13 and PM14.

  The resistor R1 is inserted so that a current mirror circuit constituted by the transistors PM11, PM12 and PM15 and a current mirror circuit constituted by the transistors PM13, PM14 and PM16 operate in two stages. This is a resistor, and one end of the resistor R1 is connected to the drain of the transistor PM13.

  The gates of the transistors PM11 and PM12 are both connected to one end of the resistor R1, and the gates of the transistors PM13 and PM14 are both connected to the other end of the resistor R1.

  The transistors NM11 to NM14 are N-channel MOSFETs, are formed on a P-type substrate, and the bulk is biased to the voltage VL. The transistors NM11 to NM14 have substantially the same characteristics.

  The transistor NM11 and the transistor NM12 are a pair of transistors constituting a current mirror circuit, and both gates as control terminals are connected to each other. The drain of the transistor NM11 is connected to the other end of the resistor R1.

  The transistor NM13 and the transistor NM14 are a pair of transistors constituting a current mirror circuit, and both gates are connected to each other. The drain of the transistor NM13 is connected to the source of the transistor NM11, and the drain of the transistor NM14 is connected to the source of the transistor NM12.

  The resistor R2 is a resistor inserted so that a current mirror circuit constituted by the transistors NM11 and NM12 and a current mirror circuit constituted by the transistors NM13 and NM14 are operated in two stages.

  One end of the resistor R2 is connected to the drain of the transistor PM14, and the gates of the transistors NM11 and NM12 are both connected to one end of the resistor R2.

  The drain of the transistor NM12 and the gates of the transistors NM13 and NM14 are both connected to the other end of the resistor R2.

  Transistors Q11-Q13 are pnp bipolar transistors, and the collectors and bases of transistors Q11-Q13 are biased to voltage VL.

  The emitter of the transistor Q11 is connected to the source of the transistor NM13 via the resistor R3, and the emitter of the transistor Q12 is connected to the source of the transistor NM14.

  The area ratio of the pn junction between the transistors Q11 and Q12 is set to m: 1 (where m is a positive natural number), and the area of the pn junction of the transistor Q11 is larger than that of the transistor Q12. ing. Accordingly, the emitter-base voltage Vbe of the transistor Q11 is smaller than that of the transistor Q12.

  The emitter of the transistor Q13 is connected to the drain of the transistor PM16 via the resistor R4. Then, the band gap circuit unit 10 outputs the voltage at the connection point between the drain of the transistor PM16 and the resistor R4 as the reference voltage Vref.

  The capacitor C1 is for stabilizing the output reference voltage Vref, and one end is connected to the drain of the transistor PM16 and the other end is connected to the voltage line of the voltage VL.

  Since the band gap circuit unit 10 may not start up in a stable state when the reference voltage Vref is 0 V, the reference voltage generation circuit may start up the band gap circuit unit 10 (not shown). Prepared).

バイポーラトランジスタがシリコンで形成されている場合、シリコンのバンドギャップ電圧Ｖegは、約１．２Ｖであり、温度係数ａは、約２ｍＶ／°Ｃである。 Next, the operation principle of the band gap circuit unit 10 configured as described above will be described.
The relationship between the forward voltage and the absolute temperature of the pn junction of bipolar transistors such as the transistors Q11 to Q13 is expressed by the following equation (1).

When the bipolar transistor is made of silicon, the band gap voltage Veg of silicon is about 1.2 V, and the temperature coefficient a is about 2 mV / ° C.

The relationship between the emitter current I of the bipolar transistor and the base-emitter voltage Vbe is expressed by the following equation (2).

  Since the transistors PM11 and PM12, the transistor PM13 and the transistor PM14, the transistor NM11 and the transistor NM12, the transistor NM13 and the transistor NM14 form a current mirror circuit, respectively, the current Iq1 flowing through the transistors NM11 and NM13, The current Iq2 flowing through the NM 14 is almost equal.

また、トランジスタＱ１１とトランジスタＱ１２とのｐｎ接合部の面積比は、ｍ：１に設定され、この分、トランジスタＱ１１のエミッタ−ベース間電圧Ｖbeは、トランジスタＱ１２と比較して小さくなる一方、トランジスタＮＭ１３のソースとトランジスタＱ１１のエミッタとの間には抵抗Ｒ３が接続されている。このため、抵抗Ｒ３両端の電位差Ｖr3は、数３より、次の数４によって表される。

数４に示すように、この抵抗Ｒ３の両端に、絶対温度に比例する電位差Ｖr3が現れることになる。 If the current Iq1 and the current Iq2 are substantially equal, the voltage at the connection point N1 between the source of the transistor NM13 and the resistor R3 and the voltage at the connection point N2 between the source of the transistor NM14 and the emitter of the transistor Q12 are also approximately equal. For this reason, the current Iq1 and the current Iq2 are expressed by the following formula 3 from the formula 2.

The area ratio of the pn junction between the transistors Q11 and Q12 is set to m: 1, and accordingly, the emitter-base voltage Vbe of the transistor Q11 becomes smaller than that of the transistor Q12, while the transistor NM13. A resistor R3 is connected between the source of the transistor and the emitter of the transistor Q11. For this reason, the potential difference Vr3 between both ends of the resistor R3 is expressed by the following equation 4 from the equation 3.

As shown in Equation 4, a potential difference Vr3 proportional to the absolute temperature appears at both ends of the resistor R3.

Next, assuming that the current amount of the current (Iq1) flowing through the transistors PM11 and PM13 and the current amount of the current (Iq2) flowing through the transistors PM12 and PM14 are ip, the current ip is expressed by the following equation (5).

Further, the current Iq3 having the same amount as the current amount ip flows through the transistors PM15 and PM16. Therefore, the potential difference (voltage drop) Vr4 between both ends of the resistor R4 is expressed by the following equation (6).

この数７に示すように、基準電圧Ｖrefは、絶対温度に比例する電位差と順バイアスされた半導体のｐｎ接合の電位差とが加算されて、半導体のバンドギャップ電圧に基づく電圧になる。 Therefore, the reference voltage Vref is expressed by the following equation (7).

As shown in Equation 7, the reference voltage Vref is a voltage based on the band gap voltage of the semiconductor by adding the potential difference proportional to the absolute temperature and the potential difference of the pn junction of the forward-biased semiconductor.

  Referring to Equation 7, the temperature coefficient between the base and emitter of a bipolar transistor such as the transistor Q13 is negative, and the base-emitter voltage Vbe13 decreases as the temperature increases.

  On the other hand, the temperature coefficient of the second term of Equation 7 is positive, and the voltage drop Vr4 at the resistor R4 increases as the temperature rises.

  In this way, since the reference voltage Vref is generated by adding parameters having different temperature coefficients, the resistance values r3 and r4 of the resistors R3 and R4 are appropriately set, so that the temperature-independent reference A voltage Vref is generated.

  Next, the constant current circuit unit 20 is for making the current supplied from the external power source of the voltage VC constant and supplying it to the band gap circuit unit 10. Thereby, the constant current circuit unit 20 prevents voltage fluctuation due to noise superimposed on the power supply line of the voltage VC.

The constant current circuit unit 20 includes transistors PM21 and PM22.
Both the transistor PM21 and the transistor PM22 are a pair of transistors composed of P-channel MOSFETs, and constitute a current mirror circuit. The transistors PM21 and PM22 are formed on the N substrate, and the bulk is biased to the voltage VC of the external power supply.

  The ratio of the channel widths of the transistors PM21 and PM22 is set corresponding to the ratio between the current path of the current control unit 30 and the current path of the band gap circuit unit 10.

  Specifically, in the bandgap circuit unit 10, three current paths are formed: a current path below the transistor PM11, a current path below the transistor PM12, and a current path below the transistor PM15. The current control unit 30 is formed with only one current path below the transistor NM31.

  Therefore, in the present embodiment, the ratio of the channel widths of the transistors PM21 and PM22 is 3 to 1, and the channel widths of the transistors PM21 and PM22 are set so as to be this ratio. When the channel width is set in this way, a current three times as large as that of the transistor PM22 flows between the source and drain of the transistor PM21.

  The gates of the transistors PM21 and PM22 serving as control terminals are both connected to the drain of the transistor PM22, and a current path is formed between the source and drain of each of the transistors PM21 and PM22.

  The sources of the transistors PM21 and PM22 are connected to the power supply line of the voltage VC. The drain of the transistor PM 21 is a current outflow end of the constant current circuit unit 20, the band gap circuit unit 10 is connected to this drain, and the constant current circuit unit 20 applies the voltage VH to the band gap circuit unit 10.

  The current control unit 30 is connected between the source and drain of the transistor PM22 so that the pair of transistors PM21 and PM22 of the constant current circuit unit 20 operate in the saturation region and the current supplied to the band gap unit 10 becomes a constant current. This is for controlling the amount of flowing drain current. The current control unit 30 includes a transistor NM31 and a transistor Q31.

  The transistor NM31 is composed of an N-channel MOSFET, is formed on a P-type substrate, and the bulk is biased to the voltage VL. The gate of the transistor NM31 is connected to the gates of the transistors NM13 and NM14 of the bandgap circuit unit 10.

  The drain of the transistor NM31 is connected to the drain of the transistor PM22 of the constant current circuit unit 20.

  The transistor Q31 is composed of a pnp bipolar transistor, its emitter is connected to the source of the transistor NM31, and its collector and base are biased to the voltage VL.

  The transistor NM31 has substantially the same characteristics as the transistors NM11 to NM14, and the transistor Q31 has substantially the same characteristics as the transistor Q12.

  The current control unit 30 configured as described above acquires a signal supplied to the gates of the transistors NM13 and NM14. The current control unit 30 operates the transistors PM21 and PM22 of the constant current circuit unit 20 in the saturation region Sp1 as shown in FIG. 2 based on the acquired signal level, that is, the gate-source voltage Vgs1. As described above, the amount of drain current flowing between the source and drain of the transistor PM22 is controlled.

In FIG. 2, L1 is a boundary line between the saturation region Sp1 and the linear region Sp2, and is expressed by the following equation (8).

  As described above, when the transistors PM21 and PM22 operate in the saturation region Sp1, if the gate-source voltage Vgs1 of the transistors PM21 and PM22 is constant, even if the drain-source voltage Vds varies, the drain current Id Does not change, and even if the voltage VC fluctuates, the current flowing through the band gap circuit section 10 becomes a constant current.

Next, the operation of the reference voltage generation circuit according to this embodiment will be described.
The constant current circuit unit 20 is supplied with the voltage VC of the external power supply and starts supplying current to the band gap circuit unit 10. When a current is supplied from the constant current circuit unit 20 to the bandgap circuit unit 10, a current Iq1 flows through the transistors PM11 and PM13 of the bandgap circuit unit 10, and a current Iq2 flows through the transistors PM12 and PM14.

  Since the current amounts ip of the currents Iq1 and Iq2 are substantially the same, and the current amounts of the currents Iq3 flowing through the transistors PM15 and PM16 are also substantially the same, the bandgap circuit unit 10 can generate the reference voltage Vref shown in Equation 7. Is output.

  On the other hand, the gate of the transistor NM31 is connected to the gates of the transistors NM13 and NM14. The characteristics of the transistor NM31 and the characteristics of the transistor Q31 are substantially the same as the characteristics of the transistors NM13 and NM14 and the characteristics of the transistor Q12, respectively. For this reason, the transistors NM31 and Q31 have a current ip having substantially the same amount of current ip as the currents Iq1 and Iq2 flowing in the transistors NM11 to NM14 and the transistors Q11 and Q12.

  Therefore, the current ip substantially flows between the source and drain of the transistor PM22 of the constant current circuit unit 20. Further, since the channel width of the transistor PM21 is set to three times that of the transistor PM22, a current three times the current amount ip flows through the transistor PM21.

  This current is distributed almost equally to the current paths of the transistors PM11, PM12, and PM15 by 1/3, and currents Iq1, Iq2, and Iq3 of the current amount ip flow through the current paths, respectively. For this reason, the constant current circuit unit 20 supplies an appropriate amount of current to the band gap circuit unit 10.

  When noise n as shown in FIG. 3 is superimposed on the power supply line of the voltage VC due to the influence of the inductance component between the wires of the external power supply, the drain-source voltage Vds of the transistor PM21 of the constant current circuit unit 20 is fluctuates due to the influence of n.

  However, even if the drain-source voltage Vds of the transistor PM21 varies, the drain current Id is constant because the transistors PM21 and PM22 of the constant current circuit unit 20 operate in the saturation region Sp1 as shown in FIG. Thus, the gate-source voltages of the transistors NM13 and NM14 do not change.

  If the gate-source voltage of the transistors NM13 and NM14 does not change, the gate-source voltage of the transistor NM31, the gate-source voltage Vgs1 of the transistors PM21 and PM22 do not change, and the drain current Id of the transistors PM21 and PM22 also changes. It becomes constant. Thus, since the current control unit 30 controls the drain current Id of the transistor PM22 of the constant current circuit unit 20, the current flowing through the bandgap circuit unit 10 is constant regardless of the fluctuation of the voltage VC.

  Therefore, even if the noise n is superimposed on the power line of the voltage VC, the constant current circuit unit 20 supplies a constant current to the band gap circuit unit 10 without being affected by the noise n, thereby preventing an increase in current consumption. can do.

  In addition, a constant current is supplied to the band gap circuit unit 10 without being affected by noise, and the reference voltage Vref is held constant.

  As described above, according to the present embodiment, the constant current circuit unit 20 includes the current mirror circuit constituted by the pair of transistors PM21 and PM22, and the transistors PM21 and PM22 are operated in the saturation region, whereby the transistors The current flowing through the PM 21 is made constant and supplied to the band gap circuit unit 10.

  In addition, a current control unit 30 is provided, and the current control unit 30 controls the amount of drain current flowing through the transistor PM22 of the constant current circuit unit 20 based on the gate voltages of the transistors NM13 and NM14 of the bandgap circuit unit 10. I made it.

  Therefore, even if noise is superimposed on the power supply line of the voltage VC due to the influence of the inductance component between the wirings of the external power supply, the constant current circuit unit 20 applies a constant current to the band gap circuit unit 10 without being affected by the noise. The reference voltage Vref generated by the band gap circuit unit 10 can be kept constant.

  Further, the ratio of the channel widths of the transistors PM21 and PM22 of the constant current circuit unit 20 is set to 3: 1, and a necessary current is supplied to each current path of the band gap circuit unit 10 through the transistor PM21. Therefore, the constant current circuit unit 20 can supply an appropriate current to the band gap circuit unit 10, and the transistors PM21 and PM22 can be reliably operated in the saturation region Sp1.

In carrying out the present invention, various forms are conceivable and the present invention is not limited to the above-described embodiment.
For example, in the above-described embodiment, the band gap circuit unit 10 includes the transistors PM11 to PM16. However, the transistors PM11, PM12, PM15, NM13, and NM14 can be provided without the transistors PM13, PM14, and PM16 and the transistors NM11 and NM12.

  Further, diodes may be used instead of the transistors Q11 to Q13. Further, a pnp bipolar transistor may be used instead of the transistors PM21 and PM22 of the constant current circuit unit 20.

It is a circuit diagram which shows the structure of the reference voltage generation circuit which concerns on embodiment of this invention. It is a figure which shows the saturation area | region and linear area | region of a transistor. It is a figure which shows the noise superimposed on the external power supply.

Explanation of symbols

10 Band gap circuit section 20 Constant current circuit section 30 Current control section

Claims (3)

A reference voltage generator that generates a reference voltage based on a band gap voltage of the semiconductor by generating a potential difference proportional to the absolute temperature, and adding the generated potential difference and a potential difference of a pn junction of a forward-biased semiconductor;
A constant current supply unit including a transistor having a current path, wherein the transistor operates in a saturation region, constant current supplied to the current path of the transistor from an external power source, and supplies the current to the reference voltage generation unit; With
A reference voltage generating circuit. The reference voltage generation unit includes a first transistor and a second transistor that form a current mirror circuit, and a current path of the first transistor includes a pn junction through a resistor. A semiconductor is connected, a second semiconductor having a pn junction is connected to the current path of the second transistor, and an area ratio of the pn junction between the first semiconductor and the second semiconductor is set. And configured to generate a potential difference proportional to absolute temperature across the resistor,
The constant current supply unit includes a third transistor and a fourth transistor constituting a current mirror circuit, and a current outflow end of a current path of the third transistor is connected to the reference voltage generation unit And
Obtaining the amount of current flowing in the current path of the second transistor of the reference voltage generation unit, and based on the obtained amount of current, the constant current supply so that the third transistor operates in a saturation region A current control unit for controlling the amount of current flowing in the current path of the fourth transistor of the unit,
The reference voltage generation circuit according to claim 1. The ratio of the channel width or the pn junction area between the third transistor and the fourth transistor of the constant current supply unit corresponds to the ratio of the current path of the current control unit and the current path of the reference voltage generation unit. Set,
The reference voltage generation circuit according to claim 2, wherein:

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