JP2003319651A - Switching power circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To enhance the power conversion efficiency of a composite resonance converter provided with a current resonance converter on the primary side thereof. <P>SOLUTION: With respect to the composite resonance converter, a partial voltage resonance circuit is combined with the current resonance converter on the primary side thereof. A secondary-side partial voltage resonance circuit is formed on the secondary side thereof. With respect to an insulating converter transformer which transfers power from the primary side to the secondary side, a primary winding and a secondary winding are wound with predetermined winding widths using a E-E-shaped or U-U-shaped core. The formation of a gap between magnet legs of the core is avoided so that the primary side and the secondary side are closely coupled with each other. <P>COPYRIGHT: (C)2004,JPO

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply in various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a switching power supply circuit that employs a switching converter such as a flyback converter or a forward converter is widely known. Since the switching operation waveforms of these switching converters are rectangular waves, there is a limit in suppressing switching noise. Also, due to its operating characteristics,
It is known that there is a limit to the improvement of power conversion efficiency. Therefore, the applicant has previously proposed various switching power supply circuits using various resonant converters.
The resonant converter can easily obtain high power conversion efficiency, and the switching operation waveform has a sinusoidal waveform to realize low noise. It also has the advantage that it can be configured with a relatively small number of parts.

The circuit diagram of FIG. 8 shows an example of a switching power supply circuit as a prior art, which can be constructed based on the invention previously proposed by the present applicant. The switching power supply circuit shown in this figure is applied to an AC adapter of, for example, a personal computer or a liquid crystal television (television), and a DC voltage EO of a low voltage of, for example, 15V is applied to the load side as described later. Is designed to be output, and is designed to support load power Po = 75 W. Further, as its configuration, a configuration as a resonance type converter in which a partial voltage resonance circuit is combined with a separately excited type current resonance type converter is adopted.

In the power supply circuit shown in this figure, first,
A full-wave rectifying / smoothing circuit including a bridge rectifying circuit Di and one smoothing capacitor Ci is provided for the commercial AC power supply AC. The full-wave rectification operation of the bridge rectification circuit Di and the smoothing capacitor Ci results in the rectification smoothing voltage Ei (DC input voltage) across the smoothing capacitor Ci. This DC input voltage Ei is
The level corresponds to the same size as the AC input voltage VAC.

As a current resonance type converter for switching by inputting the DC input voltage, as shown in the drawing, two switching elements Q by MOS-FETs are used.
1 and Q2 are connected by half bridge connection. Clamp diodes DD1 and DD2 are connected in parallel between the sources and drains of the switching elements Q1 and Q2 in the direction shown in the drawing.

A partial resonance capacitor Cp is connected in parallel between the source and drain of the switching element Q2. The capacitance of the partial resonance capacitor Cp and the leakage inductance component L of the primary winding N1
Depending on 1, the parallel resonance circuit (partial voltage resonance circuit) is formed. Then, a partial voltage resonance operation is obtained in which the voltage resonates only when the switching elements Q1 and Q2 are turned off.

In this power supply circuit, in order to drive the switching elements Q1 and Q2 in a switching manner, an oscillation drive circuit 11 composed of, for example, a general-purpose IC is provided. The oscillation drive circuit 11 applies a gate voltage as a drive signal to each gate of the switching elements Q1 and Q2. As a result, the switching elements Q1 and Q2 perform a switching operation by alternately turning on / off at a required switching frequency.

The oscillation drive circuit 11 operates by inputting a low-voltage DC voltage obtained by a rectifying circuit consisting of a low-voltage winding N4 additionally wound around the primary side of the insulating converter transformer PIT and a capacitor C4. It has a power supply.
Further, at the time of startup, the rectified and smoothed voltage Ei is input via the startup resistor Rs to start up.

Isolation Converter Transformer PIT (Power Is
(Olation Transformer) transmits the switching outputs of the switching elements Q1 and Q2 to the secondary side. One end of the primary winding N1 of the insulation transformer PIT has a resonance current detection winding N
Contact point between the source of switching element Q1 and the drain of switching element Q2 via A (switching output point)
The switching output is obtained by being connected to.

The other end of the primary winding N1 is connected to the primary side ground via a series resonance capacitor C1. And the insulation converter transformer PIT including the capacitance of the series resonance capacitor C1 and the primary winding N1.
The inductance component forms a primary side series resonance circuit for making the operation of the primary side switching converter a current resonance type. In this way, the primary side switching converter shown in this figure can obtain the current resonance type operation and the partial voltage resonance operation described above.

Here, the insulating converter transformer PIT has a structure shown in FIG. 9, for example. The insulating converter transformer PIT is provided with an EE type core in which E type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other. Then, the primary winding N1 and the secondary winding N2 are respectively wound on the bobbin B which is divided so that the winding areas on the primary side and the secondary side are independent of each other. It is wound around the area.
The primary winding N1 and the secondary winding N2 are each 60 μm.
A mmφ litz wire is wound by a glass winding. A gap G is formed on the central magnetic leg as shown in the figure. As a result, the coupling coefficient k is, for example, about 0.8 so that the loose coupling state is obtained. The gap G is defined by the E-shaped core CR1,
It can be formed by making the central magnetic leg of CR2 shorter than the two outer magnetic legs.

A secondary winding N2 is wound around the secondary side of the insulating converter transformer PIT. In this power supply circuit, a center tap is grounded to the secondary winding N2 as shown in the drawing. In the following description, the secondary winding N2 portion divided by the tap point will be referred to as "secondary winding N2a" and "secondary winding N2b" for the sake of explanation. Then, for this secondary winding N2, as shown in the drawing, rectifying diodes DO1 and D1 by Schottky diodes are provided.
A double-wave rectification circuit composed of O2 and the smoothing capacitor CO is formed, and the secondary side DC output voltage EO is obtained at both ends of the smoothing capacitor CO. This secondary side DC output voltage EO is supplied to a load side not shown and is also branched and input as a detection voltage for the control circuit 1 described next.

The control circuit 1 generates a variable DC current according to the level change of the secondary side DC output voltage EO, and supplies it to the oscillation drive circuit 11 via the photocoupler PC. In the oscillation drive circuit 11, switching drive is performed such that the switching frequencies of the switching elements Q1 and Q2 are changed according to the output of the control circuit 1 input via the photocoupler PC. By changing the switching frequencies of the switching elements Q1 and Q2 in this way,
The level of the secondary side DC output voltage EO will be stabilized. Depending on the constant voltage control operation of the control circuit 1, the level of the secondary side DC output voltage EO is maintained at the level of EO = 15V as described above.

FIG. 10 is a waveform diagram showing the operation of the main part of the power supply circuit shown in FIG. 8 by the switching cycle. In this figure, the AC input voltage VAC = 100
The operation waveforms when V and load power Po = 75 W are shown.
In this figure, the switching operation of the switching element Q2 is shown by the source-drain voltage VQ2 of the switching element Q2 and the switching current IQ2 of the switching element Q2. That is, during the period TOFF when the switching element Q2 is off, the switching current IQ
2 becomes 0 level, and the source-drain voltage VQ
For 2, the level clamped by the rectified and smoothed voltage Ei is obtained. On the other hand, during the period TON in which the switching element Q2 is on, the switching current IQ2 flows according to the waveform shown in the figure, and the source-drain voltage VQ2 becomes 0 level. The switching current IQ2 flows as the primary winding current I1 flowing through the primary winding N1 in the period TON. In addition,
Although not shown here, the switching element Q1 is
ON / OFF at alternating timing with switching element Q2
Since the switching element Q1 is off, the source-drain voltage and the switching current of the switching element Q1 have a waveform obtained by shifting the source-drain voltage VQ2 and the switching current IQ2 of the switching element Q2 by approximately 180 °. . Therefore, the waveform portion of the primary winding current I1 during the period TOFF during which the switching element Q1 side is turned on flows as the switching current of the switching element Q1.

As the secondary winding voltage V2 generated in the secondary winding N2 as the primary winding current I1 flows, a waveform corresponding to the cycle of the primary winding current I1 is obtained as shown in the figure. become. Then, as the secondary winding voltage V2, a waveform whose absolute value level is clamped at the level of the secondary side DC output voltage EO is obtained. Further, the rectification current ID1 flowing in the rectification diode DO1 and the rectification diode DO2
A waveform as shown in the figure is obtained as the rectification current ID2 flowing in the rectification current ID2. From these waveforms, it is understood that the rectification diodes DO1 and DO2 are alternately turned on / off.

As characteristics of the power supply circuit having the configuration shown in FIG. 8, AC input voltage VAC = 100V and load power P
FIG. 11 shows the AC → DC power conversion efficiency (ηAC → DC) with respect to changes of o = 15 W to 75 W. As shown in FIG. 11, the AC → DC power conversion efficiency (ηAC → DC) in the power supply circuit of FIG. 8 is 8 when the load power Po = 45 W.
The efficiency is the highest at 9.2%, and is the lowest at 82.0% when the load power Po = 45W. When the load power Po = 75 W, which is the maximum load, the AC → DC power conversion efficiency (ηAC → DC) is 88.5%, and the power loss of the entire circuit in this case reaches 9.7 W.

In order to obtain the operation shown in FIG. 10 and the characteristics shown in FIG. 11 as the power supply circuit shown in FIG. 8, each part is selected as follows. Primary winding N1 = 35T Secondary winding N2 = Secondary winding N2a + Secondary winding N2b = 8T +
8T = 16T Primary side series resonance capacitor C1 = 0.15 μF Partial resonance capacitor Cp = 680 pF

[0018]

By the way, it is preferable that the power supply circuit has as high a power conversion efficiency as possible. However, in the power supply circuit shown in FIG. 8, in order to start up by the stable operation of ZVS and ZCS, the switching frequency fs is maintained even under the condition of the minimum input voltage of the AC input voltage VAC = 90V and the maximum load power Po = 75W. The secondary side DC output voltage EO = 1 so that the required frequency higher than the resonance frequency fo1 is maintained.
It must be stabilized at 5V. For this reason, the primary side series resonance capacitor C1 is set to 0.1
Select 5μF and Isolation Converter Transformer PIT
For the above, it is necessary to form a gap of about 0.5 mm to 1 mm to obtain a loosely coupled state with a coupling coefficient k of about 0.8.

That is, according to the circuit configuration of FIG. 8, since the primary winding and the secondary winding are loosely coupled as described above, the AC → DC power conversion efficiency (ηAC → There is a limit to the improvement of DC). Specifically, for example, load power Po = 75 W, AC input voltage VAC
= AC → DC power conversion efficiency at 100V (ηAC → DC)
Is limited to about 88.5% as described with reference to FIG.

When the secondary side DC output voltage EO is set to output a relatively low voltage DC voltage of EO = 15 V as in the circuit of FIG. 8, the load power Po = the maximum load. At 75 W, a load current of 5 A is flowing on the load side. That is, as a result, the peak level of the rectification currents ID1 and ID2 flowing through the rectification diodes DO1 and DO2 becomes 6.8 Ap, which is excessive, as shown in FIG. Further, in this case, the level of the resonance current flowing on the primary side also becomes excessive, which means that the peak level of the switching current IQ2 flowing through the switching elements Q2 and (Q1) shown in FIG. 10 is 5 Ap-p. You can see from Therefore, in this case, the switching element Q
1 and Q2, and the power loss in the isolation converter transformer PIT increases.

The circuit shown in FIG. 8 is applied as an AC adapter as described above, but such an AC adapter is generally housed in a completely sealed casing. It is a thing. Therefore, when the power loss in the insulating converter transformer PIT and the semiconductor (switching elements Q1 and Q2) increases as described above, a mechanism is required to protect the circuit from heat generation due to this power loss. As a result, in the AC adapter to which the power supply circuit of FIG. 8 is applied, the housing cannot be downsized accordingly.

Further, since the insulating converter transformer PIT is in the loosely coupled state as described above, the generation level of the leakage magnetic flux from the insulating converter transformer PIT becomes high. Therefore, in actuality of the circuit, it is necessary to take countermeasures by providing a short ring of a copper plate on the insulating converter transformer PIT, which leads to an increase in cost and size of the insulating converter transformer PIT. Further, when the insulating converter transformer PIT is loosely coupled, the temperatures of the primary winding and the secondary winding near the gap G are increased due to eddy current loss due to so-called fringe magnetic flux. Is disadvantageous in terms of.

Further, in forming the gap G in the central magnetic leg of the insulating converter transformer PIT, the central magnetic leg of the E-shaped core made of, for example, a ferrite material is polished. In this case, since a polishing process is added to manufacture the insulating converter transformer PIT, there is a problem that the cost is increased accordingly.

[0024]

In view of the above problems, the present invention has a switching power supply circuit configured as follows. That is, a current resonance type switching means for performing a switching operation by intermittently inputting the input DC input voltage, and a means for transmitting the switching output of the switching means from the primary side to the secondary side, At least the primary winding and the secondary winding are wound with a required number of turns on the EE type core or the UU type core having no leg gaps, and the primary winding and the secondary winding are wound. An insulating converter transformer configured such that the winding and the winding are in a tightly coupled state with a coupling coefficient higher than a required value. Then, at least the leakage inductance component of the primary winding of the insulating converter transformer and the capacitance of the primary side series resonance capacitor connected in series to the primary winding are formed, and the operation of the switching means is a current resonance type. The primary side series resonance circuit and the capacitance of a partial resonance capacitor connected in parallel to a predetermined switching element among a plurality of switching elements forming the switching means, and the leakage inductance of the primary winding of the insulating converter transformer. A primary side partial voltage resonance circuit that performs a voltage resonance operation during the turn-off period of the plurality of switching elements that form the switching means, and further, the leakage inductance of the secondary winding of the isolation converter transformer. Ingredients and this So as comprising a secondary side parallel resonant circuit formed by the capacitance of the secondary side parallel resonant capacitor connected in parallel to the next winding. Then, the alternating voltage obtained between the two tap output terminals in the secondary winding of the insulation converter transformer is input to perform the rectification operation to generate the secondary side DC output voltage. A constant voltage configured to perform constant voltage control on the secondary side DC output voltage by varying the switching frequency of the switching means according to the output voltage generating means and the level of the secondary side DC output voltage. And a control means.

Further, in the present invention, the switching power supply circuit is configured as follows. That is, a current resonance type switching means for performing a switching operation by interrupting an input DC input voltage, and a means for transmitting the switching output of the switching means from the primary side to the secondary side, For EE type cores or U-U type cores that do not have gaps formed in the legs,
The primary winding, the first secondary winding, and the second secondary winding having a smaller number of windings than the first secondary winding are each wound with a required number of turns, and the primary winding An insulating converter transformer configured such that the wire and the secondary winding are in a tightly coupled state with a coupling coefficient higher than a required value. Then, at least the leakage inductance component of the primary winding of the insulating converter transformer and the capacitance of the primary side series resonance capacitor connected in series to the primary winding are formed, and the operation of the switching means is a current resonance type. The primary side series resonance circuit and the capacitance of a partial resonance capacitor connected in parallel to a predetermined switching element among a plurality of switching elements forming the switching means, and the leakage inductance of the primary winding of the insulating converter transformer. A primary side partial voltage resonance circuit that performs a voltage resonance operation during a turn-off period of a plurality of switching elements that form the switching means, and further includes a first secondary winding of the isolation converter transformer. Leakage inductance component of So as comprising a secondary side parallel resonant circuit formed by the capacitance of the secondary side parallel resonant capacitor connected in parallel with the first secondary winding. Then, the alternating voltage obtained in the second secondary winding is input to perform a rectifying operation to generate a secondary side DC output voltage; According to the level of the secondary side DC output voltage, by varying the switching frequency of the switching means, a constant voltage control means configured to perform constant voltage control for the secondary side DC output voltage is provided. .

According to each of the above constructions, as the composite resonance type converter, a construction in which a primary side current resonance type switching converter (switching means) and a primary side partial voltage resonance circuit are combined is adopted, and then, on the secondary side. On the other hand, a secondary side parallel resonance circuit is provided. By adopting the above-mentioned configuration as the current resonance type converter, it becomes possible to cover the electric power by the resonance circuit on the secondary side, and as the insulation converter transformer, the primary winding and the secondary winding are tightly coupled. I am trying to become. Then, while obtaining a combination as such a composite resonance type converter,
By combining with the structure of the insulating converter transformer that is tightly coupled as described above, it is possible to further improve the power conversion efficiency.

[0027]

1 shows an example of the configuration of a switching power supply circuit as an embodiment of the present invention. The power supply circuit shown in this figure is, for example, a personal computer,
Alternatively, it is applied to an AC adapter dedicated to a liquid crystal television, etc., and outputs a low-voltage DC voltage of, for example, 15 V to the load side as described later, and at the same time, up to load power Po = 75 W. It is designed to correspond to. Further, this power supply circuit is provided with a self-excited current resonance type converter as a primary side switching converter.

First, in FIG. 1, as a rectifying circuit system for generating a DC input voltage (rectified and smoothed voltage Ei) from a commercial AC power source, a bridge rectifier circuit Di and one smoothing capacitor are provided for a commercial AC power source AC. A full-wave rectifying and smoothing circuit composed of Ci is provided. Therefore, in this case, the rectified and smoothed voltage Ei (DC input voltage) is obtained across the smoothing capacitor Ci by the full-wave rectification operation. The rectified and smoothed voltage Ei has a level corresponding to the same size as the AC input voltage VAC.

The self-exciting current resonance type converter on the primary side shown in this figure comprises two switching elements Q1 and Q2 as shown in the figure. In this case, bipolar transistors are selected as the switching elements Q1 and Q2. These switching elements Q1 and Q2 are connected by a half bridge coupling method. That is, the collector of the switching element Q1 is connected to the line of the rectified and smoothed voltage Ei (the positive terminal of the smoothing capacitor Ci). The emitter of switching element Q1 is connected to the collector of switching element Q2, and the emitter of switching element Q2 is connected to the primary side ground.

Further, with respect to the base of the switching element Q1, the base current limiting resistor RB1-resonance capacitor C
A self-excited oscillation drive circuit formed by connecting B1-drive winding NB1 in series is connected. Here, the series connection of the resonance capacitor CB1 and the drive winding NB1 forms a series resonance circuit by the capacitance of the resonance capacitor CB1 and the inductance of the drive winding NB1, and switching is performed by the resonance frequency of the series resonance circuit. The frequency is determined. Further, the base current limiting resistor RB1 adjusts the base current level as a drive signal to be flown from the self-excited oscillation drive circuit to the base of the switching element Q1.

A damper diode DD1 is connected between the base and the emitter of the switching element Q1 in the direction shown in the drawing to form a reverse current path in the ON period. A starting resistor Rs1 is connected between the collector and the base of the switching element Q1 to allow a current at the time of starting to flow to the base.

Similarly, the base of the switching element Q2 is connected to a self-excited oscillation drive circuit formed by serially connecting a base current limiting resistor RB2-resonance capacitor CB2-drive winding NB2. Then, the resonance capacitor CB2-
The drive winding NB2 forms a series resonant circuit. In addition, a damper diode DD2 is provided between the base and the emitter.
Is connected, and the starting resistor Rs2 is connected between the collector and the base.

Further, the collector of the switching element Q2
A partial resonance capacitor Cp is connected in parallel between the emitters.
Are connected. Depending on the capacitance of the partial resonance capacitor Cp and the leakage inductance component L1 of the primary winding N1, a parallel resonance circuit (partial voltage resonance circuit)
To form. Then, a partial voltage resonance operation is obtained in which the voltage resonates only when the switching elements Q1 and Q2 are turned off.

Drive Transformer PRT (Power Regulati
The ng transformer) is provided for switching-driving the switching elements Q1 and Q2 and for variably controlling the switching frequency for constant voltage control. This drive transformer PRT has a drive winding NB1, NB2 and a resonance current detection winding NA wound thereon, and a saturable reactor in which a control winding Nc is wound in a direction orthogonal to each of these windings. Has been done. The drive winding NB1 and the drive winding NB2 are wound in the winding directions in which voltages of opposite polarities are excited.

Insulation Converter Transformer PIT (Power Is
(Olation Transformer) transmits the switching outputs of the switching elements Q1 and Q2 to the secondary side. By connecting the winding end end of the primary winding N1 of the insulating transformer PIT to the contact (switching output point) of the emitter of the switching element Q1 and the collector of the switching element Q2 via the resonance current detection winding NA, A switching output is provided.

As shown in the figure, the winding start end of the primary winding N1 is connected to the primary side ground via the series resonance capacitor C1. The capacitance of the series resonance capacitor C1 and the inductance component of the insulating converter transformer PIT including the primary winding N1 form a primary side series resonance circuit for making the operation of the primary side switching converter a current resonance type. . In this way, as the primary side switching converter shown in this figure, the current resonance type operation and the partial voltage resonance operation described above are obtained in a composite manner.

The switching operation of this power supply circuit is as follows, for example. First, when the commercial AC power supply AC is turned on, a base current for starting is supplied to the bases of the switching elements Q1, Q2 via the starting resistors Rs1, Rs2, for example. Here, for example, the drive windings NB1 and NB2 of the drive transformer PRT are excited with voltages of opposite polarities, so if the switching element Q1 is turned on first, the switching element Q2 is turned on.
Is controlled to be off. Then, each self-excited oscillation drive circuit of the switching elements Q1 and Q2 performs self-excited oscillation operation by resonance operation using the alternating voltage excited in these drive windings NB1 and NB2 as a source. As a result, the switching elements Q1 and Q2 are controlled to be turned on / off alternately. That is, the switching operation is performed. Then, for example, when the switching element Q1 is turned on,
As the switching output, a resonance current flows through the resonance current detection winding NA to the primary winding N1 and the series resonance capacitor C1. In the vicinity of the resonance current becoming 0, the switching element Q1 is turned off and the switching is performed. The element Q2 is turned on. As a result, the switching element Q2
A resonance current in the opposite direction to the above flows through. After that, ZV
The self-excited switching operation in which the switching elements Q1 and Q2 are alternately turned on by S and ZCS is continued.
Further, as the switching elements Q1 and Q2 are turned on / off, a current flows through the partial resonance capacitor Cp for a short period when the switching elements Q1 and Q2 are turned off.
That is, a partial voltage resonance operation can be obtained.

A secondary winding N2 is provided on the secondary side of the insulating converter transformer PIT. A secondary side parallel resonance capacitor C2 is connected in parallel to the secondary winding N2 as shown in the figure. A secondary side parallel resonance circuit is formed by the capacitance of the secondary side parallel resonance capacitor C2 and the leakage inductance L2 of the secondary winding N2. Therefore, when the alternating voltage is excited in the secondary winding N2 of the insulating converter transformer PIT, the voltage resonance operation is also obtained on the secondary side. That is, the power supply circuit shown in FIG. 1 is configured as a current resonance type converter so that current resonance operation and partial voltage resonance operation can be obtained on the primary side and voltage resonance operation can be obtained on the secondary side. become.

Further, the secondary winding N2 is provided with three tap output terminals as shown in the drawing. In the following, this secondary winding N divided by these tap points
2 parts, in order from the winding start side, the secondary windings N3b, N2
b, N2a, N3a.

Here, a structural example of the insulating converter transformer PIT provided in the power supply circuit shown in FIG. 1 will be described with reference to the sectional views of FIGS. FIG. 2 is a structural example using a pair of E-shaped cores. Insulation converter transformer PI
As the core of T, as shown in the drawing, an EE-shaped core is formed by combining two E-shaped cores CR1 and CR2 such that the ends of the magnetic legs face each other. Further, in this case, no gap is formed on the surfaces of the E-shaped cores CR1 and CR2 facing the central magnetic legs. Then, as shown in the figure, the primary / secondary split bobbin B is used, the primary winding N1 is wound on the primary winding area, and the secondary winding area is further wound on the secondary winding area. As described above, the secondary winding N2 is wound in the order of the secondary windings N3b, N2b, N2a, N3a from the inside. A ferrite material, for example, is used for the E-shaped cores CR1 and CR2.

FIG. 3 shows an example of structure using a pair of U-shaped cores. In this case, in the insulating converter transformer PIT, as its core, as shown in FIG. 3, U-shaped cores CR11 and CR12 each having two magnetic legs are combined,
It is adapted to form a U-U type core. Further, with respect to one of the magnetic legs of the U-U type core thus formed, the primary winding N1 and the secondary winding N2 (N3b, N2b, N2
a, N3a) and bobbin B wound in the same manner as in FIG.
Is attached. In addition, U- formed as described above
No gap is formed in the central magnetic leg of the U-shaped core.

As described above, in this embodiment, no gap is formed in the central magnetic leg of the core.
The gap is not formed in the magnetic leg of the core in this way, and the winding width of each of the primary winding N1 and the secondary winding N2 is narrower than that of the conventional winding, thereby coupling It is designed to obtain a tight coupling degree with a coefficient k of about 0.95.

Here, for example, in the prior art power supply circuit,
By setting the isolation converter transformer PIT in a loosely coupled state, abnormal oscillation during an intermediate load was suppressed. On the other hand, in the power supply circuit of this embodiment, the resonance operation of the parallel resonance circuit provided on the secondary side prevents abnormal oscillation from occurring at an intermediate load. In other words, by doing this, the insulation converter transformer P
Even if the IT is configured to be in the tightly coupled state, no problem occurs in the operation of the power supply circuit.

In FIG. 1, the tap that is connected to the secondary winding N2 and divides the secondary winding N2a and the secondary winding N2b is grounded to the secondary side ground. Then, as shown, the secondary winding N2a and the secondary winding N3a
Rectifier diode DO1 is attached to the tap that divides and
Further, rectifying diodes D02 are connected to the taps that divide the secondary winding N2b and the secondary winding N3b.
Further, these rectifying diodes DO1 and DO2 are respectively connected to the positive terminal of the smoothing capacitor CO. These rectifying diodes DO1 and DO2, and the smoothing capacitor CO
In some cases, a double-wave rectifying / smoothing circuit is formed, and a secondary side based on the alternating voltage obtained at the secondary winding N2a and the secondary winding N2b is provided at both ends of the smoothing capacitor CO. The DC output voltage EO can be obtained. In the case of the present embodiment, the secondary side DC output voltage EO is EO =
It is adapted to generate a DC voltage of 15V. Further, the secondary side DC output voltage EO is supplied to a load (not shown) and is also branched and input as a detection voltage for the control circuit 1 described next.

The control circuit 1 changes the level of the control current (DC current) flowing through the control winding NC in accordance with the level change of the secondary side DC output voltage EO, so that the orthogonal control transformer PR
The inductance LB of the drive winding NB wound around T is variably controlled. As a result, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the main switching element Q1 formed including the inductance LB of the drive winding NB changes. This is an operation for varying the switching frequency of the main switching element Q1, and this operation stabilizes the DC output voltage on the secondary side.

The essential parts and elements of the above-mentioned structure are selected as follows. Primary winding N1 = 33T (turn) Secondary winding N2a = Secondary winding N2b = 8T Secondary winding N3a = Secondary winding N3b = 25T Primary side series resonance capacitor C1 = 0.18 μF Partial resonance capacitor Cp = 680pF Secondary side parallel resonance capacitor C2 = 1500pF

The waveform diagram of FIG. 4 shows the operation of the main part of the power supply circuit having the configuration shown in FIG. The operation waveforms shown in this figure are AC input voltage VAC = 10
The measurement result is shown under the condition of load power Po = 75W in 0V system. In this figure, the switching element Q2 performs a switching operation so that it is turned on in the period TON and turned off in the period TOFF. And
Switching current IQ2 flowing through switching element Q2
As shown in the figure, during the period TOFF, it is 0 level, and during the period TON, first, at the start, the damper current flows from the damper diode DD2 to the positive polarity in the negative direction via the base → collector of the switching element Q2.
After this, a waveform flows through the collector-emitter.

Further, the collector of the switching element Q2
The emitter-to-emitter voltage VQ2 becomes a pulse clamped at the level of the rectified and smoothed voltage Ei (DC input voltage) in the period TOFF, and has a waveform of 0 level in the period TON. The switching element Q1 is switched with respect to the switching element Q2 at the timing of being alternately turned on / off. Therefore, the switching element Q
The switching current and the collector-emitter voltage of 1 have substantially the same waveform shapes as the switching current IQ2 and the collector-emitter voltage VQ2.
The phase is shifted by 80 °.

The primary winding current I1 flowing through the primary winding N1 as a switching output is shown in this figure. The primary winding current I1 is a resonance current obtained by the resonance operation of the series resonance circuit of the primary side series resonance circuit (C1-N1) according to the switching operation of the switching elements Q1 and Q2. Then, this primary winding current I1
The switching element Q1 is off and the switching element Q2 is
During the period TON when the switching element is turned on, the switching element Q
The second switching current IQ2 flows through the switching element Q2. Further, during the period TOFF in which the switching element Q2 is off and the switching element Q12 is on, the switching current of the switching element Q1 flows to the switching element Q1.

The secondary winding voltage V3 generated in the secondary winding N2 by the primary winding current I1 flowing in this manner is as follows.
The waveform as shown can be obtained. Further, as the voltage V2 obtained at the secondary winding N2a and the secondary winding N2b of the secondary winding N2, a waveform corresponding to the secondary winding voltage V3 is obtained as shown in the figure. Like In addition, the rectified current I flowing through the rectifier diode DO1
The waveforms shown in the figure are obtained as the rectification current ID2 flowing through D1 and the rectification diode DO2. From these waveforms,
It can be seen that the rectifier diodes DO1 and DO2 are alternately turned on / off.

Here, the AC → DC power conversion efficiency characteristics of the power supply circuit according to the present embodiment shown in FIG. 1 and the power supply circuit according to the prior art shown in FIG. 8 will be compared. FIG. 5 shows AC → DC power conversion characteristics (ηAC → DC) in the power supply circuit of the present embodiment with respect to changes in the AC input voltage VAC = 100V and the load power Po = 15 to 75.
FIG. The broken line in the figure indicates the AC → DC power conversion characteristic (ηAC → D) of the circuit of FIG. 8 under the same conditions.
It shows C). As shown in this figure, when the maximum load power Po = 75 W, the power supply circuit of the prior art shown in FIG. 8 was ηAC → DC = 88.5%, whereas the power supply circuit of FIG. ηAC → DC = 91.3%, which is an improvement of 2.8%. Further, the power loss at this time was 9.7 W in the circuit of FIG. 8, whereas it was 7.1 W in the circuit of FIG. 1, and as a result, the power loss of the circuit of FIG. 1 was reduced by 2.6 W. It will be planned.

This is also shown by comparing FIG. 4 which is a waveform diagram of the circuit of FIG. 1 and FIG. 10 which is a waveform diagram of the circuit of the prior art of FIG. First, Fig. 1
As shown in FIG. 0, in the circuit as the prior art of FIG. 8, the peak level of the primary side resonance current I1 is 5 under the conditions of the AC input voltage VAC = 100V and the load power Po = 75W.
A-Ap, the peak level of the switching current IQ2 is 2.
The peak levels of 5 Ap, rectified current ID1, and rectified current ID2 are both 6.8 Ap. On the other hand, in the circuit of FIG. 1, under the same conditions, the peak level of the primary side resonance current I1 is 3.6A-Ap and the peak level of the switching current IQ2 is 1.8Ap.
And the peak level of switching current IQ2 is 28%.
Will be reduced. Further, as shown in the figure, the rectified current I
The peak levels of D1 and the rectified current ID2 are both reduced to 5.8 Ap. Thus, these FIG.
From the comparison between FIG. 10 and FIG. 10, it can be seen that the circuit of FIG. 1 has reduced power loss and improved power conversion efficiency.

In this way, the power conversion efficiency is improved because the insulating converter transformer PIT is tightly coupled, and the secondary side parallel resonance capacitor C2 is provided to perform the voltage resonance operation also on the secondary side. It depends on what you got.

Further, in the present embodiment, in order to make the insulating converter transformer PIT tightly coupled as described above, no gap is formed in the magnetic leg of the core. Then, if the gap is not formed in the insulating converter transformer PIT as described above, the step for forming the gap is not necessary in the manufacturing, so that the manufacturing process can be simplified and the cost can be reduced. . Also, by being tightly coupled,
Since the leakage flux from the insulating converter transformer PIT is also reduced, it is not necessary to wind a short ring made of, for example, a copper plate around the insulating converter transformer PIT.
Also in this respect, the manufacturing process of the insulating converter transformer PIT is simplified, and the cost reduction is promoted. Further, since the gap is eliminated, the problem of local temperature rise of the winding of the insulating converter transformer PIT is solved, and the reliability is improved accordingly.

Here, as a modification of the circuit configuration of the power supply circuit shown in FIG. 1, the configuration on the secondary side is shown in FIG. In the configuration shown in FIG. 6, the secondary windings N2a and N2b of the secondary winding N2 described in FIG. 1 and the secondary windings N3a and N3b are separated to form separate independent windings. To be done. As a result, as shown in the figure, the secondary windings N2a and N2b of the secondary winding N2 are connected to the windings of the rectifying and smoothing circuit system for generating the secondary side DC output voltage EO, and the secondary winding N3. Are separated into windings of the secondary side parallel resonance circuit system. in this case,
These windings are wound as shown in the sectional view of FIG. That is, as shown in the drawing, first, the secondary windings N2b and N2a are wound inside the winding region on the secondary side of the bobbin B. Then, the secondary winding N3 is wound on the outer side thereof. As the core in this case, as in the case of the circuit of FIG. 1, in addition to the EE type core by the E type core CR1 and the E type core CR2 shown in the figure, a U
It is also possible to use a U-U type core formed by the type core 11 and the U type core 12.

In the circuit of FIG. 6, the secondary winding N
As for 3, the number of windings is almost equal to that of the primary winding N1. The wire used for each winding on the secondary side is 60 μφ / 18 for the secondary windings N2a and N2b.
The litz wire of 0 bundle and the litz wire of 60 μφ / 60 bundle are selected for the secondary winding N3. Even when the secondary side is configured in this way, the same effect as that of FIG. 1 can be obtained.

The switching power supply circuit according to the present invention is not limited to the configuration of each of the embodiments described above, and, for example, the constants of the component elements of the main part may be appropriately set under various conditions. It may be changed to an appropriate value depending on the situation. Further, for example, as a switching element used in the primary side switching converter, in addition to the bipolar transistor shown in each circuit diagram, a MOS-FET
Or IGBT may be adopted. In addition, MOS-
When an FET, an IGBT, or the like is adopted, it may be configured to be driven by a separately excited type.

[0058]

As described above, according to the present invention, as a composite resonance type converter, a current resonance type converter is combined with a partial voltage resonance circuit on the primary side, and a secondary side is combined with a partial voltage resonance circuit. It constitutes a secondary parallel resonant circuit. Then, for the insulation converter transformer, an EE type core or a UU type core is adopted, and after winding the primary winding and the secondary winding with a required winding width,
A gap is not formed in the magnetic leg of the core so that the primary side and the secondary side are tightly coupled. Then, with such a configuration, the power conversion efficiency is significantly improved as compared with the power supply circuit of the prior art.

By improving the power conversion efficiency as described above, the power loss in the power supply circuit is reduced, and the heat generation of each part is reduced. Further, by reducing the heat generation of each part in this way, when the switching power supply circuit of the present invention is applied as an AC adapter, for example, it is possible to make the housing smaller than conventional cases. .

Since it is not necessary to form a gap in the insulating converter transformer, the step of polishing the core for forming the gap can be omitted. Thereby, for example, the manufacturing process can be simplified, and the manufacturing cost of the insulating converter transformer can be reduced.

Further, since the primary winding and the secondary winding wound around the insulating converter transformer are tightly coupled to each other as described above, the leakage magnetic flux from the insulating converter transformer is reduced. It is not necessary to provide a short ring to the converter transformer. And
In this respect as well, the cost can be reduced and the circuit can be made smaller and lighter. Further, since the local temperature rise does not occur in the vicinity of the gap of the insulating converter transformer, the reliability of the power supply circuit is improved accordingly.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing a configuration example of a switching power supply circuit as an embodiment of the present invention.

FIG. 2 is a diagram illustrating a structural example of an insulation converter transformer provided in the power supply circuit of the embodiment.

FIG. 3 is a diagram illustrating a structural example of an insulation converter transformer provided in the power supply circuit of the embodiment.

FIG. 4 is a waveform chart showing the operation of the power supply circuit according to the embodiment.

FIG. 5 is a diagram showing a change characteristic of AC → DC power conversion efficiency with respect to a load change in the power supply circuit of the embodiment.

FIG. 6 is a circuit diagram showing a modified example of the configuration on the secondary side of the switching power supply circuit as the embodiment.

FIG. 7 is a diagram illustrating a structural example of an insulation converter transformer provided in the power supply circuit as a modified example shown in FIG. 6.

FIG. 8 is a circuit diagram showing another configuration example of a switching power supply circuit as a prior art.

9 is a cross-sectional view showing a structural example of an insulating converter transformer used in the power supply circuit shown in FIG.

10 is a waveform diagram showing an operation of the power supply circuit shown in FIG.

11 is a diagram showing a change characteristic of AC → DC power conversion efficiency with respect to a load change in the power supply circuit shown in FIG.

[Explanation of symbols]

1 Control circuit, Di bridge rectification circuit, DO1, DO2
Rectifier diode, Ci smoothing capacitor, Q1, Q2
Switching element, PIT insulation converter transformer,
N1 primary winding, N2 secondary winding, C1 primary side series resonance capacitor, Cp primary side partial resonance capacitor, C2
Secondary side parallel resonant capacitor

Claims (2)

[Claims] 1. A current resonance type switching means for performing a switching operation by interrupting an input DC input voltage, and a means for transmitting a switching output of the switching means from a primary side to a secondary side. Then, at least the primary winding and the secondary winding are wound with a required number of turns on the EE type core or the UU type core having no magnetic leg gap, and the primary winding is An insulating converter transformer configured such that the secondary winding and the secondary winding are in a tightly coupled state with a coupling coefficient higher than a required value, at least a leakage inductance component of the primary winding of the insulating converter transformer, and the primary winding And a capacitance of a primary side series resonance capacitor connected in series to the primary side series resonance circuit, which makes the operation of the switching means a current resonance type. A plurality of switching elements forming the switching means, a capacitance of a partial resonance capacitor connected in parallel to a predetermined switching element, and a leakage inductance component of the primary winding of the insulating converter transformer, A primary side partial voltage resonance circuit performing a voltage resonance operation during a turn-off period of a plurality of switching elements forming the switching means, a leakage inductance component of a secondary winding of the insulating converter transformer, and a secondary winding with respect to the secondary winding. Input the alternating voltage obtained between the secondary side parallel resonant circuit formed by the capacitance of the secondary side parallel resonant capacitor connected in parallel and the two tap output terminals in the secondary winding of the insulating converter transformer. Rectification is performed to generate the secondary side DC output voltage. By performing a constant voltage control on the secondary side DC output voltage by varying the switching frequency of the switching means according to the level of the secondary side DC output voltage and the DC output voltage generating means configured as described above. A switching power supply circuit comprising: a configured constant voltage control means. 2. A current resonance type switching means for performing a switching operation by interrupting an input DC input voltage, and for transmitting a switching output of the switching means from a primary side to a secondary side. The primary winding, the first secondary winding, and the first secondary winding with respect to the EE type core or the UU type core having no gap formed in the magnetic leg. Also, the second secondary winding having a small number of windings is wound by the required number of windings respectively, and the primary winding and the secondary winding are in a tightly coupled state with a coupling coefficient higher than the required one. Formed by at least the leakage inductance component of the primary winding of the isolation converter transformer and the capacitance of the primary side series resonance capacitor connected in series to the primary winding. A primary side series resonance circuit in which the operation of the switching means is a current resonance type, and a capacitance of a partial resonance capacitor connected in parallel to a predetermined switching element among a plurality of switching elements forming the switching means, A primary side partial voltage resonance circuit formed by the leakage inductance component of the primary winding of the insulating converter transformer and performing a voltage resonance operation during a turn-off period of a plurality of switching elements forming the switching means; A secondary side parallel resonant circuit formed by a leakage inductance component of one secondary winding and a capacitance of a secondary side parallel resonant capacitor connected in parallel to the first secondary winding; The alternating voltage obtained in the second secondary winding is input to perform rectification operation, A DC output voltage generating means configured to generate a secondary side DC output voltage, and a secondary side DC output by varying the switching frequency of the switching means according to the level of the secondary side DC output voltage. A switching power supply circuit comprising: a constant voltage control unit configured to perform constant voltage control with respect to a voltage.