JP2000166232A - Switching power circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To miniaturize a resonant power circuit and reduce cost by providing a primary parallel resonant circuit on a primary side for loose-coupling of an insulated converter transformer, and on the other hand, connecting a secondary series resonant circuit to secondary winding in series on a secondary side for formation of a series resonant circuit. SOLUTION: A voltage resonant primary parallel resonant circuit is formed out of a leakage inductance element L1 including the primary winding N1 of an insulated converter transformer PIT and the capacitance of a parallel resonant capacitor Cr for loose-coupling the insulated converter transformer PIT. On the other hand, a secondary series resonant capacitor Cs1 forms a series resonant circuit out of the capacitance itself and the leakage inductance element L2 of the secondary winding N2A so as to meet the ON/OFF operations of rectifier diodes DO1, DO2. A series resonant capacitor Cs2 forms a series resonant circuit out of the capacitance itself and the leakage inductance L2 so as to meet the rectifier diodes DO3, DO4.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a circuit employing a switching converter of a type such as a flyback converter or a forward converter is widely known. Since these switching converters have a rectangular switching operation waveform, there is a limit in suppressing switching noise. Also, due to its operating characteristics,
It has been found that there is a limit in improving the power conversion efficiency. Therefore, the present applicant has previously proposed various switching power supply circuits using various resonance type converters.
The resonance type converter can easily obtain high power conversion efficiency and realize low noise because the switching operation waveform is sinusoidal. It also has the advantage that it can be configured with a relatively small number of parts.

FIG. 6 is a circuit diagram showing an example of a voltage resonance type switching power supply circuit which can be constructed based on the invention proposed by the present applicant. In the switching power supply circuit shown in this figure, for example, a commercial AC power supply in Japan or the United States is a so-called AC 100 V system, and corresponds to a condition that the maximum load power is 120 W or more.

The switching power supply circuit shown in FIG. 1 is provided with a so-called voltage doubler rectification circuit comprising rectification diodes Di1, Di2 and smoothing capacitors Ci1, Ci2 as a rectification and smoothing circuit for rectifying and smoothing the AC power supply AC. Can be In this voltage doubler rectifier circuit, assuming that a DC input voltage corresponding to, for example, one time the peak value of the AC input voltage VAC is Ei, a DC input voltage 2Ei that is approximately twice that of the DC input voltage Ei.
Generate For example, if the AC input voltage VAC is 144V, the DC input voltage 2Ei is about 400V. As described above, the reason why the voltage doubler rectifier circuit is employed as the rectifier / smoothing circuit is that the AC input voltage is 100 V AC as described above.
The system is designed to cope with a relatively heavy load condition in which the maximum load power is 120 W or more. That is, by increasing the DC input voltage to twice the normal value, the amount of current flowing into the subsequent switching converter is suppressed, and the reliability of the components forming the switching power supply circuit is ensured. . In the voltage doubler rectifier circuit shown in this figure, an inrush current limiting resistor Ri is inserted in the rectified current path, for example, so as to suppress the inrush current flowing into the smoothing capacitor when the power is turned on. I have.

The voltage-resonant type switching converter shown in FIG. 1 has a self-excited type provided with a single switching element Q1. In this case, a high withstand voltage bipolar transistor (BJT;
Junction type transistor). The base of the switching element Q1 is connected to the positive side of the smoothing capacitor Ci1 (rectified smoothed voltage 2Ei) via a starting resistor RS so that a base current at the time of starting can be obtained from the rectified smoothing line. A self-oscillation resonance circuit including a damping resistor RB, an inductor LB, a resonance capacitor CB, and a drive winding NB is connected in series between the base of the switching element Q1 and the temporary ground. In this case, the drive winding NB is connected to the insulation converter transformer PIT (Power Is
olation Transformer) and the inductor L
Together with B, a required inductance for setting the switching frequency is obtained. Further, the base of the switching element Q1 and the negative electrode (1
A path of a damper current flowing when the switching element Q1 is turned off is formed by a clamp diode DD inserted between the secondary side ground), and the collector of the switching element Q1 is connected to the insulating converter transformer P.
It is connected to one end of the IT primary winding N1, and the emitter is grounded.

A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr
Is the combined capacitance (L1, LR) obtained by connecting the primary winding N1 of the isolated converter transformer PIT described later and the controlled winding NR of the orthogonal control transformer PRT (Power Regulating Transformer) in series. Thereby, a parallel resonance circuit of the voltage resonance type converter is formed. Although the detailed description is omitted here, when the switching element Q1 is turned off, the voltage Vd across the resonance capacitor Cr is obtained by the action of the parallel resonance circuit.
cr is actually a sinusoidal pulse waveform so that a voltage resonance type operation can be obtained.

The insulated converter transformer PIT is for transmitting the switching output of the switching element Q1 to the secondary side. In this case, the insulated converter transformer PIT is used.
One end of the primary winding N1 of T is connected to the collector of the switching element Q1, and the other end is connected in series with the controlled winding NR of the orthogonal control transformer PRT as shown in the figure.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, a parallel resonance circuit is formed by connecting the secondary side parallel resonance capacitor C2 to the secondary winding N2 in parallel. With this parallel resonance circuit,
The alternating voltage excited in the secondary winding N2 is a resonance voltage,
This resonance voltage is supplied to two sets of half-wave rectifier circuits, a half-wave rectifier circuit including a rectifier diode DO1 and a smoothing capacitor CO2, and a half-wave rectifier circuit including a rectifier diode DO2 and a smoothing capacitor CO2. Then, the DC output voltages EO1 and EO2 are obtained by these two sets of half-wave rectifier circuits, respectively. In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1.
In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1. The rectifier diodes DO1 and DO2 forming the half-wave rectifier circuit use a high-speed rectifier to rectify the alternating voltage in the switching cycle.

The control circuit 1 compares, for example, a DC voltage output on the secondary side with a reference voltage and outputs a DC current corresponding to the error.
The control winding N of the orthogonal control transformer PRT is used as the control current.
This is the error amplifier that supplies C. In this case, the DC voltage output EO1 is input to the control circuit 1 as the detection voltage, and the DC output voltage EO2 is input as the operation power supply.

For example, when the DC output voltage EO2 on the secondary side fluctuates due to the fluctuation of the AC input voltage VAC or the minimum load power, the control circuit 1 controls the control current flowing through the control winding NC to 10 mA to 40 mA, for example. Vary in range.
Thus, the inductance LR of the controlled winding NR is changed in a range of, for example, 0.1 mH to 0.6 mH.

Since the controlled winding NR forms a parallel resonance circuit for obtaining a voltage-resonant switching operation as described above, this parallel resonance circuit is not controlled for a fixed switching frequency. The resonance condition of the circuit is changed. At both ends of the parallel connection circuit of the switching element Q1 and the parallel resonance capacitor Cr, a sinusoidal resonance pulse is generated by the action of the parallel resonance circuit corresponding to the OFF period of the switching element Q1, but the resonance condition of the parallel resonance circuit , The width of the resonance pulse is variably controlled. That is, PWM (P
(ulse Width Moduration) control operation is obtained. The PWM control of the width of the resonance pulse means the control of the off period of the switching element Q1, which in other words means variably controlling the on period of the switching element Q1 under the condition of a fixed switching frequency. . By variably controlling the ON period of the switching element Q1 in this manner, the switching output transmitted from the primary winding N1 forming the parallel resonance circuit to the secondary side changes, and the DC output voltage ( The output levels of EO1, EO2) are also changed. As a result, the secondary DC voltage (EO1, EO2) can be made constant. In addition, such a constant voltage control method is herein referred to as a “primary-side voltage resonance pulse width control method”.

FIG. 7 is a circuit diagram showing another example of a voltage resonance type switching power supply circuit which can be constructed based on the invention proposed by the present applicant. In FIG. 7, the same parts as those in FIG. 6 are denoted by the same reference numerals, and the description of the parts having the same configuration will be omitted. In the power supply circuit shown in this figure, an example is shown in which the controlled winding of the orthogonal control transformer PRT is provided on the secondary side. In this case, two controlled windings NR and NR1 are wound and provided as controlled windings of the orthogonal control transformer PRT, and the controlled winding NR is connected to an end of the secondary winding N2. The rectifier diode DO1 is connected so as to be inserted in series between the anodes. The controlled winding NR1 is inserted in series between the tap output of the secondary winding N2 and the anode of the rectifier diode DO2. In such a connection form, the secondary-side parallel resonance circuit includes the controlled windings NR and NR1.
(LR, LR1).

As described above, in the case where the controlled windings (NR, NR1) of the orthogonal control transformer PRT are provided on the secondary side, the inductance of the controlled windings NR, NR1 is determined as an inductance control method. Is varied, so that the pulse width of the resonance voltage V2 of the secondary parallel resonance capacitor C2, that is, the conduction angle of the secondary rectifier diode is variably controlled. Thereby, constant voltage control is achieved by controlling the output level obtained on the secondary side (secondary side resonance voltage pulse control method).

Here, the structure of the insulating converter transformer PIT provided in the power supply circuit shown in FIGS. 6 and 7 is shown in FIG.
Is shown in cross section. Insulation converter transformer PIT
Are E-type cores CR1 and CR2 made of ferrite material, for example.
Are formed such that the magnetic legs face each other to form an EE-type core. At this time, no gap is formed in the center magnetic leg as shown in the figure. Then, the primary winding N1 (and the driving winding NB) is applied to the center magnetic leg using the bobbin B,
The secondary winding N2 is wound in a divided state. As a result, a state of loose coupling (for example, coupling coefficient k） 0.9) is obtained between the primary winding N1 and the secondary winding N2.

In the isolated converter transformer PIT, the polarities (winding direction) of the primary winding N1 and the secondary winding N2 are set.
And the connection of the rectifier diodes DO (DO1, DO2), the mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 becomes + M and -M. There is. For example, when the connection configuration shown in FIG. 9A is employed, the mutual inductance is + M, and when the connection configuration shown in FIG. 9B is employed, the mutual inductance is -M.

In the case of the orthogonal control transformer PRT of the power supply circuit shown in FIG. 6, since the controlled winding NR is inserted on the primary side, the control winding connected directly to the secondary side is controlled. Since the insulation distance from the NC must be ensured, a corresponding size is required. However, in the orthogonal control transformer PRT of the power supply circuit shown in FIG. 7, since the controlled winding NR1 is provided on the secondary side, Since the insulation distance from the control winding NC is not required, the size of the orthogonal control transformer PRT can be further reduced.

FIG. 10 shows operation waveforms of the power supply circuit shown in FIGS. 6 and 7 according to the switching cycle. FIG.
0 (a) indicates the resonance voltage Vcr obtained at both ends of the switching element Q1 // parallel resonance capacitor Cr. FIG. 10B shows a rectifier diode DO1 on the secondary side.
10 (c) shows a rectified current ID2 flowing through the rectifier diode DO1. In this figure, periods TON and TOFF indicate periods when the switching element Q1 is turned on and off, respectively, and periods DON and TON
DOFF indicates a period during which the rectifier diode DO1 is turned on and off, respectively.

The resonance voltage Vcr generated at both ends of the switching element Q1 / parallel resonance capacitor Cr is shown in FIG.
As shown in FIG. 0 (a), a sine-wave-like waveform is obtained during the period TOFF when the switching element Q1 is off, and the operation of the switching converter is of the voltage resonance type. The level of the pulse of the resonance voltage Vcr is about 1800 V at the peak as shown in the figure. This is because the impedance of the parallel resonance circuit on the primary side of the voltage resonance converter acts on the DC input voltage of 2Ei obtained by the voltage doubler rectification. As a secondary-side operation, the secondary-side rectifier diode DO1 has a period DON according to the waveform shown in FIG.
Thus, an operation in which a rectified current flows can be obtained. This is
The operation is based on + M (polarity mode) described with reference to FIG. As a result of the rectification operation, the resonance voltage V2 obtained at both ends of the secondary-side parallel resonance capacitor C2 is as shown in FIG.
As shown in FIG. 0 (b), during the period DOFF during which the rectifier diode DO1 is turned off, the DC output voltage EO (EO1, EO2)
A sinusoidal voltage having a peak level of about double to about 3.5 times is applied, and a voltage level equivalent to the DC output voltage EO (EO1, EO2) is applied during a period DON during which the rectifier diode DO1 is turned on.

[0019]

However, FIGS.
In the voltage resonance type converter having the configuration described with reference to FIG. 10, since the AC input voltage VAC corresponds to the AC 100 V system and the maximum load power is 120 W or more, the DC input voltage of 2Ei level is obtained by the voltage doubler rectification method. ing. Therefore, as shown in FIG. 10, a resonance voltage V of 1800 V is applied across the switching element Q1 and the parallel resonance capacitor Cr when the switching Q1 is off.
cr occurs. Therefore, for the switching element Q1 and the parallel resonance capacitor Cr, it is required to select a high withstand voltage product of 1800V. In this case, especially for the switching element Q1, the saturation voltage VCE (SAT), the accumulation time tSTG, the fall time tf are large, and the current amplification factor hFE is small, so that the switching loss and the drive power increase, and accordingly the power supply circuit Power loss increases.

In the configuration shown in FIG. 6 and FIG.
0 (b), the DC output voltage EO (EO1,
A secondary voltage V2 that is about twice to about 3.5 times EO2) is applied. For this reason, the rectifier diodes (DO1, DO2) used in the half-wave rectifier circuit have a DC output voltage of 135, for example.
Assuming that the voltage is about V, a high withstand voltage product of about 500 V is required, and the forward voltage drop VF and the reverse recovery time trr are correspondingly increased, which causes an increase in power loss. For example, if the switching frequency is increased, it is possible to reduce the size of various component elements. However, it is difficult to set the switching frequency fs to a high value due to the above-described circumstances. It has been found that the conversion efficiency decreases. Further, as described above, a voltage doubler rectifier circuit is required to obtain a highly reliable DC input voltage, so that two relatively large smoothing capacitors are required and the substrate area is increased.

In the power supply circuit shown in FIGS. 6 and 7,
There is a limit to the improvement of the control sensitivity of the constant voltage control circuit system. To compensate for this, the controlled winding N of the orthogonal control transformer PRT is used.
It is necessary to secure the required constant voltage control range by expanding the variable range of the inductance of R. For this purpose, it is necessary to increase the number of turns of the controlled winding NR, so that the orthogonal control transformer PRT becomes large and expensive.

[0022]

SUMMARY OF THE INVENTION Accordingly, the present invention has been made in view of the above-mentioned problems, and has been developed in view of the promotion of reduction in size and weight and cost of a resonance-type power supply circuit, as well as sensitivity to constant voltage control and power conversion efficiency. It is an object of the present invention to improve various characteristics as a resonance type power supply circuit.

Therefore, a rectifying / smoothing means for receiving a commercial AC power supply, generating a rectified smoothed voltage having a level corresponding to the same level as the commercial AC power supply level, and outputting the rectified smoothed voltage as a DC input voltage is loosely coupled. A gap is formed so as to obtain a required coupling coefficient, an insulating converter transformer provided for transmitting a primary output to a secondary side, and a switching element. A switching means configured to output to a primary winding of the transformer, and at least a leakage inductance component including a primary winding of the insulated converter transformer and a capacitance of a parallel resonance capacitor, and the operation of the switching means is controlled by a voltage. And a primary-side parallel resonance circuit of a resonance type. As a secondary-side configuration, a secondary-side series resonance capacitor is connected in series to the secondary winding of the insulating converter transformer, so that the leakage inductance component of the secondary winding of the insulating converter transformer and the secondary-side series A secondary-side series resonance circuit formed by the capacitance of the resonance capacitor, and the secondary-side series resonance capacitor inserted into the secondary-side rectified current path are formed, and are formed on the secondary winding of the insulating converter transformer. DC output voltage generating means configured to receive the input alternating voltage to obtain a secondary DC output voltage corresponding to approximately four times the input voltage level, and switching means according to the level of the secondary DC output voltage And a constant voltage control means for performing constant voltage control by variably controlling the switching frequency of the switching power supply circuit. And the.

A rectifying / smoothing means which receives a commercial AC power supply, generates a rectified smoothed voltage having a level corresponding to the same level as that of the commercial AC power supply, and outputs the rectified smoothed voltage as a DC input voltage. A gap is formed so as to obtain a coupling coefficient of, and an insulating converter transformer provided for transmitting a primary output to a secondary side, and a switching element; Switching means configured to output to the primary winding, a leakage inductance component including at least the primary winding of the insulated converter transformer, and the capacitance of a series resonant capacitor connected in series to the primary winding. And a primary side series resonance circuit that makes the operation of the switching means a current resonance type. As a secondary-side configuration, a secondary-side series resonance capacitor is connected in series to the secondary winding of the insulating converter transformer, so that the leakage inductance component of the secondary winding of the insulating converter transformer and the secondary-side series A secondary-side series resonance circuit formed by the capacitance of the resonance capacitor, and the secondary-side series resonance capacitor inserted into the secondary-side rectified current path are formed, and are formed on the secondary winding of the insulating converter transformer. DC output voltage generating means configured to receive an alternating voltage applied thereto and obtain a secondary DC output voltage corresponding to almost four times the input voltage level, and switching in accordance with the level of the secondary DC output voltage And a constant voltage control unit configured to perform constant voltage control by variably controlling a switching frequency of the unit. It was and.

According to the above configuration, the primary side is provided with a parallel resonance circuit in which the switching operation is a voltage resonance type or a series resonance circuit in which the switching operation is a current resonance type. Are loosely coupled. Then, on the secondary side, a secondary side DC output voltage is generated by a secondary side series resonance circuit and a quadruple voltage full-wave rectifier circuit to supply power to a load. That is, in the present embodiment, the resonance circuit is provided on the primary side and the secondary side, and for a required load condition, the electromagnetic energy obtained by the series resonance operation on the primary side and the resonance operation on the secondary side is obtained. Utilizing this, a DC output voltage is generated by a quadruple voltage full-wave rectifier circuit to cope with this. Accordingly, the primary side is not provided with a voltage doubler rectifier circuit but with a full-wave rectifier circuit that generates a rectified smoothed voltage corresponding to one time of the AC input voltage level.

[0026]

FIG. 1 is a circuit diagram showing a configuration example of a switching power supply circuit according to a first embodiment of the present invention. In the power supply circuit shown in this figure, a self-excited voltage-resonant switching converter using a single switching element (bipolar transistor) is provided for the primary side, as in the cases of FIGS. Is provided. In this figure, the same parts as those in FIGS. 6 and 7 are denoted by the same reference numerals, and description thereof will be omitted.
This power supply circuit is also a commercial AC power supply (AC input voltage)
Are AC100V systems in Japan and the United States, for example, and correspond to a condition of a load power of 120 W or more.

In the power supply circuit according to the present embodiment shown in this figure, as a rectifying / smoothing circuit for receiving an AC input voltage VAC and obtaining a DC input voltage, a full-wave circuit comprising a bridge rectifier circuit Di and a smoothing capacitor Ci. A rectifier circuit is provided to generate a rectified smoothed voltage Ei corresponding to a level that is one-time the AC input voltage VAC. That is, in the present embodiment, a voltage doubler rectifier circuit is not provided as in the related art. In this specification, a rectifier circuit that generates a rectified smoothed voltage Ei corresponding to one time of the level of the AC input voltage VAC as shown in FIG. 1 is also referred to as an “equal voltage rectifier circuit”. .

In the power supply circuit shown in this figure, an orthogonal drive transformer PRT (Power Regulating Transform) is used.
er). This orthogonal drive transformer PRT
Is a saturable reactor on which a detection winding ND, a drive winding NB, and a control winding NC are wound.
In addition to driving 1, the switching frequency is variably controlled as described later. Orthogonal drive transformer PRT
For example, a three-dimensional core is formed by joining the ends of the two magnetic legs of two double U-shaped cores having four magnetic legs, for example. Then, the detection winding N is applied to the predetermined two magnetic legs of the three-dimensional core in the same winding direction.
D, the drive winding NB is wound, and the control winding NC is wound in a direction orthogonal to the detection winding ND and the drive winding NB.

In this case, the orthogonal drive transformer PRT
Is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1 of the insulating converter transformer PIT, so that the switching output of the switching element Q1 is detected via the primary winding N1. It is transmitted to the winding ND. In the orthogonal drive transformer PRT, the drive winding NB is excited by the switching output obtained from the detection winding ND, and an alternating voltage is generated in the drive winding NB. This alternating voltage becomes a source of a driving voltage in the self-excited oscillation driving circuit.

When the orthogonal drive transformer PRT is provided, the following operation is obtained. The control circuit 1 changes the level of the control current (DC current) supplied to the control winding NC in accordance with the change in the secondary-side DC output voltage level, so that the drive winding wound on the orthogonal drive transformer PRT is changed. The inductance LB of the line NB is variably controlled. Thereby, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching element Q1 formed including the inductance LB of the drive winding NB changes. This is an operation of varying the switching frequency of the switching element Q1, and this operation has the effect of stabilizing the secondary DC output voltage.

As shown in FIG. 2, an insulated converter transformer PIT provided in a power supply circuit according to the present embodiment is an EE type in which E type cores CR1 and CR2 made of, for example, a ferrite material are combined such that their magnetic legs face each other. A primary winding N1 and a secondary winding N2 are provided for a center magnetic leg of the EE type core by using a divided bobbin B in which a winding portion is divided on a primary side and a secondary side. (N2A, N2B) are wound separately. And in this embodiment,
A gap G is formed for the center magnetic leg as shown in the figure. This gap G is formed by E-type cores CR1, C
R2 can be formed by forming the central magnetic leg shorter than the two outer magnetic legs. In the present embodiment,
The gap G has a required width of 1 mm or more. As a result, for example, a loose coupling with a coupling coefficient smaller than that of the insulating converter transformer PIT shown in FIG. 8 as a conventional example is achieved, so that a saturated state is hardly obtained. The coupling coefficient k in this case is, for example, k ≒ 0.85.

In this case, secondary windings N2A and N2B are provided on the secondary side of the power supply circuit according to the present embodiment. Secondary winding N of isolated converter transformer PIT of the present embodiment
2A is wound with a different number of turns from the conventional one as described later. One end of the secondary winding N2A is connected to the connection point between the anode of the rectifier diode DO1 and the cathode of the rectifier diode DO2 via the series connection of the series resonance capacitor Cs1, and the series connection of the series resonance capacitor Cs2. Is connected to the connection point between the anode of the rectifier diode DO3 and the cathode of the rectifier diode DO4.
The other end of the secondary winding N2A is connected to a connection point between the negative electrode of the smoothing capacitor CO10 and the positive electrode of the smoothing capacitor CO11. The anode of the rectifier diode DO2 and the cathode of the rectifier diode DO3 are connected to a connection point between the negative electrode of the smoothing capacitor CO10 and the positive electrode of the smoothing capacitor CO11. The smoothing capacitor CO10 and the smoothing capacitor CO11 are connected in series such that the negative electrode of the smoothing capacitor CO10 and the positive electrode of the smoothing capacitor CO11 are connected, and then the positive electrode of the smoothing capacitor CO10 is connected to the cathode of the rectifier diode DO1, And the negative electrode of CO11 are connected to the secondary side ground.

In such a connection form, as a result,
[Series resonant capacitor Cs1, rectifier diodes DO1, DO
2. A first voltage doubler full-wave rectifier circuit composed of a set of [smoothing capacitor CO10] and a second voltage doubler full-wave rectifier circuit composed of a set of [series resonance capacitor Cs2, rectifier diodes DO3, DO4, smoothing capacitor CO11]. Are formed, and these first
And the output of the second voltage doubler rectifier circuit (smoothing capacitor CO1
0, CO11) are connected in series and provided.
As a whole rectifier circuit combining the first and second voltage doubler rectifier circuits, both ends of a series-connected smoothing capacitor CO10-smoothing capacitor CO11 have an alternating voltage of the secondary winding N2A. A secondary output voltage corresponding to four times is obtained. That is, a quadruple voltage rectifier circuit is formed as an entire rectifier circuit combining the first and second voltage doubler rectifier circuits. The rectifying operation of the quadruple voltage rectifier circuit will be described later.

The series resonance capacitor Cs1 is connected in series with the on / off operation of the rectifier diodes DO1 and DO2 in the first voltage doubler rectifier circuit by its own capacitance and the leakage inductance component (L2) of the secondary winding N2A. Form a resonance circuit. Similarly, the series resonance capacitor C
s2 forms a series resonance circuit corresponding to the on / off operation of the rectifier diodes DO3 and DO4 in the second voltage doubler rectifier circuit by its own capacitance and the leakage inductance component (L2) of the secondary winding N2A. As the resonance frequency of these series resonance circuits, the parallel resonance frequency of the primary side parallel resonance circuit (N1, Cr) is fo1,
The series resonance frequency of the secondary side series resonance circuit (N2A, Cs1) is fo2, and the secondary side series resonance circuit (N2A, Cs1) is
Assuming that the series resonance frequency in 2) is fo3, fo1 ≒ fo
The capacitance of the secondary side series resonance capacitors Cs1 and Cs2 is selected so that 2 ≒ fo3.

As described above, in the power supply circuit of the present embodiment, the primary side is provided with the parallel resonance circuit for performing the voltage resonance type switching operation, and the secondary side is provided with the quadruple voltage rectification circuit, And two sets of voltage doubler rectifier circuits forming the quadruple voltage rectifier circuit are provided with two sets of series resonance circuits for obtaining current resonance operation. In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

Next, the operation of the above-described quadruple voltage rectifier circuit will be described. FIG. 3 is a waveform diagram showing an operation of a main part of the power supply circuit having the configuration shown in FIG. It should be noted that the waveform shown in this figure is such that the AC input voltage VAC = 100 V, the switching frequency fs = 100 at a maximum load power of 180 W.
Operation at KHz.

When a switching output is obtained on the primary winding N1 by the switching operation on the primary side, this switching output is excited by the secondary winding N2A. The operation on the primary side at this time is indicated by the switching output current I1 obtained in the primary winding N1 in FIG. The operation on the secondary side includes the rectification current I2 flowing commonly to the rectification diodes DO2 and DO4 in FIG. 3D, and the series resonance capacitor C in FIG.
This is shown as a resonance current (rectified current) I3 flowing through the rectifier diode DO1 via s1 and a resonance current (rectified current) I4 flowing through the rectifier diode DO1 via the series resonance capacitor Cs2 in FIG. The switching output current I1 shown in FIG. 3B has a sinusoidal waveform due to the resonance action of the primary side parallel resonance circuit.
(E) The resonance current (rectified current) I3 and the resonance current (rectified current) I4 shown in (f) are sinusoidal due to the resonance action of the series resonance circuits (N2A, Cs1) and (N2A, Cs2) on the secondary side, respectively. The following waveform is obtained. The secondary side rectified current I2 shown in FIG. 3D also has a sinusoidal waveform due to the resonance action of the secondary side series resonance circuits (N2A, Cs1) and (N2A, Cs2). Will be. Also,
Since the rectified current I2 on the secondary side is a current flowing through a rectified current path common to the first and second voltage doubler rectifier circuits,
The resonance current (rectified current) I3 and the resonance current (rectified current) I4 shown in FIGS.

Here, in describing the operation of the quadruple voltage rectifier circuit of the present embodiment, the operation of the first voltage doubler rectifier circuit composed of [series resonance capacitor Cs1, rectifier diodes DO1, DO2, smoothing capacitor CO10] will be described. . First, during a period T1 in which the rectifier diode DO1 is turned off and the rectifier diode DO2 is turned on, the secondary winding N1 operates in the depolarized mode in which the polarity of the primary winding N1 and the secondary winding N2A is -M. Due to the series resonance effect of the leakage inductance of the line N2A and the series resonance capacitor Cs1, an operation of charging the series resonance capacitor Cs1 with the rectified current I2 rectified by the rectifier diode DO2 is obtained. Then, during the period T2 in which the rectifier diode DO2 is turned off and the rectifier diode DO1 is turned on and the rectification operation is performed, the polarity of the primary winding N1 and the secondary winding N2A is + M, and the secondary polarity mode is set. The smoothing capacitor CO10 is charged in a state where series resonance occurs in which the potential of the series resonance capacitor Cs1 is added to the voltage induced in the winding N2A. As described above, the rectifying operation is performed by using both the positive polarity mode (+ M; forward operation) and the negative polarity mode (-M; flyback operation), so that the smoothing capacitor CO10 has: A DC voltage (rectified and smoothed voltage) corresponding to almost twice the induced voltage of the secondary winding N2A is obtained. The same operation is performed by the second voltage doubler full-wave rectifier circuit composed of a set of [series resonance capacitor Cs2, rectifier diodes DO3 and DO4, and smoothing capacitor CO11]. A DC voltage corresponding to almost twice the induced voltage of the line N2A is obtained.

As described above, the voltage doubler rectification operation is performed by each of the first and second voltage doubler rectifier circuits. As a result, a secondary capacitor is connected across the smoothing capacitor CO10 and the smoothing capacitor CO11 connected in series. A secondary side DC output voltage EO1 corresponding to approximately four times the induced voltage of the winding N2A is obtained.

As described above with reference to FIG. 2, the configuration for obtaining the voltage doubler full-wave rectification operation in each of the first and second voltage doubler rectifier circuits is the same as that of the isolated converter transformer PIT. On the other hand, the gap G is formed and loose coupling is performed by a required coupling coefficient, thereby achieving a state in which the state is more unlikely to be saturated. For example, when the gap G is not provided for the insulating converter transformer PIT as in the related art, there is a high possibility that the insulating converter transformer PIT becomes saturated during flyback operation and the operation becomes abnormal. It is difficult to desirably perform the voltage doubling rectification operation as described in the first embodiment.

In the secondary winding N2B, after a center tap provided on the secondary winding N2B is grounded, an ordinary full-wave rectifier circuit composed of rectifier diodes D05 and D06 and a smoothing capacitor CO2 is formed. As a result, the secondary side DC output voltage EO2 is obtained.

According to the configuration of the power supply circuit described above, as the quadruple voltage rectifier circuit on the secondary side, two sets of voltage doubler full-wave rectifier circuits are in a state where the mutual inductance is in the operation mode of + M and -M. Is used to perform voltage doubler full-wave rectification. The voltage doubler rectification operation is performed by transmitting electromagnetic energy transmitted from the primary winding to the secondary winding of the isolated converter transformer PIT, which has been loosely coupled (coupling coefficient k = 0.85), to a primary-side parallel resonance circuit ( Voltage resonance circuit) and the series resonance circuit (current resonance circuit) on the secondary side, which can reduce the resonance impedance of the resonance circuit between the primary side and the secondary side. It also operates to amplify the secondary side resonance current. Then, the power supplied to the load side also increases due to the amplifying action of the secondary side resonance current, thereby increasing the maximum load power.

By increasing the maximum load power as described above, in the present embodiment, it is necessary to use a double voltage rectification method as a rectifying and smoothing circuit for generating a DC input voltage to cover the load power. There is no. For this reason, as described with reference to FIG. 1, for example, the configuration of a normal equal-voltage rectifier circuit using a bridge rectifier circuit can be employed. Thereby, for example, the rectified smoothed voltage Ei when the AC input voltage VAC = 144 V becomes lower than 200 V. Switching element Q1 / secondary parallel resonance capacitor C shown in FIG.
The resonance voltage Vcr obtained at both ends of the r parallel connection circuit is
When the primary side parallel resonance circuit acts on the rectified smoothed voltage Ei, the switching element Q1 is generated when the switching element Q1 is turned off. In the present embodiment, the rectified smoothed voltage Ei is generated as described above.
Is about half of that at the time of voltage doubler rectification, so that the resonance voltage Vcr is about the resonance voltage Vcr (1800 V) generated in the power supply circuit as the conventional example shown in FIG. 6 or FIG. It will be kept below 1/2. That is, the resonance voltage Vcr is suppressed to about 700 V at the peak. Therefore, in the present embodiment, for the switching element Q1 and the parallel resonance capacitor Cr,
For example, a withstand voltage product of about 900 V may be selected.

Further, on the secondary side, a quadruple voltage rectifier circuit comprising two sets of voltage doubler full-wave rectifier circuits for performing a rectification operation during both positive and negative excitation voltages of the secondary winding N2A is provided. In this embodiment, as can be seen from the waveform of the voltage V5 across the rectifier diode DO4 shown in FIG. 3 (g), for example, the voltage applied to each secondary rectifier diode is
It is clamped to a level corresponding to 1/2 of the secondary side DC output voltage EO. As a result, as the rectifier diode forming the quadruple voltage rectifier circuit on the secondary side, a withstand voltage product corresponding to a level corresponding to 1/2 of the secondary side DC output voltage EO may be selected.

Further, since the secondary DC output voltage is obtained by the quadruple voltage rectifier circuit, for example, a level equivalent to the secondary DC output voltage obtained by the equal voltage rectifier circuit can be obtained. For example, the secondary winding N of the present embodiment
For 2A, the number of turns is only 1/4 of the conventional number. This reduction in the number of turns leads to a reduction in the size and weight of the insulating converter transformer PIT and a reduction in cost.

Further, in the present embodiment, as described above, the switching element Q1, the parallel resonance capacitor C
As for the rectifier diode on the secondary side, a low breakdown voltage product can be used as compared with a conventional rectifier diode, so that the element is less expensive. Therefore, for example, the switching element Q
1 and secondary side rectifier diodes having improved characteristics (the switching element Q1 can be used to obtain the saturation voltage VCE
(SAT), accumulation time tSTG, fall time tf, current amplification factor hFE
A rectifier diode having good characteristics such as a forward voltage drop VF and a reverse recovery time trr can be selected. Due to such improvement in characteristics, in the present embodiment, the switching frequency can be set higher than in the conventional case, and accordingly, reduction of power loss and reduction in size and weight of various components are promoted. That is, it is possible to improve various characteristics such as power conversion efficiency as compared with the related art, and to promote reduction in size, weight, and cost.

Further, from the viewpoint of reducing the size and weight of the power supply circuit, the conventional configuration including the voltage doubler rectifier circuit for generating the DC input voltage requires two sets of rectifier diodes and a smoothing capacitor. However, in the present embodiment, for example, a full-wave rectifier circuit using a normal bridge rectifier circuit can be used, so that a set of a block-type smoothing capacitor and a bridge rectifier diode can be employed. Therefore, cost reduction and downsizing of parts can be achieved.

Further, since the primary side rectifier circuit is the same-size voltage rectifier circuit, the number of turns of the primary winding N1 is reduced as compared with the conventional case, and as a result, the orthogonal drive transformer PR
Even if the variable range of the inductance by T is narrower than in the past, the constant voltage control is performed stably. As a result, the number of windings of the control winding NC and the like wound around the orthogonal drive transformer PRT is reduced, and the orthogonal drive transformer PR is reduced.
T can be reduced in size and weight and cost can be reduced.

Further, as a result of variably controlling the switching frequency as a constant voltage control operation on the primary side, the resonance impedance of the parallel resonance circuit on the primary side and the resonance impedance of two series resonance circuits on the secondary side are controlled simultaneously. Operation is obtained. Therefore, the control sensitivity for the constant voltage control is improved as compared with the conventional case, and for example, even under the condition of the same switching frequency variable width (control current variable width) as in the conventional case, the variation in the AC input voltage and the load power is suppressed. The control range is expanded. This can also contribute to downsizing of the above-described orthogonal drive transformer PRT.

The actual experimental results show that the AC input voltage V
At the time of AC = 100 V, the power conversion efficiency is improved to 93%, and the constant voltage control range is expanded even when no load is applied as the minimum load power. The switching frequency at no load is, for example, 195 KHz.

FIG. 4 is a circuit diagram showing a configuration example of a switching power supply circuit according to a second embodiment of the present invention.
This power supply circuit employs a self-excited current resonance type converter. This power supply circuit also has a configuration in which a commercial AC power supply (AC input voltage) is, for example, an AC 100 V system such as in Japan or the United States and the load power is 120 W or more. In this figure, the same parts as those in FIGS. 1, 6, and 7 are denoted by the same reference numerals, and description thereof will be omitted.

The current resonance type switching converter of this power supply circuit has two switching elements Q 1,
After Q2 is half-bridge-coupled, it is connected so as to be inserted between the connection point on the positive electrode side of the smoothing capacitor Ci and ground. The switching elements Q1, Q
The start resistor RS between each collector and base
1, RS2 is inserted. Also, the switching elements Q1,
Clamp diodes DD1 and DD2 are inserted between each base and emitter of Q2. A series connection circuit including a resonance capacitor CB1, a base current limiting resistor RB1, and a driving winding NB1 is inserted between the base of the switching element Q1 and the collector of the switching element Q2. The resonance capacitor CB1 has its own capacitance,
A series resonance circuit for self-excited oscillation is formed together with the inductance LB1 of the drive winding NB1 described below, thereby determining the switching frequency of the switching element Q1. Similarly, a series connection circuit including a resonance capacitor CB2, a base current limiting resistor RB2, and a driving winding NB2 is inserted between the base of the switching element Q2 and the primary side ground. A series resonance circuit for self-excited oscillation is formed together with the inductance LB2 of the drive winding NB2 to determine the switching frequency of the switching element Q2.

One end of the primary winding N1 of the insulating converter transformer PIT is connected to the contact point between the emitter of the switching element Q1 and the collector of the switching element Q2 via the resonance current detection winding ND, so that a switching output is obtained. To be. The other end of the primary winding N1 is grounded to the primary side ground via a series resonance capacitor C1. In this case, the series resonance capacitor C1 and the primary winding N1 are connected in series, but the capacitance of the series resonance capacitor C1 and the leakage inductance of the insulating converter transformer PIT including the primary winding N1 (series resonance winding) are included. (Leakage inductance) L1 forms a series resonance circuit for making the operation of the switching converter a current resonance type. It should be noted that the insulation converter transformer PIT shown in this figure employs the same structure as that described with reference to FIG. 2 so that a loosely coupled state with a coupling coefficient k = 0.85 can be obtained. Has been.

In the power supply circuit shown in FIG.
An orthogonal drive transformer PRT is provided. This orthogonal drive transformer PRT also has a switching element Q
1. While driving Q2, constant voltage control is performed as described later. In the case of the orthogonal drive transformer PRT shown in this figure, the drive windings NB1, NB2 and the resonance current detection winding N
D is wound, and a control winding N is provided for each of these windings.
It is configured as an orthogonal saturable reactor in which C is wound in the orthogonal direction. Also in this case, the driving winding N
B1 and the drive winding NB2 are wound so that mutually opposite voltages are generated.

As a switching operation of the power supply circuit having the above-described configuration, when a commercial AC power supply is first turned on, for example, the switching elements Q1, Q2 are connected via the starting resistors RS1, RS2.
2 is supplied with a starting current. For example, if the switching element Q1 is turned on first,
The switching element Q2 is controlled to be turned off.
As an output of the switching element Q1, a resonance current flows through the resonance current detection winding ND → the primary winding N1 → the series resonance capacitor C1, and when the resonance current becomes zero, the switching element Q2 is turned on and the switching element Q1 is turned on. It is controlled to be off. Then, a resonance current flows in the opposite direction through the switching element Q2. Or later,
A self-excited switching operation in which the switching elements Q1 and Q2 are turned on alternately is started. As described above, by using the terminal voltage of the smoothing capacitor Ci as the operating power source, the switching elements Q1 and Q2 alternately open and close,
A drive current close to the resonance current waveform is supplied to the primary winding N1 of the insulating converter transformer, and an alternating output is obtained at the secondary winding N2.

The constant voltage control by the orthogonal drive transformer PRT is performed as follows. For example, the secondary output voltage EO1 is changed by the fluctuation of the AC input voltage or the load power.
Is varied, the control circuit 1 variably controls the level of the control current flowing through the control winding NC in accordance with the variation of the secondary output voltage EO1. Under the influence of the magnetic flux generated in the orthogonal drive transformer PRT by the control current, the state of the saturation tendency in the orthogonal drive transformer PRT changes, which acts to change the inductance of the drive windings NB1 and NB2. Thus, the condition of the self-excited oscillation circuit is changed so that the switching frequency is changed. In the power supply circuit shown in FIG.
Although the switching frequency is set in a frequency region higher than the resonance frequency of the series resonance circuit of the primary winding N1 and the primary winding N1, for example, when the switching frequency is increased, the switching frequency is separated from the resonance frequency of the series resonance circuit. To be. Thereby, the resonance impedance of the primary side series resonance circuit with respect to the switching output increases. By increasing the resonance impedance in this way, the drive current supplied to the primary winding N1 of the primary side series resonance circuit is suppressed, and as a result, the secondary side output voltage is suppressed, and the constant voltage Control will be achieved.
That is, constant voltage control by the switching frequency control method is performed.

In the power supply circuit shown in FIG. 4, a small-capacity ceramic capacitor Cc1 is connected between each collector and emitter of the switching elements Q1 and Q2.
Cc2 is connected in parallel. The ceramic capacitors Cc1 and Cc2 are also provided to absorb the switching noise of the switching elements Q1 and Q2. Here, the ceramic capacitors Cc1 and Cc2 correspond to the switching frequency that changes relatively widely by the constant voltage control operation. Also, it has an operation for obtaining a zero-voltage switching operation when the switching elements Q1 and Q2 are turned off.

The configuration on the secondary side is the same as that of the power supply circuit shown in FIG. That is, the secondary winding N
To 2A, a quadruple voltage rectifier circuit including two sets of voltage doubler rectifier circuits having a series resonance circuit is connected. Therefore,
As a power supply circuit according to the present embodiment, a series resonance circuit for providing a switching converter with current resonance type operation is provided on the primary side, and two sets forming a quadruple voltage rectification circuit are provided on the secondary side. As a complex resonance type switching converter provided with the series resonance circuit described above.

FIG. 5 shows operation waveforms of main parts in the switching cycle of the power supply circuit shown in FIG. In this figure, the periods TON and TOFF are respectively defined by the switching element Q2
Indicates a period during which is turned on and off. In the period T1, the rectifier diodes DO2 and DO4 are on and the rectifier diodes DO and
The period T2 is a period in which the rectifier diodes DO1 and DO3 are on and the rectifier diodes DO2 and DO4 are in a period T2.
Indicates a period in which is turned off.

As the operation on the switching element Q2 side when the switching operation by the switching elements Q1 and Q2 is performed, the switching voltage Vcr obtained between the collector and the emitter of the switching element Q2 is as shown in FIG.
As shown in FIG. 3A, a waveform is obtained in which a pulse is generated during the period TOFF when the switching element Q2 is off. In the present embodiment, the switching voltage V
The pulse of cr is substantially rectified and smoothed voltage Ei (for example, about 14
4V). As shown in FIG. 3C, the collector current Icp flowing through the switching element Q2 is at the 0 level during the period TOFF, and a waveform having a sinusoidal waveform component as shown in FIG. Can be Then, as shown in FIG. 3B, a sine-wave waveform corresponding to the switching cycle is obtained as the series resonance current I1 flowing through the primary winding N1 as a switching output.

FIG. 3D shows the operation on the secondary side.
(E) The rectified current I2, the resonant current I3,
The waveforms of I4 and the voltage V5 across the rectifier diode DO4 are:
After setting the cycle substantially corresponding to the switching cycle on the primary side, the same operation as that of the power supply circuit of the first embodiment shown in FIG. 3 is obtained.

With such a configuration, the secondary parallel resonance circuit (N2A, C2) performs a resonance operation, thereby increasing the maximum load power and reducing the primary-side parallel resonance circuit as in the first embodiment. The control range is expanded by simultaneous control of the resonance impedance on the primary side and the secondary side accompanying the switching frequency control, and as a result, the same effect as in the first embodiment is obtained.

Incidentally, the power supply circuit shown in FIG.
It is a so-called single-ended type voltage resonance type converter in which a switching operation is performed by one switching element. In a voltage resonance type converter, two switching elements are alternately turned on / off. There is also known a configuration using a so-called push-pull system in which a switching operation is performed. The current resonance type converter shown in FIG. 4 employs a configuration in which two switching elements are half-bridge-coupled.
There is also known a configuration in which four switching elements are full-bridge-coupled to alternately turn on / off a set of two switching elements. The present invention is also applicable to the above-described voltage-resonant converter using the push-pull method and to a current-resonant converter using a full-bridge coupling configuration. That is, the resonance converter having the above configuration is provided with a series resonance circuit for the secondary side.
By providing the voltage doubler rectifier circuit, the same effect as that obtained by the configuration shown in FIG. 1 or 4 can be obtained. In such a configuration, the load power that can be handled increases compared to the configuration shown in FIG. 1 or FIG.
This is suitable for a case where a maximum load power of 0 W or more is supported.

It should be noted that the power supply circuit of the present invention may be appropriately modified in accordance with actual use conditions in addition to the configuration shown in FIGS. For example, in each of the above embodiments, a configuration of switching drive by a self-excited system is adopted, but the present invention is also applicable to a configuration in which a switching element is driven by a separately excited system. Further, as the switching element, other component elements other than the bipolar transistor and the MOS-FET may be employed.

[0065]

As described above, according to the present invention, as a switching power supply circuit, a voltage resonance type converter (primary side parallel resonance circuit) or a current resonance type converter (primary side series resonance circuit) is provided on the primary side. The operation mode in which the mutual inductances of the primary winding and the secondary winding have opposite polarities by loosely coupling the insulating converter transformer (+
(M / -M). And on the secondary side, a secondary side series resonance circuit is connected in series to the secondary winding to form a series resonance circuit, and a quadruple voltage full-wave rectifier circuit using this series resonance circuit is provided. Thus, a secondary side DC output voltage corresponding to four times the alternating voltage (excitation voltage) obtained in the secondary winding is obtained.

As a result of supplying power to the load by the above-described configuration on the secondary side, according to the present invention, it is possible to increase the maximum load power that can be handled, as compared with the related art. Along with this, the primary side is not a voltage doubler rectifier circuit, but a full-wave rectifier circuit is used to input a rectified smoothed voltage at a level corresponding to the AC input voltage level. It will be able to respond.

The following can be said from the above configuration. For example, conventionally, in the case where the above condition is satisfied, the AC input voltage level of 2 is controlled by the voltage doubler rectifier circuit.
It is necessary to obtain a rectified and smoothed voltage corresponding to the doubled voltage.Therefore, it is necessary to select a withstand voltage product corresponding to the switching voltage generated according to the rectified and smoothed voltage level for the switching element and the parallel resonance capacitor on the primary side. Was.

On the other hand, in the present invention, since the switching voltage depending on the rectified smoothing voltage level is about 1/2 of the conventional one, the withstand voltage of the switching element and the resonance capacitor on the primary side is about 1/2 of the conventional one. Goods can be used. On the secondary side, the quadruple voltage rectifier circuit performs a rectification operation in which the alternating voltage performs the rectification operation in both the positive and negative periods. As a result, the voltage applied to the rectifier diode is almost equal to the rectified smooth voltage level. Since the rectifier diode is suppressed to about ２, a rectifier diode having a lower withstand voltage than the conventional rectifier diode can also be selected. As a result, it is possible to reduce costs for the switching element, the primary-side parallel resonance capacitor, the secondary-side rectifier diode, and the like. In addition, it is also possible to easily select a switching element and a secondary rectifier diode having improved characteristics and set a high switching frequency, thereby improving the power conversion efficiency. Also, it is possible to reduce the size and weight of circuit components around the switching element. Further, as described above, since the circuit for obtaining the rectified smoothed voltage from the commercial AC power supply is a normal equal-voltage rectifier circuit, for example, a normal set of a block-type smoothing capacitor and a bridge rectifier diode are used. Therefore, the cost and the circuit scale can be reduced in this respect as well.

Further, in the present invention, a quadruple voltage rectifier circuit is employed for the rectifier circuit provided on the secondary side, so that a DC output voltage of the same level as that obtained when, for example, an equal voltage rectifier circuit is provided is obtained. Then, the number of turns of the secondary winding can be reduced to about 1/4 of the conventional number.

Further, in the circuit configuration of the present invention, regardless of whether the primary side employs a voltage resonance type converter or a current resonance type converter, the primary side performs constant voltage control by variably controlling the switching frequency. At this time, the resonance impedance of the primary side series resonance circuit and the resonance impedance of the secondary side resonance circuit (series resonance circuit or parallel resonance circuit) are changed according to the switching frequency being varied. Since the control operation can be performed at the same time, the constant voltage control range is expanded. Further, thereby, for example, the number of turns of the control winding and the like in the orthogonal transformer can be reduced, and reduction in size, weight, and cost can be achieved. In addition, since the primary side rectifier circuit is a unity voltage rectifier circuit, the number of turns of the primary winding N1 can be reduced as compared with the conventional case, so that the number of turns of the winding can be reduced. Is achieved.

As described above, according to the present invention, the cost, size, and weight of the resonance type power supply circuit and the improvement of various characteristics such as the power conversion efficiency and the constant voltage control range are promoted.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration example of a power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a cross-sectional view illustrating a configuration of an insulating converter transformer of the power supply circuit according to the present embodiment.

FIG. 3 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

FIG. 4 is a circuit diagram illustrating a configuration example of a power supply circuit according to a second embodiment;

5 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

FIG. 6 is a circuit diagram showing a configuration of a power supply circuit as a conventional example.

FIG. 7 is a circuit diagram showing a configuration of a power supply circuit as a conventional example.

FIG. 8 is a cross-sectional view illustrating a configuration of an insulating converter transformer as a conventional example.

FIG. 9 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

FIG. 10 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIGS. 6 and 7.

[Explanation of symbols]

1 control circuit, Ci smoothing capacitor, Cs1, Cs2
Secondary side series resonance capacitor, Di bridge rectifier circuit,
DO1, DO2, DO3, DO4 Rectifier diode, PRT orthogonal drive transformer, PIT isolation converter transformer, NC control winding, Q1, Q2 switching element

Claims (6)

[Claims] A rectifying / smoothing means for receiving a commercial AC power supply, generating a rectified smoothed voltage having a level corresponding to the same level as the commercial AC power supply level, and outputting the rectified smoothed voltage as a DC input voltage; And an insulating converter transformer provided with a gap so as to obtain the coupling coefficient of the above, and provided for transmitting the primary side output to the secondary side; and a switching element. A switching means configured to output to the primary winding of the converter, and at least a leakage inductance component including the primary winding of the insulated converter transformer and a capacitance of a parallel resonance capacitor, and the operation of the switching means is controlled by a voltage. A secondary parallel resonance circuit to be a resonance type, and a secondary By connecting a series resonance capacitor in series, a secondary series resonance circuit formed by a leakage inductance component of a secondary winding of the insulating converter transformer and a capacitance of the secondary series resonance capacitor, The alternating voltage obtained by inserting the secondary series resonance capacitor into the side rectified current path and obtained in the secondary winding of the insulated converter transformer is input to correspond to almost four times the input voltage level. DC output voltage generating means configured to obtain a secondary DC output voltage; and performing constant voltage control by variably controlling a switching frequency of the switching means according to the level of the secondary DC output voltage. And a constant voltage control means. 2. The switching power supply circuit according to claim 1, wherein said switching means includes a self-excited oscillation drive circuit for performing a switching operation in a self-excited manner. 3. The constant voltage control means comprises: at least a drive winding forming the self-excited oscillation drive circuit; and a winding wound in a direction orthogonal to the drive winding, and 3. The switching power supply circuit according to claim 2, further comprising an orthogonal transformer as a saturable reactor wound with a control winding to which a control current of a level varied according to the fluctuation is supplied. 4. A rectifying / smoothing means for receiving a commercial AC power supply, generating a rectified smoothed voltage having a level corresponding to the same level as the commercial AC power supply, and outputting the rectified smoothed voltage as a DC input voltage. And an insulating converter transformer provided with a gap so as to obtain the coupling coefficient of the above, and provided for transmitting the primary side output to the secondary side; and a switching element. A switching means configured to output to a primary winding of at least a leakage inductance component including a primary winding of the insulating converter transformer, and a capacitance of a series resonance capacitor connected in series to the primary winding. And a primary side series resonance circuit that makes the operation of the switching means a current resonance type. By connecting the secondary side series resonance capacitor to the secondary winding of the inverter transformer in series, the leakage inductance component of the secondary winding of the insulating converter transformer and the capacitance of the secondary side series resonance capacitor A secondary-side series resonance circuit formed, and the secondary-side rectified current path is formed by inserting the secondary-side series resonance capacitor into the secondary-side rectified current path. And a DC output voltage generating means configured to obtain a secondary DC output voltage corresponding to approximately four times the input voltage level; and the switching means according to the level of the secondary DC output voltage. And a constant voltage control means for performing constant voltage control by variably controlling the switching frequency. circuit. 5. The switching power supply circuit according to claim 4, wherein said switching means includes a self-excited oscillation drive circuit for performing a switching operation in a self-excited manner. 6. The constant voltage control means comprises: at least a driving winding forming the self-excited oscillation driving circuit; and a winding wound in a direction orthogonal to the driving winding, and 6. The switching power supply circuit according to claim 5, further comprising: a quadrature transformer as a saturable reactor wound with a control winding to which a control current of a level varied according to the fluctuation is supplied.