JP2000125552A - Voltage-resonance type switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To improve power conversion efficiency and to miniaturize and to make a circuit light in weight. SOLUTION: A secondary side DC output voltage can be obtained by a full- wave rectifying circuit (a set of [rectifying diodes D01 and D02 and a smoothing capacitor C01] and a set of [rectifying diodes D03 and D04 and a smoothing capacitor C02]), where a parallel resonance capacitor C2 is connected to increase the maximum load power at a secondary side as a voltage-resonance type switching power supply which corresponds to relatively high-load conditions exceeding 150 W in an AC input voltage AC 100 V system. A primary side is composed so that a rectification smoothing voltage at a level, corresponding to an AC input voltage level can be inputted by a normal full-wave rectifying circuit, instead of a double-voltage rectifying circuit. Then, a boost circuit (NR, a diode DB for boosting, and a smoothing capacitor CiB), including coil winding NR to be controlled of a cross-type control transformer PRT is provided at the primary side to obtain a boost smoothing voltage. Then, by the constant voltage control operation of the cross-type control transformer PRT, control is made so that a boost smoothing voltage becomes constant, thus reducing current that flows to the coil winding NR to be controlled.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a voltage resonance type switching power supply circuit provided as a power supply for various electronic devices.

[0002]

2. Description of the Related Art As a so-called soft switching power supply circuit, a switching power supply circuit having a voltage resonance type switching converter is known. The voltage resonance type switching converter has a low noise because a smooth waveform is obtained for the switching output voltage pulse and the switching output current flowing into the insulating converter transformer, and can be configured with a relatively small number of parts.

FIG. 10 is a circuit diagram showing an example of a voltage resonance type switching power supply circuit which can be constructed based on the invention proposed by the present applicant. In the switching power supply circuit shown in this figure, for example, a commercial AC power supply in Japan or the United States is a so-called AC 100 V system, and corresponds to a condition where the maximum load power is 150 W or more.

The switching power supply circuit shown in FIG. 1 is provided with a so-called voltage doubler rectification circuit comprising rectification diodes Di1, Di2 and smoothing capacitors Ci1, Ci2 as a rectification and smoothing circuit for rectifying and smoothing the AC power supply AC. Can be In this voltage doubler rectifier circuit, assuming that a DC input voltage corresponding to, for example, one time the peak value of the AC input voltage VAC is Ei, a DC input voltage 2Ei that is approximately twice that of the DC input voltage Ei.
Generate For example, if the AC input voltage VAC is 144V, the DC input voltage 2Ei is about 400V. As described above, the reason why the voltage doubler rectifier circuit is employed as the rectifier / smoothing circuit is that the AC input voltage is 100 V AC as described above.
The system is designed to cope with a relatively heavy load condition in which the maximum load power is 150 W or more. That is, by increasing the DC input voltage to twice the normal value, the amount of current flowing into the subsequent switching converter is suppressed, and the reliability of the components forming the switching power supply circuit is ensured. . In the voltage doubler rectifier circuit shown in this figure, an inrush current limiting resistor Ri is inserted in the rectified current path, for example, so as to suppress the inrush current flowing into the smoothing capacitor when the power is turned on. I have.

The voltage-resonant type switching converter shown in FIG. 1 has a self-excited type provided with a single switching element Q1. In this case, a high withstand voltage bipolar transistor (BJT;
Junction type transistor). The base of the switching element Q1 is connected to the positive side of the smoothing capacitor Ci1 (rectified smoothed voltage 2Ei) via a starting resistor RS so that a base current at the time of starting can be obtained from the rectified smoothing line. A self-excited oscillation resonance circuit including an inductor LB, a drive winding NB, a resonance capacitor CB, and a damping resistor RB is connected in series between the base of the switching element Q1 and the temporary ground. In this case, the drive winding NB is connected to an insulation converter transformer PIT (Power Isol
ation Transformer) and the inductor LB
At the same time, a required inductance for setting the switching frequency is obtained. A clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a damper current flowing when the switching element Q1 is off. The collector of the switching element Q1 is an isolated converter transformer PI
T is connected to one end of the primary winding N1 and the emitter is grounded.

A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr
Is the combined capacitance (L1, LR) obtained by connecting the primary winding N1 of the isolated converter transformer PIT described later and the controlled winding NR of the orthogonal control transformer PRT (Power Regulating Transformer) in series. Thereby, a parallel resonance circuit of the voltage resonance type converter is formed. Although the detailed description is omitted here, when the switching element Q1 is turned off, the voltage Vd across the resonance capacitor Cr is obtained by the action of the parallel resonance circuit.
cr is actually a sinusoidal pulse waveform so that a voltage resonance type operation can be obtained.

The insulated converter transformer PIT is for transmitting the switching output of the switching element Q1 to the secondary side. In this case, the insulated converter transformer PIT is used.
One end of the primary winding N1 of T is connected to the collector of the switching element Q1, and the other end is connected in series with the controlled winding NR of the orthogonal control transformer PRT as shown in the figure.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, a parallel resonance circuit is formed by connecting the secondary side parallel resonance capacitor C2 to the secondary winding N2 in parallel. With this parallel resonance circuit,
The alternating voltage excited in the secondary winding N2 is a resonance voltage,
This resonance voltage is supplied to two sets of half-wave rectifier circuits, a half-wave rectifier circuit including a rectifier diode DO1 and a smoothing capacitor CO2, and a half-wave rectifier circuit including a rectifier diode DO2 and a smoothing capacitor CO2. Then, the DC output voltages EO1 and EO2 are obtained by these two sets of half-wave rectifier circuits. The rectifier diodes DO1 and DO2 forming the half-wave rectifier circuit use a high-speed rectifier to rectify the alternating voltage in the switching cycle.

The control circuit 1 compares, for example, a DC voltage output on the secondary side with a reference voltage and outputs a DC current corresponding to the error.
The control winding N of the orthogonal control transformer PRT is used as the control current.
This is the error amplifier that supplies C. In this case, the DC voltage output EO1 is input to the control circuit 1 as the detection voltage, and the DC output voltage EO2 is input as the operation power supply.

For example, when the DC output voltage EO2 on the secondary side fluctuates due to the fluctuation of the AC input voltage VAC or the minimum load power, the control circuit 1 controls the control current flowing through the control winding NC to 10 mA to 40 mA, for example. Vary in range.
Thus, the inductance LR of the controlled winding NR is changed in a range of, for example, 0.1 mH to 0.6 mH. Here, FIG. 16 shows an inductance DC superposition characteristic of the orthogonal control transformer PRT. In this figure,
The horizontal axis represents the current IR flowing through the controlled winding NR, and the vertical axis represents the inductance LR of the controlled winding NR. The parameter is a control current IC flowing through the control winding NC. In this case, the inductance LR tends to increase as the level of the control current IC decreases.

Since the controlled winding NR forms a parallel resonance circuit for obtaining a voltage-resonant switching operation as described above, this parallel resonance circuit is not controlled for a fixed switching frequency. The resonance condition of the circuit is changed. At both ends of the parallel connection circuit of the switching element Q1 and the parallel resonance capacitor Cr, a sinusoidal resonance pulse is generated by the action of the parallel resonance circuit corresponding to the OFF period of the switching element Q1, but the resonance condition of the parallel resonance circuit , The width of the resonance pulse is variably controlled. That is, PWM (P
(ulse Width Moduration) control operation is obtained. The PWM control of the width of the resonance pulse means the control of the off period of the switching element Q1, which in other words means variably controlling the on period of the switching element Q1 under the condition of a fixed switching frequency. . By variably controlling the ON period of the switching element Q1 in this manner, the switching output transmitted from the primary winding N1 forming the parallel resonance circuit to the secondary side changes, and the DC output voltage ( The output levels of EO1, EO2) are also changed. As a result, the secondary DC voltage (EO1, EO2) can be made constant. Note that such a constant voltage control method is hereinafter referred to as an inductance control method.

FIG. 11 is a circuit diagram showing another example of a voltage resonance type switching power supply circuit which can be configured based on the invention proposed by the present applicant. Note that FIG.
In FIG. 10, the same parts as those in FIG. In the power supply circuit shown in this figure, an example is shown in which the controlled winding NR of the orthogonal control transformer PRT is provided on the secondary side. In the case shown in this figure, the controlled winding NR
Are connected in series between the end of the secondary winding N2 and the anode of the rectifier diode DO2.

In the circuit shown in FIG. 11, a winding N3 is inserted in series between the tap output of the secondary winding N2 and the anode of the rectifier diode DO1. In this case, the winding N3 is connected to the orthogonal control transformer PRT.
It is wound in a winding direction orthogonal to the control winding NC.

Thus, even if the controlled winding NR of the orthogonal control transformer PRT is provided on the secondary side,
Constant voltage control is achieved by the same operation as described above.

Here, the structure of the insulated converter transformer PIT provided in the power supply circuit shown in FIGS. 10 and 11 is shown in cross section by FIG. Insulation converter transformer P
IT is, for example, E type cores CR1 and C made of ferrite material.
An EE-type core is formed by combining R2 with their magnetic legs facing each other. At this time, no gap is formed in the center magnetic leg as shown in the figure. Then, the primary winding N1 (and the driving winding N
B), the secondary winding N2 is wound in a divided state. Thereby, the primary winding N1 and the secondary winding N2
With, a state of loose coupling (for example, coupling coefficient k ≒ 0.9) is obtained.

In the isolated converter transformer PIT, the polarities (winding direction) of the primary winding N1 and the secondary winding N2 are set.
And the connection of the rectifier diodes DO (DO1, DO2), the mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 becomes + M and -M. There is. For example, when the connection configuration shown in FIG. 13A is employed, the mutual inductance is + M, and when the connection configuration shown in FIG. 13B is employed, the mutual inductance is -M.

Next, the structure of the orthogonal control transformer PRT provided in the power supply circuit shown in FIG. 10 or 11 is shown in FIG.
Is shown in a perspective view. The orthogonal control transformer PR shown in FIG.
T has cores CR11 and CR12 made of a ferrite material having four magnetic legs.
Are assembled such that their magnetic legs face each other to form a core body. In this case, the core body is 23 ×
23 × 23 (mm) size, each magnetic leg is 7 × 7m
m. The control winding NC is wound around the two magnetic legs of the four magnetic legs, and the controlled winding is performed using the two magnetic legs so that the winding direction is orthogonal to the control winding NC. The line NR is wound. Further, among the four magnetic legs, for example, the gaps G are formed on the opposing surfaces of the two pairs of magnetic legs shown in the figure, so that the saturable reactor is formed.

FIG. 15 shows operation waveforms of the power supply circuit shown in FIGS. 10 and 11 according to the switching cycle.
FIGS. 15A and 15B respectively show the switching element Q1 of the power supply circuit shown in FIG. 10, the voltage Vcr across the parallel resonance capacitor Cr, and the collector of the switching element Q1 via the controlled winding NR and the primary winding N1. 2 shows the switching current I1 flowing through the switch. FIGS. 15C and 15D respectively show the switching element Q1 of the power supply circuit shown in FIG. 11, the voltage Vcr across the parallel resonance capacitor Cr, and the primary winding N1.
Shows the switching current I2 flowing through the collector of the switching element Q1 through the switching element I1. The periods TON and TOFF indicate periods when the switching element Q1 is turned on and off, respectively.

As can be seen from this figure, FIG. 10 and FIG.
In any of the power supply circuits shown in FIG. 1, during a period TOFF when the switching element Q1 is turned off, a resonance voltage Vcr of about 1800 V is generated at a peak. This is because the impedance of the parallel resonance circuit on the primary side of the voltage resonance converter acts on the DC input voltage of 2Ei obtained by the voltage doubler rectification.

[0020]

The above-mentioned FIG.
In the voltage resonance type converter having the configuration described with reference to FIG. 16, since the AC input voltage VAC corresponds to the AC 100 V system and the maximum load power is 150 W or more, the DC input voltage of the level of 2Ei is obtained by the voltage doubler rectification method. I have to. Therefore, as shown in FIG. 15, a resonance voltage Vcr of 1800 V is generated across the switching element Q1 and the parallel resonance capacitor Cr when the switching Q1 is off. Therefore, for the switching element Q1 and the parallel resonance capacitor Cr, it is required to select a high withstand voltage product of 1800V. In this case, especially for the switching element Q1, the saturation voltage VCE (SAT), the accumulation time tSTG, the fall time tf are large, and the current amplification factor hFE is small, so that the switching loss and the drive power increase, and accordingly the power supply circuit Power loss increases.

In the configuration shown in FIGS. 10 and 11,
Although the DC output voltages (EO1, EO2) are obtained by the half-wave rectifier circuit, the rectifier diodes (DO1, DO) of the half-wave rectifier circuit are obtained.
In the non-conducting state of 2), as shown in FIGS. 16D and 16I, a secondary side voltage V2 of about 2 to 3.5 times the DC output voltage EO (EO1, EO2) is applied. You. Therefore, the rectifier diodes (DO1, DO) used in the half-wave rectifier circuit
As for 2), for example, assuming that the DC output voltage is about 135 V, a high withstand voltage product of about 500 V is required, which increases the forward voltage drop VF and the reverse recovery time trr, which causes an increase in power loss. For example, if the switching frequency is increased, it is possible to reduce the size of various component elements. However, it is difficult to set the switching frequency fs to be high due to the above-described circumstances.
It has been found that when s is equal to or higher than 50 KHz, the power conversion efficiency is rapidly reduced. Further, as described above, a voltage doubler rectifier circuit is required to obtain a highly reliable DC input voltage, so that two relatively large smoothing capacitors are required and the substrate area is increased.

[0022]

SUMMARY OF THE INVENTION In view of the above-mentioned problems, the present invention provides a voltage-resonant switching power supply circuit, for example, capable of accommodating a maximum load power of 150 W in an AC 100 V system, thereby improving power conversion efficiency, and An object is to reduce the size and weight of the substrate area.

Therefore, as the voltage resonance type switching power supply circuit, rectifying and smoothing means for generating a rectified and smoothed voltage of a level corresponding to the input commercial AC power supply level by full-wave rectification, and a required coupling which is loosely coupled. A gap is formed so that a coefficient can be obtained, an insulating converter transformer provided for transmitting a primary output to a secondary side, and a rectifying and smoothing voltage output from a rectifying and smoothing means intermittently connected to the primary of the insulating converter transformer. A switching means configured to output to a winding, a primary resonance circuit formed by at least an inductance component including a primary winding of an insulated converter transformer and a capacitance of a resonance capacitor to make the operation of the switching means a voltage resonance type; And the inductance component of the secondary winding of the isolated converter transformer, A secondary side resonance circuit formed on the secondary side by the capacitance of a resonance capacitor, and an alternating voltage obtained in a secondary winding of an insulating converter transformer, and a DC that obtains a secondary side DC output voltage by full-wave rectification. An output voltage generating means, a controlled winding provided so as to function as an inductance component of the primary resonance circuit, and a control winding whose winding direction is orthogonal to the controlled winding. A quadrature control transformer is provided, and a variable control current is supplied to the control winding in accordance with the level of the DC output voltage to change the inductance of the controlled winding, thereby controlling the secondary side DC output voltage. A constant voltage control unit configured to perform constant voltage control, and a boost voltage generated by using a switching output of a switching unit is superimposed on a rectified smoothed voltage. A boosted rectified smoothed voltage is obtained, and the boosted rectified smoothed voltage can be controlled to be constant by changing the inductance of the controlled winding by including the controlled winding. Means. The boost means includes a boost winding formed by winding up a primary winding, and the controlled winding having one end connected to a connection point between the boost winding and the primary winding. A series connection circuit in which a boost rectifier diode for rectifying the alternating voltage obtained in the boost winding is connected in series, and the rectified output obtained by this series connection circuit is charged, so that the boost voltage is applied to both ends thereof. And a boosting smoothing capacitor provided so as to generate the voltage, and the negative electrode side of the boosting smoothing capacitor is connected to the positive electrode side of the smoothing capacitor forming the rectifying / smoothing means.

According to the above configuration, the insulation converter transformer is loosely coupled, and on the secondary side, a secondary-side DC output voltage is generated by the full-wave rectifier circuit to supply power to the load. In other words, a required load condition is handled by providing a full-wave rectifier circuit on the secondary side, and accordingly, the primary side is not a voltage doubler rectifier circuit, but one time of the AC input voltage level. Is provided with a full-wave rectifier circuit that generates a rectified smoothed voltage corresponding to. In addition, a boost circuit is provided on the primary side to obtain a boost smoothed voltage by superimposing a boost voltage on the rectified smoothed voltage, thereby increasing an apparent DC input voltage level to the switching converter. The boost circuit is configured to stabilize the boost smoothed voltage in accordance with constant voltage control having an orthogonal control transformer.

[0025]

FIG. 1 shows an example of the configuration of a voltage resonance type switching converter according to a first embodiment of the present invention. In the switching power supply circuit shown in this figure, similar to the case of FIG. 10 or FIG.
A self-excited voltage resonance type switching converter using one switching element (bipolar transistor) is employed. In this figure, FIG. 10 or FIG.
The same parts as those described above are denoted by the same reference numerals and description thereof will be omitted.

In the power supply circuit according to the embodiment shown in FIG. 1, a full-wave rectifying / smoothing circuit comprising a bridge rectifier circuit Di and a smoothing capacitor Ci serves as a rectifying / smoothing circuit for receiving an AC input voltage VAC to obtain an AC input voltage. A rectifier circuit is provided to generate a rectified smoothed voltage Ei corresponding to a level that is one-time the AC input voltage VAC. That is, in the present embodiment, a voltage doubler rectifier circuit is not provided as in the related art. In this specification, the rectified and smoothed voltage E corresponding to one time the level of the AC input voltage VAC is used.
The full-wave rectifier circuit that generates i is also referred to as an “equal voltage rectifier circuit”.

[0027] Insulated converter transformer P of the present embodiment
In the IT, a winding N3 is provided to wind up the primary winding N1 in addition to the primary winding N1, the secondary winding N2, and the driving winding NB. This insulation converter transformer PI
T is, for example, as shown in FIG.
An EE-type core is provided in which the mold legs CR1 and CR2 are combined so that their magnetic legs face each other. The primary winding N1 (and N3, N3, NB) and the secondary winding N2 are wound separately. In the present embodiment, a gap G is formed for the center magnetic leg as shown in the figure. As a result, for example, a loose coupling is achieved by a coupling coefficient smaller than that of the insulating converter transformer PIT shown in FIG. 12 as a conventional example, so that a saturated state is hardly obtained. The coupling coefficient k in this case is, for example, k ≒ 0.85.

In the present embodiment, an end of a winding N3 formed by winding up a primary winding N1 of the insulating converter transformer PIT is connected to a positive electrode of a smoothing capacitor CiB for generating a boost voltage described later. . The negative electrode of the smoothing capacitor CiB is the positive electrode (E
i line). In this embodiment, a boost diode DB is provided. The anode of the boost diode DB has a smoothing capacitor CiB.
Is connected to a connection point (Ei line) between a negative electrode of the smoothing capacitor Ci and a positive electrode of the smoothing capacitor Ci.
It is connected to the connection point between the primary winding N1 and the winding N3 through the series connection of the controlled winding NR of T. According to such a connection configuration, the switching output voltage obtained in the winding N3 is rectified by the boost diode DB and smoothed by the smoothing capacitor CiB, thereby generating a boost voltage VB across the smoothing capacitor CiB. A circuit will be formed. However, as described above, a controlled winding is inserted in series in this boost circuit. In the present embodiment, the voltage resonance type switching converter including the switching element Q1 performs switching using the boost smoothed voltage EB obtained by superimposing the boost voltage VB on the rectified smoothed voltage Ei as an operation power supply. To be. That is, in the present embodiment, the apparent DC input voltage level to be supplied to the voltage resonance type switching converter is increased by the boost circuit.

On the secondary side of the power supply circuit according to the present embodiment, a center tap is provided for the secondary winding N2, and the rectifier diodes DO1, DO2, DO3, DO4 and the smoothing capacitors CO1, CO2 are provided. By connecting as shown in the figure,
A set of [rectifier diodes DO1, DO2, smoothing capacitor C01] and [rectifier diodes DO3, DO4, smoothing capacitor C0]
O2], two sets of full-wave rectifier circuits are provided.
A full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] generates a DC output voltage EO1, and a full-wave rectifier circuit composed of [rectifier diodes DO3, DO4, smoothing capacitor CO2] generates a DC output voltage EO2. I do. That is, in the present embodiment, a full-wave rectifier circuit is provided to obtain a DC output voltage on the secondary side. In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1.

As described above with reference to FIG. 13, in the isolated converter transformer PIT, the mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 becomes + M and -M It has been described that there are cases where the secondary winding N2
Rectifier diode D when the alternating voltage obtained at
The operation in which the rectified current flows through O1 and DO3 can be regarded as a + M operation mode (forward mode). Conversely, when the alternating voltage obtained in the secondary winding N2 has a negative polarity, it flows through the rectified diodes DO2 and DO4. The rectified current can be considered to be in the -M operation mode (flyback method). That is, in the present embodiment, each time the alternating voltage obtained in the secondary winding becomes positive / negative, the mutual inductance operates in the + M / -M mode.

In the control circuit 1 in this case, the resistors R3 and R4 are connected in series between the DC output voltage EO1 and the secondary side ground, and the control terminal of the shunt regulator Q3 is connected to this connection point (voltage division point). Connected. The anode of the shunt regulator Q3 is grounded, and the cathode is connected to the base of the transistor Q2, which is an amplifying element, and to the DC output voltage EO2 via the resistor R1. In this case, the DC output voltage EO2 is used as the operation power supply of the control circuit 1. The connection point between the cathode and the base of the transistor Q2 is connected to the connection point between the resistors R3 and R4 via a series connection of the capacitor C11 and the resistor R2. The emitter of transistor Q2 is connected to the line of DC output voltage EO2. The collector is connected to the ground via the control winding NC.

The control circuit 1 formed by the above connection forms a function as an error amplifier having the DC output voltage EO1 as a detection input. That is, the DC output voltage EO1 is changed to the resistance R
3. The voltage divided by R4 is input to the control terminal of the shunt regulator Q3 as a control voltage. Therefore, in the shunt regulator Q3, a current having a level corresponding to the DC output voltage EO1 is supplied to the transistor Q2.
, The current level flowing to the collector of the transistor Q2 is variably controlled. That is, the control current level flowing through the control winding NC is variably controlled.

Further, in the power supply circuit having the above-described configuration, as described above, the boost circuit provided on the primary side generates the boost voltage VB across the smoothing capacitor CiB. A boosted smoothing voltage EB obtained by superimposing the boosted voltage VB on the rectified smoothed voltage Ei is obtained at both ends of the series-connected smoothing capacitor CiB-smoothing capacitor Ci. ,

(Equation 1) Can be represented by Here, the relationship of L3 = 1.2 × L1 is obtained as the inductance of the winding N3 and the primary winding N1, and LR = about the inductance LR of the controlled winding NR and the inductance L1 of the primary winding N1. Assuming that a relationship of 0.1 × L1 to 1.2 × L1 is obtained, based on the above (Equation 1),

(Equation 2) Is obtained. That is, in the present embodiment, by changing the inductance LR of the controlled winding NR of the orthogonal transformer PRT in the range of 0.1 × L1 to 1.2 × L1, the boost smoothing voltage EB is changed to Ei to
It is possible to vary within a range of 2Ei (Ei corresponds to a rectified smoothed voltage level obtained at both ends of the smoothing capacitor Ci).

The control circuit 1 according to the present embodiment employs an orthogonal transformer PRT so that the DC output voltage EO1 is constant.
A control current having a level corresponding to the fluctuation of the DC output voltage EO1 is supplied to the control winding NC to vary the inductance LR of the controlled winding NR. Since the controlled winding NR is also connected to the primary winding N1, the DC output voltage can be stabilized by the above-described inductance control method. As a result, the boost represented by (Equation 1) is obtained. The smoothing voltage EB is also controlled to be constant. That is, in the present embodiment, by controlling the DC output voltage on the secondary side to be constant, the boost smoothed voltage EB on the primary side is also controlled to be constant.

FIG. 3 shows the above-mentioned stabilization characteristic of the boosted smoothed voltage EB on the primary side in comparison with a conventional example (the power supply circuit shown in FIG. 10 or FIG. 11). In this figure, the horizontal axis is the AC input voltage VAC, and the vertical axis is the rectified smoothed voltage Ei (2
Ei) or the level of the boost smoothing voltage EB. As can be seen from this figure, in the conventional example, the rectified and smoothed voltage at the level of 2Ei obtained by the voltage doubler rectifier circuit has the maximum load power Pomax and the minimum load power Pom.
At the time of in, it tends to increase with the increase of the AC input voltage VAC. Conventionally, in order to maintain a constant DC output voltage on the secondary side obtained as an input voltage with a rectified smoothed voltage having such characteristics, a configuration is adopted in which the inductance of the primary side parallel resonance circuit is varied, and constant voltage control is performed. There was a corresponding burden on the system. For this reason, for example, since the current flowing through the controlled winding NR is relatively large, the orthogonal transformer PRT has to be somewhat large in size as shown in FIG.

On the other hand, in the present embodiment, the AC input voltage AC 100 V
~ 120 and AC100 ~ 140 in region B
In both cases, the following characteristics are obtained. In other words, the rectified smoothed voltage Ei obtained at both ends of the smoothing capacitor Ci rises as the AC input voltage VAC rises. On the other hand, the boost smoothed voltage EB obtained by superimposing the boost voltage VB is: As described above, by varying within the range of Ei to 2Ei, as shown in the figure, at the time of the maximum load power Pomax and at the time of the minimum load power Pomi.
At the time of n, control is performed so as to be substantially constant irrespective of the fluctuation of the AC input voltage. In the case of the present embodiment, the level of the boost voltage VB is controlled using the constant voltage control operation of the orthogonal control transformer PRT in order to keep the boost smoothed voltage EB constant as shown in FIG. I have. That is, the level of the boost voltage VB is varied by inserting the controlled winding NR that is a variable inductance into the boost circuit that generates the boost voltage VB. For this reason, the controlled winding N
In the present embodiment, the amount of current flowing through the controlled winding NR may be smaller than in the configuration in which R is inserted in series only into the primary-side parallel resonance circuit. Therefore, in the present embodiment, the size of the orthogonal control transformer PRT can be made smaller than before.

FIG. 4 is a waveform chart showing the operation of the main part of the power supply circuit according to the present embodiment during the switching cycle.
FIG. 4A shows the resonance voltage Vcr obtained at both ends of the parallel connection of the switching element Q1 / parallel resonance capacitor Cr, and FIG. 4B shows the voltage VR between the end of the controlled winding NR and the primary side ground. 4 (c) is the voltage VDB across the boost diode DB, FIG. 4 (d) is the current IR flowing through the controlled winding NR, and FIG. 4 (e) is the boost circuit from the positive electrode of the smoothing capacitor CiB to the winding N3. Current I3 flowing in FIG.
(F) shows the current I1 flowing through the primary winding N1. The periods TON and TOFF indicate the periods when the switching element Q1 is turned on and off, respectively.

As can be seen from this figure, the pulse waveform appearing as the resonance voltage Vcr shown in FIG. 4A is a voltage resonance type operation, and has a smooth waveform shape close to a sine wave instead of a rectangular waveform. are doing. Also, the controlled winding NR
(FIG. 4D) has only a positive polarity.
The current I1 flowing in the primary winding N1 (FIG. 4 (f)) is
The current IR (FIG. 4D) and the current I3 (FIG. 4E) flowing through the winding N3 are combined. FIG. 4 (a)
Has a peak of about 1200 V, the reason for which will be described later.

In the present embodiment, by increasing the apparent AC input voltage by the boost circuit as described above, the maximum equivalent to the conventional case where a rectified smoothed voltage is obtained by voltage doubler rectification as in the prior art. It is possible to cope with the load power. In addition to this, when the secondary side parallel resonance circuit is formed by providing the secondary side parallel resonance capacitor C2 as in the power supply circuit of the present embodiment, for example, the operation of the secondary side parallel resonance circuit Since power is supplied to the load side, the maximum load power increases as compared with the case where the secondary side parallel resonance capacitor C2 is not provided. At this time, if the rectifying / smoothing circuit connected to the secondary parallel resonance circuit is a half-wave rectifying circuit, a rectified current cannot be obtained when the alternating voltage generated in the secondary winding N2A has a negative polarity. The load power increase rate is about 50%. On the other hand, when a full-wave rectifier circuit is connected to the secondary-side parallel resonance circuit as in the present embodiment, as described above, the mutual inductance is in both operation modes of + M / -M. A rectified current flows alternately. In other words, since the alternating voltage allows the rectified output to be obtained during both the positive and negative periods, the power supplied to the load increases accordingly.
The rate of increase of the maximum load power is also improved. For example, compared to a configuration in which the secondary parallel resonance capacitor C2 is not provided and a half-wave rectifier circuit is connected, in the present embodiment, the maximum load power is increased by about twice.

By increasing the maximum load power as described above, in the present embodiment, the rectifying and smoothing circuit for generating the DC input voltage (rectified and smoothed voltage) adopts the double voltage rectification method to reduce the load power. No need to cover. As a result, in the present embodiment, as described with reference to FIG. 1, for example, a configuration of a normal equal-voltage rectifier circuit using a bridge rectifier circuit can be adopted. Thus, for example, in the present embodiment, the rectified smoothed voltage Ei at the time of the AC input voltage VAC = 144 V becomes about 200 V. Resonant voltage V
The cr is generated when the switching element Q1 is turned off by the action of the primary parallel resonance circuit on the rectified smoothed voltage Ei. In the present embodiment, the rectified smoothed voltage Ei is double-voltage rectified as described above. It will be about 1/2 of the time. However, in the present embodiment, the boost smoothed voltage EB is superimposed on the rectified smoothed voltage Ei by superimposing the boosted voltage VB.
Is generated, the resonance voltage Vcr becomes equal to the boost smoothed voltage E.
Although it depends on the level of B, the resonance voltage V
cr is the conventional 18 cr as shown in FIG.
The voltage can be suppressed from about 00V to about 1200V. Therefore, in the present embodiment, the switching element Q1 and the parallel resonance capacitor Cr are 1200
It is only necessary to select a withstand voltage product of V.

As described above, the full-wave rectifier circuit is provided on the secondary side so that the rectified current flows during both the positive and negative periods of the alternating voltage of the secondary winding N2A. , The secondary side resonance voltage V2 is suppressed to the same level as the rectified smoothed voltage Ei in both the positive and negative periods. As a result, as the rectifier diodes (DO1 to D04) forming the secondary-side full-wave rectifier circuit, a withstand voltage product substantially corresponding to the level of the rectified smoothed voltage Ei can be selected.

As described above, in the present embodiment, the switching element Q1, the parallel resonance capacitor Cr, and the rectifier diode forming the full-wave rectifier circuit on the secondary side have a lower breakdown voltage than those provided in the conventional example. Can be used, so that the element becomes less expensive. Therefore, for example, the switching element Q1 and a rectifying diode forming a full-wave rectifier circuit on the secondary side have improved characteristics (for the switching element Q1, the saturation voltage VCE (SAT ), Good characteristics such as accumulation time tSTG, fall time tf, and current amplification factor hFE, and rectifier diodes having good characteristics such as forward voltage drop VF and reverse recovery time trr). Therefore, reduction of power loss is promoted accordingly. That is, it is possible to improve the power conversion efficiency at a lower cost or almost the same cost as the conventional one. Further, by improving the power conversion efficiency, for example, a radiator plate or the like which has been required conventionally becomes unnecessary.

Further, if the switching frequency is set higher than that of the conventional example in the configuration of the voltage resonance type converter, the above-mentioned various components can be reduced in size and weight. Here, if the winding N3 is selected such that the boost voltage VB is optimized according to the maximum load power to be actually handled, further reduction in size and weight of various components can be realized.
Furthermore, from the viewpoint of reducing the size and weight of the power supply circuit, the conventional configuration including the voltage doubler rectifier circuit for generating the DC input voltage requires two sets of rectifier diodes and a smoothing capacitor. However, in the present embodiment, for example, since a full-wave rectifier circuit using a normal bridge rectifier circuit is used, a set of a block-type smoothing capacitor and a bridge rectifier diode can be employed. The cost and the size of parts can be reduced.

As an actual experimental result, the maximum load power is 1
When the conventional power supply circuit shown in FIG. 10 or FIG. 11 and the power supply circuit of the present embodiment shown in FIG. 1 are compared under the condition of 60 W, the switching frequency fs = 50 KHz in the conventional power supply circuit , Smoothing capacitor Ci1,
Ci2 = 1000 .mu.F / 180 V, parallel resonance capacitor Cr = 0.022 .mu.F / 1800 V, secondary side parallel resonance capacitor C2 = 0.014 .mu.F, ferrite core of insulation converter transformer PIT (cores CR1 and CR in FIG. 12).
2) The ferrite cores (cores CR11 and CR12 in FIG. 14) of the EE-40 type and orthogonal control transformer PRT are 23
With a configuration of two sets of × 23 × 23 mm, the power conversion efficiency is set to 84%. On the other hand, in the power supply circuit of the present embodiment shown in FIG. 1, the switching frequency fs = 100K
Hz, smoothing capacitor Ci = 1000 μF / 180 V,
The smoothing capacitor CiB = 1000 .mu.F / 100 V, the parallel resonance capacitor Cr = 6800 pF / 1200 V, the secondary side parallel resonance capacitor C2 = 0.01 .mu.F, and the ferrite cores (cores CR1 and CR2 shown in FIG. 2) of the insulating converter transformer PIT are EE-35. The power conversion efficiency is 90% in a configuration in which the ferrite cores of the type and orthogonal type control transformer PRT are two sets of 16 × 16 × 16 mm. That is, in the present embodiment, various components including the insulating converter transformer PIT and the quadrature control transformer PRT are reduced in size as compared with the conventional example, and the power conversion efficiency is improved by 6%. . It should be noted that the various constants, the standard size, and the like in the above configuration are merely examples.

FIG. 5 is a circuit diagram showing a configuration example of a voltage resonance type switching power supply circuit according to a second embodiment of the present invention. In this figure, the same parts as those in FIG.

In the power supply circuit shown in FIG.
As switching element Q1, bipolar transistors (BJT) Q11 and Q12, clamp diodes DD1 and DD
2. A Darlington circuit formed by connecting the resistors R11 and R12 as shown in the figure is provided. In this case, as the connection mode of the Darlington circuit, the collector of the transistor Q11 is connected to the collector of the transistor Q12, the emitter of the transistor Q11 is connected to the emitter of the transistor Q12, and the emitter of the transistor Q12 is grounded. . Further, the anode of the damper diode DD1 is connected to the emitter of the transistor Q11, and the cathode of the damper diode DD1 is connected to the transistor Q11 via the resistor R11.
Connected to the base. The anode of the damper diode DD2 is connected to the emitter of the transistor Q12, and the cathode is connected to the collector of the transistor Q12. The resistor R12 is connected in parallel between the base and the emitter of the transistor Q12. In the Darlington circuit thus formed, the base of the transistor Q11 is connected to the switching element Q shown in each of the above embodiments.
1 and the collector contacts of the transistors Q11 and Q12 become equivalent to the collector of the switching element Q1. Further, the emitter of the transistor Q12 is equivalent to the emitter of the switching element Q1.

In this case, a self-excited oscillation circuit for driving the switching element in a self-excited manner is omitted.
Instead, a configuration in which the switching drive by the separately-excited system including the oscillation circuit 2 and the drive circuit 3 is performed is adopted. For this reason, in the present embodiment, the windings N4A and N4B are provided in the insulating converter transformer PIT. Then, a +12 V DC voltage is generated by a half-wave rectifier circuit including the winding N4A, the rectifier diode D1, and the capacitor CA, and a -12V DC voltage is generated by a half-wave rectifier circuit including the winding N4B, the rectifier diode D2, and the capacitor CB. Is generated. The DC voltage of +12 V is supplied as an operating power to the oscillation circuit 2, and the DC voltage of +12 V and −1 is supplied to the drive circuit 3.
A DC voltage of 2 V is supplied as an operation power supply.

The oscillating circuit 2 is started by a starting resistor RS and has a required switching frequency fs
An oscillation signal having a cycle (for example, fs = 100 KHz) is generated and output to the drive circuit 3. In the drive circuit 3, the input oscillation signal and the above + 12V /-
Using a 12V operating power supply, a positive (on) / negative (off) switching drive current is generated for each switching cycle as shown in the figure and output to the base terminal of the switching element Q1. As a result, the switching element Q1 is driven to perform a switching operation at a required switching frequency. When a Darlington circuit is employed for the switching element Q1 as in the present embodiment, higher power conversion efficiency can be obtained than when the switching element Q1 is a single bipolar transistor, for example.

On the secondary side of the power supply circuit shown in this figure, two sets of secondary windings N2 and N2A are provided.
In this case, the parallel resonance capacitor C2 is connected only to the secondary winding N2, but may be connected to the secondary winding N2A. Here, the secondary winding N2
, A bridge rectifier circuit DO is connected, and a rectifier circuit including the bridge rectifier circuit DO and the smoothing capacitor C01 performs full-wave rectification to obtain a DC output voltage EO1. On the secondary winding N2A side, a center tap is provided to the secondary winding N2A and grounded, and a center tap full-wave rectification system including two rectifier diodes DO3 and DO4 and a smoothing capacitor C02 is used. The DC output voltage EO2 is obtained by the rectifier circuit.

FIG. 6 shows a configuration as a modification of the present embodiment. In this figure, only the isolated converter transformer PRT and the circuit on the secondary side are extracted and shown, and the configuration of the other parts is the same as that of FIG. 1 or FIG. In FIG. 6, a full-wave voltage rectification system is employed as a rectifier circuit for obtaining a DC output voltage EO on the secondary side. That is, a voltage doubler rectifier circuit formed by two smoothing capacitors CO11 and CO12 and rectifier diodes DO1 and DO2 is provided for the secondary winding N2A.
In this circuit, during the period in which the alternating voltage obtained in the secondary winding N2A is positive, a rectified current flows through the route of the secondary winding N2A → the rectifier diode DO1 → the smoothing capacitor CO11 → the secondary winding N2A, and the smoothing capacitor CO11. Charge the battery. In addition, during the period when the alternating voltage obtained in the secondary winding N2A is negative, the secondary winding N
A rectified current flows through the path of 2A → smoothing capacitor CO12 → rectifier diode DO2 → secondary winding N2A to charge the smoothing capacitor CO12. This operation is repeated every cycle. By this operation, the smoothing capacitor CO1
1, a DC voltage level corresponding to the voltage level obtained at the secondary winding N2A is obtained at CO12. Therefore, both ends of the series-connected smoothing capacitors CO11-CO12 are
A DC voltage corresponding to twice the voltage level obtained at the secondary winding N2A is obtained, and this DC voltage becomes the DC output voltage EO. In such a configuration, for example, in order to obtain a DC output voltage EO of the same level as the DC output voltage EO1 in the power supply circuits of the first and second embodiments described above, the number of turns of the secondary winding N2A Can be reduced, for example, to about 1/2 of the number of turns of the secondary winding N2 in the first and second embodiments (shown as N2A = N2 / 2 in the figure).

In the description of the embodiments above,
The case where a single bipolar transistor (BJT) or a Darlington circuit having two bipolar transistors is employed as the switching element Q1 has been described as an example, but the embodiment of the present invention will be described below. The switching element can be employed instead of the switching element Q1.

FIG. 7 shows a MOS-FET (MOS type field effect transistor) instead of the switching element Q1.
An example using a metal oxide semiconductor is shown. MO
When an S-FET is used, a Zener diode ZD for forming a current path during the switching-off period is connected in parallel in the direction shown in the drawing between the drain and the source. That is, the anode is connected to the source of the MOS-FET, and the cathode is connected to the drain of the Zener diode ZD. In this case, the base, the collector, and the emitter of the switching element Q1 shown in the above embodiment are respectively the gate, the drain, and the MOS-FET.
Will be replaced by the source.

FIG. 8 shows that the switching element Q1 is replaced with
An example using an IGBT (insulated gate bipolar transistor) is shown. A diode D for forming a current path during switching off is connected in parallel between the collector and the emitter of the IGBT. Here, the anode and cathode of the diode D are I
It is connected to the collector and emitter of the GBT.
In this circuit, the base, collector and emitter of the switching element Q1 shown in each of the above embodiments are respectively
Replaced by the gate, collector and emitter of the IGBT.

FIG. 9 shows an example in which the switching element Q1 is replaced by
An example using an SIT (static induction thyristor) is shown. Even between the collector and emitter of this SIT,
A diode D for forming a current path during switching off is connected in parallel, and an anode and a cathode of the diode D are connected to a cathode and an anode of the SIT, respectively. In this circuit, the base, collector and emitter of the switching element Q1 shown in each of the above embodiments are replaced with the gate, anode and cathode of the SIT, respectively.

In this embodiment, even if any of the configurations shown in FIGS. 7 to 9 is employed, it is possible to further increase the efficiency. In the case where the configuration shown in FIGS. 7 to 9 is employed, although not shown here, the switching element Q
The configuration of the drive circuit is changed to conform to the configuration to be adopted instead of 1. Further, the details of the configuration shown in each of the above-described embodiments may be changed in accordance with actual use conditions and the like.

[0056]

As described above, according to the present invention, an insulation converter transformer is loosely coupled as a voltage resonance type switching power supply circuit corresponding to a relatively high load condition of 150 W or more with an AC input voltage of 100 V AC, for example. so,
An operation mode (+ M / -M) in which the mutual inductances of the primary winding and the secondary winding have opposite polarities is obtained. In addition, by providing the full-wave rectifier circuit on the secondary side, the secondary-side DC output voltage is obtained from the alternating voltage obtained in the secondary winding. That is, as a result of supplying power to the load by the full-wave rectifier circuit on the secondary side, in the present invention, the maximum load power that can be handled is smaller than in the conventional case where the secondary-side DC output voltage is obtained by the half-wave rectifier circuit. Can be improved. Along with this, the primary side is not a voltage doubler rectifier circuit, but a full-wave rectifier circuit is used to input a rectified smoothed voltage at a level corresponding to the AC input voltage level. It will be able to respond.

For example, conventionally, when the above conditions are satisfied, it is necessary to obtain a rectified smoothed voltage corresponding to twice the AC input voltage level by a voltage doubler rectifier circuit. It is necessary to select a withstand voltage product corresponding to the switching voltage generated according to the rectified smoothing voltage level for the parallel resonance capacitor. In the related art, a DC output voltage is generated by a half-wave rectifier circuit on the secondary side.
Since a voltage about twice to 3.5 times is applied, a withstand voltage product corresponding to this voltage level has been selected.

On the other hand, in the present invention, since the switching voltage depending on the rectified smoothing voltage level is の of the conventional one, the withstand voltage of the switching element and the resonance capacitor on the primary side is much lower than the conventional one. Component elements can be used. On the secondary side, an alternating voltage (resonance voltage) obtained in a secondary winding forming a parallel resonance circuit
Is rectified by a full-wave rectifier circuit to obtain a DC output voltage. As a result, after increasing the maximum load power that can be handled, the voltage applied to the rectifier diode is suppressed to approximately the same level as the rectification smoothed voltage level. Things can be selected. As a result, it is possible to reduce costs for the switching element, the primary-side parallel resonance capacitor, the secondary-side rectifier diode, and the like. In addition, if the switching element and the secondary rectifier diode having improved characteristics are selected, the power conversion efficiency can be improved. Further, if the characteristics of the switching element are improved, it becomes easy to increase the switching frequency. Therefore, if the switching frequency is increased, it is possible to reduce the size and weight of circuit components around the switching element. . Further, since a circuit for obtaining a rectified smoothed voltage from a commercial AC power supply is a normal full-wave rectifier circuit, for example, a normal set of a block-type smoothing capacitor and a bridge rectifier diode can be employed. However, cost and circuit scale can be reduced.

Further, according to the present invention, a boost circuit (boost means) for obtaining a boost smoothed voltage by superimposing a boost voltage on a rectified smoothed voltage is provided. This also provides the increase in the maximum load power described above.
Here, in the present invention, the controlled winding of the orthogonal control transformer provided for the constant voltage control is inserted into the boost circuit, so that the boost smoothing voltage also acts to be constant. For this reason, for example, as compared with the conventional case where the controlled winding is inserted into the parallel resonance circuit on the primary side,
According to the present invention, it is possible to reduce the current to be passed through the controlled winding. This makes it possible to reduce the size of the orthogonal control transformer. The boost circuit is composed of a series connection circuit in which a controlled winding and a rectifying diode for boost are connected in series to a boost winding formed by winding up a primary winding, and a rectified output of the series connection circuit. By forming a simple circuit including a boosting smoothing capacitor that obtains a boost voltage by charging, an increase in circuit scale is avoided. In particular, with regard to the boosting smoothing capacitor, a rectified smoothed voltage (DC input voltage) is obtained by a normal full-wave rectification method instead of a voltage doubler rectification method.
It is sufficient to select a withstand voltage product of about V, and an inexpensive and small product can be adopted.

Further, as the present invention, a rectifier circuit provided on the secondary side employs a full-wave rectifier circuit using a bridge rectifier circuit, a center-tap type full-wave rectifier circuit, and a doubled voltage full-wave rectifier circuit. Although it is possible to have a certain degree of freedom as a circuit form, the number of turns of the secondary winding should be reduced to about 1/2 of the conventional one, especially when a voltage doubler full-wave rectifier circuit is adopted. Also becomes possible.

The switching means may be constituted by a Darlington circuit formed with a bipolar transistor, or a MOS type field effect transistor, an insulated gate bipolar transistor, or an electrostatic induction thyristor. For example, the power conversion efficiency can be further improved as compared with the case where the switching means is formed by a single bipolar transistor.

As described above, according to the present invention, the power supply circuit including the voltage resonance type converter can be reduced in cost, reduced in size and weight, and improved in power conversion efficiency.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration example of a power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a cross-sectional view illustrating a configuration of an insulating converter transformer of the power supply circuit according to the present embodiment.

FIG. 3 shows an AC input voltage and a DC input voltage (rectified smoothed voltage,
FIG. 5 is an explanatory diagram showing a relationship with a boost smoothed voltage.

FIG. 4 is a waveform diagram illustrating an operation of a main part of the power supply circuit according to the first embodiment.

FIG. 5 is a circuit diagram showing a configuration example of a power supply circuit according to a second embodiment of the present invention.

FIG. 6 is a circuit diagram showing a configuration as a modification of the embodiment of the present invention.

FIG. 7 is a circuit diagram showing a configuration as a modification of the embodiment of the present invention.

FIG. 8 is a circuit diagram showing a configuration as a modification of the embodiment of the present invention.

FIG. 9 is a circuit diagram showing a configuration as a modification of the embodiment of the present invention.

FIG. 10 is a circuit diagram showing a configuration of a power supply circuit as a conventional example.

FIG. 11 is a circuit diagram showing a configuration of a power supply circuit as a conventional example.

FIG. 12 is a circuit diagram showing a configuration of an insulating converter transformer of a power supply circuit as a conventional example.

FIG. 13 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

FIG. 14 is a perspective view showing a configuration of an orthogonal control transformer provided in a power supply circuit as a conventional example.

FIG. 15 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIGS. 10 and 11;

FIG. 16 is an explanatory diagram showing inductance superposition characteristics of a constant voltage control system in the power supply circuits shown in FIGS. 10 and 11;

[Explanation of symbols]

1 control circuit, 2 oscillation circuit, 3 drive circuit, C
i, CiB smoothing capacitor, Cr parallel resonance capacitor, C2 (secondary side) parallel resonance capacitor, Di, DO
Bridge rectifier circuit, DO1, DO2, DO3, DO4 rectifier diode, PIT insulation converter transformer, PRT orthogonal control transformer, NC control winding, NR controlled winding, Q1
Switching element, DB boost diode, N3
Winding (for boost)

Claims (8)

[Claims] 1. A rectifying / smoothing means for generating a rectified / smoothed voltage having a level corresponding to an input commercial AC power supply level by full-wave rectification, and a gap formed so as to obtain a required coupling coefficient which is loosely coupled. An insulating converter transformer provided for transmitting a primary output to a secondary side; and a rectifying and smoothing voltage output from the rectifying and smoothing means being intermittently output to a primary winding of the insulating converter transformer. Switching means, at least a primary-side resonance circuit formed by an inductance component including a primary winding of the insulated converter transformer and a capacitance of a resonance capacitor to make the operation of the switching means a voltage resonance type; and Of the secondary winding and the capacitance of the secondary side parallel resonant capacitor And a secondary-side resonance circuit formed on the secondary side, and an alternating voltage obtained in a secondary winding of the insulating converter transformer, and a DC output voltage generation for obtaining a secondary-side DC output voltage by full-wave rectification. Means, a controlled winding provided so as to function as an inductance component of the primary side resonance circuit, and a control winding whose winding direction is orthogonal to the controlled winding. An orthogonal control transformer having a variable control current flowing through the control winding according to the level of the DC output voltage to change the inductance of the controlled winding.
Constant voltage control means configured to perform constant voltage control on the secondary side DC output voltage; and superimposing a boost voltage generated using the switching output of the switching means on the rectified smoothed voltage, thereby boosting. A boost configured to obtain a rectified smoothed voltage and including the above-described controlled winding, so as to be configured to be able to control the boost rectified smoothed voltage to be constant by a change in inductance of the controlled winding. Means for boosting, wherein the boosting means is formed by winding up the primary winding, and the casing has one end connected to a connection point between the boosting winding and the primary winding. A series connection circuit in which a control winding and a boost rectifier diode for rectifying an alternating voltage obtained in the boost winding are connected in series; A boosting smoothing capacitor provided at both ends thereof to generate a boost voltage by being charged with a rectified output obtained from the boosting smoothing capacitor. A voltage resonant switching power supply circuit formed by connecting to a positive electrode side. 2. The voltage resonance type switching power supply circuit according to claim 1, wherein said DC output voltage generation means is configured as a full-wave rectification circuit having a bridge rectification circuit. 3. The voltage-resonant switching power supply circuit according to claim 1, wherein said DC output voltage generating means is provided with a rectifier circuit based on a center tap full-wave rectification method. 4. The voltage resonance type switching power supply circuit according to claim 1, wherein said DC output voltage generation means includes a voltage doubler full-wave rectifier circuit. 5. The voltage resonance type switching power supply circuit according to claim 1, wherein said switching means is a Darlington circuit formed with a bipolar transistor. 6. The voltage resonance type switching power supply circuit according to claim 1, wherein said switching means is a MOS field effect transistor. 7. The voltage resonance type switching power supply circuit according to claim 1, wherein said switching means is an insulated gate bipolar transistor. 8. The voltage resonance type switching power supply circuit according to claim 1, wherein said switching means is an electrostatic induction thyristor.