JP2001186763A - Switching power source circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To ensure a power-factor capable of coping with a practical usage condition to a load and AC input voltage change, and suppress ripple component. SOLUTION: This switching power source circuit is provided with a power- factor improving circuit having a magnetic coupling transformer MCT to a switching frequency control system complex resonance type converter of a divided voltage push-pull type. A switching output obtained in a primary side resonance circuit is fed back by a magnetic coupling system via a primary side parallel resonance capacitor.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit having a power factor correction circuit.

[0002]

2. Description of the Related Art The applicant has previously proposed various power supply circuits having a resonance type converter on the primary side. Further, various power supply circuits including a power factor improving circuit for improving the power factor of the resonant converter have been proposed.

FIG. 10 is a circuit diagram showing an example of a switching power supply circuit constructed based on the invention previously filed by the present applicant. This power supply circuit has a configuration in which a power factor improvement circuit for improving a power factor is provided for a self-excited current resonance type switching converter.

The power supply circuit shown in FIG. 1 includes a bridge rectifier circuit Di for full-wave rectification of a commercial AC power supply AC. In this case, the rectified output rectified by the bridge rectification circuit Di is charged to the smoothing capacitor Ci via the power factor correction circuit 20, and both ends of the smoothing capacitor Ci correspond to the level of one time of the AC input voltage VAC. The rectified smoothed voltage Ei is obtained. Also, in the rectifying / smoothing circuit (Di, Ci), an inrush current limiting resistor Ri is inserted in the rectification current path so as to suppress an inrush current flowing into the smoothing capacitor when the power is turned on, for example. ing.

In the power factor correction circuit 20 shown in FIG. 1, a filter choke coil LN, a high-speed recovery type diode D1 and a choke coil LS are provided between a positive output terminal of a bridge rectifier circuit Di and a positive terminal of a smoothing capacitor Ci. They are connected in series and inserted. The filter capacitor CN is inserted between the anode side of the high-speed recovery diode D1 and the positive terminal of the smoothing capacitor Ci to form a normal mode low-pass filter together with the filter choke coil LN.

In the power factor correction circuit 20, an end of a primary side series resonance circuit, which will be described later, is connected to a connection point between the cathode of the high-speed recovery type diode D1 and the choke coil LS. The switching output obtained in the series resonance circuit is fed back. The power factor improving operation by the power factor improving circuit 20 will be described later.

This power supply circuit includes a self-excited current resonance type converter using a rectified smoothed voltage Ei, which is a voltage across a smoothing capacitor Ci, as an operating power supply. In this current resonance type converter, as shown in the figure, switching elements Q1 and Q2 of two bipolar transistors are half-bridge-coupled and then inserted between the connection point on the positive side of the smoothing capacitor Ci and ground. It is connected. Starting resistors RS1 and RS2 are inserted between the collectors and the bases of the switching elements Q1 and Q2, respectively. The resistors RB1 and RB2 connected to the bases of the switching elements Q1 and Q2 set the base current (drive current) of the switching elements Q1 and Q2. Further, clamp diodes DD1 and Q2 are connected between the base and the emitter of the switching elements Q1 and Q2, respectively.
DD2 is inserted. The clamp diodes DD1 and DD2 are
A current path for a clamp current flowing through the base-emitter is formed during a period in which the switching elements Q1 and Q2 are turned off. And the resonance capacitors CB1, C
B2 is a drive winding N of a drive transformer PRT described below.
A series resonance circuit (self-excited oscillation drive circuit) for self-excited oscillation is formed together with B1 and NB2, and the switching elements Q1, Q2
Determine the switching frequency of 2.

[0008] Drive transformer PRT (Power Regulati
ng Transformer) is provided to drive the switching elements Q1 and Q2 and to perform constant voltage control by variably controlling the switching frequency. In this case, the drive windings NB1 and NB2 and the resonance current detection are performed. Winding N
D is wound, and a control winding N is provided for each of these windings.
C is an orthogonal saturable reactor wound in the orthogonal direction. One end of the drive winding NB1 of the drive transformer PRT is connected to the base of the switching element Q1 via a series connection of the resistor RB1 and the resonance capacitor CB1, and the other end is connected to the emitter of the switching element Q1. One end of the drive winding NB2 is grounded, and the other end is connected to the base of the switching element Q2 via a series connection of a resistor RB2 and a resonance capacitor CB2. The drive winding NB1 and the drive winding NB2 are wound so that voltages of opposite polarities are generated.

Insulated converter transformer PIT (Power Is
olation Transformer) is the switching element Q1, Q2
Is transmitted to the secondary side. One end of the primary winding N1 of the insulation converter transformer PIT is connected to the contact point (switching output point) between the emitter of the switching element Q1 and the collector of the switching element Q2 via the resonance current detection winding ND, thereby providing a switching output. Is obtained.

The other end of the primary winding N1 is connected to a connection point between the cathode of the high-speed recovery type diode D1 and the choke coil LS in the power factor correction circuit 20 via the series resonance capacitor C1. ing.

In this case, the series resonance capacitor C1 and the primary winding N1 are connected in series, but the capacitance of the series resonance capacitor C1 and the primary winding N1 are connected.
Converter transformer PIT including (series resonance winding)
The primary side series resonance circuit for making the operation of the switching converter a current resonance type is formed by the leakage inductance (leakage inductance) component.

On the secondary side of the insulated converter transformer PIT shown in FIG. 1, a center tap is provided for the secondary winding N2, and the rectifier diodes DO1, DO2, D
By connecting O3, DO4 and smoothing capacitors CO1, CO2 as shown in the figure, a set of [rectifier diodes DO1, DO2, smoothing capacitor CO1] and a set of [rectifier diodes DO3, DO4, smoothing capacitor CO2] are obtained. Two sets of full wave rectifier circuits are provided. The full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] has a DC output voltage EO1.
And a full-wave rectifier circuit composed of [rectifier diodes DO3, DO4, smoothing capacitor CO2] generates a DC output voltage EO2. In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1.

The control circuit 1 supplies a DC current whose level varies in accordance with the level of the DC voltage output EO1 on the secondary side to the control winding NC of the drive transformer PRT as a control current, as will be described later. The constant voltage control is performed as described above.

The switching operation of the power supply circuit according to the above configuration is as follows. First, when a commercial AC power supply is turned on, the switching elements Q1, Q2 are activated via, for example, start-up resistors RS1, RS2.
2 is supplied with a starting current. For example, if the switching element Q1 is turned on first,
The switching element Q2 is controlled to be turned off.
Then, as an output of the switching element Q1, a resonance current flows through the resonance current detection winding ND → the primary winding N1 → the series resonance capacitor C1. When the resonance current becomes zero, the switching element Q2 is turned on, and It is controlled to be off. Then, a resonance current flows in the opposite direction through the switching element Q2. Or later,
A self-excited switching operation in which the switching elements Q1 and Q2 are turned on alternately is started. As described above, by using the terminal voltage of the smoothing capacitor Ci as the operating power source, the switching elements Q1 and Q2 alternately open and close,
A drive current close to the resonance current waveform is supplied to the primary winding N1 of the insulating converter transformer PIT, and an alternating output is obtained at the secondary winding N2.

The constant voltage control by the drive transformer PRT is performed as follows. For example, assuming that the secondary output voltage EO1 fluctuates so as to increase with the AC input voltage level or load fluctuation, the level of the control current flowing through the control winding NC as described above is also changed to the secondary output voltage EO1.
Is controlled so as to increase in accordance with the rise of. Under the influence of the magnetic flux generated in the drive transformer PRT by this control current, the drive transformer PRT tends to approach a saturation state, and acts to reduce the inductance of the drive windings NB1 and NB2. The switching frequency is controlled so as to increase by changing the condition of the oscillation circuit. In this power supply circuit, the switching frequency is set in a frequency region higher than the resonance frequency of the series resonance circuit of the series resonance capacitor C1 and the primary winding N1 (upper side control). Then, the switching frequency is separated from the resonance frequency of the series resonance circuit.
Thereby, the resonance impedance of the series resonance circuit with respect to the switching output increases.

By increasing the resonance impedance in this way, the drive current supplied to the primary winding N1 of the primary series resonance circuit is suppressed, and as a result, the secondary output voltage is suppressed. , Constant voltage control is achieved. Hereinafter, the constant voltage control method using the above method will be referred to as “switching frequency control method”.

The power factor improving operation by the power factor improving circuit 20 is as follows. In the configuration of the power factor correction circuit 20 shown in this figure, the switching output supplied to the series resonance circuit (N1, C1) has a rectified current path via an inductive reactance (magnetic coupling) assumed to be included in the choke coil LS itself. It is made to return to.

The switching output fed back as described above causes the alternating voltage of the switching cycle to be superimposed on the rectified current path. The superimposed voltage of the switching cycle causes the high speed recovery type diode D1 to be superimposed. In this case, an operation of interrupting the rectified current in the switching cycle is obtained, and the apparent inductance of the filter choke coil LN and choke coil LS also increases due to the intermittent action. This allows the charging current to flow to the smoothing capacitor Ci even during a period in which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, and the conduction angle of the AC input current is increased. As a result, the power factor is improved.

FIG. 11 is a circuit diagram showing another example of the configuration of a switching power supply circuit that can be configured based on the invention previously proposed by the present applicant. This power supply circuit is also provided with a current resonance type converter in which two switching elements are half-bridge-coupled, and the drive system is separately excited. Also in this case, a power factor improving circuit for improving the power factor is provided. Note that the same parts as those in FIG. 10 are denoted by the same reference numerals and description thereof will be omitted.

The primary-side current resonance type converter shown in FIG. 1 includes two switching elements Q11 and Q12, for example, MOS-FETs. here,
By connecting the drain of the switching element Q11 to the line of the rectified and smoothed voltage Ei, connecting the source of the switching element Q11 and the drain of the switching element Q12, and connecting the source of the switching element Q12 to the primary side ground, it is compatible with the separately excited type. A half-bridge connection is obtained. These switching elements Q11 and Q12 are switching-driven by the oscillation drive circuit 2 so that on / off operations are alternately repeated, and intermittently output the rectified smoothed voltage Ei as a switching output. In this case, a clamp diode DD connected between the drain and source of each of the switching elements Q11 and Q12 in the direction shown in FIG.
1, DD2 are provided.

In this case, the switching element Q
11. By connecting one end of the primary winding N1 of the isolated converter transformer PIT to the source-drain connection point (switching output point) of Q12, the switching output is supplied to the primary winding N1. Is done. The other end of the primary winding N1 is connected via a series resonance capacitor C1 to a filter choke coil LN of a power factor improving circuit 21 described below.
And the anode of the high speed recovery type diode D1.

Also in this case, a series resonance circuit for making the switching power supply circuit a current resonance type is formed by the capacitance of the series resonance capacitor C1 and the leakage inductance component of the insulating converter transformer PIT including the primary winding N1. .

In this case, the control circuit 1 outputs a control signal having a level corresponding to the fluctuation of the DC output voltage EO1 to the oscillation drive circuit 2, for example. The oscillation drive circuit 2 changes the frequency of the switching drive signal supplied from the oscillation drive circuit 2 to each gate of the switching elements Q11 and Q12 based on the control signal supplied from the control circuit 1,
The switching frequency is made variable. In the power supply circuit shown in FIG. 11, as in the power supply circuit shown in FIG. 10, the switching frequency is set as a region higher than the series resonance frequency. And
For example, when the DC output voltage EO1 rises, the control circuit 1 controls the oscillation drive circuit 2 to increase the switching frequency according to the level. Thus, the constant voltage control is performed in the same manner as described with reference to FIG. The starting circuit 3 is provided for detecting the voltage or current obtained on the rectifying / smoothing line immediately after the power is turned on, and for starting the oscillation drive circuit 2. Is input as an operation power supply.

In the power factor correction circuit 21 shown in this figure, the positive output terminal of the bridge rectification circuit Di and the smoothing capacitor Ci
Of the filter choke coil LN−
A fast recovery diode D1 is inserted in series. Here, the filter capacitor CN is provided in parallel with the series connection circuit of the filter choke coil LN and the high-speed recovery type diode D1. Also in this connection form, the filter capacitor CN forms a normal mode low-pass filter together with the filter choke coil LN. The resonance capacitor C3
Are provided in parallel with the high-speed recovery type diode D1. Although a detailed description is omitted here, for example, the resonance capacitor C3 forms a parallel resonance circuit together with, for example, the filter choke coil LN and the like, so that its resonance frequency is substantially equal to the resonance frequency of the series resonance circuit described later. Is set. This has the effect of suppressing an increase in the rectified smoothed voltage Ei when the load is reduced. As described above, the power factor improving circuit 21 is connected to the end of the series resonance circuit (N1, C1) with respect to the connection point between the filter choke coil LN and the anode of the fast recovery type diode D1. Is connected.

In such a connection form, the switching output obtained on the primary winding N1 is connected to the series resonance capacitor C1.
The switching output is fed back to the rectified current path through the capacitive coupling of the switch. In this case, the switching output is applied to the connection point between the filter choke coil LN and the anode of the high speed recovery type diode D1 so that the resonance current obtained in the primary winding N1 flows. become.

As the switching output is fed back as described above, the alternating voltage of the switching cycle is superimposed on the rectified current path. In the diode D1, an operation of interrupting the rectified current in the switching cycle is obtained, and the apparent inductance of the filter choke coil LN also increases due to the interrupting action. Further, a voltage is generated at both ends of the resonance capacitor C3 when a current of a switching period flows through the resonance capacitor C3. However, the level of the rectified smoothed voltage Ei is reduced by the voltage across the resonance capacitor C3. Accordingly, even during the period when the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci, the smoothing capacitor C
The charging current to i flows. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, the conduction angle of the AC input current is expanded, and the power factor is also improved.

As described above, in the power supply circuits shown in FIGS. 10 and 11, the power factor can be improved by providing the power factor improvement circuits (20, 21). Since the power factor improving circuit is formed with a small number of components, it has the advantage that the power factor can be improved with high efficiency, low noise, small size, light weight, and low cost.

[0028]

FIG. 12 shows the relationship between the load power Po and the power factor PF of the power supply circuit shown in FIGS. Here,
The condition when the AC input voltage VAC is 230 V is shown.
According to this figure, it can be seen that the characteristic that the power factor PF decreases as the load power Po decreases.

FIG. 13 shows the relationship between the AC input voltage VAC and the power factor PF. Here, the maximum load power Pomax = 200 W and the minimum load power Pomin = 50
The characteristics under each condition at the time of W are shown. As shown in this figure, it can be seen that the power factor PF decreases proportionally as the AC input voltage VAC increases. Further, as the power factor PF under the condition of the minimum load power Pomin = 50 W, the maximum load power Pomax = 20
The power factor is lower than 0W. That is, FIG.
As described above, the characteristic that the power factor PF is lower when the load power is smaller is also obtained here.

An operation waveform diagram corresponding to the characteristic shown in FIG. 13 is as shown in FIG.
Here, FIGS. 14A and 14B show that the AC input voltage VAC =
FIG. 14C shows the AC input voltage VAC and the AC input current IAC when the maximum load power Pomax = 200 W at 230 V.
(D) shows the AC input voltage VAC and the AC input current IAC when the AC input voltage VAC is 230 V and the minimum load power Pomin is 50 W. Here, assuming that the half cycle of the AC input voltage VAC is 10 ms, the maximum load power Pomax =
At 200 W, the conduction period τ of the AC input current IAC is actually about 5 ms, and the power factor is PF = 0.85
Becomes On the other hand, the minimum load power Pomin = 50W
At times, the conduction period τ of the AC input current IAC is reduced to about 2.5 ms, and the power factor is reduced to about PF = 0.65. The value of the power factor PF obtained when the minimum load power Pomin = 50 W may not satisfy the value as the power factor required for practical use.

As described above, the fact that the power factor is reduced by the fluctuation of the AC input voltage and the fluctuation of the load power means that the AC input voltage and the load conditions for these power supply circuits are limited. become. That is, there is a problem that devices that can be used as the power supply circuit are limited. Specifically, for example, it can be adopted in a color television receiver in which an AC input voltage and a load condition are specified,
Office equipment and information equipment cannot be adopted.

In the configuration for improving the power factor shown in FIGS. 10 and 11, the series resonance circuit on the primary side is connected to the rectified current path of the commercial AC power supply. On the other hand, the commercial AC power supply cycle (50Hz / 60H
It has been found that the ripples of z) overlap. The superimposed level of the ripple component increases as the load power increases. If, for example, required components are selected so as to maintain a power factor of about PF = 0.8 under predetermined measurement conditions corresponding to practicality, the power factor improving circuit is provided. It is known that the ripple voltage level appearing on the secondary side DC output voltage at the time of maximum load power increases to about 3 to 4 times as compared with the case where it is not possible.

In order to suppress the increase of the ripple component as described above, for example, in the actual power supply circuit shown in FIGS. 10 and 11, the gain of the control circuit 1 is improved, and the capacitance of the primary-side smoothing capacitor Ci is increased. However, in this case, the cost of the component element is increased, and the switching operation is liable to abnormally oscillate.

Further, in the power supply circuit shown in FIGS. 10 and 11, the impedance of the choke coil LS and the resonance capacitor C3 is inserted into the primary side series resonance circuit of the current resonance type converter, so that the maximum load power Pomax is reduced. descend. Therefore, there is a disadvantage that the primary winding N1 and the series resonance capacitor C1 of the insulating converter transformer PIT need to be redesigned.

[0035]

In view of the above-mentioned problems, the present invention is configured as a switching power supply circuit as follows. That is, rectifying and smoothing means for rectifying the input commercial AC power supply, dividing the smoothed voltage obtained between both ends of two smoothing capacitors connected in series to output first and second DC input voltages, and A gap is formed so as to obtain a required coupling coefficient to be coupled, an insulating converter transformer provided for transmitting a primary output to a secondary side, and the first and second DC input voltages, respectively. First and second switching means intermittently output by a switching element to the primary winding of the insulating converter transformer, and at least a leakage inductance component including the primary winding of the insulating converter transformer and a primary parallel circuit The operation of the first and second switching means is formed as a voltage resonance type by being formed by the capacitance of a resonance capacitor. And a primary-side resonant circuit.
Further, as the power factor improving circuit, a magnetic coupling transformer for magnetically coupling a first winding connected to the primary winding, a second winding inserted to the rectification current path, and a magnetic coupling transformer inserted to the rectification current path. And at least a switching element. A secondary resonance circuit formed on the secondary side by a leakage inductance component of a secondary winding of the insulating converter transformer and a capacitance of a secondary resonance capacitor, and a secondary resonance circuit formed including the secondary resonance circuit. DC output voltage generation means configured to input an alternating voltage obtained to a secondary winding of the insulating converter transformer and perform a rectification operation to generate a secondary DC output voltage; and Constant voltage control means configured to perform constant voltage control on the secondary DC output voltage by variably controlling the switching frequency of the switching means according to the level of the side DC output voltage. And

According to the above configuration, the power supply circuit, which is a so-called composite resonant converter of the voltage dividing push-pull system, is provided with the power factor improving circuit having the magnetic coupling transformer, so that the switching output obtained in the primary side resonant circuit is obtained. But,
It is fed back to the rectified current path by the magnetic coupling method.

[0037]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to an embodiment of the present invention.
In this figure, the same parts as those in FIG. 10 or FIG.

A voltage resonance type switching converter (voltage resonance type converter) is provided on the primary side of the power supply circuit shown in FIG. A power factor improving circuit 10 is provided for the voltage resonance type converter. The power supply circuit shown in FIG.
= 150V or more (so-called AC200V system).

The power supply circuit shown in this figure employs a so-called voltage division push-pull system. For this reason,
As a smoothing capacitor for obtaining a rectified smoothed voltage, 2
The smoothing capacitors Ci1 and Ci2 are provided. These smoothing capacitors Ci1 and Ci2 are provided so as to be connected in series between the positive output line of the bridge rectifier circuit Di and the primary side ground as shown in the figure. In addition,
Since the series connection circuit of the smoothing capacitor Ci1 and the smoothing capacitor Ci2 shown by the broken line can be regarded as one smoothing circuit for generating the rectified smoothing voltage Ei, the series connection circuit of the smoothing capacitor Ci1 and the smoothing capacitor Ci2 is as follows. It is simply referred to as a smoothing capacitor Ci.

In this case, one end of the primary winding N1A is connected to the positive electrode side of the smoothing capacitor Ci1 via the magnetic coupling transformer MCT of the power factor improvement circuit 10. The connection point between the negative electrode of the smoothing capacitor Ci1 and the positive electrode of the smoothing capacitor Ci2 is connected via a choke coil CH to the connection point between the end of the primary winding N1B and the emitter of the switching element Q1.

As this power supply circuit, a quadrature drive transformer PRT is used to drive two sets of switching elements Q1 and Q2 by a push-pull operation and variably control the switching frequency of the switching elements Q1 and Q2. Is provided.

As the structure of the orthogonal drive transformer PRT, for example, a three-dimensional core is formed by joining the ends of the magnetic legs of two double U-shaped cores having four magnetic legs. The detection winding ND and the drive windings NB1 and NB2 are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction, and the control winding NC is connected to the detection winding. By winding in the direction orthogonal to ND and the drive windings NB1 and NB2, a saturable reactor is formed.

In this case, the driving windings NB1 and NB2 are
As shown in the figure, they are wound independently on the orthogonal control transformer PRT. One end of the drive winding NB1 is connected to a connection point between the smoothing capacitors Ci1 and Ci2 via the choke coil CH, and the other end is connected to the resonance capacitor CB1.
The switching element Q via the base current limiting resistor RB1
Connected to the base of one. One end of the driving winding NB2 is grounded to the primary side ground, and the other end is a resonance capacitor C.
B2--connected to the base of switching element Q2 via base current limiting resistor RB2. That is, a self-excited oscillation drive circuit for the switching element Q1 is formed by the drive winding NB1-the resonance capacitor CB1-the base current limiting resistor RB1.
The drive winding NB2-the resonance capacitor CB2-the base current limiting resistor RB2 forms a self-excited oscillation drive circuit for the switching element Q2.

In the detection winding ND, an alternating voltage corresponding to a switching output is detected by a switching operation described later. In the drive windings NB1 and NB2, alternating voltages having opposite polarities different from each other by 180 ° can be obtained in accordance with the switching output detected by the detection winding ND.

As the two switching elements Q1 and Q2 provided for the push-pull operation, a high breakdown voltage bipolar transistor (BJT: junction type transistor) is employed.

The drive circuit (drive winding NB1-base current limiting resistor RB1-resonance capacitor CB1), clamp diode DD1, and parallel resonance capacitor Cr1 are connected to the switching element Q1 as shown in FIG. Is connected to a drive circuit (drive winding NB2-base current limiting resistor RB2-resonance capacitor CB2), clamp diode DD2, and parallel resonance capacitor Cr2 as shown in the figure.

Here, the clamp diodes DD1, DD2
Are connected in parallel with each other between the base and collector of the switching elements Q1 and Q2. The parallel resonance capacitor Cr1 is connected between the collector and the emitter of the switching element Q1. Also the parallel resonance capacitor Cr
2 is connected between the collector and the emitter of the switching element Q2. Further, the emitter of the switching element Q2 is connected to the primary side ground.

The starting resistor Rs is connected between the positive electrode of the smoothing capacitor Ci2 and the switching element Q2. The starting resistor Rs is inserted to supply a starting current for starting a switching operation to the switching element Q2 at the time of starting.

The insulating converter transformer PIT for transmitting the switching outputs of the switching elements Q1 and Q2 to the secondary side is, as shown in FIG. 2, an E-shaped core CR1, CR2 made of, for example, a ferrite material.
An EE-type core is provided so that the magnetic legs of the EE-type core are opposed to each other. A primary winding N1 (N1A, N1B) The secondary winding N2 is wound in a divided state. A gap G is formed for the center magnetic leg as shown in the figure. As a result, loose coupling with a required coupling coefficient can be obtained. Gap G is
The E-shaped cores CR1 and CR2 can be formed by forming the central magnetic legs shorter than the two outer magnetic legs. Also,
As the coupling coefficient k, for example, a loosely coupled state of k ≒ 0.85 is obtained, so that a saturated state is hardly obtained.

In this case, the insulation converter transformer PIT
Is divided into primary windings N1A and N1B, one end of the primary winding N1A is connected to the collector of the switching element Q1, and the other end is smoothed via the magnetic coupling transformer MCT of the power factor improvement circuit 10. Connected to the positive electrode side of capacitor Ci1. One end of primary winding N1B is connected to the collector of switching element Q2 via detection winding ND, and the other end is inductance winding L of choke coil CH.
c is connected to the positive electrode of the smoothing capacitor Ci2 via a series connection.

In this case, the above-described parallel resonance capacitor C
r1 is the leakage inductance component of the primary winding N1A (L1
A) and the combined inductance (L1A + Lc) of the inductance winding Lc form a parallel resonance circuit for causing the switching element Q1 to operate in a voltage resonance type. Similarly, the parallel resonance capacitor Cr2 is connected to the primary winding N1B.
And the combined inductance (L1B + Lc) of the inductance winding Lc and the leakage inductance component (L1B), form a parallel resonance circuit for causing the switching element Q2 to operate in a voltage resonance type. Although not described in detail here, when each of the switching elements Q1 and Q2 is turned off, the voltage across the resonance capacitors Cr1 and Cr2 actually becomes a sinusoidal pulse waveform due to the action of these parallel resonance circuits. A resonance type operation is obtained.

In such a configuration on the primary side, the driving winding N
Since alternating voltages of opposite polarities are obtained in B1 and the driving winding NB2, the self-excited oscillation driving circuit including the driving winding NB1 and the self-excited oscillation driving circuit including the driving winding NB2 respectively have opposite polarities. , A drive current (base current) as an alternating current flows through each base of the switching elements Q1 and Q2. Thereby, the switching element Q1,
As for Q2, an operation of alternately turning on / off at a switching frequency determined by a constant of the self-excited oscillation driving circuit is obtained. That is, a switching operation by push-pull is obtained in a voltage resonance type.

When the switching elements Q1 and Q2 perform the push-pull operation, the switching element Q1
Performs the switching operation by inputting the voltage between both ends of the smoothing capacitor Ci1, and the switching element Q2 performs the switching operation by inputting the voltage between both ends of the smoothing capacitor Ci2. That is, the switching elements Q1, Q2
Are switched by inputting a DC voltage having a level of 1/2 Ei, respectively. Then, depending on the switching operation on the switching element Q1, the switching output current flows through the path of the smoothing capacitor Ci1, the primary winding N1A, the collector of the switching element Q1, the emitter of the switching element Q1, and the choke coil CH. Depending on the switching operation, the smoothing capacitor Ci2 → the choke coil CH → the primary winding N1B
A switching output current flows through a path from the collector of the switching element Q2 to the emitter of the switching element Q2. By configuring so that such an operation can be obtained,
For example, even under the condition of AC200V system, AC100V
The switching operation by push-pull can be performed under the same conditions as those of the system.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. With this parallel resonance circuit,
The alternating voltage excited by the secondary winding N2 becomes a resonance voltage.
That is, a voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, the primary side is provided with a parallel resonance circuit for making the switching operation a voltage resonance type, and the secondary side is also provided with a parallel resonance circuit for obtaining the voltage resonance operation. . In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

In this case, for the secondary parallel resonance circuit formed as described above, a tap is provided for the secondary winding N2, and the rectifier diodes DO1, DO2, DO are provided.
3, DO4 and the smoothing capacitors CO1, CO2 are connected as shown in the figure, so that a set of [rectifier diodes DO1, DO2, smoothing capacitor CO1] and a set of [rectifier diodes DO3, DO4, smoothing capacitor CO2] are obtained. Two sets of full wave rectifier circuits are provided. The full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] has a DC output voltage EO1.
And a full-wave rectifier circuit composed of [rectifier diodes DO3, DO4, smoothing capacitor CO2] generates a DC output voltage EO2. In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1.

The control circuit 1 supplies a DC current, the level of which is varied according to the level of the DC voltage output EO1 on the secondary side, to the control winding NC of the drive transformer PRT as a control current, as will be described later. The constant voltage control is performed as follows.

Incidentally, the insulation converter transformer PIT
, The polarity (winding direction) of the primary winding N1 and the secondary winding N2 and the rectifier diode DO (DO1, DO2, DO3, DO4)
, The mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 may be + M or -M. For example, when the connection configuration shown in FIG. 3A is adopted, the mutual inductance is + M (additive polarity: forward method), and when the connection configuration shown in FIG. 3B is employed, the mutual inductance is −M (depolarization: Flyback method). If this is made to correspond to the operation of the secondary side of the power supply circuit shown in FIG. 1, for example, when the alternating voltage obtained in the secondary winding N2 has a positive polarity, the rectifier diode DO1 (DO3)
Can be regarded as an operation mode of + M (forward system). Conversely, when the alternating voltage obtained in the secondary winding N2 has a negative polarity, the rectifier diode DO2
The operation in which the rectified current flows through (DO4) can be regarded as the -M operation mode (flyback method). That is, in this power supply circuit, each time the alternating voltage obtained in the secondary winding becomes positive / negative, the mutual inductance becomes + M / -M
Mode.

The control circuit 1 varies the level of the control current (DC current) flowing through the control winding NC according to the change in the secondary-side DC output voltage level (EO1), thereby winding the control coil PRT on the orthogonal control transformer PRT. The inductances LB1 and LB2 of the mounted drive windings NB1 and NB2 are variably controlled. Thereby, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching elements Q1 and Q2 formed including the inductances LB1 and LB2 of the drive windings NB1 and NB2 changes. This is an operation of changing the switching frequency of the switching elements Q1 and Q2. This operation has an effect of stabilizing the secondary-side DC output voltage.

Next, the configuration of the power factor correction circuit 10 will be described. The power factor correction circuit 10 shown in FIG. 1 includes a filter capacitor CN, a high speed recovery type diode D1, and a magnetic coupling transformer MCT. Here, the magnetic coupling transformer MCT is configured by winding a primary winding (first winding) Np and a secondary winding (second winding) Ns in a tightly coupled state, for example. Further, in the present embodiment, on the assumption that the inductance Ls of the primary winding Np and the inductance Ls of the secondary winding Ns are larger than the inductance Lp, A predetermined inductance value is selected. For this purpose, for example, the number of turns (turn ratio) of the primary winding Np and the secondary winding Ns is set according to the inductance value to be actually selected.

The winding start end of the primary winding Np of the magnetic coupling transformer MCT is connected to the primary winding N1A of the insulating converter transformer PIT, and the winding end is a smoothing capacitor Ci.
Connected to one positive terminal. That is, the primary winding N1
Connected in series with A. Thereby, the switching element Q1 is connected to the primary winding Np of the magnetic coupling transformer MCT.
Is transmitted via the primary winding N1A.

The winding start end of the secondary winding Ns of the magnetic coupling transformer MCT is connected to the cathode of the high-speed recovery type diode D1. Here, the anode of the fast recovery diode D1 is connected to the positive output terminal of the bridge rectifier circuit Di. The winding end of the secondary winding Ns is connected to the positive terminal of the smoothing capacitor Ci1. In other words, a series connection circuit of the high-speed recovery type diode D1 and the secondary winding Ns is inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci1 (that is, a rectified current path). The primary winding Np and the secondary winding Ns of the magnetic coupling transformer MCT shown in FIG.
3 (a), the operation is in the polarity (+ M) mode.

In this case, the filter capacitor CN
Is inserted between the anode side of the high-speed recovery type diode D1 and the primary side ground to form, for example, a normal mode low-pass filter together with the secondary winding Ns of the magnetic coupling transformer MCT.

The power factor improving operation of the power factor improving circuit 10 having such a configuration is basically as follows. In the power factor correction circuit 10, first, the isolation converter transformer P
The switching output of switching element Q1 obtained in IT primary winding N1A is transmitted to primary winding Np of magnetically coupled transformer MCT. Then, via the magnetic coupling in the magnetic coupling transformer MCT, the switching output obtained in the primary winding Np is excited with respect to the secondary winding Ns. That is, the switching output is fed back to the rectified current path by the magnetic coupling of the magnetic coupling transformer MCT.

By the switching output thus fed back, the fast recovery type diode D1 operates so that the rectified current is intermittently switched in the switching cycle. Due to this intermittent operation, the inductance LS of the secondary winding Ns functioning as a choke coil also increases, and the charging current to the smoothing capacitor Ci also increases during the period when the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci. Flows. Therefore, also in this case, as a result, the conduction angle of the AC input current is enlarged, and the power factor is improved.

FIG. 4 shows operation waveforms of the power supply circuit shown in FIG. Here, FIGS. 4A to 4E show the operation of each unit in the commercial power supply cycle, and FIGS. 4F to 4J show the operation of each unit in the switching cycle. This operation is performed under the condition that the AC input voltage VAC = 230 V and the maximum load power 2
The operation at the time of 00W is performed.

Here, the frequency of the commercial power supply is 50 Hz, and the AC input voltage VAC has a sinusoidal waveform having a half cycle of 10 ms as shown in FIG. 4A.
As the rectified current flowing through the bridge rectifier circuit Di, when the AC input current IAC flows as shown in FIG. 4B, the high-speed recovery type diode D1 switches so as to intermittently switch the current. The switching current ID having the waveform shown in FIG.

Here, the operation of the switching cycle during the period when the AC input voltage VAC is high and the AC input current IAC flows is as follows. For example, assuming that the switching element Q2 performs a switching operation during this period, a sine wave is applied to both ends of the parallel resonance capacitor Cr2 in a period Toff (3 μs) during which the switching element Q2 is turned off as shown in FIG. A parallel resonance voltage Vcp that becomes a wave-like pulse is generated. Then, during a period Ton (7 μs) during which the switching element Q1 is turned on, a voltage between the collector and the emitter of the switching element Q1 is
As shown in FIG. 4 (g), the switching output current Ic
Although p flows, the switching output current Icp flows from the filter capacitor CN in the power factor correction circuit 10 to the smoothing capacitor Ci via the high speed recovery type diode D1 → secondary winding Ns. At this time, the switching current ID flowing through the high-speed recovery type diode D1 is as shown in FIG.
As shown in FIG. 7, the waveform has a level of 11 Ap on a substantially sine wave.

Further, as shown in FIG. 4 (i), the voltage VL across the secondary winding Ns in the switching cycle becomes a sinusoidal waveform of 100 Vp in the positive direction during the period Toff, and during the period Ton. In the start period, a waveform of 60 Vp in the negative direction is obtained. FIG. 4 (j) shows the secondary winding Ns in the switching cycle.
-The voltage V1 across the series connection circuit of the smoothing capacitor Ci is shown, and this voltage V1 is
A sinusoidal waveform of 460 Vp is obtained. In the period Ton, a sinusoidal waveform of 300 Vp in the reverse direction is obtained in the start period, and thereafter, 350 V is maintained. In the configuration shown in FIG. 1, this voltage V1 is supplied as an operating power supply for the primary-side switching converter as a DC input voltage.

Then, as an operation based on the commercial power supply cycle,
In a period in which the AC input voltage VAC is low and the AC input current IAC does not flow (a period in which the high-speed recovery type diode D1 does not perform a switching operation), the isolation converter transformer is connected to the primary winding Np of the magnetic coupling transformer. A primary-side switching current I1 flowing through the primary winding N1 of the PIT flows, thereby generating an excitation voltage in the secondary winding Ns of the magnetic coupling transformer MCT. Thereby, the voltage VL across the secondary winding Ns as viewed in the commercial power supply cycle is
FIG. 4D shows that a substantially constant voltage level is constantly obtained. The voltage VL across the secondary winding Ns is superimposed on the rectified and smoothed voltage Ei (the voltage across the smoothing capacitor Ci), so that the secondary winding Ns
A voltage V1 across the series connection circuit of the smoothing capacitor Ci;
In the commercial power cycle, a waveform shown in FIG. 4E is obtained. As can be seen from the waveform shown in FIG. 4E, in the present embodiment, the secondary winding N of the magnetic coupling transformer MCT
It operates to increase the level of the rectified and smoothed voltage Ei (DC input voltage) by the voltage excited by s. As described above, this operation is set such that the inductance Ls of the primary winding Np of the magnetic coupling transformer MCT and the inductance Ls of the secondary winding Ns are larger than the inductance Lp. It is obtained by doing.

As described above, by increasing the level of the DC input voltage, for example, the cathode potential V1 of the high-speed recovery type diode D1 is changed to the input voltage (the high-speed recovery type diode D1 of the high-speed recovery type diode D1). The high-speed recovery diode D1 operates so as to continue the switching operation even during a period that is lower than the anode side potential). In other words, the switching period of the high-speed recovery type diode D1 is extended every half cycle of the commercial power supply cycle. By such an operation, for example, for a half cycle of the commercial power supply of 10 ms, FIG.
As shown in (b), the conduction angle of the AC input current IAC is 6 ms.
Thus, a higher power factor can be obtained as compared with the circuits shown in FIGS. 9 and 10.

Here, FIGS. 5, 6 and 7 show experimental results of the power supply circuit shown in FIG. In addition,
In obtaining the experimental results shown in these figures, the EI type core of EI-25 was used for the magnetic coupling transformer MCT, and the inductance Lp = 2 for the primary winding Np.
5 μH, the inductance Ls =
It is 210 μH. And the filter capacitor C
N = 1μF, parallel resonance capacitor Cr (Cr1, Cr2)
= 2700 pF, switching elements Q1 and Q2 are VCBO>
800V is selected. The operating conditions are as follows: load power Po = 200 W to 0 W, AC input voltage VAC = 16
0 V to 288 V.

First, FIG. 5 shows that the AC input voltage VAC = 230
7 shows the relationship between the load and the power factor under the condition that V is constant. As shown in the figure, in the present embodiment, the load power Po is about 40 W or more, and the power factor PF is kept about 0.75 or more. Then, the characteristic that the power factor gradually increases until the power factor PF becomes approximately 0.8 at the load power Po = 200 W is obtained. Thus, in the present embodiment, a high power factor can be obtained over a wide range of load conditions.

FIG. 6 shows that the load power Po = 200 W
2 shows the relationship between the AC input voltage VAC and the power factor under the condition that is constant. As can be seen from this figure, as the power factor PF, a high power factor of about PF = about 0.9 or more has already been obtained near the AC input voltage VAC = 160 V.
As the AC input voltage VAC fluctuates so as to increase, the power factor PF decreases, but even at the AC input voltage VAC = 280 V, a value close to PF = 0.8 is maintained. Is what it is.

As described above, in the power supply circuit of the present embodiment, a high power factor can be maintained even when the AC input voltage and the load fluctuate. For this reason, the power supply circuit according to the present embodiment is not limited to a television receiver or the like in which the AC input voltage or the load condition is specified, and is applied to office equipment such as an office equipment or a personal computer in which the load condition varies. It is practically possible to mount it.

Further, in the power supply circuit of the present embodiment, when the AC input voltage VAC = 100 V and the load power Po = 200 W, the DC input voltage (in the case of FIG.
As a result, about 384 V can be obtained. This is because the DC input voltage when the power factor correction circuit 10 is not provided in the circuit of FIG. Is what it is. This allows
In the present embodiment, the power conversion efficiency increases, for example, 90.5% in the case where the power factor improvement circuit 10 is not provided, whereas in the present embodiment, it is improved to 91.8%. . Further, the AC input power is reduced by 3.5 W as compared with the case where the power factor improvement circuit 10 is not provided.

As an experimental result, feedback of the switching current I1 from the primary winding Np to the secondary winding Ns by changing the ratio of the inductance values between the windings (Np, Ns) of the magnetic coupling transformer MCT. It has been confirmed that the power factor characteristics can be changed by changing the amount. That is, in the present embodiment, the winding (N
By changing the inductance value of (p, Ns), the adjustment can be performed so that the power factor characteristics under conditions suitable for actual use can be obtained. In order to change the ratio of the inductance value of the primary winding Np / secondary winding Ns of the magnetic coupling transformer MCT as described above, for example, the winding ratio of the primary winding Np / secondary winding Ns may be changed. It is.

Here, as the inductance value of the primary winding Np / secondary winding Ns in the magnetic coupling transformer MCT,
For the primary winding Np, the inductance Lp = 13 μH,
For the secondary winding Ns, the inductance Ls = 160 μ
FIG. 7 shows operation waveforms of the respective parts obtained when the turns ratio is set so as to be H.

As shown in FIG. 7A, assuming that the AC input voltage VAC is input at a commercial power supply cycle of, for example, 50H, the high-speed recovery type diode D1
The switching operation is performed as indicated by the switching current ID in (c). The voltage VL across the secondary winding Ns of the magnetic coupling transformer MCT at this time is as shown in FIG.
As shown in, the AC input voltage VAC is assumed to be low.
The alternating waveform corresponding to the switching cycle is obtained substantially corresponding to the period other than the period, and the voltage (DC input voltage) V1 obtained in the series connection circuit of the secondary winding Ns and the smoothing capacitor Ci is the AC voltage. The waveform substantially the same as the voltage VL is superimposed on the voltage level of 330 Vp during the τ period in which the input voltage VAC is assumed to be high, and 240 V during periods other than the τ period.

Further, when the winding ratio of the magnetic coupling transformer MCT is changed as described above, the AC input voltage VAC =
FIG. 8 shows the relationship between the load and the power factor under a condition where the voltage is constant at 230 V. For example, in comparison with FIG. 5, in the case of FIG. 8, when the load power Po = 50 W or more, the value of the power factor PF becomes lower, but the power factor PF is almost irrespective of the fluctuation of the load power.
= 0.75, and the load power Po = 25
Below about W, the characteristic that the power factor increases is obtained.

FIG. 9 shows a load power Po = 200 in a configuration in which the winding ratio of the magnetic coupling transformer MCT is also changed.
The relationship between the AC input voltage VAC and the power factor under the condition where the power is constant at W is shown. Also in this case, as for the power factor PF, when the AC input voltage VAC is about 200 V or more, the power factor PF becomes almost constant at about 0.75, and VAC = 200 V
The characteristic that the power factor PF becomes higher when the power factor is lower than the level is obtained.

In the configuration of the switching power supply circuit shown in FIG. 1, primary winding N of magnetically coupled transformer MCT
Regarding the relationship between p and the winding direction of the secondary winding Ns, the relationship shown in FIG. 3B, that is, even when the magnetic coupling transformer MCT is configured to operate with a reduced polarity (−M: flyback method), The characteristics shown in FIGS. 5, 6, 8, and 9 can be obtained. In this case, when the switching element Q1 is turned off, the switching current ID flows through the high-speed recovery type diode D1, and the phases of the high-frequency waveforms of the voltage VL and the voltage V1 are opposite.

Although the embodiments have been described above, the present invention may have various modifications. For example, the present applicant has already proposed a configuration provided with a quadruple voltage rectifier circuit using a secondary-side series resonance circuit as a composite resonance type switching converter, but such a configuration is also a modification of the present embodiment. It can be established as an example. That is, the present embodiment is not particularly limited as the configuration of the secondary-side resonance circuit and the rectifier circuit.

In the above-described embodiment, the configuration in which the switching output of switching element Q1 is fed back to power factor correction circuit 10 has been described as an example.
It is conceivable that the switching output of No. 2 is fed back to the power factor correction circuit 10.

In the present embodiment, the orthogonal control transformer is used as a control transformer for performing constant voltage control under a configuration having a self-excited resonance converter for the primary side. Instead of the orthogonal control transformer, an oblique control transformer previously proposed by the present applicant can be employed. Although the illustration of the structure of the oblique control transformer is omitted here, for example, as in the case of the orthogonal control transformer, a three-dimensional structure is obtained by combining two sets of double U-shaped cores having four magnetic legs. Form a mold core. Then, the control winding NC and the driving winding NB are wound around the three-dimensional core. At this time, the winding directions of the control winding and the driving winding obliquely intersect. To be. Specifically, the control winding N
C and one of the drive windings NB are wound around two magnetic legs adjacent to each other among the four magnetic legs, and the other winding is wound diagonally. It is assumed that there is a positional relationship 2
It is wound around the magnetic legs of the book. When such an oblique control transformer is provided, even when the AC current flowing through the drive winding changes from a negative current level to a positive current level, the operation tendency that the inductance of the drive winding increases. Is obtained. Thereby, the current level in the negative direction for turning off the switching element increases, and the accumulation time of the switching element is shortened.Accordingly, the fall time at the time of turning off the switching element is also shortened, The power loss of the switching element can be further reduced.

[0086]

As described above, the present invention relates to a switching power supply circuit provided with a power factor improvement circuit for a complex resonance type converter of a divided voltage push-pull type. Is provided, and the switching output supplied to the primary winding (first winding) of the magnetic coupling transformer is transmitted to the secondary winding (second winding) of the magnetic coupling transformer, so that the switching output is rectified current. Power is fed back to the route. The switching output fed back to the rectified current path causes the high-speed recovery diode (switching element) to perform a switching operation for interrupting the rectified current, thereby improving the power factor. As a result, the power factor can be improved in the power supply circuit of the divided voltage push-pull system. That is, it is possible to provide a power supply circuit that is compatible with the AC 200 V system and that has an improved power factor.

In the magnetic coupling transformer, the first
By setting the inductance of the second winding to be larger than the inductance of the winding, the DC input voltage increases. As a result, as a result, the AC input voltage and the load power are reduced. Even with fluctuations, a power factor sufficient for practical use is maintained. Therefore, for example, it is suitable as a power factor improving power supply circuit for office equipment and information equipment having a large load variation as an AC200V power supply circuit. The characteristics of the power factor include, for example, adjusting the ratio between the inductance of the first winding and the inductance of the second winding to obtain an AC input voltage,
Since it can be made substantially constant with respect to the fluctuation of the load power, it is possible to suppress the ripple component. In addition, the power conversion efficiency is improved by increasing the DC input voltage.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a side sectional view showing a structure of an insulating converter transformer employed in the power supply circuit of the present embodiment.

FIG. 3 is an explanatory diagram showing each operation when the mutual inductance is + M / -M.

FIG. 4 is a waveform chart showing an operation of the switching power supply circuit shown in FIG.

FIG. 5 is a characteristic diagram showing a relationship between load power and power factor for the switching power supply circuit shown in FIG.

6 is a characteristic diagram showing a relationship between an AC input voltage and a power factor for the switching power supply circuit shown in FIG.

FIG. 7 is a waveform chart showing an operation of the switching power supply circuit shown in FIG. 1 when the winding ratio of the magnetic coupling transformer is changed.

FIG. 8 is a characteristic diagram showing a relationship between load power and power factor of the switching power supply circuit shown in FIG. 1 when the winding ratio of the magnetic coupling transformer is changed.

9 is a characteristic diagram showing a relationship between an AC input voltage and a power factor of the switching power supply circuit shown in FIG. 1 when a winding ratio of a magnetic coupling transformer is changed.

FIG. 10 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 11 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 12 is a characteristic diagram showing a relationship between load power and power factor for a power supply circuit of the prior art.

FIG. 13 is a characteristic diagram illustrating a relationship between an AC input voltage and a power factor in a power supply circuit according to the related art.

FIG. 14 is a waveform diagram showing the operation of a prior art power supply circuit with respect to the input of a commercial AC power supply according to load power.

[Explanation of symbols]

1 control circuit, 10 power factor improvement circuit, Di bridge rectifier circuit, Ci smoothing capacitor, D1 high speed recovery type diode, MCT magnetic coupling transformer, Cr1, Cr
2 parallel resonance capacitor, C3 parallel resonance capacitor,
C2 Secondary parallel resonance capacitor, PRT orthogonal control transformer, PIT isolation converter transformer, Q1, Q2
Switching element

Claims (3)

[Claims] 1. A rectifying / smoothing means for rectifying an inputted commercial AC power supply, dividing a smoothed voltage obtained between both ends of two series-connected smoothing capacitors, and outputting first and second DC input voltages. An insulating converter transformer provided with a gap so as to obtain a required coupling coefficient that is loosely coupled, and provided to transmit a primary output to a secondary side; and the first and second DC input voltages. First and second switching means, each of which is intermittently output by a switching element to output to the primary winding of the insulating converter transformer, and at least a leakage inductance component including the primary winding of the insulating converter transformer. And the capacitance of the primary side parallel resonance capacitor, and controls the operation of the first and second switching means in a voltage resonance type. A magnetic coupling transformer that magnetically couples a primary winding connected to the primary winding, a first winding connected to the primary winding, a second winding inserted into the rectifying current path, and a magnetic coupling transformer inserted into the rectifying current path A power factor improving means for performing a power factor improving operation by including at least a switching element; a leakage inductance component of a secondary winding of the insulating converter transformer; and a capacitance of a secondary side resonance capacitor, formed on a secondary side. A secondary-side resonance circuit, which is formed including the secondary-side resonance circuit, receives an alternating voltage obtained in a secondary winding of the insulated converter transformer, performs a rectification operation, and performs a secondary-side DC output. DC output voltage generating means configured to generate a voltage; and variably controlling a switching frequency of the switching means according to a level of the secondary DC output voltage. And
A switching power supply circuit comprising: constant voltage control means configured to perform constant voltage control on a secondary-side DC output voltage. 2. The switching power supply circuit according to claim 1, wherein in the magnetic coupling transformer, the second winding has a predetermined inductance larger than that of the first winding. . 3. A turns ratio between the first winding and the second winding so that a required value is obtained for each of the inductance values of the first winding and the second winding of the magnetic coupling transformer. The switching power supply circuit according to claim 1, wherein the switching power supply can be changed and set.