JP2001136745A - Switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To maintain power factor sufficient for actual application conditions for a load, an AC input voltage variation which corresponds to a wide range of input voltage. SOLUTION: A switching output voltage, obtained at a primary side resonance circuit, is fed back with an electrostatic coupling system or a magnetic coupling system to a power factor improving circuit provided in the power supply circuit called a 'push-pull type switching frequency control system composite resonance type converter', and push-pull operation and voltage-divided push-pull operation are changed over, depending on the AC input voltage with a switching operation change-over means.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit having a power factor correction circuit.

[0002]

2. Description of the Related Art Conventionally, an AC input voltage of 100 V system and 2
A wide range power factor improving soft switching power supply commonly used for the 00V system has various circuits configured by combining a current resonance type converter and an electrostatic coupling or magnetic coupling type power factor improving circuit. First, the configurations and problems of various switching power supply circuits will be described with reference to FIGS.

FIGS. 8 and 9 show a power factor improving soft switching power supply in which a conventional current resonance type converter and a power feedback type power factor improving circuit are combined, and FIG. 8 shows an electrostatic coupling type power factor improving circuit. FIG. 9 shows a method employing a magnetic coupling type power factor improving circuit 21.

The power supply circuits shown in FIGS. 8 and 9 are provided with a bridge rectifier circuit Di for performing full-wave rectification on a commercial AC power supply AC. In this case, the bridge rectifier circuit D
The rectified output rectified by i is charged to the smoothing capacitor Ci via the power factor correction circuit 20 or 21, and a rectified smoothed voltage Ei corresponding to a level that is one time of the AC input voltage VAC is applied across the smoothing capacitor Ci. Is obtained.
Also, in the rectifying / smoothing circuit (Di, Ci), an inrush current limiting resistor Ri is inserted in the rectification current path so as to suppress an inrush current flowing into the smoothing capacitor when the power is turned on, for example. ing.

The switching power supply circuits of FIGS. 8 and 9 each include a self-excited current resonance type converter using a rectified smoothed voltage Ei, which is a voltage across a smoothing capacitor Ci, as an operating power supply. In this current resonance type converter, the switching elements Q100 and Q200 of two bipolar transistors are half-bridge-coupled as shown in the figure, and then inserted between the connection point on the positive side of the smoothing capacitor Ci and the ground. Connected.
Note that clamp diodes DD100 and DD200 are connected in parallel between the emitter and collector of switching elements Q100 and Q200, respectively. And switching element Q10
0 and Q200 each perform a switching operation at a required switching frequency based on a signal from the control circuit 1.

[0006] Insulated converter transformer PIT (Power Is
olation Transformer) is the switching element Q100, Q2
The switching output of 00 is transmitted to the secondary side. That is,
The switching elements Q100 and Q200 alternately open and close with the terminal voltage of the smoothing capacitor Ci as the operating power supply, thereby forming the primary winding N1 of the insulating converter transformer PIT.
To supply a drive current close to the resonance current waveform to obtain an alternating output to the secondary winding N2. This insulation converter transformer P
One end of the primary winding N1 of the IT is connected to the switching element Q100.
Is connected to the contact point (switching output point) between the emitter of the switching element Q200 and the collector of the switching element Q200, so that a switching output can be obtained.

The other end of the primary winding N1 is connected via a series resonance capacitor C100 to a power factor improving circuit 20 (or 2).
1). In this case, the series resonance capacitor C100 and the primary winding N1 are connected in series, but the capacitance of the series resonance capacitor C100 and the leakage inductance of the insulating converter transformer PIT including the primary winding N1 (series resonance winding) A leakage inductance component forms a primary side series resonance circuit for making the operation of the switching converter a current resonance type.

On the secondary side of the insulated converter transformer PIT in FIGS. 8 and 9, a center tap is provided for the secondary winding N2, and the rectifier diodes DO1, DO2 are provided.
By connecting DO2 and smoothing capacitor CO as shown in the figure, a full-wave rectifier circuit is formed, and a DC output voltage EO is generated. The control circuit 1 is, for example, a DC voltage output E on the secondary side.
The constant voltage control is performed by controlling the switching frequency of the switching elements Q100 and Q200 such that the level is varied according to the level of O.

The power factor improving circuit 20 in the case of FIG.
A filter choke coil L is connected between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci.
N- A fast recovery diode D1 is inserted in series. Here, the filter capacitor CN is provided in parallel with the series connection circuit of the filter choke coil LN and the high-speed recovery type diode D1. Also in this connection form, the filter capacitor CN forms a normal mode low-pass filter together with the filter choke coil LN. The resonance capacitor C3 is provided in parallel with the high-speed recovery type diode D1. Although a detailed description is omitted here, for example, the resonance capacitor C3 is, for example, a filter choke coil L
A parallel resonance circuit is formed together with N and the like, and its resonance frequency is set to be substantially equal to the resonance frequency of a series resonance circuit described later. This has the effect of suppressing an increase in the rectified smoothed voltage Ei when the load is reduced.

The power factor improving circuit 20 includes a filter choke coil LN and a high speed recovery type diode D1.
The primary side series resonance circuit (N
1, C100) are connected.

In such a connection form, the switching output obtained from the primary winding N1 is connected to the series resonance capacitor C1.
The switching output is fed back to the rectified current path through the capacitive coupling of the switch. In this case, the switching output is applied to the connection point between the filter choke coil LN and the anode of the fast recovery diode D1 so that the resonance current obtained in the primary winding N1 flows.

When the switching output is fed back as described above, the alternating voltage of the switching cycle is superimposed on the rectified current path. In the diode D1, an operation of interrupting the rectified current in the switching cycle is obtained, and the apparent inductance of the filter choke coil LN also increases due to the interrupting action. Further, a voltage is generated at both ends of the resonance capacitor C3 when a current of a switching period flows through the resonance capacitor C3. However, the level of the rectified smoothed voltage Ei is reduced by the voltage across the resonance capacitor C3. Accordingly, even during the period when the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci, the smoothing capacitor C
The charging current to i flows. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, the conduction angle of the AC input current is enlarged, and the power factor is improved.

The power factor improving circuit 21 in the case of FIG.
The filter choke coil LN, the high-speed recovery type diode D1, and the choke coil LS are connected in series between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. The filter capacitor CN is inserted between the anode side of the high-speed recovery diode D1 and the positive terminal of the smoothing capacitor Ci to form a normal mode low-pass filter together with the filter choke coil LN. The resonance capacitor C3 is provided in series with the high-speed recovery type diode D1.

For the power factor correction circuit 21, the cathode of the high-speed recovery type diode D1 and the choke coil L
The primary side series resonance circuit (N1, C10
0) is connected. That is, the series resonance circuit (N
1, C100) is fed back to the rectified current path via an inductive reactance (magnetic coupling) assumed to be possessed by the choke coil LS itself.

The switching output fed back as described above causes the alternating voltage of the switching cycle to be superimposed on the rectified current path. The superimposed voltage of the switching cycle causes the high speed recovery type diode D1 to be superimposed. In this case, an operation of interrupting the rectified current in the switching cycle is obtained, and the apparent inductance of the filter choke coil LN and choke coil LS also increases due to the intermittent action. This allows the charging current to flow to the smoothing capacitor Ci even during a period in which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, and the conduction angle of the AC input current is increased. As a result, the power factor is improved.

FIG. 10 shows an AC input voltage VAC = 100 V in the switching power supply circuits of FIGS.
And the operation waveform of the AC input current when VAC = 230 V. FIG. 11 shows the power factor PF-AC input voltage VAC characteristic. As can be seen from these figures, even when the power factor PF = 0.85 when the AC input voltage VAC = 100 V, the power factor PF =
It drops to 0.65. For example, the switching power supply circuit shown in FIGS. 8 and 9 is not suitable as a wide range power factor improving soft switching power supply commonly used for 100 V and 200 V systems. .

On the other hand, FIG. 12 shows a series resonance circuit (L
30 and C30), so that the series resonance current I01 of the choke coil L30 and the resonance capacitor C30 is fed back to the parallel connection point of the high-speed recovery type diode D1 and the capacitor C3 of the power factor correction circuit 20. is there. In the following description of each drawing, the same portions as those of the power supply circuit in the drawings already described are denoted by the same reference numerals, and description thereof is omitted.

In this power supply circuit, the switching element Q10
The series resonance circuit (N1, C100) provided to make the switching operation of Q1, Q102 a current resonance type has one end of a primary winding N1 connected to the source-drain of switching elements Q11, Q12 via a resonance capacitor C100. A switching output is supplied by being connected to a connection point (switching output point) and being grounded to the primary side ground at the other end. Further, a series resonance circuit (L30, C30) is added as described above, and the series resonance current I01 is fed back to the power factor improvement circuit 20 with power.

The switching elements Q101 and Q102 are
A drive pulse having a required switching frequency is supplied from the drive / oscillation circuit 2 to the gate based on a trigger from the start circuit 3 to alternately turn on / off.
Performs off switching. Further, the control circuit 1 adjusts the pulse frequency (switching frequency) supplied from the drive / oscillation circuit 2 to the switching elements Q101 and Q201 so that the level can be varied according to the level of the DC voltage output EO on the secondary side. By controlling, constant voltage control is performed.

The power factor PF of the power supply circuit shown in FIG.
-AC input voltage VAC characteristics and power factor PF-Load power Po
The characteristics are shown in FIGS. FIGS. 13A and 13B
As can be seen from FIG.
= AC input voltage VAC for fluctuations of 113W to 47W =
In both the case of 100 V and the case of AC input voltage VAC = 230 V, the power factor PF is about 0.7 or more, which is sufficient. However, in this power supply circuit, the primary side series resonance current I02 and the power feedback series resonance current I01 are superimposed and flow through the switching elements Q101 and Q102, so that the switching loss of the switching elements Q101 and Q102Q1 increases and the power conversion efficiency increases. descend. Therefore, it can be said that this method can be applied only to a light load having a load power Po of about 100 W or less.

When the load power Po is a heavy load of 200 W or more, as shown in FIG. 14, a power feedback type power factor correction circuit is inserted into the AC line, and four sets of rectifier diodes (D101 to D1) are provided.
04) and two sets of electrolytic capacitors for smoothing (Ci1, Ci2)
And a doubler / full-wave rectifier circuit switching method using the electromagnetic power relay RY, the AC input voltage VAC is 100 V and 200 V.
V-system wide-range support is possible.

That is, in the power supply circuit of FIG. 14, in the power factor correction circuit 22, a filter choke coil LN is inserted in series with the positive input line of the commercial AC power supply AC, and is connected in parallel to the commercial AC power supply AC. A normal mode low-pass filter is formed together with the filter capacitor CN to prevent a harmonic current from flowing to the commercial AC power supply AC. In this case, two parallel resonance capacitors C31, C31,
C32 is provided, the parallel resonance capacitor C31 is inserted in parallel with the rectifier diode D101, and the parallel resonance capacitor C32 is inserted in parallel with the rectifier diode D102.
The parallel resonance capacitors C31 and C32 form a parallel resonance circuit together with, for example, the inductance of the filter choke coil LN, and the resonance frequency of the parallel resonance circuit is substantially the same as the resonance frequency of a series resonance circuit described later, for example. It is set as follows.

In this case, four rectifying diodes D101 to
As a bridge rectifier circuit using D104, a high-speed recovery type is used in response to a high-frequency current of a switching cycle flowing through a rectified current path along with a power factor improving operation as described later. The primary winding N1 of the insulated converter transformer PIT is connected to the connection point of the rectifier diodes D101 and D102 via the series resonance capacitor C100, and the switching of the switching converter (described later) obtained on the primary winding N1 is performed. Output is a series resonance capacitor C10
The signal is fed back to the rectified current path via the zero capacitive coupling.

In this power supply circuit, two smoothing capacitors Ci1 and Ci2 are connected in series and provided so as to be inserted between the positive output terminal of the bridge rectifier circuit and the primary side ground. The connection point of the smoothing capacitors Ci1 and Ci2 is connected via a switch S to the negative input terminal of the bridge rectifier circuit. The rectified and smoothed voltage obtained at both ends of the smoothing capacitors Ci1 and Ci2 connected in series is input to a later-stage separately excited current resonance type converter. The current resonance type converter performs a switching operation based on the input rectified smoothed voltage, and finally outputs a stabilized secondary-side DC output voltage Eo.

The switch S is provided for switching between a voltage doubler rectifying / smoothing operation and a normal rectifying / smoothing operation, and is controlled on / off by an electromagnetic relay RY. Electromagnetic relay RY
Are driven by the relay drive circuit 40. The relay drive circuit 40 includes a half-wave rectifier circuit including a rectifier diode D105 for half-wave rectifying the commercial AC power supply AC and a smoothing capacitor C33. A resistance is provided between the output of the half-wave rectifier circuit and the primary side ground. R1 and R2 are connected in series. A zener diode ZD is inserted between the voltage dividing points of the resistors R1 and R2 and the base of the transistor Q300. In this case, when the AC input voltage VAC supplied to the commercial AC power supply AC is equal to or higher than AC 150 V, the resistors R1, R
The Zener diode ZD depends on the voltage value divided by 2.
It is assumed that each of the above-mentioned components is selected so as to conduct. That is, a voltage detection circuit for detecting whether or not the AC input voltage level is equal to or higher than AC 150 V is formed by each of the above components. Transistor Q300 drives electromagnetic relay RY. A resistor R3 and a capacitor C34 are connected between the base of the transistor Q300 and the primary side ground. The collector of the transistor Q300 is grounded to the primary side ground. The emitter is an electromagnetic relay RY
Through a tertiary winding N3, a rectifier diode D300 and a smoothing capacitor C
It is connected to the line of the low-voltage DC voltage obtained by 101. A protection diode D5 for flowing a reverse current is connected in parallel to the electromagnetic relay RY.

For example, as an AC100V system, AC150
When the AC input voltage VAC equal to or lower than V is supplied, the Zener diode ZD does not conduct, so that the base current of the transistor Q300 is caused to flow through the resistor R3, and the transistor Q300 is turned on. Thereby, the emitter current is conducted to the electromagnetic relay RY. And the electromagnetic relay RY
The switch S is turned on by the exciting action of. Thus, the connection point between the smoothing capacitors Ci1 and Ci2 and the negative input terminal of the bridge rectifier circuit are connected via the switch S. In such a connection form, when the AC input voltage VAC is positive, a rectification current path for charging the smoothing capacitor Ci1 with the commercial AC power supply AC rectified by the rectifier diode D102 is formed.
A voltage across the level corresponding to the AC input voltage of the AC 100 V system is generated in the smoothing capacitor Ci1. On the other hand, when the AC input voltage VAC is negative, the rectifier diode D
The commercial AC power supply AC rectified at 101 is converted to a smoothing capacitor Ci.
By forming a rectified current path for charging the capacitor 2, a voltage across the level corresponding to the AC input voltage of the AC 100 V system is also generated in the smoothing capacitor Ci 2. Therefore, 100 is applied to both ends of the smoothing capacitors Ci1-Ci2 connected in series.
The double voltage rectifying and smoothing operation generates a rectified and smoothed voltage Ei having a level corresponding to almost twice that of the V system.

When an AC input voltage VAC of 150 V AC or more is supplied as an AC 200 V system, the Zener diode ZD is turned on, so that the base potential of the transistor Q 300 is raised to a predetermined value or more so that the base current does not flow. , The transistor Q300 is turned off. Therefore, the emitter current of the transistor Q300 does not flow through the electromagnetic relay RY, and the switch S is turned off. In this case, the commercial AC power supply A
C is the bridge rectifier circuit (rectifier diodes D101 to D1)
04) to perform full-wave rectification and smoothing capacitors Ci1-Ci
Thus, a full-wave rectification smoothing operation for charging the series connection of the two is obtained, and a rectified smoothed voltage Ei of an AC 200 V system corresponding to the AC input voltage VAC is obtained.

The switching converter shown in this figure is a separately excited current resonance type converter. In this current resonance type converter, for example, two switching elements Q101 and Q201 formed by MOS-FETs are provided in a half-bridge connection. These switching elements Q10
1, Q201 is alternately turned on / off by the oscillation drive circuit 2.
The switching drive is performed so that the OFF operation is repeated, and the rectified smoothed voltage Ei is intermittently turned into a switching output. Note that damper diodes D101 and D201 are provided in parallel with the direction shown in the drawing between the drain and source of each of the switching elements Q101 and Q201.

Sources of the switching elements Q101 and Q201
For the drain connection point (switching output point),
One end of a primary winding N1 of the insulated converter transformer PIT is connected to supply a switching output to the primary winding N1. The primary winding N1 of the insulating converter transformer PIT is connected in series with the series resonance capacitor C1.
And the leakage inductance component of the insulated converter transformer PIT including the primary winding N1,
A series resonance circuit for forming the switching power supply circuit into a current resonance type is formed.

The insulating converter transformer PIT transfers the alternating voltage obtained by the switching output supplied to the primary winding N1 to the secondary side. On the secondary side of the insulating converter transformer PIT, a DC output voltage Eo is obtained in the same manner as in the case of FIGS.

The starting circuit 3 is provided for detecting the voltage or current obtained on the rectifying / smoothing line immediately after the power is turned on, and for starting the oscillation drive circuit 2. The starting circuit 3 has an operating power supply. Tertiary winding N3 and rectifier diode D30 provided in the insulated converter transformer PIT
0 and the low-voltage DC voltage supplied by the smoothing capacitor C101.

As described above, the power factor improving circuit 22 returns the switching output obtained from the primary winding N1 of the insulating converter transformer PIT to the rectified current path via the capacitive coupling of the series resonance capacitor C100. Has been. The switching output thus fed back acts to superimpose an alternating voltage (switching voltage) of a switching cycle on the rectified output voltage via the inductance of the filter choke coil LN. Rectifier diode D
101 and D102 operate so that the rectified current is interrupted in the switching cycle. Since the rectifier diodes D101 and D102 are in the path of the rectified current in both the case of voltage doubler rectification and the case of full wave rectification, the above-described operation is performed in both the case of voltage doubler rectification and the case of full wave rectification. Will be

By this operation, for example, during the voltage doubler rectification operation, the rectified output voltage is charged to the smoothing capacitors Ci1 and Ci2 in a state where the switching voltage is superimposed. The voltages at both ends of Ci1 and Ci2 are reduced in the switching cycle. For this reason, the smoothing capacitors Ci1 and C2 also operate during the period in which the rectified output voltage level is lower than the voltage across each of the smoothing capacitors Ci1 and Ci2.
The charging current to i2 is caused to flow. In the full-wave rectification operation, the rectified output voltage charges the smoothing capacitors Ci1 and Ci2 connected in series by the rectified output voltage on which the switching voltage is superimposed. The voltage (rectified smoothed voltage) across the smoothed capacitors Ci1-Ci2 is reduced in the switching cycle. For this reason,
Smoothing capacitor C with rectified output voltage level connected in series
The charging current is allowed to flow also in a period that is lower than the voltage between i1 and Ci2.

As a result, in either case of the voltage doubler rectifying operation or the full-wave rectifying operation, the average waveform of the AC input current IAC is made to approach the waveform of the AC input voltage VAC, and the AC input current IAC is reduced. The conduction angle will be enlarged. Thus, in the power supply circuit shown in this figure, the power factor is improved both in the voltage doubler rectification operation and in the full-wave rectification operation.

FIGS. 15A and 15B show a power factor PF-AC input voltage VAC characteristic and a power factor PF-load power Po characteristic of the power supply circuit of FIG. FIG. 15 (a)
As can be seen from (b), in the case of this power supply circuit, the AC input voltage VAC = 100 V and the AC input voltage VAC = 230 V with respect to the fluctuation of the load power PO = 250 W to 150 W.
In both cases, the power factor PF is sufficient.
However, in the circuit shown in FIG. 15, since the component elements for improving the power factor, such as the normal mode low-pass filters (LN, CN) and the parallel resonance capacitors C31 and C32, are provided in the AC input line, for example, worldwide It is necessary to adopt a safety certified product, which is disadvantageous in terms of cost. Also, the detection circuit of the AC input voltage VAC may malfunction due to an instantaneous power failure or disturbance noise, so that measures must be taken to prevent malfunction.

FIG. 16 shows four switching elements (Q41 to Q41).
Combining the full-bridge coupled current resonance type converter according to 44) and the magnetic coupling type power feedback type power factor correction circuit, a series resonance circuit of the primary inductance L30 and C100 of the magnetic coupling transformer (MCT) is added, and the secondary inductance L
Power is fed back via R.

In this power supply circuit, a separately-excited current resonance type converter in which four switching elements Q41 to Q44 are full-bridge-coupled is provided as a switching converter. Such a separately excited current resonance type converter by full bridge coupling is configured such that a set of switching elements Q41 and Q42 and a set of switching elements Q43 and Q44 are driven by push-pull. At this time, damper diodes DD41 and DD44 are provided between the drains and the sources of the switching elements Q41 to Q44.

In this case, drive circuit 2B outputs a switching drive signal to each gate of switching elements Q41 and Q42, and drive circuit 2C outputs a switching drive signal to each gate of switching elements Q43 and Q44. It is configured as follows. In this case, the drive circuits 2B and 2C alternately switch the set of [switching elements Q41, Q44] and [switching elements Q42, Q43] at a required switching frequency based on the oscillation outputs of the opposite phases from the oscillation circuit 2A. A switching drive signal is output so as to perform an on / off switching operation.

When the AC input voltage VAC is detected to be 150 V or more by the VAC detection circuit 2D for detecting the charging voltage of the smoothing capacitor Ci, the VAC detection circuit 2D
If the switching element Q43 is turned off and Q44 is turned on under the control of D, the half-bridge coupled current resonance type converter operates. That is, a full-bridge / half-bridge coupling switching system is used to enable a wide range.

The power factor improving circuit 23 has a magnetic coupling transformer MCT corresponding to the fact that the current resonance type converter employs a full bridge coupling type circuit form, and is supplied to the series resonance circuit by the magnetic coupling action. The switching output is fed back to the rectified current path. In the power factor correction circuit 23, a filter choke coil LN, a high-speed recovery type diode D1, and a winding NR are connected in series between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. Although the filter capacitor CN is provided in parallel with the series connection circuit of the filter choke coil LN and the winding NR, a low-pass filter is formed together with the filter choke coil LN by such a connection form.

The magnetic coupling transformer is formed by winding the MCT so that the winding NR and the winding N30 are magnetically tightly coupled to the core. One end of the winding N30 is connected to the connection point of the switching elements Q41 and Q42, and the other end is connected to the switching element Q43,
Connected to the connection point of Q44. This connection point is connected to one end of the primary winding N1 via the series resonance capacitor C1. With this connection form, switching elements Q41 to Q44
Is supplied to a series resonance circuit on the primary winding N1 side, and a switching output is also obtained for a winding N30 connected in series to the series resonance circuit. Become.

The power factor improving circuit 2 thus formed
In 3, the switching output obtained at the winding N30 is transmitted to the winding LR through the magnetic coupling action of the magnetic coupling transformer MCT. As a result, a voltage of the switching period (switching voltage) is generated in the winding NR. However, since the winding NR is inserted in the rectification current path, the operation is performed such that the switching voltage is superimposed on the rectification output voltage. I do. Then, the superimposed portion of the switching output voltage causes the high-speed recovery type diode D1 to operate such that the rectified current is intermittently switched in the switching cycle, thereby increasing the conduction angle of the AC input current and improving the power factor. .

FIG. 17 shows a power factor PF-AC input voltage VAC characteristic of the power supply circuit of FIG. This FIG.
As can be seen from FIG.
= 192 W to 84 W, a power factor PF that clears the regulation value is obtained. However, there are disadvantages that the circuit configuration becomes complicated, the number of components increases, and the printed circuit board mounting area increases.

[0044]

Various power supply circuit examples have been described with reference to FIGS. 8 to 17. The problems of these conventional power supply circuits are summarized as follows.

When an LC series resonance circuit is provided in parallel with the series resonance circuit of the half-bridge coupled current resonance type converter and combined with the power feedback type power factor correction circuit, the switching current of the two switching elements increases and the power conversion Efficiency decreases. The load power satisfying the harmonic distortion regulation is about 113 W to 47 W. -In the case of the half-bridge coupled current resonance type converter with AC input voltage rectification switching method, the maximum load power is improved to about 250W.
The components are connected to the line, and it is necessary to select a product approved by safety standards, and it is necessary to take measures against malfunction of the rectification method switching circuit due to momentary power failure or disturbance noise in the case of AC200V system.・ AC full bridge coupled current resonance type converter
In the case of a full-bridge or half-bridge switching system in which a 100 V system and a half-bridge coupled current resonance type converter are operated on an AC 200 V system, the drive stages of the switch elements are two sets and the circuit configuration is complicated. Therefore, the number of components is large and the board area is increased.

[0046]

In view of the above-mentioned problems, the present invention is configured as a switching power supply circuit as follows. That is, the switching power supply circuit of the present invention rectifies the input commercial AC power and
Rectifying and smoothing means for dividing the smoothed voltage obtained at both ends of the two smoothing capacitors to output the first and second DC input voltages, and a gap for obtaining a required coupling coefficient which is loosely coupled. An isolated converter transformer formed and provided for transmitting the primary side output to the secondary side;
First and second switching means configured to intermittently output the first DC input voltage by a push-pull operation and output the first DC input voltage to a primary winding of the insulating converter transformer, and at least a primary winding of the insulating converter transformer; A first and second primary resonance circuit formed by the leakage inductance component and the capacitance of the primary-side parallel resonance capacitor to make the operation of the first and second switching means a voltage resonance type; The switching output that is inserted and obtained in one of the first and second primary-side resonance circuits is fed back via the primary-side parallel resonance capacitor in one of the primary-side resonance circuits. A power factor improving means for improving a power factor by intermittently commutating a rectified current based on an output; and a commercial AC power supply voltage. Accordingly, the first and second DC input voltages are intermittently driven by the divided voltage push-pull operation by the first and second switching means and output to the primary winding of the insulating converter transformer. A secondary-side resonance circuit formed on the secondary side by means of a switching operation switching means that can be performed, a leakage inductance component of a secondary winding of the insulating converter transformer, and a capacitance of a secondary-side resonance capacitor; DC output voltage generation means formed to include a circuit and configured to generate a secondary side DC output voltage by inputting an alternating voltage obtained in a secondary winding of the insulating converter transformer and performing a rectification operation, Constant voltage control configured to perform constant voltage control on the secondary DC output voltage according to the level of the secondary DC output voltage To make and a stage.

Further, the power factor improving means is connected in parallel with a high-speed recovery type diode for interrupting the rectified current and a capacitor, and is connected to one of the first and second primary side resonance circuits. A primary side parallel resonance capacitor is connected to a connection point between the high speed recovery type diode and the capacitor. Alternatively, in the power factor improving means, a high-speed recovery diode for interrupting a rectified current and an inductance are connected in series, and one of the first and second primary side resonance circuits is provided. A primary side parallel resonance capacitor is connected to a connection point between the high speed recovery type diode and the inductance.

One of the first and second primary-side resonance circuits has a primary-side parallel resonance capacitor, which is connected to the first and second primary-side resonance capacitors.
And a switching output of one of the first and second switching means is fed back to the power factor improving means via a connection point of the first and second capacitors. To do.

According to the above configuration, the switching output voltage obtained in the primary side resonance circuit is electrostatically coupled to the power factor improvement circuit provided in the power supply circuit called a push-pull type switching frequency control type composite resonance type converter. In addition, when the AC voltage is 150 V or more, for example, the operation is switched to the divided voltage push-pull type operation by the above-mentioned switching operation switching means so that the input operation can be performed over a wide range of input voltages. The power factor can be kept constant between the AC100V system and the AC200V system.

[0050]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to an embodiment of the present invention.
The switching power supply circuit of FIG. 1 is provided with a voltage resonance type converter having two switching elements Q1 and Q2 and performing a switching operation by a so-called push-pull method by a self-excited method. A power factor improving circuit 10 is provided for the voltage resonance type converter.

The power supply circuit shown in this figure has, for example, an AC input voltage Vac = 150 V or more (so-called AC 200 V).
System), the push-pull operation by the switching elements Q1 and Q2 can be switched to a so-called partial voltage push-pull operation. Therefore, two smoothing capacitors Ci1 and Ci2 are used as a smoothing capacitor for obtaining a rectified smoothed voltage.
Is provided. These smoothing capacitors Ci1 and Ci2 are:
As shown in the drawing, a positive output line of the bridge rectifier circuit Di and a primary side ground are connected in series.

Switches S1 and S2 are provided, and the connection point of the smoothing capacitors Ci1 and Ci2 is t
Connected to terminal a. The tb terminal of the switch S1 is connected to the primary side ground. The switch S1 is a switch S2
, And the tb terminal of the switch S2 is connected to the positive electrode of the smoothing capacitor Ci1.

The switches S1 and S2 are linked by an electromagnetic relay RY.
By being connected to the terminals, the switching elements Q1, Q2
, The switching operation of the voltage division push-pull method is performed, while the switches S1 and S2
By connecting to the terminal b, the switching elements Q1, Q
2 to perform a push-pull switching operation.

In the power supply circuit shown in this figure, a bridge rectification circuit Di is used as a rectification smoothing circuit for receiving a commercial AC power supply (AC input voltage VAC) and obtaining a DC input voltage.
And a full-wave rectifier circuit composed of a smoothing capacitor Ci, and generates a rectified smoothed voltage Ei corresponding to a level that is one time the AC input voltage VAC. Further, in the rectifying / smoothing circuit, an inrush current limiting resistor Ri is inserted into the rectifying current path so as to suppress, for example, an inrush current flowing into the smoothing capacitor when the power is turned on.

One end of the primary winding N1A of the insulating converter transformer PIT is connected to the positive electrode side of the smoothing capacitor Ci1 via the detection winding ND of the orthogonal drive transformer PRT and the choke coil CH1. To enable the push-pull operation of the divided voltage, the switch S
When S1 and S2 are connected to the ta terminal, the primary winding N is connected to the connection point between the negative electrode of the smoothing capacitor Ci1 and the positive electrode of the smoothing capacitor Ci2 via the choke coil CH2.
One end of 1B is connected.

As this power supply circuit, two sets of switching elements Q1 and Q2 are switched and driven by a push-pull operation (and a divided voltage push-pull operation), and the switching frequency of the switching elements Q1 and Q2 is variably controlled. An orthogonal drive transformer PRT is provided.

As the structure of the orthogonal drive transformer PRT, for example, a three-dimensional core is formed by joining the ends of the two magnetic legs of two double U-shaped cores having four magnetic legs. The detection winding ND and the drive windings NB1 and NB2 are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction, and the control winding NC is connected to the detection winding. By winding in the direction orthogonal to ND and the drive windings NB1 and NB2, a saturable reactor is formed.

In this case, the driving windings NB1 and NB2 are
As shown in the figure, they are wound independently on the orthogonal control transformer PRT. One end of the driving winding NB1 is grounded to the primary side ground when the switch S1 is connected to the tb terminal, and is connected to the smoothing capacitor Ci1-Ci2 when the switch S1 is connected to the ta terminal. Connected to. The other end of the drive winding NB1 is connected to the base of the switching element Q1 via the resonance capacitor CB1 and the base current limiting resistor RB1.

One end of the drive winding NB2 is grounded to the primary side ground, and the other end is connected to the base of the switching element Q2 via the resonance capacitor CB2-base current limiting resistor RB2. That is, a self-excited oscillation driving circuit for the switching element Q1 is formed by the driving winding NB1-the resonance capacitor CB1-the base current limiting resistor RB1.
The resonance capacitor CB2 and the base current limiting resistor RB2 form a self-excited oscillation drive circuit for the switching element Q2.

In the detection winding ND, an alternating voltage corresponding to a switching output is detected by a switching operation described later. In the drive windings NB1 and NB2, alternating voltages having opposite polarities different from each other by 180 ° can be obtained in accordance with the switching output detected by the detection winding ND.

Two switching elements Q1, provided for push-pull operation and voltage division push-pull operation,
Q2 has a high withstand voltage bipolar transistor (BJT;
Junction type transistor).

The drive circuit (drive winding NB1-base current limiting resistor RB1-resonance capacitor CB1), clamp diode DD1, and parallel resonance capacitor Cr1 are connected to the switching element Q1 as shown in FIG. Is connected to a drive circuit (drive winding NB2-base current limiting resistor RB2-resonance capacitor CB2), clamp diode DD2, and parallel resonance capacitor Cr2 as shown in the figure.

Here, the clamp diodes DD1, DD2
Are connected in parallel with each other between the base and collector of the switching elements Q1 and Q2. The parallel resonance capacitor Cr1 is connected between the collector and the emitter of the switching element Q1. However, the other parallel resonance capacitor Cr2 is disposed between the collector of the switching element Q2 and the anode side of the high-speed recovery type diode in the power factor correction circuit 10.

When the switch S1 is connected to the tb terminal, the emitter of the switching element Q1 is grounded to the primary side ground (during push-pull operation).
Is connected to the terminal ta, the smoothing capacitor Ci1
-Connected to the connection point of Ci2 (during divided voltage push-pull operation). Further, the emitter of the switching element Q2 is connected to the primary side ground.

The starting resistor Rs is connected between the positive electrode of the smoothing capacitor Ci1 and the switching element Q2. The starting resistor Rs is inserted to supply a starting current for starting a switching operation to the switching element Q2 at the time of starting.

The insulating converter transformer PIT for transmitting the switching outputs of the switching elements Q1 and Q2 to the secondary side is, as shown in FIG. 2, for example, E-type cores CR1 and CR2 made of ferrite material, as shown in FIG.
An EE-type core is provided so that the magnetic legs of the EE-type core are opposed to each other. A primary winding N1 (N1A, N1B) The secondary winding N2 is wound in a divided state. A gap G is formed for the center magnetic leg as shown in the figure. As a result, loose coupling with a required coupling coefficient can be obtained. Gap G is
The E-shaped cores CR1 and CR2 can be formed by forming the central magnetic legs shorter than the two outer magnetic legs. Also,
As the coupling coefficient k, for example, a loosely coupled state of k ≒ 0.85 is obtained, so that a saturated state is hardly obtained.

In this case, the insulation converter transformer PIT
Is divided into primary windings N1A and N1B, one end of the primary winding N1A is connected to the collector of the switching element Q1, the other end is connected in series with the inductance winding Lc1 of the choke coil CH1, and is orthogonal. Capacitor C via the detection winding ND of the type drive transformer PRT
It is connected to the positive electrode side of i1. One end of primary winding N1B is connected to the collector of switching element Q2. The other end of the primary winding N1B is connected to the smoothing capacitor Ci1 via the series connection of the inductance winding Lc2 of the choke coil CH2 when the switch S2 is connected to the tb terminal.
(During the push-pull operation), and when the switch S2 is connected to the terminal ta, it is connected to the positive electrode of the smoothing capacitor Ci2 via a series connection of the inductance winding Lc2 of the choke coil CH2. (During voltage division push-pull operation).

In this case, the above-described parallel resonance capacitor C
r1 is the leakage inductance component of the primary winding N1A (L1
A) and a combined inductance (L1A + Lc1) of the inductance winding Lc1 form a parallel resonance circuit for causing the switching element Q1 to operate in a voltage resonance type. Similarly, the parallel resonance capacitor Cr2 is a composite inductance (L1B + Lc) of the leakage inductance component (L1B) of the primary winding N1B and the inductance winding Lc2.
2) forms a parallel resonance circuit for turning the switching element Q2 into a voltage resonance type operation. Although detailed description is omitted here, the switching elements Q1, Q2
When each is turned off, the voltage across the resonance capacitors Cr1 and Cr2 is actually a sinusoidal pulse waveform by the action of these parallel resonance circuits, so that a voltage resonance type operation can be obtained.

In such a configuration on the primary side, the driving winding N
Since alternating voltages of opposite polarities are obtained in B1 and the driving winding NB2, the self-excited oscillation driving circuit including the driving winding NB1 and the self-excited oscillation driving circuit including the driving winding NB2 respectively have opposite polarities. , A drive current (base current) as an alternating current flows through each base of the switching elements Q1 and Q2. Thereby, the switching element Q1,
As for Q2, an operation of alternately turning on / off at a switching frequency determined by a constant of the self-excited oscillation driving circuit is obtained. That is, a switching operation by push-pull is obtained in a voltage resonance type. Switching element Q1
Is supplied to the primary winding N1A, and the switching output of the switching element Q2 is supplied to the primary winding N1B.

When the switches S1 and S2 are connected to the tb terminal and the switching elements Q1 and Q2 perform the push-pull operation, the switching elements Q1 and Q2 input the voltage across the smoothing capacitor Ci (Ci1 + Ci2) to perform the switching operation. Perform That is, the switching elements Q1, Q2
Are switched by inputting a DC input voltage Ei. When the switches S1 and S2 are connected to the terminal ta and the switching elements Q1 and Q2 perform a voltage division push-pull operation, the switching element Q1 performs a switching operation by inputting a voltage between both ends of the smoothing capacitor Ci1. The element Q2 is a smoothing capacitor Ci2
, The switching operation is performed by inputting the voltage between both ends. That is, the switching elements Q1 and Q2 are each switched by inputting a DC voltage of 1/2 Ei level.

With such a configuration, when the AC input voltage is AC
The push-pull operation is performed for the 100 V system, and the divided voltage push-pull operation is performed for the AC 200 V system.
Even under the condition of the 00V system, the switching operation by push-pull can be performed under the same conditions as those of the AC100V system.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. With this parallel resonance circuit,
The alternating voltage excited by the secondary winding N2 becomes a resonance voltage.
That is, a voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, the primary side is provided with a parallel resonance circuit for making the switching operation a voltage resonance type, and the secondary side is also provided with a parallel resonance circuit for obtaining the voltage resonance operation. . In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

In this case, for the secondary parallel resonance circuit formed as described above, a tap is provided for the secondary winding N2, and then the rectifier diodes DO1, DO2, DO are provided.
3, DO4 and the smoothing capacitors CO1, CO2 are connected as shown in the figure, so that a set of [rectifier diodes DO1, DO2, smoothing capacitor CO1] and a set of [rectifier diodes DO3, DO4, smoothing capacitor CO2] are obtained. Two sets of full wave rectifier circuits are provided. The full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] has a DC output voltage EO1.
And a full-wave rectifier circuit composed of [rectifier diodes DO3, DO4, smoothing capacitor CO2] generates a DC output voltage EO2. In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1.

The control circuit 1 supplies a DC current whose level varies in accordance with the level of the DC voltage output EO1 on the secondary side to the control winding NC of the drive transformer PRT as a control current, for example. The constant voltage control is performed as follows.

By the way, the insulation converter transformer PIT
, The polarity (winding direction) of the primary winding N1 and the secondary winding N2 and the rectifier diode DO (DO1, DO2, DO3, DO4)
, The mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 may be + M or -M. For example, when the connection configuration shown in FIG. 3A is adopted, the mutual inductance is + M (additive polarity: forward method), and when the connection configuration shown in FIG. 3B is employed, the mutual inductance is −M (depolarization: Flyback method). If this is made to correspond to the operation of the secondary side of the power supply circuit shown in FIG. 1, for example, when the alternating voltage obtained in the secondary winding N2 has a positive polarity, the rectifier diode DO1 (DO3)
Can be regarded as an operation mode of + M (forward system). Conversely, when the alternating voltage obtained in the secondary winding N2 has a negative polarity, the rectifier diode DO2
The operation in which the rectified current flows through (DO4) can be regarded as the -M operation mode (flyback method). That is, in this power supply circuit, each time the alternating voltage obtained in the secondary winding becomes positive / negative, the mutual inductance becomes + M / -M
Mode.

In the control circuit 1, the level of the control current (DC current) flowing through the control winding NC is varied according to the change in the secondary DC output voltage level (EO1), so that the winding is wound around the orthogonal control transformer PRT. The inductances LB1 and LB2 of the mounted drive windings NB1 and NB2 are variably controlled. Thereby, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching elements Q1 and Q2 formed including the inductances LB1 and LB2 of the drive windings NB1 and NB2 changes. This is an operation of changing the switching frequency of the switching elements Q1 and Q2. This operation has an effect of stabilizing the secondary-side DC output voltage.

Next, the configuration of the power factor improving circuit 10 will be described. In the power factor correction circuit 10, a choke coil Ls and a high-speed recovery type diode D1 are inserted in series between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. The filter capacitor CN is provided in parallel with the series connection circuit of the choke coil Ls and the high-speed recovery type diode D1, thereby forming a normal mode low-pass filter together with the choke coil Ls.

The parallel resonance capacitor C3 is provided in parallel with the high speed recovery type diode D1. Although detailed description is omitted here, for example, the parallel resonance capacitor C3 forms a parallel resonance circuit together with, for example, the choke coil Ls. This has the effect of suppressing an increase in the rectified smoothed voltage Ei when the load is reduced. In the power factor correction circuit 10, the above-described parallel resonance capacitor Cr2 is connected to a connection point of the choke coil Ls, the anode of the high-speed recovery type diode D1, and the parallel resonance capacitor C3, and switching is performed. The switching output (voltage resonance pulse voltage) obtained in the primary parallel resonance circuit on the element Q2 side is fed back.

The power factor improving operation by the power factor improving circuit 10 is basically as follows. In the configuration of the power factor correction circuit 10 shown in this figure, the switching output obtained in the primary side parallel resonance circuit on the switching element Q2 side is fed back to the rectified current path via the capacitive coupling of the parallel resonance capacitor Cr2. Become.

The switching output fed back in this way causes the alternating voltage of the switching cycle to be superimposed on the rectified current path. An operation of interrupting the rectified current at the switching cycle is obtained, and the apparent inductance of the choke coil Ls also increases due to the interrupting action. In addition, a voltage is generated at both ends of the parallel resonance capacitor C3 when a current of a switching period flows through the capacitor.
The level of the rectified smoothed voltage Ei is reduced by the voltage across the parallel resonance capacitor C3. This allows the charging current to flow to the smoothing capacitor Ci even during a period in which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci (Ci is a generic term for Ci1 + Ci2; the same applies hereinafter). Is done. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, and the conduction angle of the AC input current is increased. As a result, the power factor is improved.

As described above, the parallel resonance capacitor Cr2 is connected to the cathode of the high-speed recovery type diode D1 of the power factor correction circuit 10. This means that the parallel resonance capacitor Cr2 and the parallel resonance capacitor C3 are connected in series, that is, the parallel resonance capacitor Cr2
Is divided by the capacitance ratio of the parallel resonance capacitor Cr2 and the parallel resonance capacitor C3. Then, a circuit system is formed as a voltage feedback system in which the voltage is fed back to the smoothing capacitor Ci via the parallel resonance capacitor C3 connected in parallel with the high-speed recovery type diode D1.

The capacitance of the parallel resonance capacitors Cr1, Cr2 is Cr1 <Cr2, and the capacitance of the parallel resonance capacitor C3 is sufficiently larger than Cr2. In particular, when the capacitance of the capacitor C3 is increased, the power factor PF is improved. That is, the switching frequency fs is controlled to be high during the period when the AC input voltage VAC is high, and the switching frequency fs is controlled to be low during the period when the AC input voltage VAC is low. The resonance pulse voltage is not fed back to the power factor correction circuit 10, and the AC input current IAC from the AC power supply AC is changed from the bridge rectifier circuit Di to the choke coil L.
s → The smoothing capacitor Ci is charged via the high-speed recovery type diode D1. Then, as the AC input voltage VAC decreases, the power factor improving circuit 10 of the voltage resonance pulse voltage increases.
The amount of return to is increased. The switching elements Q1, Q
2, the peak value of the voltage resonance pulse voltage generated at each collector is equal to the pulse width at the time of off.

In this example, as described above, the switch S
1, the push-pull operation and the divided voltage push-pull operation are switched by S2. And switches S1, S2
Are switched according to the AC input voltage value, so that a push-pull operation is performed in the case of the AC input voltage 100 V system, and a divided voltage push-pull operation is performed in the case of the AC input voltage 200 V system. That is, in this example, the relay drive circuit 4, the electromagnetic relay RY, and the switches S1 and S2 form a switching operation switching circuit that switches between the push-pull operation and the divided voltage push-pull operation according to the commercial AC power supply voltage.

As described above, the switches S1 and S2 are on / off controlled in conjunction with each other by the electromagnetic relay RY. The electromagnetic relay RY is driven by the relay drive circuit 4. The relay drive circuit 4 is configured by connecting resistors R1 to R3, a thyristor Q11, a Zener diode ZD, a capacitor C5, and a diode D5 as shown in the figure.

In this relay drive circuit 4, a resistor R is connected between the positive electrode of the smoothing capacitor Ci1 and the primary side ground.
1, R2 are connected in series. And these resistors R1, R
Zener diode Z between the voltage dividing point 2 and thyristor Q11
D is inserted. In this case, when the rectified and smoothed voltage appearing in the smoothing capacitor Ci is 150 V AC or more, the above components are selected so that the Zener diode ZD conducts by the voltage value divided by the resistors R1 and R2. You. That is, a voltage detection circuit for detecting whether or not the AC input voltage level is equal to or higher than AC 150 V is formed by each of the above components. Then, the electromagnetic relay RY is driven by turning on / off the thyristor Q11. Electromagnetic relay RY
Protection diode D for flowing a reverse current
5 are connected in parallel.

As an operating power supply for operating the relay drive circuit 4, a tertiary winding N5 is wound around the insulating converter transformer PIT, and the alternating voltage excited by the tertiary winding N5 is rectified. A DC voltage obtained by rectification by a half-wave rectification circuit including a diode D6 and a capacitor C6 is used.

When an AC input voltage VAC of AC 150 V or less is supplied by the relay drive circuit 4 as a 100 V AC system, for example, the Zener diode Z
Since D does not conduct, thyristor Q11 is turned off. At this time, the switches S1 and S2 switched by the electromagnetic relay RY are in a state of being connected to the tb terminal side, respectively. That is, the state becomes the push-pull operation state.

On the other hand, when the AC 200 V
When the AC input voltage VAC is supplied, the Zener diode ZD is turned on and the thyristor Q11 is turned on, so that the switches S1 and S2 are respectively switched to the ta terminal side by the exciting action of the electromagnetic relay RY. The divided voltage push-pull operation state is set.

FIG. 4A shows operation waveforms of the AC input current IAC at the time of the maximum load power and the minimum load power of the switching power supply circuit when the AC input voltage VAC is 100 V, and FIG. 4B shows operation waveforms of the AC input current IAC at the time of the maximum load power and the minimum load power when the AC input voltage VAC is 230 V.

In the switching power supply circuit of this example, the filter capacitor CN = 1 μF and the choke coil LS = 7
5 μH, parallel resonance capacitor Cr1 = 4700 pF, parallel resonance capacitor Cr2 = 5600 pF, parallel resonance capacitor C3 = 0.022 μF, and maximum load power P
The experiment was performed under the input / output conditions of Omax = 250 W, minimum load power POmin = 0 W, and AC input voltage VAC = 80 V to 288 V. As a result, FIGS.
Are obtained. FIG. 5 shows load power PO = 0 W to 2
Shows the characteristics of the power factor PF with respect to a load change of 50 W,
FIG. 6 shows the characteristics of the power factor PF with respect to the fluctuation in the range of the AC input voltage VAC = 80 V to 280 V. FIG. 6
The vicinity of 150 V indicated by a broken line in FIG. 7 is a boundary where the push-pull operation and the divided voltage push-pull operation are switched.

As can be seen from FIGS. 5 and 6, in the switching power supply circuit of the present embodiment, the load power PO ranges from 250 W to 50 W.
For a load change of W, the power factor PF is 0.89 to 0.8.
81, and AC input voltage VAC = 80V-2
The load power PO = 250 even in the range of 80V.
Under each condition of W to 50 W, a sufficient power factor (PF = 0.8 or more) was obtained.

That is, according to the present embodiment, the power factor PF becomes very stable with respect to changes in the load power PO and the AC input voltage VAC, as compared with the conventional power feedback system. Further, the fluctuation range of the load power PO is greatly expanded. Since the high power factor can be maintained even when the AC input voltage and the load fluctuate in this manner, the present invention is not limited to a television receiver or the like in which the AC input voltage and the load condition are specified, and for example, office equipment in which the load condition fluctuates. It is practically possible to mount the power supply circuit according to the present embodiment on office equipment such as personal computers and personal computers. Further, the detection of the AC input voltage VAC for switching between the push-pull operation and the divided voltage push-pull operation is performed from the DC input voltage appearing on the positive electrode of the smoothing capacitor Ci, so that there is no erroneous detection due to instantaneous power failure or disturbance noise. Therefore, the switching operation of the push-pull operation / minute voltage push-pull operation can be reliably performed without malfunction.

Next, a second embodiment of the present invention will be described with reference to FIG. FIG. 7 is a circuit diagram showing a configuration of a power supply circuit according to a second embodiment of the present invention. In this figure, the same parts as those in FIG.

The power supply circuit shown in FIG. 7 is also provided with a voltage resonance type converter having two switching elements Q1 and Q2 as in FIG. 1 and performing a switching operation by a so-called push-pull system by a self-excited system. You. Further, as in FIG. 1, the relay driving circuit 4 and the electromagnetic relay R
A push-pull operation and a divided voltage push-pull operation are switched by the operation of the switching operation switching circuit by Y and switches S1 and S2.

In the case of FIG. 7, the switching element Q2
Between the emitter and collector of the capacitor Cr21, C
r22 are connected in series, and the capacitors Cr21 and Cr22 connected in series function as parallel resonance capacitors.

Further, in the power factor improving circuit 11 shown in this figure, the power supply between the positive terminal of the bridge rectifier circuit Di and the positive terminals of the smoothing capacitors Ci (Ci1, Ci2) is provided.
The filter choke coil LN, the fast recovery type diode D1, and the choke coil LS are connected in series and inserted. The filter capacitor CN is inserted between the anode side of the high-speed recovery type diode D1 and the positive terminal of the smoothing capacitor Ci, so that the filter choke coil L
Together with N, a normal mode low-pass filter is formed.

For this power factor improving circuit 11,
Capacitors Cr21 and Cr connected in series that constitute the above-described parallel resonance capacitor are connected to the connection point between the cathode of the fast recovery type diode D1 and the choke coil LS.
The connection point 22 is connected so that the switching output (voltage resonance pulse voltage) obtained in the primary side parallel resonance circuit is fed back.

The power factor improving operation by the power factor improving circuit 11 is basically as follows. In the configuration of the power factor improving circuit 11 shown in this figure, the switching output supplied to the primary side parallel resonance circuit is fed back to the rectified current path via an inductive reactance (magnetic coupling) assumed to be included in the choke coil LS itself. To be.

The alternating voltage of the switching cycle is superimposed on the rectified current path by the switching output fed back as described above. The superimposed voltage of the switching cycle causes the high speed recovery type diode D1 to be superimposed. In this case, an operation of interrupting the rectified current in the switching cycle is obtained, and the apparent inductance of the filter choke coil LN and choke coil LS also increases due to the intermittent action. This allows the charging current to flow to the smoothing capacitor Ci even during a period in which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, and the conduction angle of the AC input current is increased. As a result, the power factor is improved.

Here, in this example, as described above, the parallel resonance capacitor is formed by connecting the capacitors Cr21 and Cr22 in series, and the capacitors Cr21 and Cr2 are connected in series.
The connection point 2 is connected to the cathode of the high-speed recovery type diode D1 of the power factor correction circuit 11. Accordingly, the voltage resonance pulse voltage that appears as the voltage across the resonance capacitors (Cr21, Cr22) is
A circuit is formed as a voltage feedback system in which voltage is divided by the capacitance ratio of r22 and fed back to the connection point between the high-speed recovery type diode D1 and the choke coil LS.

The capacitance of the capacitors Cr21 and Cr22 is such that Cr21 <Cr22. In particular, when the capacitance of the capacitor Cr22 is increased, the power factor PF is improved. That is, during the period when the AC input voltage VAC is high,
Since the switching frequency fs is controlled to be high and the switching frequency fs is controlled to be low during a period when the AC input voltage VAC is low, the voltage resonance pulse voltage is fed back to the power factor correction circuit 11 near the peak value of the AC input voltage VAC. Instead, the AC input current IAC from the AC power supply AC is charged to the smoothing capacitor Ci via the bridge rectifier circuit Di → the filter choke coil LN → the fast recovery type diode D1 → the choke coil LS. And the AC input voltage VAC
Becomes smaller, the amount of feedback of the voltage resonance pulse voltage to the power factor improving circuit 11 increases.

In such a power supply circuit, as in the example described with reference to FIG. 1, a high power factor can be maintained even when the AC input voltage and the load fluctuate.

On the secondary side of the power supply circuit shown in FIG. 7, one end of the secondary winding N2 is connected to the secondary side ground, and the other end is connected to a rectifier diode via a series connection of a series resonance capacitor Cs. Anode of DO1 and rectifier diode D
Connected to the O2 cathode connection point. The cathode of the rectifier diode DO1 is connected to the positive electrode of the smoothing capacitor CO1, and the anode of the rectifier diode DO2 is connected to the secondary side ground. The negative side of the smoothing capacitor CO1 is connected to the secondary side ground.

As a result, in such a connection form, a voltage doubler half-wave rectifier circuit comprising a set of [series resonance capacitor Cs, rectifier diodes D01, D02, smoothing capacitor C01] is provided. Here, the series resonance capacitor Cs
Forms a series resonance circuit corresponding to the on / off operation of the rectifier diodes DO1 and DO2 by its own capacitance and the leakage inductance component of the secondary winding N2. That is, the power supply circuit of the present embodiment is provided with a parallel resonance circuit on the primary side for performing a voltage resonance type switching operation, and a series resonance circuit for obtaining a voltage doubled half-wave rectification operation on the secondary side. A complex resonance type switching converter provided with a circuit is employed.

Here, the [series resonance capacitors Cs,
The double voltage half-wave rectification operation by the combination of the rectifier diodes DO1, DO2 and the smoothing capacitor CO1] is as follows. When a switching output is obtained on the primary winding N1 by the switching operation on the primary side, this switching output is excited by the secondary winding N2. The voltage doubler rectifier circuit performs a rectifying operation by inputting the obtained alternating voltage to the secondary winding N2. In this case, first, during the period when the rectifier diode DO1 is turned off and the rectifier diode DO2 is turned on, the primary winding N
The polarity (mutual inductance M) between 1 and the secondary winding N2 is-
M operates in the depolarization mode, and the operation of charging the rectified current rectified by the rectifier diode D02 to the series resonance capacitor Cs by the leakage inductance of the secondary winding N2 and the series resonance action of the series resonance capacitor Cs. can get.

During the period when the rectifier diode DO2 is turned off and the rectifier diode DO1 is turned on to perform the rectification operation, the polarity (mutual inductance M) of the primary winding N1 and the secondary winding N2 is + M. The operation is in the additive polarity mode, in which the smoothing capacitor C01 is charged in a state where series resonance occurs in which the potential of the series resonance capacitor Cs is added to the voltage induced in the secondary winding N2. As described above, the insulated converter transformer PIT operates in the positive polarity mode (+ M; forward operation) and the depolarization mode (-M).
M; flyback operation), the DC output voltage EO1 (rectified smoothed voltage) corresponding to almost twice the induced voltage of the secondary winding N2 in the smoothing capacitor CO1.
Is obtained. By performing the double voltage half-wave rectification in this way, the secondary side DC output voltage Eo1 is obtained.

In this embodiment, a secondary winding N2A is wound independently of the secondary winding N2, a center tap is grounded to the secondary winding N2A, and a rectifying diode A DC output voltage EO2 is generated by connecting a full-wave rectifier circuit composed of DO3, DO4 and a smoothing capacitor CO2.

Although the embodiments have been described above, the present invention may have various modifications. For example, the present applicant has already proposed a configuration provided with a quadruple voltage rectifier circuit using a secondary-side series resonance circuit as a composite resonance type switching converter, but such a configuration is also a modification of the present embodiment. It can be established as an example. That is, the present embodiment is not particularly limited as the configuration of the secondary-side resonance circuit and the rectifier circuit.

Further, a switching element for performing a switching operation is a Darlington-connected transistor or MOS.
The present invention is also applicable to a case of a separately excited oscillation type constituted by an FET and driven by an IC.

[0111]

As described above, according to the present invention, the switching output obtained in the primary side resonance circuit is different from the power factor improvement circuit provided in the power supply circuit called the push-pull type switching frequency control type composite resonance type converter. The voltage is fed back by the electrostatic coupling method or the magnetic coupling method, and the switching operation switching means switches between the push-pull operation and the divided voltage push-pull operation according to the AC input voltage. As a result, there is an effect that a wide range of input voltage can be supported, and a sufficient power factor can be maintained over a wide range with respect to fluctuations of the AC input voltage and the load power. For example, when the load is 200 W or more and the load power is 200 W or more and the load power is 200 W or more, the change of the power factor is small and the harmonic distortion regulation value is satisfied for a wide range of load fluctuation of 250 W to 50 W. It is suitable as a power factor improving power supply circuit for office equipment and information equipment having a large value.

Also, since there is little increase in the 50 Hz ripple voltage component of the DC output voltage, no special countermeasures against ripples are required, thereby improving the gain of the control circuit and increasing the capacity of the electrolytic capacitor. is there.

Further, since the maximum load power that can be handled by the power supply circuit does not decrease, it is not necessary to redesign the composite resonance type converter. In addition, the power conversion efficiency is slightly improved before the power factor is improved, and the operation waveform is a sine wave for both the voltage and the current.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a side sectional view showing a structure of an insulating converter transformer employed in the power supply circuit of the present embodiment.

FIG. 3 is an explanatory diagram showing each operation when the mutual inductance is + M / -M.

FIG. 4 is a waveform chart showing an operation of the switching power supply circuit according to the first embodiment.

FIG. 5 is a characteristic diagram illustrating a relationship between load power and a power factor for the switching power supply circuit according to the first embodiment.

FIG. 6 is a characteristic diagram illustrating a relationship between an AC input voltage and a power factor of the switching power supply circuit according to the first embodiment.

FIG. 7 is a circuit diagram illustrating a configuration of a switching power supply circuit according to a second embodiment.

FIG. 8 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 9 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 10 is a waveform chart showing an operation of the power supply circuit of FIGS. 8 and 9;

FIG. 11 is a characteristic diagram showing a relationship between an AC input voltage and a power factor of the power supply circuits of FIGS. 8 and 9;

FIG. 12 is a circuit diagram showing a configuration of a power supply circuit according to the prior art.

13 is a characteristic diagram showing a relationship between load power and a power factor and a relationship between an AC input voltage and a power factor for the power supply circuit of FIG.

FIG. 14 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

15 is a characteristic diagram showing a relationship between load power and a power factor and a relationship between an AC input voltage and a power factor for the power supply circuit of FIG. 14;

FIG. 16 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

17 is a characteristic diagram showing a relationship between an AC input voltage and a power factor for the power supply circuit of FIG.

[Explanation of symbols]

1 control circuit, 4 relay drive circuit, 4A switching operation switching circuit, 10, 11 power factor improvement circuit, Di bridge rectifier circuit, Ci smoothing capacitor, D1 high speed recovery type diode, Cr1, Cr2 parallel resonance capacitor, C3 parallel resonance capacitor, C2 secondary parallel resonance capacitor, Cs secondary series resonance capacitor, PRT
Quadrature control transformer, PIT isolation converter transformer, Q1, Q2, Q21, Q22 switching element, RY
Electromagnetic relay, S switch

──────────────────────────────────────────────────続 き Continued on the front page (51) Int.Cl. ⁷ Identification symbol FI Theme coat ゛ (Reference) H02M 3/337 H02M 3/337 C 7/06 7/06 P N

Claims (4)

[Claims] An input commercial AC power supply can be rectified and a first and a second DC input voltage can be output by dividing a smoothed voltage obtained between both ends of two smoothing capacitors connected in series. Rectifying / smoothing means, an insulating converter transformer having a gap formed so as to obtain a required coupling coefficient that is loosely coupled, and provided for transmitting a primary output to a secondary side, and the first DC input voltage First and second switching means which are intermittently output by a push-pull operation and output to the primary winding of the insulated converter transformer, and at least a leakage inductance component including the primary winding of the insulated converter transformer and a primary The operation of the first and second switching means is a voltage resonance type, formed by the capacitance of the side parallel resonance capacitor. A first and second primary-side resonance circuit, and a switching output that is inserted into a rectified current path and is obtained by one of the first and second primary-side resonance circuits. Power factor improving means for improving the power factor by intermittently commutating a rectified current based on the returned switching output; and A switching operation in which the first and second DC input voltages are intermittently output by the divided voltage push-pull operation by the first and second switching means and output to the primary winding of the insulating converter transformer; Switching means, the leakage inductance component of the secondary winding of the insulating converter transformer, and the capacitance of the secondary resonance capacitor. A secondary-side resonance circuit formed on the secondary side, and an alternating voltage formed on the secondary winding of the isolation converter transformer, which is formed including the secondary-side resonance circuit, performs rectification operation. DC output voltage generating means configured to generate a secondary side DC output voltage, and performing constant voltage control on the secondary side DC output voltage according to the level of the secondary side DC output voltage. And a constant voltage control means. 2. The power factor improving means is connected in parallel with a high-speed recovery type diode for interrupting a rectified current and a capacitor, and includes one of the first and second primary side resonance circuits. Primary parallel resonance capacitor,
2. The switching power supply circuit according to claim 1, wherein the switching power supply circuit is connected to a connection point between the fast recovery diode and the capacitor. 3. The power factor improving means includes a series connection of a high-speed recovery type diode for interrupting a rectified current and an inductance, and the first and second primary side resonance circuits. 2. The switching power supply circuit according to claim 1, wherein one of the primary parallel resonance capacitors is connected to a connection point between the high-speed recovery type diode and the inductance. 4. A primary parallel resonance capacitor of one of the first and second primary resonance circuits is formed by connecting a first and a second capacitor in series, and the first and second primary resonance circuits are connected in series.
2. A switching output of one of said switching means is fed back to said power factor improving means via a connection point of said first and second capacitors.
3. The switching power supply circuit according to item 1.