JP2001095247A - Switching power circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To eliminate the need for improving the breakdown voltage of a smoothing capacitor on primary side by suppressing a rise in DC input voltage at no-load. SOLUTION: The power-factor improved circuit of a composite resonance converter is formed, so that a switching pulse is fed back from a tertiary winding wound around an insulated converter transfomer via a series resonance capacitor CA. Although the feedback output permits rectified current to be interrupted by means of a high-speed recovery diode D2 to improve power-factor, and even while the high-speed recovery diode D2 is not being conducted due to the resonance operation of a series resonance circuit which includes the series resonance capacitor CA, resonance current is fed to charge a smoothing capacitor.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit having a power factor correction circuit.

[0002]

2. Description of the Related Art The applicant has previously proposed various power supply circuits having a resonance type converter on the primary side. Further, various power supply circuits including a power factor improving circuit for improving the power factor of the resonant converter have been proposed.

FIG. 6 is a circuit diagram showing an example of a switching power supply circuit constructed based on the invention previously filed by the present applicant. This power supply circuit has a configuration in which a power factor improving circuit for improving a power factor is provided for a self-excited voltage resonance type switching converter.

In the switching power supply circuit shown in FIG. 1, a common mode choke coil CMC and an across capacitor CL are provided as a noise filter for removing a common mode noise from the AC power supply AC.
The AC power supply AC is full-wave rectified by a bridge rectifier circuit Di composed of four diodes, and the rectified output is charged to a smoothing capacitor Ci via a power factor correction circuit 20. The configuration and operation of the power factor correction circuit 20 will be described later.

In FIG. 1, a voltage-resonant type switching converter is provided with, for example, a single switching element Q1 as a high withstand voltage bipolar transistor. That is, a so-called single-end system is used.
The base of the switching element Q1 is connected to the positive electrode side of the smoothing capacitor Ci via a starting resistor RS so that a base current at the time of starting can be obtained from a rectifying and smoothing line. The base of the switching element Q1 is also connected to the switching drive circuit 2. The switching drive circuit section 2 includes a self-excited oscillation drive circuit system for driving the switching element Q1 by a self-excited oscillation method, and a variable oscillation frequency (that is, a switching frequency) in the self-excited oscillation drive circuit system to make the voltage constant. And a switching frequency control system for achieving the above. As a specific configuration of such a switching drive circuit section 2, as seen in various power supply circuits applied for by the present applicant, for example, the self-excited oscillation drive circuit system includes a drive winding and a resonance capacitor. And a detection winding for transmitting an alternating voltage to the drive winding. The detection winding is, for example, not shown here, but is actually connected in series to the primary winding N1. That is, the switching element Q1 is switched by the resonance output of the resonance circuit of the self-excited oscillation drive circuit, and the resonance frequency is the switching frequency. The switching frequency control system employs a configuration for varying the resonance frequency. For this purpose, for example, a control transformer PR for varying the inductance of the drive winding is used.
T is provided. The control transformer PRT includes, for example, a transformer-coupled drive winding and a detection winding, and a control winding wound so that the winding directions of the drive winding and the detection winding are not the same. The structure is wound.
A DC control current output from the control circuit 1 is supplied to the control winding.

In the control circuit 1, the secondary side DC output voltage Eo
Is output to the control winding. In the control transformer PRT, the inductance of the drive winding is varied according to the level of the control current flowing through the control winding. If the inductance of the drive winding changes, the resonance frequency of the self-excited oscillation drive circuit system,
That is, the switching frequency is variably controlled. The constant voltage operation by such switching frequency control will be described later.

[0007] The collector of the switching element Q1 is connected to the positive terminal of the smoothing capacitor Ei via the primary winding N1 of the insulating converter transformer PRT, and the emitter is grounded. In this case, a clamp diode DD is connected between the collector and the emitter of the switching element Q1 to form a path for a damper current flowing when the switching element Q1 is turned off.

The first resonance capacitor Cr forms a parallel resonance circuit mainly with the leakage inductance of the primary winding N1 of the insulating converter transformer PIT together with the second resonance capacitor Cr1 in the power factor correction circuit 2 described later. ing. By the operation of the parallel resonance circuit, a voltage resonance type operation is obtained as the switching operation of the switching element Q1. Correspondingly, a sinusoidal pulse waveform is obtained as the voltage VCP between the collector and the emitter of the switching element Q1 during the period when the switching element is off.

The insulating converter transformer PIT for transmitting the switching output of the switching element Q1 to the secondary side is, as shown in FIG. 12, an E-shaped core CR1, CR2 made of, for example, a ferrite material.
An EE-type core is provided so that the magnetic legs of the EE-type core are opposed to each other. A primary winding N1 and a secondary winding N2 are attached to the center magnetic leg of the EE-type core by using a divided bobbin B. Each is wound in a divided state. A gap G is formed for the center magnetic leg as shown in the figure. As a result, loose coupling with a required coupling coefficient can be obtained. Gap G is E-type core CR
1 and CR2 can be formed by forming the center magnetic leg shorter than the two outer magnetic legs. The coupling coefficient k is set to a loosely coupled state of, for example, k ≒ 0.85, so that a saturated state is hardly obtained.

One end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1, and the other end is connected to the positive electrode (rectified smoothing voltage Ei) of the smoothing capacitor Ci.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. The alternating voltage excited by the secondary winding N2 by this parallel resonance circuit becomes a resonance voltage. That is, a voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, the primary side is provided with a parallel resonance circuit for performing a voltage resonance type switching operation, and the secondary side is also provided with a parallel resonance circuit for obtaining a voltage resonance operation in a rectifier circuit system. Is provided. In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

With respect to the secondary parallel resonance circuit formed as described above, a center tap is provided for the secondary winding N2, and a half-wave composed of a rectifier diode DO and a smoothing capacitor Co is provided. A rectifier circuit is provided. This half-wave rectifier circuit receives the resonance voltage supplied from the secondary parallel resonance circuit and outputs it as a DC output voltage EO.

By the way, the insulation converter transformer PIT
, The mutual inductance between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 depends on the relationship between the polarity (winding direction) of the primary winding N1 and the secondary winding N2 and the connection of the rectifier diode DO. About M, +
There are cases where it becomes M and cases where it becomes -M. For example, FIG.
When the connection form shown in FIG. 3A is adopted, the mutual inductance becomes + M (forward method), and when the connection form shown in FIG. 13B is adopted, the mutual inductance becomes -M (flyback method). If this is made to correspond to the operation of the secondary side of the power supply circuit shown in FIG.
The operation in which a rectified current flows through the rectifier diode DO when the alternating voltage obtained in 2 has a positive polarity can be regarded as a + M operation mode (forward mode). That is, the power supply circuit shown in FIG. 6 operates in the mode of the mutual inductance of + M (forward system) each time the alternating voltage obtained in the secondary winding becomes positive / negative.

In such a configuration, power is supplied to the load side increased by the action of the secondary side parallel resonance circuit.
As a result, the power supplied to the load side increases accordingly, and the rate of increase of the maximum load power also increases. In response to such a load condition, as described above with reference to FIG. 12, the gap G is formed in the insulated converter transformer PIT and loose coupling is performed by a required coupling coefficient, thereby further saturating. This is achieved by obtaining a difficult state. For example, when the gap G is not provided for the insulating converter transformer PIT, there is a high possibility that the insulating converter transformer PIT will be in a saturated state during flyback operation and the operation will be abnormal. It is difficult to want to be done.

In the circuit shown in FIG. 6, the switching frequency is varied for the purpose of constant voltage control. The switching frequency variable operation is such that the period during which the switching element Q1 is turned off is fixed and then the switching element Q1 is turned on. The operation of variably controlling the period of time is obtained. In other words, in this power supply circuit, as a constant voltage control operation, an operation is performed to variably control the switching frequency, thereby performing resonance impedance control on the switching output, and at the same time, controlling the conduction angle of the switching element in the switching cycle (PWM). Control). This complex control operation is realized by a set of control circuit systems. Here, as the switching frequency control, when the secondary output voltage increases due to, for example, a tendency toward light load, the control is performed so as to suppress the secondary output by increasing the switching frequency. It is supposed to be done.

The power factor is improved by a power factor improving circuit 20. In the power factor correction circuit 20 shown in the figure, a series connection circuit of a choke coil Ls and a high-speed recovery type diode D2 is inserted between the positive output of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci, as shown in the figure. Is done. The fast recovery diode D2 has an anode connected to the choke coil Ls and a cathode connected to the positive terminal side of the smoothing capacitor Ci. The choke coil Ls functions as a load of a switching output that is fed back as described later. A filter capacitor CN is connected in parallel to the series connection of the choke coil Ls and the high-speed recovery type diode D2. Here, a normal mode LC low-pass filter is formed by the choke coil Ls and the filter capacitor CN to prevent high frequency noise of the switching frequency from flowing into the AC line. The connection point (voltage division point) of the series connection of the first parallel resonance capacitor Cr and the second parallel resonance capacitor Cr1 is connected to the connection point of the choke coil Ls and the high-speed recovery type diode D2. .

Here, the first parallel resonance capacitor Cr = 8
200 pF, second parallel resonance capacitor Cr1 = 0.02
It is assumed that 7 μF, choke coil Ls = 75 μH, and filter capacitor CN = 1 μF are selected.

In the power factor improving circuit 20 having such a connection configuration, when the switching element Q1 is performing a switching operation, the resonance pulse voltage Vcp obtained when the switching element Q1 is turned off is equal to the first parallel resonance capacitor Cr-second parallel resonance. The voltage is divided by the series connection of the capacitor Cr1, and the divided voltage is applied so as to be fed back to the connection point between the choke coil Ls and the high-speed recovery type diode D2. For example, the resonance pulse voltage Vcp is 600 V
If p is obtained, a voltage of about 150 Vp, which is divided about 3: 1, is fed back to the connection point between the choke coil Ls and the high-speed recovery type diode D2.

Here, at the timing when the AC input voltage VAC is near the positive and negative peaks, the high-speed recovery type diode D
2 conducts. At this time, a steeply rising pulse waveform current flows from the output terminal of the bridge rectifier circuit Di via the choke coil Ls and the high-speed recovery type diode D2 to the smoothing capacitor Ci to perform charging. On the other hand, during a period other than when the AC input voltage VAC is near the positive or negative peak, the high-speed recovery diode D2 repeats the switching operation by the resonance pulse voltage fed back as the voltage V2 as described above. In this switching operation, the high-speed recovery type diode D
2 is turned off, the second parallel resonance capacitor Cr1, the choke coil Ls, and the filter capacitor CN
A parallel resonance current flows through the circuit consisting of the following, and at the timing of turning on, a high-frequency charging current flows from the AC input voltage VAC to the smoothing capacitor Ci via the choke coil Ls.
In this way, by using the primary-side voltage resonance pulse fed back to the rectification current path, the operation of alternating the current to be passed through the rectification current path is obtained by increasing the frequency, thereby conducting the AC input current IAC. The corners are enlarged to improve the power factor.

FIG. 7 shows, as characteristics of the power supply circuit having the configuration shown in FIG. 6, a change characteristic of a power factor and a DC input voltage (rectified smoothed voltage Ei) with respect to a load change. Further, in this figure, the characteristics (solid line) of the circuit provided with the power factor correction circuit 20 shown in FIG. 6 are compared with the characteristics of a circuit not provided with the power factor correction circuit 20 as the circuit shown in FIG. Is shown. According to this figure, when the load power Po is in the range of 0 W to 200 W, the configuration having the power factor improvement circuit 20 shown in FIG. 6 is more powerful than the circuit having no power factor improvement circuit as the power factor PF. It can also be seen that the power factor has improved. In particular, the circuit shown in FIG. 6 has a characteristic in which the power factor has a peak near the load power Po = 50 W. It can also be seen that the level of the rectified smoothed voltage Ei tends to increase as the load power Po decreases.

FIG. 8 shows the change characteristics of the power factor and the DC input voltage (rectified smoothed voltage Ei) with respect to the fluctuation of the AC input voltage VAC. This figure also shows, as a comparison, the characteristics (solid line) of the circuit provided with the power factor correction circuit 20 shown in FIG. 6 and the characteristics of the circuit not provided with the power factor correction circuit 20 as the circuit shown in FIG. . As shown in this diagram, the power factor is reduced in the circuit configuration in which the power factor is not improved as the AC input voltage VAC increases in the range of 80 V to 140 V, whereas the circuit shown in FIG. In addition to the improvement of the power factor PF, the characteristic that the power factor PF increases as the AC input voltage VAC increases is obtained. The rectified smoothed voltage E
i has a characteristic of increasing with an increase in the AC input voltage VAC.

Next, the circuit diagram of FIG. 9 shows another example of a switching power supply circuit constructed based on the invention previously filed by the present applicant. This power supply circuit also has a configuration in which a power factor improving circuit for improving a power factor is provided for a self-excited voltage resonance type switching converter. In this figure, the same parts as those in FIG.

The power supply circuit shown in FIG. 1 includes a power factor improving circuit 21. The power factor improving circuit 21 is different from the power factor improving circuit 20 shown in FIG. 6 in that the connection between the high-speed recovery type diode D2 and the choke coil Ls is reversed. That is, the anode of the fast recovery diode D2 is connected to the positive output terminal of the bridge rectifier circuit Di, and the cathode is connected to one end of the choke coil Ls. The other end of the choke coil Ls is connected to a positive terminal of the smoothing capacitor Ci. Then, the connection point between the high-speed recovery type diode D2 and the choke coil Ls is connected to the first parallel resonance capacitor Cr-second connection.
The connection configuration is such that the voltage resonance pulse Vcp divided by the parallel resonance capacitor Cr1 is applied.

Even in the case of such a configuration, at the timing when the AC input voltage VAC is near the positive and negative peaks,
The high-speed recovery type diode D2 conducts, and the high-speed recovery type diode D
Through the 2-choke coil Ls, a steeply rising pulse waveform current is charged in the smoothing capacitor Ci.

In this case, when the absolute value level of the AC input voltage VAC drops to a certain level, the high-speed recovery type diode D2 is temporarily turned off, and at this time, the second parallel resonance capacitor Cr1 // the choke coil Ls is formed. Voltage resonance is caused by the parallel resonance circuit. Due to this voltage resonance, a sine wave pulse voltage is superimposed on the cathode potential V2 (divided resonance pulse voltage) of the fast recovery diode D2. Then, the high speed recovery type diode D2 repeats the switching operation due to the potential difference between the cathode potential V2 and the potential V1 on the anode side of the high speed recovery type diode D2. In this switching operation, during the period in which the high-speed recovery type diode D2 is turned on, the filter capacitor CN is switched to the smoothing capacitor C2.
The charging current to i flows. By this operation, the conduction angle of the AC input current IAC is expanded, and the power factor is improved.

FIG. 10 shows the characteristics of the power supply circuit having the configuration shown in FIG. 9 described above, showing the power factor and the change characteristics of the DC input voltage (rectified smoothed voltage Ei) with respect to a load change. FIG. 11 shows the change characteristics of the power factor and the DC input voltage (rectified smoothed voltage Ei) with respect to the AC input voltage fluctuation. Note that, in these figures, the second
0.033μ for the constant of the parallel resonance capacitor Cr1
The characteristics are shown for a case where F is selected and a case where 0.043 μF is selected.

First, as can be seen from FIG.
As long as the load power Po is in a practical range of 50 W to 200 W, PF> 0.70 can be maintained. The rectified smoothed voltage Ei also tends to increase as the load power Po decreases. Further, according to the characteristic shown in FIG. 11, the power factor P varies with a change in the AC input voltage VAC = 80 V to 140 V.
It can be seen that F is maintained at 0.7 or more, and the rectified smoothed voltage Ei increases as the AC input voltage VAC increases.

The provision of the power factor improvement circuits 20 and 21 as shown in FIGS. 6 and 9 makes it possible to improve the power factor PF.
Nos. 0 and 21 are known to increase the ripple component superimposed on the DC input voltage (rectified smoothed voltage Ei) because the switching output is fed back to the rectified current path. For example, in the case of the circuit shown in FIG.
The ripple component ΔEi to be superimposed on the power factor correction circuit is ΔEi = 9.2 V.
Ei increases to 35.3V. In particular, when there is no load, ΔEi rises to about 26V. The same can be said for the power supply circuit having the configuration shown in FIG.

For example, taking the configuration shown in FIG. 9 as an example, if the first parallel resonance capacitor Cr = 8200 pF, the second parallel resonance capacitor Cr1 = 0.027 μF, and the choke coil Ls = 75 μH are selected, the load power Po = 2
Although the power factor PF can be maintained at 0.73 or more in the range of 5 W to 200 W, it increases to ΔEi = 31.8 V. Therefore, in the circuit shown in FIG. 9, the second parallel resonance capacitor Cr1 = 0.033 μF or Cr1 =
0.043 μF, the first parallel resonance capacitor Cr−
If the voltage division ratio by the second parallel resonance capacitor Cr1 is changed to adjust (reduce) the feedback amount of the voltage resonance pulse, Cr
When 1 = 0.033 μF, it decreases to ΔEi = 25.3 V, and when Cr 1 = 0.043 μF, ΔEi is further reduced
It decreases to Ei = 9.1V.

If the feedback amount of the voltage resonance pulse is reduced as described above, ΔEi can be suppressed.
However, when the feedback amount of the voltage resonance pulse is reduced, the power factor P
F decreases. For example, this characteristic is shown in FIGS.
The better power factor characteristics are obtained when Cr1 is 0.043 μF than when Cr1 is 0.033 μF. Therefore, for example, the circuit shown in FIG. 9 is adjusted so as to limit the ripple voltage ΔEi and obtain a power factor PF sufficient for practical use, with a limit of Cr 1 = 0.043 μF. The same can be said for the circuit shown in FIG.

[0032]

In the circuits shown in FIGS. 6 and 9, as shown in FIGS. 7 and 10, as the load power Po decreases, the DC input voltage ( As the level of the rectified and smoothed voltage Ei) increases, and particularly as the condition approaches a no-load condition, the above-described ripple ΔEi increases, and the rate of increase increases. This means that the voltage fluctuation rate with respect to the load fluctuation increases. For this reason, when adopting a configuration in which the withstand voltage of the smoothing capacitor Ci that generates the DC input voltage is 200 V when the power factor is not improved when the power factor is not improved when the configuration is adapted to the AC 100 V system, 250V
And must be. In addition, when the power factor correction is not performed, the voltage is 400 V when the power factor correction is not performed, and when the power factor correction is performed, the voltage is 500 V.
V. For this reason, one problem is that the size of the smoothing capacitor Ci increases, which hinders a reduction in circuit size and cost.

As the smoothing capacitor Ci, for example, an electrolytic capacitor is used. If a capacitor having a higher withstand voltage after making the capacitance of the electrolytic capacitor equal is selected, the amount of self-heating also increases because the equivalent internal resistance increases. To increase. As a result, the degree of deterioration of the electrolytic capacitor due to aging increases, and the reliability decreases accordingly.

[0034]

In view of the above-mentioned problems, the present invention is configured as a switching power supply circuit as follows. That is, rectifying and smoothing means for receiving a commercial AC power supply, generating a rectified smoothed voltage, and outputting the rectified smoothed voltage as a DC input voltage;
A gap is formed so as to obtain a required coupling coefficient that is loosely coupled, and an insulating converter transformer provided for transmitting a primary output to a secondary side and a DC input voltage obtained by intermittent switching by a switching element. A switching unit configured to output a switching output to a primary winding of the insulated converter transformer, and formed by at least a leakage inductance component including a primary winding of the insulated converter transformer and a capacitance of a parallel resonance capacitor; A primary-side parallel resonance circuit that operates the means in a voltage resonance type. In addition, a power factor improving unit configured to perform a power factor improving operation by returning the switching output obtained in the primary winding to a rectified current path is provided. And a secondary resonance circuit formed on the secondary side by a leakage inductance component of the secondary winding of the insulating converter transformer and a capacitance of the secondary resonance capacitor, and a secondary resonance circuit, DC output voltage generation means configured to input an alternating voltage obtained to the secondary winding of the insulating converter transformer and perform a rectification operation to generate a secondary DC output voltage, and a secondary DC output A constant-voltage control unit configured to perform constant-voltage control on the secondary-side DC output voltage in accordance with the voltage level. Then, the power factor improving means rectifies the tertiary winding wound around the insulating converter transformer by winding up the primary winding, and the switching output transmitted to the tertiary winding via its own capacitance. A resonance capacitor inserted so as to return to the current path, a switching element inserted into the rectified current path so as to perform a switching operation based on the returned switching output, and an inductor inserted into the rectified current path. , At least.

According to the above configuration, as a power factor improving circuit provided in a power supply circuit called a composite resonance type converter, a switching output obtained from a primary winding of an insulated converter transformer is transmitted to a tertiary winding. A configuration is employed in which feedback is made from the winding to the rectified current path via a resonance capacitor. In such a configuration, the switching output transmitted through the tertiary winding generates a period in which the resonance circuit formed by the resonance capacitor and the inductor in the power factor correction unit performs a resonance operation.

[0036]

FIG. 1 is a circuit diagram showing a configuration example of a power supply circuit according to an embodiment of the present invention. The power supply circuit shown in this figure is also provided with the insulating converter transformer PIT having the structure shown in FIG. 12 to form a composite resonance type converter. In this figure, the same parts as those in FIGS. 6 and 9 are denoted by the same reference numerals, and description thereof will be omitted.

In the power supply circuit shown in this figure, the tertiary winding N3 is connected to the primary side of the isolated converter transformer PIT.
Is wound. In this case, the tertiary winding N3 is provided so as to wind up the winding start side of the primary winding. The end of the tertiary winding N3 (the winding start end of the primary winding) is connected to the series resonance capacitor CA of the power factor correction circuit 10 described below.

The configuration of the power factor improving circuit 10 shown in FIG. The choke coil Ls and the fast recovery type diode D2 are connected in series to form a series connection circuit. And this series connection circuit is a bridge rectifier circuit Di.
And the positive electrode terminal of the smoothing capacitor Ci. In this case, the fast recovery diode D2
Has an anode connected to the positive terminal of the bridge rectifier circuit Di via the choke coil Ls, and a cathode connected to the positive terminal of the smoothing capacitor Ci. The filter capacitor CN is connected in parallel with the series connection of the choke coil Ls and the fast recovery diode D2. In this case, a series resonance capacitor CA is provided. This series resonance capacitor CA is inserted between the connection point of the choke coil Ls and the high-speed recovery type diode D2 and the end of the tertiary winding N3 (the winding start end of the primary winding). Then, for example, a series resonance circuit is formed by at least the capacitance of the series resonance capacitor CA and each inductance of the choke coil Ls and the tertiary winding N3.

In the configuration shown in FIG. 1, the primary winding N1 = 45T, the tertiary winding N3 = 4T, the series resonance capacitor CA = 0.027 μF, and the choke coil Ls = 1.
It is assumed that 3 μH has been selected.

In this case, the parallel resonance capacitor Cr
Is connected in parallel with the collector-emitter of the switching element Q1, and has a parallel resonance for making the switching operation a voltage resonance type by its own capacitance and the leakage inductance of the primary winding N1 of the insulating converter transformer PIT. Form a circuit.

According to such a configuration, the resonance pulse voltage Vcp generated while the switching element Q1 in the switching operation is off is transmitted to the tertiary winding N3 via the primary winding N1. Then, the alternating voltage as the resonance pulse voltage Vcp transmitted to the tertiary winding N3 is fed back to the connection point of the choke coil Ls and the high-speed recovery type diode D2 via the series resonance capacitor CA. .

FIG. 2 is a waveform diagram showing the operation of the main part of the power supply circuit having the structure shown in FIG. The operation shown in this figure is an operation when the AC input voltage VAC is 100 V and the maximum load power is 200 W.

Here, the frequency of the commercial power supply is 50 Hz, and the AC input voltage VAC has a sinusoidal waveform having a half cycle of 10 ms as shown in FIG. 2A.
Then, when a rectified current is output from the bridge rectifier circuit Di in response to the AC input current IAC flowing as shown in FIG. 2B, the high-speed recovery diode D2 performs a switching operation so as to interrupt the rectified current. I do.

In the case of the present embodiment, the high-speed recovery type diode D2 performs the switching operation only during a period of 5 ms in which the AC input voltage VAC is assumed to be high and the AC input current IAC flows. Said to be low
During a period during which the AC input current IAC does not flow, the fast recovery diode D2 does not perform a switching operation.
For this reason, the switching current ID flowing through the fast recovery diode D2 is as shown in FIG. Here, during the period when the AC input voltage VAC is assumed to be high,
The charging current flowing through the smoothing capacitor Ci is made to flow through the filter capacitor CN → the choke coil Ls → the high speed recovery type diode D2. by this,
As for the AC input current IAC shown in FIG. 2 (b), the conduction angle is actually enlarged and the power factor is improved. Further, in response to this operation, the cathode potential V2 of the fast recovery type diode D2 becomes an alternating voltage based on the envelope shown in FIG.

Here, as the operation of the high-speed recovery type diode D2, during the period in which the high-speed recovery type diode D2 is off, it is formed by the series resonance capacitor CA, the choke coil Ls, the filter capacitor CN, and the tertiary winding N3 (inductance L3). As a result of the resonance operation of the series resonance circuit being performed, the alternating current Ic flowing through the series resonance capacitor CA becomes a resonance current flowing through the path of the choke coil Ls → the filter capacitor CN → the tertiary winding N3. On the other hand, during the period when the high-speed recovery type diode D2 is turned on, the above-described resonance operation of the series resonance circuit is not performed, and the alternating current Ic becomes the high-speed recovery type diode D2.
Flows to the tertiary winding N3. In response to such an operation, the resonance current Ic flowing through the series resonance capacitor CA and the current IL flowing through the choke coil Ls are as shown in FIGS. 2E and 2F, respectively. The anode potential V1 of the high-speed recovery type diode D2 is shown in FIG.
The waveform shown in (c) is obtained. By such an operation, in the present embodiment, the voltage fluctuation characteristic of the rectified smoothed voltage Ei with respect to the load fluctuation is improved as described later.

Here, experimental results of the power supply circuit shown in FIG. 1 are shown in FIGS. In order to obtain the experimental results shown in these figures, the operating conditions were as follows: load power Po = 200 W to 0 W, AC input voltage VAC
= 85V to 144V. 3 and 4
For comparison, a circuit configuration including the power factor correction circuit 10 shown in FIG. 1 (with power factor correction) and a basic configuration (power factor correction circuit) without the power factor correction circuit 10 in the configuration shown in FIG. And none).

First, FIG. 3 shows that the AC input voltage VAC = 100
7 shows the relationship between the load and the power factor under the condition that V is constant. As shown in this figure, in the circuit with the power factor improvement, the load power Po = 50 compared to the circuit without the power factor improvement.
It can be seen that the power factor PF is maintained at 0.7 or more in the range of W to 200 W, and is greatly improved. Here, a characteristic is obtained in which the power factor PF increases as the load power Po increases.

Also, as compared with the prior art, the rectified smoothed voltage Ei with respect to the load power fluctuation is suppressed from rising steeply when there is no load. Further, in the circuit with the power factor improvement, the voltage is constantly higher than the circuit without the power factor improvement by about 5 V irrespective of the load power value. It is said that there is no need to select the one with the higher value. That is,
Regarding the smoothing capacitor Ci, a withstand voltage product equivalent to a circuit without power factor improvement can be selected even in a circuit with power factor improvement. Further, in the circuit having the power factor improvement as described above, the level of the rectified and smoothed voltage Ei is increased by about 5 V, so that the power conversion efficiency is improved in this case. It has been confirmed that the power conversion efficiency is improved by about 0.3%.

FIG. 4 shows the relationship between the power factor and the rectified smoothed voltage Ei level with respect to the AC input voltage change when the load power Po is constant at 200 W. According to this figure, in the circuit with the power factor improvement, the power factor PF is maintained at 0.8 or more in the range of the AC input voltage VAC = 85 V to 144 V with respect to the circuit without the power factor improvement. You can see that it is. Also in this figure, the rectified smoothed voltage Ei is steadily higher by about 5 V in the circuit with power factor improvement than in the circuit without power factor improvement regardless of the change in the AC input voltage VAC. It is shown.

FIG. 5 is a circuit diagram showing a configuration example of a power supply circuit according to another embodiment of the present invention. In this figure, the same parts as those in FIG. The power factor correction circuit 11 shown in FIG. 1 includes a bridge rectifier circuit DiF for full-wave rectification of the commercial AC power supply AC. 4 which forms this bridge rectifier circuit DiF
The diodes D3, D4, D5, and D6 employ a high-speed recovery type. The diodes D3, D4, D5, D6 function as switching elements for improving the power factor.

In response to the provision of the bridge rectifier circuit DiF in the power factor correction circuit as described above,
A choke coil Ls1 is inserted in series with the positive line of the commercial AC power supply AC, and a choke coil Ls2 is inserted in series with the negative line. The inductance values of the choke coils Ls1 and Ls2 are selected such that Ls1 = Ls2 = Ls / 2 as a relationship with the inductance value of the choke coil Ls provided in the power factor improvement circuit 10 of FIG. It is assumed that. In this case, the across capacitor CL functions in the power factor correction circuit 11 as, for example, the filter capacitor CN previously shown in FIG.

Further, CA1,
Two of CA2 are provided. The series resonance capacitor CA1 is inserted between the winding start end of the tertiary winding N3 and the positive input terminal of the bridge rectifier circuit DiF (the connection point between D3 and D5). Also, the other series resonance capacitor CA2
Is the winding start end of the tertiary winding N3 and the bridge rectifier circuit D
It is inserted between the negative input terminal of iF (the connection point of D4 and D6).

In such a configuration, the voltage resonance pulse Vcp generated when the switching element Q1 performing the switching operation is turned off is transmitted from the primary winding N1 to the tertiary winding N3, and the transmitted voltage resonance pulse is connected in series. The bridge rectifier circuit D is connected via the resonance capacitors CA1 and CA2, respectively.
Feedback is made to each line corresponding to the positive / negative side of the rectified current path including iF. Then, the diodes D3, D4, D5, and D6 forming the bridge rectifier circuit DiF perform the switching operation by the feedback switching output. By this operation, the conduction angle of the AC input current IAC is enlarged, and the power factor is improved. Also, as the circuit characteristics, the same characteristics as those in FIGS. 3 and 4 corresponding to the above embodiment can be obtained. Therefore, in this case as well, an electrolytic capacitor as the smoothing capacitor Ci having a higher breakdown voltage is selected. You don't have to.

When the circuit configuration shown in FIG. 5 is compared with the circuit configuration shown in FIG. 1, in the circuit shown in FIG. 5, the high-speed recovery type diode D2 and the filter capacitor CN are omitted. become. Here, if the bridge rectifier circuit DiF composed of the high-speed recovery type diodes D3, D4, D5, and D6 is modularized,
The number of parts can be reduced. In this case, since the high-speed recovery type diode D2 is omitted, the power loss due to the high-speed recovery type diode D2 is eliminated, and as a result, the power conversion is about 0.8% as compared with the circuit shown in FIG. The efficiency is improved. Also,
Accordingly, the input power is also reduced by about 1.8 W.

On the secondary side of the power supply circuit shown in this figure, one end of the secondary winding N2 is connected to the secondary side ground, and the other end is connected to the rectifier diode DO1 via the series connection of the series resonance capacitor Cs. Is connected to the connection point between the anode of the rectifier diode DO2 and the anode of the rectifier diode DO2. The cathode of the rectifier diode DO1 is connected to the positive electrode of the smoothing capacitor CO1, and the anode of the rectifier diode DO2 is connected to the secondary side ground. The negative side of the smoothing capacitor CO1 is connected to the secondary side ground.

As a result of such a connection form, a voltage doubler full-wave rectifier circuit comprising a set of [series resonance capacitor Cs, rectifier diodes DO1, DO2, smoothing capacitor CO1] is provided. Here, the series resonance capacitor Cs
Forms a series resonance circuit corresponding to the on / off operation of the rectifier diodes DO1 and DO2 by its own capacitance and the leakage inductance component of the secondary winding N2. That is, the power supply circuit of the present embodiment is provided with a parallel resonance circuit on the primary side for performing a voltage resonance type switching operation, and a series resonance circuit for obtaining a voltage doubler full-wave rectification operation on the secondary side. A complex resonance type switching converter provided with a circuit is employed.

Here, the above [series resonance capacitors Cs,
The voltage doubler full-wave rectification operation by the combination of the rectifier diodes DO1, DO2 and the smoothing capacitor CO1] is as follows. When a switching output is obtained on the primary winding N1 by the switching operation on the primary side, this switching output is excited by the secondary winding N2. During the period when the rectifier diode DO1 is turned off and the rectifier diode DO2 is turned on, the rectifier diode DO1 operates in the depolarization mode in which the polarity (mutual inductance M) of the primary winding N1 and the secondary winding N2 is -M. Rectifier diode DO2 due to the series resonance effect of the leakage inductance of the secondary winding N2 and the series resonance capacitor Cs.
Accordingly, an operation of charging the rectified rectified current to the series resonance capacitor Cs is obtained. During the period in which the rectifier diode DO2 is turned off and the rectifier diode DO1 is turned on and the rectifying operation is performed, the polarity mode (mutual inductance M) between the primary winding N1 and the secondary winding N2 is + M. Thus, the smoothing capacitor C01 is charged in a state where series resonance occurs in which the potential of the series resonance capacitor Cs is added to the voltage induced in the secondary winding N2. As described above, the additional polarity mode (+
M: forward operation) and the depolarization mode (-M: flyback operation) to perform the rectification operation, so that in the smoothing capacitor CO1, the induced voltage of the secondary winding N2 is substantially reduced. DC output voltage EO1 corresponding to double
Is obtained.

As described above, on the secondary side of the circuit shown in FIG. 5, by utilizing the state in which the mutual inductance is in the operation mode of + M and -M, the double-sided full-wave rectification is performed, so that the secondary-side DC output voltage is increased. I'm trying to get That is, since the electromagnetic energy due to the primary side current resonance action and the secondary side current resonance action is simultaneously supplied to the load side, the power supplied to the load side further increases accordingly, and the maximum load A large increase in power will be achieved.

Further, by obtaining the secondary side DC output voltage by the voltage doubler full-wave rectifier circuit, it is possible to obtain a level equivalent to the secondary side DC output voltage obtained by the unity voltage rectifier circuit, for example. In this case, the secondary winding N2 of the present embodiment requires only half the number of turns of the conventional winding.
This reduction in the number of turns leads to a reduction in the size and weight of the insulating converter transformer PIT and a reduction in cost.

Incidentally, the applicant of the present invention has a structure in which a full-wave rectification circuit is provided for a secondary-side parallel resonance circuit as a complex resonance type switching converter, and a quadruple voltage rectification circuit using a secondary-side series resonance circuit. Has already been proposed, but such a configuration can also be realized as a modification of the present embodiment. That is, the present embodiment is not particularly limited as the configuration of the secondary-side resonance circuit and the rectifier circuit.

In each of the above embodiments, a self-excited configuration is adopted as the primary-side voltage resonance type converter, but the present invention can be applied to a separately-excited configuration. is there. In this case, for example, an oscillation / drive circuit using an IC (integrated circuit) may be provided instead of the self-excited oscillation drive circuit, and the switching element of the voltage resonance type converter may be driven by the oscillation / drive circuit. In this case, as the constant voltage control, the drive signal waveform generated by the oscillation / drive circuit is variably controlled according to the secondary output voltage level. As the control, the drive signal waveform may be generated such that the period during which the switching element is turned off is constant, and the period during which the switching element is turned on is shortened in accordance with an increase in the secondary-side output voltage level. When such a separately excited configuration is employed, the orthogonal control transformer PRT is omitted.

In the case of employing the separately excited configuration as described above, a single bipolar transistor (BJT)
Instead of the switching element Q1
It is possible to employ a Darlington circuit in which a stone bipolar transistor (BJT) is connected in Darlington. Furthermore, one bipolar transistor (BJT)
MOS-FE instead of the switching element Q1
It is possible to use T (MOS field effect transistor; metal oxide semiconductor), IGBT (insulated gate bipolar transistor), or SIT (static induction thyristor), and these Darlington circuits or any of the above elements When is used as a switching element, it is possible to further increase the efficiency.

[0063]

As described above, the present invention provides a switching power supply in which a power factor improving circuit is provided for a composite resonant converter having a voltage resonance type converter on the primary side and a resonance circuit on the secondary side. A tertiary winding for transmitting the output of the primary winding to the insulating converter transformer is provided as a circuit. In addition, a resonance capacitor is provided in a path for returning the output of the tertiary winding to the rectification current path. Thus, the voltage resonance pulse obtained by the switching operation of the switching element is fed back from the tertiary winding to the rectified current path via the resonance capacitor. The switching element performs a switching operation of interrupting the rectified current by the switching output fed back to the rectified current path, thereby improving the power factor. In the present invention, a resonance circuit is formed by the resonance capacitor and the inductor in the power factor correction circuit. During a period when the switching element is not conducting, the resonance operation of the resonance circuit is obtained and the resonance operation is performed. The current is allowed to flow.

As a result of this operation, as a result, the DC input voltage (rectified smoothed voltage) at light load or no load
It is possible to suppress the rise of
As a smoothing capacitor for obtaining a DC input voltage, it is possible to select a withstand voltage product equivalent to a smoothing capacitor used in a circuit having no configuration for improving a power factor.
That is, it is not necessary to increase the withstand voltage of the smoothing capacitor in accordance with the provision of the power factor improving circuit. Thus, first,
Since the size of the smoothing capacitor cannot be increased, the circuit size does not increase. Also, an increase in cost can be avoided. In addition, since aging (deterioration) due to an increase in the withstand voltage of the smoothing capacitor is small, the reliability of the power supply circuit in this respect is also improved.

Further, according to the present invention, if a diode of a rectifier circuit for rectifying a commercial AC power supply is used as a switching element in the power factor correction circuit, the number of components for power factor improvement can be reduced. Therefore, a greater effect can be expected in terms of suppressing cost increase and avoiding an increase in circuit size. Furthermore, since it is not necessary to additionally provide a switching element in addition to the diode of the rectifier circuit, the power conversion efficiency can be improved accordingly.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to an embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of the switching power supply circuit shown in FIG.

FIG. 3 is a characteristic diagram showing a relationship between a power factor and a DC input voltage level with respect to a load power variation in the switching power supply circuit shown in FIG.

4 is a characteristic diagram showing a relationship between a power factor and a DC input voltage level with respect to an AC input voltage fluctuation in the switching power supply circuit shown in FIG.

FIG. 5 is a circuit diagram showing a configuration of a switching power supply circuit according to another embodiment.

FIG. 6 is a circuit diagram showing a configuration of a switching power supply circuit as a prior art.

FIG. 7 is a characteristic diagram illustrating a relationship between a power factor and a DC input voltage level with respect to a load power variation in the switching power supply circuit illustrated in FIG. 6;

8 is a characteristic diagram showing a relationship between a power factor and a DC input voltage level with respect to an AC input voltage change in the switching power supply circuit shown in FIG. 6;

FIG. 9 is a circuit diagram showing another configuration of a switching power supply circuit as a prior art.

FIG. 10 shows the switching power supply circuit shown in FIG.
FIG. 4 is a characteristic diagram illustrating a relationship between a power factor and a DC input voltage level with respect to a load power variation.

FIG. 11 shows the switching power supply circuit shown in FIG. 9;
FIG. 4 is a characteristic diagram illustrating a relationship between a power factor and a DC input voltage level with respect to AC input voltage fluctuation.

FIG. 12 is a side sectional view showing a structure of an insulating converter transformer employed in the power supply circuit of the present embodiment.

FIG. 13 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

[Explanation of symbols]

1 control circuit, 2 switching drive circuit section, 10, 1
1 Power factor improvement circuit, Di, DiF bridge rectifier circuit,
Ci smoothing capacitor, D2, D3, D4, D5, D6 fast recovery type diode, Cr parallel resonance capacitor,
CA, CA1, CA2 series resonance capacitor, C2 secondary parallel resonance capacitor, Cs secondary series resonance capacitor, P
IT isolation converter transformer, Q1 switching element, N3 tertiary winding

Claims (2)

[Claims] A rectifying / smoothing means for generating a rectified / smoothed voltage by inputting a commercial AC power supply and outputting the rectified / smoothed voltage as a DC input voltage; Converter for providing side output to the secondary side; and switching for outputting a switching output obtained by intermittently applying the DC input voltage by a switching element to a primary winding of the insulating converter transformer. A primary-side parallel resonance circuit formed by at least a leakage inductance component including a primary winding of the insulating converter transformer and a capacitance of a parallel resonance capacitor to make the operation of the switching means a voltage resonance type; Power factor improving means for performing an improving operation; and A secondary-side resonance circuit formed on the secondary side by a leakage inductance component and a capacitance of the secondary-side resonance capacitor; and a secondary-side resonance circuit formed including the secondary-side resonance circuit. DC output voltage generating means configured to input the obtained alternating voltage and perform a rectifying operation to generate a secondary DC output voltage, and a secondary DC output voltage according to the level of the secondary DC output voltage. Constant-voltage control means configured to perform constant-voltage control on the side DC output voltage, wherein the power factor improving means includes a tertiary coil wound around the insulating converter transformer by winding up the primary winding. Winding, the switching output transmitted from the primary winding to the tertiary winding,
A resonance capacitor inserted so as to return to the rectification current path via its own capacitance, a switching element inserted into the rectification current path so as to perform a switching operation based on the returned switching output, and a rectifier. A switching power supply circuit, comprising: at least an inductor inserted into a current path. 2. The switching element provided in the power factor improving means is a diode element forming a rectifying circuit for rectifying a commercial AC power supply in the rectifying and smoothing means. Item 2. The switching power supply circuit according to item 1.