JP2001112248A - Switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To maintain the power factor of a switching power supply circuit by the portion capable of dealing with its actual use conditions with respect to the fluctuations of its AC input voltage and its load, and suppress its ripple components. SOLUTION: To a power-factor improving circuit provided in the power supply circuit called a push-pull-form, switching-frequency-controlling-mode, and composite-resonance-type converter, the switching output obtained from its primary-side resonance circuit is fed back via a primary-side parallel- resonance capacitor and by an electrostatically coupling method or a magnetically coupling method.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit having a power factor correction circuit.

[0002]

2. Description of the Related Art The applicant has previously proposed various power supply circuits having a resonance type converter on the primary side. Further, various power supply circuits including a power factor improving circuit for improving the power factor of the resonant converter have been proposed.

FIG. 10 is a circuit diagram showing an example of a switching power supply circuit constructed based on the invention previously filed by the present applicant. This power supply circuit has a configuration in which a power factor improvement circuit for improving a power factor is provided for a self-excited current resonance type switching converter.

The power supply circuit shown in FIG. 1 includes a bridge rectifier circuit Di for full-wave rectification of a commercial AC power supply AC. In this case, the rectified output rectified by the bridge rectification circuit Di is charged to the smoothing capacitor Ci via the power factor correction circuit 20, and both ends of the smoothing capacitor Ci correspond to the level of one time of the AC input voltage VAC. The rectified smoothed voltage Ei is obtained. Also, in the rectifying / smoothing circuit (Di, Ci), an inrush current limiting resistor Ri is inserted in the rectification current path so as to suppress an inrush current flowing into the smoothing capacitor when the power is turned on, for example. ing.

In the power factor correction circuit 20 shown in FIG. 1, a filter choke coil LN, a high-speed recovery type diode D1 and a choke coil LS are provided between a positive output terminal of a bridge rectifier circuit Di and a positive terminal of a smoothing capacitor Ci. They are connected in series and inserted. The filter capacitor CN is inserted between the anode side of the high-speed recovery diode D1 and the positive terminal of the smoothing capacitor Ci to form a normal mode low-pass filter together with the filter choke coil LN.

In the power factor correction circuit 20, an end of a primary side series resonance circuit, which will be described later, is connected to a connection point between the cathode of the high-speed recovery type diode D1 and the choke coil LS. The switching output obtained in the series resonance circuit is fed back. The power factor improving operation by the power factor improving circuit 20 will be described later.

This power supply circuit includes a self-excited current resonance type converter using a rectified smoothed voltage Ei, which is a voltage across a smoothing capacitor Ci, as an operating power supply. In this current resonance type converter, as shown in the figure, switching elements Q1 and Q2 of two bipolar transistors are half-bridge-coupled and then inserted between the connection point on the positive side of the smoothing capacitor Ci and ground. It is connected. Starting resistors RS1 and RS2 are inserted between the collectors and the bases of the switching elements Q1 and Q2, respectively. The resistors RB1 and RB2 connected to the bases of the switching elements Q1 and Q2 set the base current (drive current) of the switching elements Q1 and Q2. Further, clamp diodes DD1 and Q2 are connected between the base and the emitter of the switching elements Q1 and Q2, respectively.
DD2 is inserted. The clamp diodes DD1 and DD2 are
A current path for a clamp current flowing through the base-emitter is formed during a period in which the switching elements Q1 and Q2 are turned off. And the resonance capacitors CB1, C
B2 is a drive winding N of a drive transformer PRT described below.
A series resonance circuit (self-excited oscillation drive circuit) for self-excited oscillation is formed together with B1 and NB2, and the switching elements Q1, Q2
Determine the switching frequency of 2.

[0008] Drive transformer PRT (Power Regulati
ng Transformer) is provided to drive the switching elements Q1 and Q2 and to perform constant voltage control by variably controlling the switching frequency. In this case, the drive windings NB1 and NB2 and the resonance current detection are performed. Winding N
D is wound, and a control winding N is provided for each of these windings.
C is an orthogonal saturable reactor wound in the orthogonal direction. One end of the drive winding NB1 of the drive transformer PRT is connected to the base of the switching element Q1 via a series connection of the resistor RB1 and the resonance capacitor CB1, and the other end is connected to the emitter of the switching element Q1. One end of the drive winding NB2 is grounded, and the other end is connected to the base of the switching element Q2 via a series connection of a resistor RB2 and a resonance capacitor CB2. The drive winding NB1 and the drive winding NB2 are wound so that voltages of opposite polarities are generated.

Insulated converter transformer PIT (Power Is
olation Transformer) is the switching element Q1, Q2
Is transmitted to the secondary side. One end of the primary winding N1 of the insulation converter transformer PIT is connected to the contact point (switching output point) between the emitter of the switching element Q1 and the collector of the switching element Q2 via the resonance current detection winding ND, thereby providing a switching output. Is obtained.

The other end of the primary winding N1 is connected to a connection point between the cathode of the high-speed recovery type diode D1 and the choke coil LS in the power factor correction circuit 20 via the series resonance capacitor C1. ing.

In this case, the series resonance capacitor C1 and the primary winding N1 are connected in series, but the capacitance of the series resonance capacitor C1 and the primary winding N1 are connected.
Converter transformer PIT including (series resonance winding)
The primary side series resonance circuit for making the operation of the switching converter a current resonance type is formed by the leakage inductance (leakage inductance) component.

On the secondary side of the insulated converter transformer PIT shown in FIG. 1, a center tap is provided for the secondary winding N2, and the rectifier diodes DO1, DO2, D
By connecting O3, DO4 and smoothing capacitors CO1, CO2 as shown in the figure, a set of [rectifier diodes DO1, DO2, smoothing capacitor CO1] and a set of [rectifier diodes DO3, DO4, smoothing capacitor CO2] are obtained. Two sets of full wave rectifier circuits are provided. The full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] has a DC output voltage EO1.
And a full-wave rectifier circuit composed of [rectifier diodes DO3, DO4, smoothing capacitor CO2] generates a DC output voltage EO2. In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1.

The control circuit 1 supplies a DC current whose level varies in accordance with the level of the DC voltage output EO1 on the secondary side to the control winding NC of the drive transformer PRT as a control current, as will be described later. The constant voltage control is performed as described above.

The switching operation of the power supply circuit according to the above configuration is as follows. First, when a commercial AC power supply is turned on, the switching elements Q1, Q2 are activated via, for example, start-up resistors RS1, RS2.
2 is supplied with a starting current. For example, if the switching element Q1 is turned on first,
The switching element Q2 is controlled to be turned off.
Then, as an output of the switching element Q1, a resonance current flows through the resonance current detection winding ND → the primary winding N1 → the series resonance capacitor C1. When the resonance current becomes zero, the switching element Q2 is turned on, and It is controlled to be off. Then, a resonance current flows in the opposite direction through the switching element Q2. Or later,
A self-excited switching operation in which the switching elements Q1 and Q2 are turned on alternately is started. As described above, by using the terminal voltage of the smoothing capacitor Ci as the operating power source, the switching elements Q1 and Q2 alternately open and close,
A drive current close to the resonance current waveform is supplied to the primary winding N1 of the insulating converter transformer PIT, and an alternating output is obtained at the secondary winding N2.

The constant voltage control by the drive transformer PRT is performed as follows. For example, assuming that the secondary output voltage EO1 fluctuates so as to increase with the AC input voltage level or load fluctuation, the level of the control current flowing through the control winding NC as described above is also changed to the secondary output voltage EO1.
Is controlled so as to increase in accordance with the rise of. Under the influence of the magnetic flux generated in the drive transformer PRT by this control current, the drive transformer PRT tends to approach a saturation state, and acts to reduce the inductance of the drive windings NB1 and NB2. The switching frequency is controlled so as to increase by changing the condition of the oscillation circuit. In this power supply circuit, the switching frequency is set in a frequency region higher than the resonance frequency of the series resonance circuit of the series resonance capacitor C1 and the primary winding N1 (upper side control). Then, the switching frequency is separated from the resonance frequency of the series resonance circuit.
Thereby, the resonance impedance of the series resonance circuit with respect to the switching output increases.

By increasing the resonance impedance in this way, the drive current supplied to the primary winding N1 of the primary series resonance circuit is suppressed, and as a result, the secondary output voltage is suppressed. , Constant voltage control is achieved. Hereinafter, the constant voltage control method using the above method will be referred to as “switching frequency control method”.

The power factor improving operation by the power factor improving circuit 20 is as follows. In the configuration of the power factor correction circuit 20 shown in this figure, the switching output supplied to the series resonance circuit (N1, C1) has a rectified current path via an inductive reactance (magnetic coupling) assumed to be included in the choke coil LS itself. It is made to return to.

The switching output fed back as described above causes the alternating voltage of the switching cycle to be superimposed on the rectified current path. The superimposed voltage of the switching cycle causes the high speed recovery type diode D1 to be superimposed. In this case, an operation of interrupting the rectified current in the switching cycle is obtained, and the apparent inductance of the filter choke coil LN and choke coil LS also increases due to the intermittent action. This allows the charging current to flow to the smoothing capacitor Ci even during a period in which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, and the conduction angle of the AC input current is increased. As a result, the power factor is improved.

FIG. 11 is a circuit diagram showing another example of the configuration of a switching power supply circuit that can be configured based on the invention previously proposed by the present applicant. This power supply circuit is also provided with a current resonance type converter in which two switching elements are half-bridge-coupled, and the drive system is separately excited. Also in this case, a power factor improving circuit for improving the power factor is provided. Note that the same parts as those in FIG. 10 are denoted by the same reference numerals and description thereof will be omitted.

The primary-side current resonance type converter shown in FIG. 1 includes two switching elements Q11 and Q12, for example, MOS-FETs. here,
By connecting the drain of the switching element Q11 to the line of the rectified and smoothed voltage Ei, connecting the source of the switching element Q11 and the drain of the switching element Q12, and connecting the source of the switching element Q12 to the primary side ground, it is compatible with the separately excited type. A half-bridge connection is obtained. These switching elements Q11 and Q12 are switching-driven by the oscillation drive circuit 2 so that on / off operations are alternately repeated, and intermittently output the rectified smoothed voltage Ei as a switching output. In this case, a clamp diode DD connected between the drain and source of each of the switching elements Q11 and Q12 in the direction shown in FIG.
1, DD2 are provided.

In this case, the switching element Q
11. By connecting one end of the primary winding N1 of the isolated converter transformer PIT to the source-drain connection point (switching output point) of Q12, the switching output is supplied to the primary winding N1. Is done. The other end of the primary winding N1 is connected via a series resonance capacitor C1 to a filter choke coil LN of a power factor improving circuit 21 described below.
And the anode of the high speed recovery type diode D1.

Also in this case, a series resonance circuit for making the switching power supply circuit a current resonance type is formed by the capacitance of the series resonance capacitor C1 and the leakage inductance component of the insulating converter transformer PIT including the primary winding N1. .

In this case, the control circuit 1 outputs a control signal having a level corresponding to the fluctuation of the DC output voltage EO1 to the oscillation drive circuit 2, for example. The oscillation drive circuit 2 changes the frequency of the switching drive signal supplied from the oscillation drive circuit 2 to each gate of the switching elements Q11 and Q12 based on the control signal supplied from the control circuit 1,
The switching frequency is made variable. In the power supply circuit shown in FIG. 11, as in the power supply circuit shown in FIG. 10, the switching frequency is set as a region higher than the series resonance frequency. And
For example, when the DC output voltage EO1 rises, the control circuit 1 controls the oscillation drive circuit 2 to increase the switching frequency according to the level. Thus, the constant voltage control is performed in the same manner as described with reference to FIG. The starting circuit 3 is provided for detecting the voltage or current obtained on the rectifying / smoothing line immediately after the power is turned on, and for starting the oscillation drive circuit 2. Is input as an operation power supply.

In the power factor correction circuit 21 shown in this figure, the positive output terminal of the bridge rectification circuit Di and the smoothing capacitor Ci
Of the filter choke coil LN−
A fast recovery diode D1 is inserted in series. Here, the filter capacitor CN is provided in parallel with the series connection circuit of the filter choke coil LN and the high-speed recovery type diode D1. Also in this connection form, the filter capacitor CN forms a normal mode low-pass filter together with the filter choke coil LN. The resonance capacitor C3
Are provided in parallel with the high-speed recovery type diode D1. Although a detailed description is omitted here, for example, the resonance capacitor C3 forms a parallel resonance circuit together with, for example, the filter choke coil LN and the like, so that its resonance frequency is substantially equal to the resonance frequency of the series resonance circuit described later. Is set. This has the effect of suppressing an increase in the rectified smoothed voltage Ei when the load is reduced. As described above, the power factor improving circuit 21 is connected to the end of the series resonance circuit (N1, C1) with respect to the connection point between the filter choke coil LN and the anode of the fast recovery type diode D1. Is connected.

In such a connection form, the switching output obtained on the primary winding N1 is connected to the series resonance capacitor C1.
The switching output is fed back to the rectified current path through the capacitive coupling of the switch. In this case, the switching output is applied to the connection point between the filter choke coil LN and the anode of the high speed recovery type diode D1 so that the resonance current obtained in the primary winding N1 flows. become.

As the switching output is fed back as described above, the alternating voltage of the switching cycle is superimposed on the rectified current path. In the diode D1, an operation of interrupting the rectified current in the switching cycle is obtained, and the apparent inductance of the filter choke coil LN also increases due to the interrupting action. Further, a voltage is generated at both ends of the resonance capacitor C3 when a current of a switching period flows through the resonance capacitor C3. However, the level of the rectified smoothed voltage Ei is reduced by the voltage across the resonance capacitor C3. Accordingly, even during the period when the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci, the smoothing capacitor C
The charging current to i flows. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, the conduction angle of the AC input current is expanded, and the power factor is also improved.

As described above, in the power supply circuits shown in FIGS. 10 and 11, the power factor can be improved by providing the power factor improvement circuits (20, 21). Since the power factor improving circuit is formed with a small number of components, it has the advantage that the power factor can be improved with high efficiency, low noise, small size, light weight, and low cost.

FIG. 12 is a circuit diagram showing another example of the configuration of a switching power supply circuit which can be configured based on the invention previously proposed by the present applicant. This power supply circuit is an example of a case where the maximum load power POmax is to cope with a heavy load of 150 W or more. The power supply circuit shown in this figure is provided with a self-excited current resonance type converter in which four switching elements are full-bridge-coupled.

The current-resonant converter of the full-bridge coupling type shown in FIG. 1 includes switching elements Q1, Q2, Q3, and Q4 formed of four bipolar transistors. As shown in this figure, the switching elements Q1 and Q2 are connected in series via the respective collectors and emitters between the positive electrode of the smoothing capacitor Ci and the ground. The switching elements Q3 and Q4 are also connected in the same manner as described above.

As a drive circuit system for the switching element Q1, a starting resistor RS1 is inserted between the collector and the base, and a clamp diode DD1 is inserted between the base and the emitter. Then, a drive circuit for self-excited oscillation (resonant capacitor CB1-drive winding NB1) is connected to the base of switching element Q1 via a base current limiting resistor RB1. Similarly, switching elements Q2, Q3, Q4
[Starting resistances (RS2, RS3, RS4) and clamp diodes (DD2, DD2,
DD3, DD4), base current limiting resistors (RB2, RB3, R
B4), the resonance capacitors (CB2, CB3, CB4) and the drive windings (NB2, NB3, NB4) form a drive circuit system.

The drive transformer PRT has driving windings NB1 to NB4 and a resonance current detecting winding ND formed by winding up the driving winding NB1 corresponding to the full bridge coupling system. The control winding NC is wound so that the winding directions are orthogonal to each other, thereby constituting a saturable reactor.

The series resonance circuit shown in FIG.
1 and a series resonance capacitor C1. The power factor correction circuit 22 is different from the power factor correction circuit 20 of FIG.
Of the winding Ns. To the power factor improving circuit 22, an end of the primary side series resonance circuit is connected to the winding Np of the magnetic coupling transformer MCT, and a switching output obtained in the series resonance circuit is fed back. I am trying to.

The switching operation of the current resonance type having the above-described configuration is, for example, that a set of switching elements [Q1, Q4] and a set of switching elements [Q2, Q3] are alternately turned on / off.
An off operation is performed.

The power factor improving operation of the power factor improving circuit 22 includes a switching output supplied to the series resonance circuit (N1, C1) through a rectifying current path via the inductive reactance (magnetic coupling) of the winding Ns. It is made to return to.
With the switching output thus fed back, the power factor can be improved as in the example described with reference to FIG.

[0035]

FIG. 13 shows the relationship between the load power Po and the power factor PF of the power supply circuit shown in FIGS. Here,
The condition when the AC input voltage VAC = 100 V is shown.
According to this figure, it can be seen that the characteristic that the power factor PF decreases as the load power Po decreases.

FIG. 14 shows the relationship between the AC input voltage VAC and the power factor PF. Here, the maximum load power Pomax = 120 W and the minimum load power Pomin = 40
The characteristics under each condition at the time of W are shown. As shown in this figure, it can be seen that the power factor PF decreases proportionally as the AC input voltage VAC increases. As the power factor PF under the condition of the minimum load power Pomin = 40 W, the maximum load power Pomax = 12
The power factor is lower than 0W. That is, FIG.
As described above, the characteristic that the power factor PF is lower when the load power is smaller is also obtained here.

The characteristic shown in FIG. 14 is shown as an operation waveform diagram as shown in FIG. Here, FIGS. 15A and 15B show the AC input voltage VA when the AC input voltage VAC = 100 V and the maximum load power Pomax = 120 W.
C and the AC input current IAC are shown. FIGS. 14C and 14D show the minimum load power Pomi at the AC input voltage VAC = 100 V.
The AC input voltage VAC and the AC input current IAC when n = 40 W are shown. Here, the half cycle of the AC input voltage VAC is 1
0 ms, the maximum load power Pomax = 120 W
Sometimes, the conduction period τ of the AC input current IAC is actually 5 m
s, and the power factor is PF = 0.85.
On the other hand, when the minimum load power Pomin = 40W,
The conduction period τ of the AC input current IAC is reduced to about 2.5 ms, and the power factor is reduced to about PF = 0.65. The value of the power factor PF obtained when the minimum load power Pomin = 40 W may not satisfy the value as the power factor required for practical use.

As described above, the fact that the power factor is reduced by the fluctuation of the AC input voltage and the fluctuation of the load power means that the AC input voltage and the load conditions for these power supply circuits are restricted. become. That is, there is a problem that devices that can be used as the power supply circuit are limited. Specifically, for example, it can be adopted in a color television receiver in which an AC input voltage and a load condition are specified,
Office equipment and information equipment cannot be adopted.

In the configuration for improving the power factor shown in FIGS. 10 and 11, the series resonance circuit on the primary side is connected to the rectified current path of the commercial AC power supply. On the other hand, the commercial AC power supply cycle (50Hz / 60H
It has been found that the ripples of z) overlap. The superimposed level of the ripple component increases as the load power increases. If, for example, required components are selected so as to maintain a power factor of about PF = 0.8 under predetermined measurement conditions corresponding to practicality, the power factor improving circuit is provided. It is known that the ripple voltage level appearing on the secondary side DC output voltage at the time of maximum load power increases to about 3 to 4 times as compared with the case where it is not possible.

In order to suppress the increase of the ripple component as described above, for example, in the actual power supply circuit shown in FIGS. 10 and 11, the gain of the control circuit 1 is improved, and the capacitance of the primary-side smoothing capacitor Ci is increased. However, in this case, the cost of the component element is increased, and the switching operation is liable to abnormally oscillate.

Further, in the power supply circuit shown in FIGS. 10 and 11, the impedance of the choke coil LS and the resonance capacitor C3 is inserted into the primary side series resonance circuit of the current resonance type converter, so that the maximum load power Pomax is reduced. descend. Therefore, there is a disadvantage that the primary winding N1 and the series resonance capacitor C1 of the insulating converter transformer PIT need to be redesigned.

In the power supply circuit shown in FIG.
It is the same that the power factor is reduced by the fluctuation of the AC input voltage or the fluctuation of the load power. Further, in this case, there is a problem that the circuit configuration is complicated and the number of components increases.

[0043]

In view of the above-mentioned problems, the present invention is configured as a switching power supply circuit as follows. That is, a gap is formed so as to obtain a required coupling coefficient which is loosely coupled, and a rectifying / smoothing means for receiving a commercial AC power supply to generate a rectified smoothed voltage and outputting the rectified smoothed voltage as a DC input voltage. An insulating converter transformer provided for transmission to a secondary side, and first and second switching means adapted to intermittently output the DC input voltage by a push-pull operation and output the DC input voltage to a primary winding of the insulating converter transformer. The first and second switching means are formed by at least a leakage inductance component including a primary winding of the insulated converter transformer and a capacitance of a primary side parallel resonance capacitor, and operate the first and second switching means in a voltage resonance type. Two primary-side resonance circuits and one of the first and second primary-side resonance circuits inserted into the rectified current path. Power factor improving means for improving the power factor by intermittently commutating a rectified current based on the returned switching output, wherein the obtained switching output is fed back via the primary side parallel resonance capacitor in one of the primary side resonance circuits. A secondary resonance circuit formed on the secondary side by the leakage inductance component of the secondary winding of the insulating converter transformer and the capacitance of the secondary resonance capacitor; and a secondary resonance circuit formed including the secondary resonance circuit. DC output voltage generating means configured to input an alternating voltage obtained to a secondary winding of the insulating converter transformer and perform a rectification operation to generate a secondary DC output voltage; and Constant voltage control means configured to perform constant voltage control on the secondary side DC output voltage according to the level of the output voltage Constitute a mode power supply circuit.

The power factor improving means is connected in parallel with a high-speed recovery type diode for intermitting a rectified current and a capacitor, and is connected to one of the first and second primary side resonance circuits. The primary parallel resonance capacitor is connected to the anode of the fast recovery diode. Alternatively, the power factor improving means includes a high-speed recovery type diode for interrupting a rectified current and an inductance connected in series,
One of the first and second primary-side resonance circuits, the primary-side parallel resonance capacitor is connected to a connection point between the high-speed recovery diode and the inductance.

According to the above configuration, the switching output obtained in the primary-side resonance circuit is different from that of the primary-side parallel resonance circuit in the power-factor improvement circuit provided in the power supply circuit called a push-pull type switching frequency control type complex resonance converter. By passing through the capacitor, feedback is performed by an electrostatic coupling method or a magnetic coupling method.

[0046]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to an embodiment of the present invention.
In this figure, the same parts as those in FIGS. 10, 11, and 12 are denoted by the same reference numerals, and description thereof will be omitted.

The switching power supply circuit shown in FIG. 1 is provided with a voltage resonance type converter having two switching elements Q1 and Q2 and performing a switching operation by a so-called push-pull method by a self-excited method. A power factor improving circuit 10 is provided for the voltage resonance type converter.

In the power supply circuit shown in this figure, a bridge rectification circuit Di is used as a rectification and smoothing circuit for receiving a commercial AC power supply (AC input voltage VAC) and obtaining a DC input voltage.
And a full-wave rectifier circuit composed of a smoothing capacitor Ci, and generates a rectified smoothed voltage Ei corresponding to a level that is one time the AC input voltage VAC. Further, in the rectifying / smoothing circuit, an inrush current limiting resistor Ri is inserted into the rectifying current path so as to suppress, for example, an inrush current flowing into the smoothing capacitor when the power is turned on.

Further, the power supply circuit shown in FIG. 1 is designed to drive two sets of switching elements Q1 and Q2, which will be described later, by push-pull operation and to variably control the switching frequency of the switching elements Q1 and Q2. A type drive transformer PRT is provided. As a structure of the orthogonal drive transformer PRT, for example, a three-dimensional core is formed by joining the ends of two magnetic legs of two double U-shaped cores having four magnetic legs. The detection winding ND and the drive windings NB1 and NB2 are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction, and the control winding NC is connected to the detection winding. ND, drive winding N
A saturable reactor is formed by winding the coil in a direction orthogonal to B1 and NB2.

In this case, one end of the detection winding ND is connected to the positive electrode of the smoothing capacitor Ci via the choke coil CH (inductance Lc) and the primary winding N1B, and the other end is connected to the collector of the switching element Q1. Is done.

The drive windings NB1 and NB2 are formed by dividing one winding by center tapping the ground with respect to the ground. The end of the drive winding NB1 is connected to the base of the switching element Q1 via a series connection of the resonance capacitor CB1 and the base current limiting resistor RB1. The end of the drive winding NB2 is connected to the base of the switching element Q2 via the resonance capacitor CB2 and the base current limiting resistor RB2. That is, the driving winding NB1 and the resonance capacitor C
Switching element Q by B1 and base current limiting resistor RB1
A self-excited oscillation drive circuit for the switching element Q2 is formed by the drive winding NB2, the resonance capacitor CB2, and the base current limiting resistor RB2. In the detection winding ND, an alternating voltage corresponding to a switching output is detected by a switching operation described later.
The drive windings NB1 and NB2 are connected to each other in accordance with the switching output detected by the resonance current detection winding ND.
° Alternating voltages of opposite polarities with different phases can be obtained.

In this power supply circuit, two switching elements Q1 and Q2 are provided for a push-pull operation. In this case, a high withstand voltage bipolar transistor (BJT: junction type transistor) is employed for the switching elements Q1 and Q2.

The driving circuit (drive winding NB1, base current limiting resistor RB1, resonance capacitor CB1), clamp diode DD1, and parallel resonance capacitor Cr1 are connected to the switching element Q1 as shown in FIG. Is connected to a drive circuit (drive winding NB2, base current limiting resistor RB2, resonance capacitor CB2), clamp diode DD2, and parallel resonance capacitor Cr2 as shown in the figure.

Here, the clamp diodes DD1, DD2
Are connected in parallel with each other between the base and collector of the switching elements Q1 and Q2. The parallel resonance capacitor Cr1 is connected between the collector and the emitter of the switching element Q1. However, the other parallel resonance capacitor Cr2 is arranged between the collector of the switching element Q2 and the anode side of the high-speed recovery type diode in the power factor correction circuit 10. Also, the switching element Q1,
The emitter of Q2 is connected to the primary side ground.

The starting resistor Rs is provided with a smoothing capacitor Ci.
And the switching element Q2. The starting resistor Rs is inserted to supply a starting current for starting a switching operation to the switching element Q2 at the time of starting.

The insulating converter transformer PIT for transmitting the switching outputs of the switching elements Q1 and Q2 to the secondary side is, as shown in FIG. 2, for example, E-type cores CR1 and CR2 made of ferrite material, as shown in FIG.
An EE-type core is provided so that the magnetic legs of the EE-type core are opposed to each other. A primary winding N1 (N1A, N1B) The secondary winding N2 is wound in a divided state. A gap G is formed for the center magnetic leg as shown in the figure. As a result, loose coupling with a required coupling coefficient can be obtained. Gap G is
The E-shaped cores CR1 and CR2 can be formed by forming the central magnetic legs shorter than the two outer magnetic legs. Also,
As the coupling coefficient k, for example, a loosely coupled state of k ≒ 0.85 is obtained, so that a saturated state is hardly obtained.

In this case, the insulation converter transformer PIT
Is divided into primary windings N1A and N1B. One end of the primary winding N1A is connected to the collector of the switching element Q1 via the detection winding ND, and the other end is connected to the positive electrode of the smoothing capacitor Ci via the series connection of the inductance winding Lc of the choke coil CH. One end of the primary winding N1B is connected to the collector of the switching element Q2, and the other end is connected to the positive electrode of the smoothing capacitor Ci via a series connection of the inductance winding Lc of the choke coil CH.

In this case, the above-described parallel resonance capacitor C
r1 is the leakage inductance component of the primary winding N1A (L1
A) and the combined inductance (L1A + Lc) of the inductance winding Lc form a parallel resonance circuit for causing the switching element Q1 to operate in a voltage resonance type. Similarly, the parallel resonance capacitor Cr2 is connected to the primary winding N1B.
And the combined inductance (L1B + Lc) of the inductance winding Lc and the leakage inductance component (L1B), form a parallel resonance circuit for causing the switching element Q2 to operate in a voltage resonance type.

In such a configuration on the primary side, the driving winding N
Since alternating voltages of opposite polarities are obtained in B1 and the driving winding NB2, the self-excited oscillation driving circuit including the driving winding NB1 and the self-excited oscillation driving circuit including the driving winding NB2 respectively have opposite polarities. , A drive current (base current) as an alternating current flows through each base of the switching elements Q1 and Q2. Thereby, the switching element Q1,
As for Q2, an operation of alternately turning on / off at a switching frequency determined by a constant of the self-excited oscillation driving circuit is obtained. That is, a switching operation by push-pull is obtained in a voltage resonance type. Switching element Q1
Is supplied to the primary winding N1A via the detection winding ND, and the switching output of the switching element Q2 is supplied to the primary winding N1B. Then, the current flows to the smoothing capacitor Ci via the inductance winding Lc of the choke coil CH. By providing the voltage resonance type converter performing the push-pull operation in this way, for example,
It is possible to cope with a maximum load power of 00 W or more.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. With this parallel resonance circuit,
The alternating voltage excited by the secondary winding N2 becomes a resonance voltage.
That is, a voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, the primary side is provided with a parallel resonance circuit for making the switching operation a voltage resonance type, and the secondary side is also provided with a parallel resonance circuit for obtaining the voltage resonance operation. . In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

In this case, for the parallel resonance circuit on the secondary side formed as described above, taps are provided for the secondary winding N2, and the rectifier diodes DO1, DO2, DO are provided.
3, DO4 and the smoothing capacitors CO1, CO2 are connected as shown in the figure, so that a set of [rectifier diodes DO1, DO2, smoothing capacitor CO1] and a set of [rectifier diodes DO3, DO4, smoothing capacitor CO2] are obtained. Two sets of full wave rectifier circuits are provided. The full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] has a DC output voltage EO1.
And a full-wave rectifier circuit composed of [rectifier diodes DO3, DO4, smoothing capacitor CO2] generates a DC output voltage EO2. In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1.

The control circuit 1 supplies a DC current whose level varies in accordance with the level of the DC voltage output EO1 on the secondary side to the control winding NC of the drive transformer PRT as a control current, for example. The constant voltage control is performed as follows.

Incidentally, the insulation converter transformer PIT
, The polarity (winding direction) of the primary winding N1 and the secondary winding N2 and the rectifier diode DO (DO1, DO2, DO3, DO4)
, The mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 may be + M or -M. For example, when the connection configuration shown in FIG. 3A is adopted, the mutual inductance is + M (additive polarity: forward method), and when the connection configuration shown in FIG. 3B is employed, the mutual inductance is −M (depolarization: Flyback method). If this is made to correspond to the operation of the secondary side of the power supply circuit shown in FIG. 1, for example, when the alternating voltage obtained in the secondary winding N2 has a positive polarity, the rectifier diode DO1 (DO3)
Can be regarded as an operation mode of + M (forward system). Conversely, when the alternating voltage obtained in the secondary winding N2 has a negative polarity, the rectifier diode DO2
The operation in which the rectified current flows through (DO4) can be regarded as the -M operation mode (flyback method). That is, in this power supply circuit, each time the alternating voltage obtained in the secondary winding becomes positive / negative, the mutual inductance becomes + M / -M
Mode.

The control circuit 1 varies the level of the control current (DC current) flowing through the control winding NC according to the change in the secondary-side DC output voltage level (EO1), thereby winding the control coil PRT on the orthogonal control transformer PRT. The inductances LB1 and LB2 of the mounted drive windings NB1 and NB2 are variably controlled. Thereby, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching elements Q1 and Q2 formed including the inductances LB1 and LB2 of the drive windings NB1 and NB2 changes. This is an operation of changing the switching frequency of the switching elements Q1 and Q2. This operation has an effect of stabilizing the secondary-side DC output voltage.

Next, the configuration of the power factor improving circuit 10 will be described. In the power factor correction circuit 10, a choke coil Ls and a high-speed recovery type diode D1 are inserted in series between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. The filter capacitor CN is provided in parallel with the series connection circuit of the choke coil Ls and the high-speed recovery type diode D1, thereby forming a normal mode low-pass filter together with the choke coil Ls.

The parallel resonance capacitor C3 is provided in parallel with the high speed recovery type diode D1. Although detailed description is omitted here, for example, the parallel resonance capacitor C3 forms a parallel resonance circuit together with, for example, the choke coil Ls. This has the effect of suppressing an increase in the rectified smoothed voltage Ei when the load is reduced. In the power factor correction circuit 10, the above-described parallel resonance capacitor Cr2 is connected to a connection point of the choke coil Ls, the anode of the high-speed recovery type diode D1, and the parallel resonance capacitor C3, and switching is performed. The switching output (voltage resonance pulse voltage) obtained in the primary parallel resonance circuit on the element Q2 side is fed back.

The power factor improving operation by the power factor improving circuit 10 is basically as follows. In the configuration of the power factor correction circuit 10 shown in this figure, the switching output obtained in the primary side parallel resonance circuit on the switching element Q2 side is fed back to the rectified current path via the capacitive coupling of the parallel resonance capacitor Cr2. Become.

The switching output fed back in this way causes the alternating voltage of the switching cycle to be superimposed on the rectified current path. An operation of interrupting the rectified current at the switching cycle is obtained, and the apparent inductance of the choke coil Ls also increases due to the interrupting action. In addition, a voltage is generated at both ends of the parallel resonance capacitor C3 when a current of a switching period flows through the capacitor.
The level of the rectified smoothed voltage Ei is reduced by the voltage across the parallel resonance capacitor C3. This allows the charging current to flow to the smoothing capacitor Ci even during a period in which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, and the conduction angle of the AC input current is increased. As a result, the power factor is improved.

As described above, the parallel resonance capacitor Cr2 is connected to the cathode of the high-speed recovery type diode D1 of the power factor correction circuit 10. This means that the parallel resonance capacitor Cr2 and the parallel resonance capacitor C3 are connected in series, that is, the parallel resonance capacitor Cr2
The voltage resonance pulse voltage, which appears as the voltage across
The voltage is divided by the capacitance ratio of the parallel resonance capacitor Cr2 and the parallel resonance capacitor C3. Then, a circuit system is formed as a voltage feedback system in which the voltage is fed back to the smoothing capacitor Ci via the parallel resonance capacitor C3 connected in parallel with the high-speed recovery type diode D1.

The capacitance of the parallel resonance capacitors Cr1 and Cr2 is Cr1 <Cr2, and the capacitance of the parallel resonance capacitor C3 is sufficiently larger than Cr2. In particular, when the capacitance of the capacitor C3 is increased, the power factor PF is improved. That is, the switching frequency fs is controlled to be high during the period when the AC input voltage VAC is high, and the switching frequency fs is controlled to be low during the period when the AC input voltage VAC is low. The resonance pulse voltage is not fed back to the power factor correction circuit 10, and the AC input current IAC from the AC power supply AC is changed from the bridge rectifier circuit Di to the choke coil L.
s → The smoothing capacitor Ci is charged via the high-speed recovery type diode D1. Then, as the AC input voltage VAC decreases, the power factor improving circuit 10 of the voltage resonance pulse voltage increases.
The amount of return to is increased. The switching elements Q1, Q
2, the peak value of the voltage resonance pulse voltage generated at each collector is equal to the off-order pulse width.

FIGS. 4 and 5 show such a power factor improving circuit 1.
7 shows the state of the power factor improved by 0. FIG. 4 shows an AC input voltage VAC = 100 V and a load power PO = 25.
The AC input current IAC with a power factor PF = 0.85 at 0 W also shows the operation waveform, and FIG. 5 shows that the load power PO = 10
At the time of W, the AC input current IAC with the power factor PF = 0.8 also shows the operation waveform.

In the switching power supply circuit of this embodiment, the filter capacitor CN = 1 μF and the choke coil LS = 6
8 μH, parallel resonance capacitor Cr1 = 4700 pF, parallel resonance capacitor Cr2 = 5600 pF, parallel resonance capacitor C3 = 0.033 μF, and the maximum load power P
The experiment was performed under the conditions of Omax = 250 W, minimum load power POmin = 0 W, and AC input voltage VAC = 80 V to 144 V. As a result, as shown in FIG. 6, the power factor PF became approximately 0.85 with respect to the load fluctuation of the load power PO = 250 W to 20 W, and was kept constant. Also, as shown in FIG. 7, when the AC input voltage VAC fluctuates in the range of
Under each condition of load power PO = 250 W to 50 W, almost the same power factor (PF = 0.85) was obtained.

As described above, in the power supply circuit of the present embodiment, a high power factor can be maintained even when the AC input voltage and the load fluctuate. For this reason, the power supply circuit according to the present embodiment is not limited to a television receiver or the like in which the AC input voltage or the load condition is specified, and is applied to office equipment such as an office equipment or a personal computer in which the load condition varies. It is practically possible to mount it.

The secondary side DC output voltage levels EO1, E02
Of the 50 Hz ripple voltage component is only about twice as large as that when the power factor correction circuit 10 is not provided, and is within a range in which there is no practical problem as a power supply circuit used for, for example, a color television.

Further, in the case of the voltage period method as in this example, the control range does not become narrow at the time of the maximum load power POmax, and it is not necessary to redesign the composite resonance type converter.

Next, a second embodiment of the present invention will be described with reference to FIG. FIG. 8 is a circuit diagram showing a configuration of a power supply circuit according to a second embodiment of the present invention. In this figure, the same parts as those in FIG. 1, and FIGS. 10, 11, and 12 are denoted by the same reference numerals, and description thereof will be omitted.

The power supply circuit shown in FIG. 8 is also provided with a voltage resonance type converter having two switching elements Q1 and Q2 as in FIG. 1 and performing a switching operation by a so-called push-pull system by a self-excited system. You. The power supply circuit shown in FIG.
1 and Q2 are switched by a push-pull operation, and the orthogonal drive transformer PR is used to variably control the switching frequency of the switching elements Q1 and Q2.
T is provided.

Further, the parallel resonance capacitor Cr1 disposed between the emitter and the collector of the switching element Q1 is
The combined inductance (L1A + L) of the leakage inductance component (L1A) of the primary winding N1A and the inductance winding Lc.
c) forms a parallel resonance circuit for causing the switching element Q1 to operate in a voltage resonance type.

Also on the switching element Q2 side, the parallel resonance capacitor Cr2 sets the switching element Q2 to a voltage resonance type by the combined inductance (L1B + Lc) of the leakage inductance component (L1B) of the primary winding N1B and the inductance winding Lc. A parallel resonance circuit for operation is formed. However, in this example, capacitors Cr21 and Cr are provided between the emitter and collector of the switching element Q2.
22 are connected in series, and this series-connected capacitor C
r21 and Cr22 function as the parallel resonance capacitor Cr2.

Further, in the power factor correction circuit 11 shown in this figure, a filter choke coil LN, a high-speed recovery type diode D1 and a choke coil LS are provided between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. Are connected in series and inserted. The filter capacitor CN is inserted between the anode side of the high-speed recovery diode D1 and the positive terminal of the smoothing capacitor Ci to form a normal mode low-pass filter together with the filter choke coil LN.

For the power factor improving circuit 11,
Capacitors Cr21 and Cr connected in series that constitute the above-described parallel resonance capacitor are connected to the connection point between the cathode of the fast recovery type diode D1 and the choke coil LS.
The connection point 22 is connected so that the switching output (voltage resonance pulse voltage) obtained in the primary side parallel resonance circuit is fed back.

The power factor improving operation by the power factor improving circuit 11 is basically as follows. In the configuration of the power factor improving circuit 11 shown in this figure, the switching output supplied to the primary side parallel resonance circuit is fed back to the rectified current path via an inductive reactance (magnetic coupling) assumed to be included in the choke coil LS itself. To be.

The alternating voltage of the switching cycle is superimposed on the rectified current path by the switching output fed back as described above. The high-speed recovery type diode D1 is superimposed on the alternating voltage of the switching cycle. In this case, an operation of interrupting the rectified current in the switching cycle is obtained, and the apparent inductance of the filter choke coil LN and choke coil LS also increases due to the intermittent action. This allows the charging current to flow to the smoothing capacitor Ci even during a period in which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, and the conduction angle of the AC input current is increased. As a result, the power factor is improved.

Here, in this example, as described above, the parallel resonance capacitor Cr2 is formed by connecting the capacitors Cr21 and Cr22 in series, and
The connection point of Cr22 is connected to the cathode of the high-speed recovery type diode D1 of the power factor correction circuit 11. Therefore,
A voltage resonance pulse voltage that appears as a voltage across the resonance capacitor Cr2 (Cr21, Cr22) is divided by the capacitance ratio of the capacitors Cr21, Cr22, and is fed back to the connection point between the high-speed recovery diode D1 and the choke coil LS. A circuit system as a feedback system is formed.

The capacitance of the capacitors Cr21 and Cr22 is set to satisfy Cr21 <Cr22. In particular, when the capacitance of the capacitor Cr22 is increased, the power factor PF is improved. That is, during the period when the AC input voltage VAC is high,
Since the switching frequency fs is controlled to be high and the switching frequency fs is controlled to be low during a period when the AC input voltage VAC is low, the voltage resonance pulse voltage is fed back to the power factor correction circuit 11 near the peak value of the AC input voltage VAC. Instead, the AC input current IAC from the AC power supply AC is charged to the smoothing capacitor Ci via the bridge rectifier circuit Di → the filter choke coil LN → the fast recovery type diode D1 → the choke coil LS. And the AC input voltage VAC
Becomes smaller, the amount of feedback of the voltage resonance pulse voltage to the power factor improving circuit 11 increases.

In such a power supply circuit, as in the example described with reference to FIG. 1, a high power factor can be maintained even when the AC input voltage and the load fluctuate, and the TV in which the AC input voltage and the load conditions are specified The present invention is not limited to the John receiver and the like, but is practically sufficient for office equipment such as office equipment and personal computers in which the load condition varies.

On the secondary side of the power supply circuit shown in FIG. 8, one end of the secondary winding N2 is connected to the secondary side ground, and the other end is connected to the rectifier diode via the series connection of the series resonance capacitor Cs1. It is connected to the connection point between the anode of DO1 and the cathode of rectifier diode DO2. The cathode of the rectifier diode DO1 is connected to the positive electrode of the smoothing capacitor CO1, and the anode of the rectifier diode DO2 is connected to the secondary side ground. The negative side of the smoothing capacitor CO1 is connected to the secondary side ground.

As a result, in such a connection form, a voltage doubler full-wave rectifier circuit including a set of [series resonance capacitor Cs1, rectifier diodes DO1, DO2, and smoothing capacitor CO1] is provided. Here, the series resonance capacitor C
s1 is a rectifier diode DO1, DO2 due to its own capacitance and the leakage inductance component of the secondary winding N2.
To form a series resonance circuit corresponding to the on / off operation.
That is, the power supply circuit of the present embodiment is provided with a parallel resonance circuit on the primary side for performing a voltage resonance type switching operation, and a series resonance circuit for obtaining a voltage doubler full-wave rectification operation on the secondary side. A complex resonance type switching converter provided with a circuit is employed.

Here, the [series resonance capacitor Cs]
1, rectifier diodes DO1, DO2, smoothing capacitor CO1]
The double voltage full-wave rectification operation by the set is as follows. When a switching output is obtained on the primary winding N1 by the switching operation on the primary side, this switching output is excited by the secondary winding N2. And a rectifier diode DO
During the period when 1 is off and the rectifier diode DO2 is on, the secondary winding N1 operates in the depolarization mode in which the polarity (mutual inductance M) of the primary winding N1 and the secondary winding N2 is -M. The operation of charging the series resonant capacitor Cs1 with the rectified current IC2 rectified by the rectifier diode DO2 is obtained by the series resonance effect of the leakage inductance of the line N2 and the series resonant capacitor Cs1.

During the period when the rectifier diode DO2 is turned off and the rectifier diode DO1 is turned on and the rectifying operation is performed, the polarity (mutual inductance M) of the primary winding N1 and the secondary winding N2 is + M. The operation becomes an additional polarity mode, in which the smoothing capacitor C01 is charged in a state where series resonance occurs in which the potential of the series resonance capacitor Cs1 is added to the voltage induced in the secondary winding N2. As described above, the rectifying operation is performed by using both the adding polarity mode (+ M; forward operation) and the decreasing polarity mode (-M; flyback operation).
In the smoothing capacitor CO1, a DC output voltage EO1 corresponding to almost twice the induced voltage of the secondary winding N2 is obtained.

According to the above configuration, on the secondary side of the circuit shown in FIG. 8, by utilizing the state in which the mutual inductance is in the operation mode of + M and -M, the voltage doubler full-wave rectification is performed, so that the secondary side DC Output voltage is obtained. That is, since the electromagnetic energy due to the primary side current resonance action and the secondary side current resonance action is simultaneously supplied to the load side, the power supplied to the load side further increases accordingly, and the maximum load A large increase in power will be achieved.

Further, since the secondary DC output voltage is obtained by the voltage doubler full-wave rectifier circuit, for example, a level equivalent to the secondary DC output voltage obtained by the equal voltage rectifier circuit may be obtained. In this case, the secondary winding N2 of the present embodiment requires only half the number of turns of the conventional winding.
This reduction in the number of turns leads to a reduction in the size and weight of the insulating converter transformer PIT and a reduction in cost. In this case, a secondary winding N2A is wound independently of the secondary winding N2, a center tap is grounded to the secondary winding N2A, and a rectifier diode DO3 is connected to the secondary winding N2A. , DO4 and a smoothing capacitor CO2 are connected to generate a DC output voltage EO2.

Next, a third embodiment of the present invention will be described with reference to FIG. In FIG. 9, the same parts as those in FIGS. 1, 8, 10, 11, and 12 are denoted by the same reference numerals, and description thereof will be omitted.

In the power supply circuit shown in FIG. 9, the voltage resonance type converter provided on the primary side employs a separately-excited structure, for example, a switching element Q2 using a MOS-FET.
1, Q22 is provided.

The drain of the switching element Q21 is connected to the positive electrode of the smoothing capacitor Ci via the primary winding N1A and the choke coil CH, and the source is connected to the primary side ground. The parallel resonance capacitor Cr1 is connected between the drain and the source of the switching element Q21.
Further, a clamp diode DD1 is connected in parallel between the drain and source of the switching element Q21.

The drain of the switching element Q22 is connected to the positive electrode of the smoothing capacitor Ci via the primary winding N1B and the choke coil CH, and the source is connected to the primary side ground. The parallel resonance capacitor Cr2 is connected between the drain and the source of the switching element Q22, and a parallel resonance capacitor Cr20 is connected in series with the parallel resonance capacitor Cr2. Further, a clamp diode DD is connected in parallel between the drain and source of the switching element Q22. In this case, the drain of the switching element Q22 is connected to the connection point between the high-speed recovery type diode D1 in the power factor correction circuit 12 and the choke coil LS via the parallel resonance capacitor Cr20. In addition, the parallel resonance capacitor Cr
The capacitance of the power factor correction circuit 12 is set to be a voltage period type for the power factor correction circuit 12 because the capacitance of the capacitor 2 is sufficiently larger than that of the parallel resonance capacitor Cr22.

The switching elements Q21 and Q22 are switched by the oscillation / drive circuit 2 so that the switching operation of the push-pull method described above with reference to FIG. 1 is obtained. The control circuit 1 supplies a current or a voltage of a level fluctuated according to the fluctuation of the secondary DC output voltage E01 to the oscillation / drive circuit 2. In the oscillation / drive circuit 2, a switching drive signal (voltage) whose cycle is varied according to the output level from the control circuit 1 is switched to the switching element Q21 so that the secondary side DC output voltage E01 is stabilized. And to the gate of Q22.

In this case, the starting circuit 3 is supplied with the rectified and smoothed voltage Ei obtained from the smoothing capacitor Ci as an operating power source.
The start-up circuit 3 performs an operation for starting the oscillation / drive circuit 2 by the start-up voltage obtained in the winding N4 additionally wound around the IT.

The power factor improving circuit 12 shown in FIG.
A choke coil Ls and a high-speed recovery type diode D1 are connected in series and inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. The filter capacitor CN is provided in parallel with the series connection circuit of the choke coil Ls and the high-speed recovery type diode D1, thereby forming a normal mode low-pass filter together with the choke coil Ls. And, as described above, the drain of the switching element Q22
The switching output (voltage resonance pulse voltage) obtained in the primary side parallel resonance circuit on the switching element Q22 side by being connected to the connection point between the fast recovery diode D1 and the choke coil LS via the parallel resonance capacitor Cr20. Are fed back by the electrostatic coupling method.

With such a configuration, as in the examples described with reference to FIGS. 1 and 8, a high power factor can be maintained even with fluctuations in the AC input voltage and load, and the AC input voltage and load conditions are specified. The present invention is not limited to a television receiver and the like, but is practically sufficient for office equipment such as office equipment or a personal computer in which the load condition varies.

Incidentally, the secondary side of the power supply circuit shown in FIG. 9 includes a secondary side parallel resonance circuit formed by providing a secondary side parallel resonance capacitor C2 for the secondary winding N2. After that, a rectifying / smoothing circuit including a bridge rectifying circuit DBR and a smoothing capacitor CO1 is provided for the secondary winding N2 to obtain a secondary output voltage EO1. That is, in this configuration, a full-wave rectification operation is obtained by the bridge rectification circuit DBR on the secondary side.

In this case, on the secondary side, another secondary winding N2A is provided independently of the secondary winding N2.
And a center tap is applied thereto, and the rectifier diodes DO3, DO4 and the smoothing capacitor CO2 are connected as shown in the figure, so that the secondary-side output voltage EO2 is obtained by full-wave rectification.
I'm trying to get However, no parallel resonance capacitor is provided for the secondary winding N2A.

Although the embodiments have been described above, the present invention may have various modifications. For example, the present applicant has already proposed a configuration provided with a quadruple voltage rectifier circuit using a secondary-side series resonance circuit as a composite resonance type switching converter, but such a configuration is also a modification of the present embodiment. It can be established as an example. That is, the present embodiment is not particularly limited as the configuration of the secondary-side resonance circuit and the rectifier circuit.

[0105]

As described above, according to the present invention, the switching output obtained in the primary side resonance circuit is different from the power factor improvement circuit provided in the power supply circuit called the push-pull type switching frequency control type composite resonance type converter. Is fed back by the electrostatic coupling method or the magnetic coupling method through the primary side parallel resonance capacitor. This allows
There is an effect that the power factor is kept constant over a wide range with respect to fluctuations in the AC input voltage and the load power. For example, AC
Under a heavy load with a load power of 200 W or more in a 100 V system, the power factor is maintained substantially constant over a wide range with respect to fluctuations in the AC input voltage and the load power. For this reason, it becomes suitable as a power factor improving power supply circuit for office equipment and information equipment having a large load variation. Further, the circuit configuration does not become complicated as in the full bridge push-pull system.

Also, since there is little increase in the 50 Hz ripple voltage component of the DC output voltage, no special countermeasures against ripples are required, thereby improving the gain of the control circuit and increasing the capacity of the electrolytic capacitor. is there.

Further, since the maximum load power that can be handled by the power supply circuit does not decrease, it is not necessary to redesign the composite resonance type converter. In addition, the power conversion efficiency is slightly improved before the power factor is improved, and the operation waveform is a sine wave for both the voltage and the current.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a side sectional view showing a structure of an insulating converter transformer employed in the power supply circuit of the present embodiment.

FIG. 3 is an explanatory diagram showing each operation when the mutual inductance is + M / -M.

FIG. 4 is a waveform chart showing an operation of the switching power supply circuit according to the first embodiment.

FIG. 5 is a waveform chart showing an operation of the switching power supply circuit according to the first embodiment.

FIG. 6 is a characteristic diagram illustrating a relationship between load power and a power factor of the switching power supply circuit according to the first embodiment.

FIG. 7 is a characteristic diagram illustrating a relationship between an AC input voltage and a power factor of the switching power supply circuit according to the first embodiment.

FIG. 8 is a circuit diagram illustrating a configuration of a switching power supply circuit according to a second embodiment.

FIG. 9 is a circuit diagram showing a configuration of a switching power supply circuit according to a third embodiment.

FIG. 10 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 11 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 12 is a circuit diagram showing a configuration of a power supply circuit according to the prior art.

FIG. 13 is a characteristic diagram showing a relationship between load power and power factor for a power supply circuit of the prior art.

FIG. 14 is a characteristic diagram showing a relationship between an AC input voltage and a power factor in a power supply circuit according to the related art.

FIG. 15 is a waveform diagram showing an operation of a prior art power supply circuit with respect to an input of a commercial AC power supply according to load power.

[Explanation of symbols]

1 control circuit, 10, 11, 12 power factor improvement circuit, Di
Bridge rectifier circuit, Ci smoothing capacitor, D1 fast recovery diode, Cr1, Cr2 parallel resonance capacitor, C3 parallel resonance capacitor, C2 secondary parallel resonance capacitor, Cs1 secondary series resonance capacitor, PRT orthogonal control transformer, PIT insulation Converter transformer, Q1, Q2, Q21, Q22 switching element

Claims (3)

[Claims] A gap is formed between a rectifying / smoothing means that receives a commercial AC power supply to generate a rectified smoothed voltage and outputs the rectified smoothed voltage as a DC input voltage, and a required coupling coefficient that is loosely coupled. An insulating converter transformer provided for transmitting a primary side output to a secondary side; and first and second insulating converter transformers configured to intermittently output the DC input voltage by a push-pull operation to output to a primary winding of the insulating converter transformer. 2 switching means, and at least a leakage inductance component including a primary winding of the insulated converter transformer and a capacitance of a primary side parallel resonance capacitor, and the operation of the first and second switching means is a voltage resonance type. And the first and second primary side resonance circuits inserted into the rectified current path and The switching output obtained in one of the circuits is fed back through the primary parallel resonance capacitor in the one primary resonance circuit, and the rectified current is intermittently connected based on the returned switching output, thereby obtaining a power factor. A secondary-side resonance circuit formed on the secondary side by a leakage inductance component of the secondary winding of the insulating converter transformer, and a capacitance of the secondary-side resonance capacitor; A DC output voltage formed including a side resonance circuit, and configured to input an alternating voltage obtained to a secondary winding of the insulating converter transformer and perform a rectification operation to generate a secondary side DC output voltage. Generating means, and a constant voltage configured to perform constant voltage control on the secondary DC output voltage in accordance with the level of the secondary DC output voltage A switching power supply circuit, comprising: control means; 2. The power factor improving means is connected in parallel with a high-speed recovery type diode for interrupting a rectified current and a capacitor, and includes one of the first and second primary side resonance circuits. 2. The switching power supply circuit according to claim 1, wherein the primary parallel resonance capacitor is connected to an anode of the fast recovery diode. 3. The power factor improving means includes a series connection of a high-speed recovery type diode for interrupting a rectified current and an inductance, and the first and second primary side resonance circuits. 2. The switching power supply circuit according to claim 1, wherein one of the primary-side parallel resonant capacitors is connected to a connection point between the high-speed recovery type diode and an inductance. 3.