JP2000166235A - Switching power circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To miniaturize a resonant power circuit and reduce cost by forming a primary parallel resonant circuit out of a leakage inductance element including the primary winding of an insulated converter transformer and the capacitance of a parallel resonant capacitor so that a switching means may be operated on a voltage resonant type. SOLUTION: The primary winding of an insulated converter transformer PIT is divided into primary windings N1A, N1B by a center tap, the end of the primary winding N1A is connected to the collector of a switching element Q1, and the end of the primary winding NIB is connected to the collector of a switching element Q2. A parallel resonant capacitor Cr1 forms a parallel resonant circuit out of combined inductances L1A+Lc consisting of the leakage inductance element L1A of the primary winding N1A and inductance winding Lc so that a switching element Q1 may be operated on a voltage resonant type. A parallel resonant capacitor Cr2 also forms a parallel resonant circuit.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a circuit employing a switching converter of a type such as a flyback converter or a forward converter is widely known. Since these switching converters have a rectangular switching operation waveform, there is a limit in suppressing switching noise. Also, due to its operating characteristics,
It has been found that there is a limit in improving the power conversion efficiency. Therefore, the present applicant has previously proposed various switching power supply circuits using various resonance type converters.
The resonance type converter can easily obtain high power conversion efficiency and realize low noise because the switching operation waveform is sinusoidal. It also has the advantage that it can be configured with a relatively small number of parts.

FIG. 6 is a circuit diagram showing an example of a voltage resonance type switching power supply circuit which can be constructed based on the invention proposed by the present applicant. In the switching power supply circuit shown in this figure, for example, a commercial AC power supply in Japan or the United States is a so-called AC 100 V system, and corresponds to a condition that the maximum load power is 120 W or more.

The switching power supply circuit shown in FIG. 1 is provided with a so-called voltage doubler rectification circuit comprising rectification diodes Di1, Di2 and smoothing capacitors Ci1, Ci2 as a rectification and smoothing circuit for rectifying and smoothing the AC power supply AC. Can be In this voltage doubler rectifier circuit, assuming that a DC input voltage corresponding to, for example, one time the peak value of the AC input voltage VAC is Ei, a DC input voltage 2Ei that is approximately twice that of the DC input voltage Ei.
Generate For example, if the AC input voltage VAC is 144V, the DC input voltage 2Ei is about 400V. As described above, the reason why the voltage doubler rectifier circuit is employed as the rectifier / smoothing circuit is that the AC input voltage is 100 V AC as described above.
The system is designed to cope with a relatively heavy load condition in which the maximum load power is 120 W or more. That is, by increasing the DC input voltage to twice the normal value, the amount of current flowing into the subsequent switching converter is suppressed, and the reliability of the components forming the switching power supply circuit is ensured. . In the voltage doubler rectifier circuit shown in this figure, an inrush current limiting resistor Ri is inserted in the rectified current path, for example, so as to suppress the inrush current flowing into the smoothing capacitor when the power is turned on. I have.

The voltage-resonant type switching converter shown in FIG. 1 has a self-excited type provided with a single switching element Q1. In this case, a high withstand voltage bipolar transistor (BJT;
Junction type transistor). The base of the switching element Q1 is connected to the positive side of the smoothing capacitor Ci1 (rectified smoothed voltage 2Ei) via a starting resistor RS so that a base current at the time of starting can be obtained from the rectified smoothing line. A self-oscillation resonance circuit including a damping resistor RB, an inductor LB, a resonance capacitor CB, and a drive winding NB is connected in series between the base of the switching element Q1 and the temporary ground. In this case, the drive winding NB is connected to the insulation converter transformer PIT (Power Is
olation Transformer) and the inductor L
Together with B, a required inductance for setting the switching frequency is obtained. Further, the base of the switching element Q1 and the negative electrode (1
A path of a damper current flowing when the switching element Q1 is turned off is formed by a clamp diode DD inserted between the secondary side ground), and the collector of the switching element Q1 is connected to the insulating converter transformer P.
It is connected to one end of the IT primary winding N1, and the emitter is grounded.

A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr
Is the combined capacitance (L1, LR) obtained by connecting the primary winding N1 of the isolated converter transformer PIT described later and the controlled winding NR of the orthogonal control transformer PRT (Power Regulating Transformer) in series. Thereby, a parallel resonance circuit of the voltage resonance type converter is formed. Although the detailed description is omitted here, when the switching element Q1 is turned off, the voltage Vd across the resonance capacitor Cr is obtained by the action of the parallel resonance circuit.
cr is actually a sinusoidal pulse waveform so that a voltage resonance type operation can be obtained.

The insulated converter transformer PIT is for transmitting the switching output of the switching element Q1 to the secondary side. In this case, the insulated converter transformer PIT is used.
One end of the primary winding N1 of T is connected to the collector of the switching element Q1, and the other end is connected in series with the controlled winding NR of the orthogonal control transformer PRT as shown in the figure.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, a parallel resonance circuit is formed by connecting the secondary side parallel resonance capacitor C2 to the secondary winding N2 in parallel. With this parallel resonance circuit,
The alternating voltage excited in the secondary winding N2 is a resonance voltage,
This resonance voltage is supplied to two sets of half-wave rectifier circuits, a half-wave rectifier circuit including a rectifier diode DO1 and a smoothing capacitor CO2, and a half-wave rectifier circuit including a rectifier diode DO2 and a smoothing capacitor CO2. Then, the DC output voltages EO1 and EO2 are obtained by these two sets of half-wave rectifier circuits, respectively. In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1.
In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1. The rectifier diodes DO1 and DO2 forming the half-wave rectifier circuit use a high-speed rectifier to rectify an alternating voltage in a switching cycle.

The control circuit 1 compares, for example, a DC voltage output on the secondary side with a reference voltage and outputs a DC current corresponding to the error.
The control winding N of the orthogonal control transformer PRT is used as the control current.
This is the error amplifier that supplies C. In this case, the DC voltage output EO1 is input to the control circuit 1 as the detection voltage, and the DC output voltage EO2 is input as the operation power supply.

For example, when the DC output voltage EO2 on the secondary side fluctuates due to the fluctuation of the AC input voltage VAC or the minimum load power, the control circuit 1 controls the control current flowing through the control winding NC to 10 mA to 40 mA, for example. Vary in range.
Thus, the inductance LR of the controlled winding NR is changed in a range of, for example, 0.1 mH to 0.6 mH.

Since the controlled winding NR forms a parallel resonance circuit for obtaining a voltage-resonant switching operation as described above, this parallel resonance circuit is not controlled for a fixed switching frequency. The resonance condition of the circuit is changed. At both ends of the parallel connection circuit of the switching element Q1 and the parallel resonance capacitor Cr, a sinusoidal resonance pulse is generated by the action of the parallel resonance circuit corresponding to the OFF period of the switching element Q1, but the resonance condition of the parallel resonance circuit , The width of the resonance pulse is variably controlled. That is, PWM (P
(ulse Width Moduration) control operation is obtained. The PWM control of the width of the resonance pulse means the control of the off period of the switching element Q1, which in other words means variably controlling the on period of the switching element Q1 under the condition of a fixed switching frequency. . By variably controlling the ON period of the switching element Q1 in this manner, the switching output transmitted from the primary winding N1 forming the parallel resonance circuit to the secondary side changes, and the DC output voltage ( The output levels of EO1, EO2) are also changed. As a result, the secondary DC voltage (EO1, EO2) can be made constant. In addition, such a constant voltage control method is herein referred to as a “primary-side voltage resonance pulse width control method”.

FIG. 7 is a circuit diagram showing another example of a voltage resonance type switching power supply circuit which can be constructed based on the invention proposed by the present applicant. In FIG. 7, the same parts as those in FIG. 6 are denoted by the same reference numerals, and the description of the parts having the same configuration will be omitted. In the power supply circuit shown in this figure, an example is shown in which the controlled winding of the orthogonal control transformer PRT is provided on the secondary side. In this case, two controlled windings NR and NR1 are wound and provided as controlled windings of the orthogonal control transformer PRT, and the controlled winding NR is connected to an end of the secondary winding N2. The rectifier diode DO1 is connected so as to be inserted in series between the anodes. The controlled winding NR1 is inserted in series between the tap output of the secondary winding N2 and the anode of the rectifier diode DO2. In such a connection form, the secondary-side parallel resonance circuit includes the controlled windings NR and NR1.
(LR, LR1).

As described above, in the case where the controlled windings (NR, NR1) of the orthogonal control transformer PRT are provided on the secondary side, the inductance of the controlled windings NR, NR1 is determined as an inductance control method. Is varied, so that the pulse width of the resonance voltage V2 of the secondary parallel resonance capacitor C2, that is, the conduction angle of the secondary rectifier diode is variably controlled. Thereby, constant voltage control is achieved by controlling the output level obtained on the secondary side (secondary side resonance voltage pulse control method). In this power supply circuit, the switching frequency is fixed.

Here, the structure of the insulating converter transformer PIT provided in the power supply circuit shown in FIGS. 6 and 7 is shown in FIG.
Is shown in cross section. Insulation converter transformer PIT
Are E-type cores CR1 and CR2 made of ferrite material, for example.
Are formed such that the magnetic legs face each other to form an EE-type core. At this time, no gap is formed in the center magnetic leg as shown in the figure. Then, the primary winding N1 (and the driving winding NB) is applied to the center magnetic leg using the bobbin B,
The secondary winding N2 is wound in a divided state. As a result, a state of loose coupling (for example, coupling coefficient k） 0.9) is obtained between the primary winding N1 and the secondary winding N2.

In the isolated converter transformer PIT, the polarities (winding direction) of the primary winding N1 and the secondary winding N2 are set.
And the connection of the rectifier diodes DO (DO1, DO2), the mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 becomes + M and -M. There is. For example, when the connection configuration shown in FIG. 10A is employed, the mutual inductance is + M, and when the connection configuration shown in FIG. 10B is employed, the mutual inductance is -M.

In the case of the orthogonal control transformer PRT of the power supply circuit shown in FIG. 6, since the controlled winding NR is inserted on the primary side, the control winding connected directly to the secondary side is controlled. Since the insulation distance from the NC must be ensured, a corresponding size is required. However, in the orthogonal control transformer PRT of the power supply circuit shown in FIG. 7, since the controlled winding NR1 is provided on the secondary side, Since the insulation distance from the control winding NC is not required, the size of the orthogonal control transformer PRT can be further reduced.

FIG. 11 shows operating waveforms of the power supply circuit shown in FIGS. 6 and 7 according to the switching cycle. FIG.
1 (a) shows the resonance voltage Vcr obtained at both ends of the switching element Q1 // parallel resonance capacitor Cr. FIG. 11B shows a rectifier diode DO1 on the secondary side.
11C shows a rectified current ID2 flowing through the rectifier diode DO1. In this figure, periods TON and TOFF indicate periods when the switching element Q1 is turned on and off, respectively, and periods DON and TON
DOFF indicates a period during which the rectifier diode DO1 is turned on and off, respectively.

The resonance voltage Vcr generated at both ends of the switching element Q1 / parallel resonance capacitor Cr is shown in FIG.
As shown in FIG. 1A, a sinusoidal pulse waveform is obtained during the period TOFF when the switching element Q1 is off, and the operation of the switching converter is of the voltage resonance type. The level of the pulse of the resonance voltage Vcr is about 1800 V at the peak as shown in the figure. This is because the impedance of the parallel resonance circuit on the primary side of the voltage resonance converter acts on the DC input voltage of 2Ei obtained by the voltage doubler rectification. In addition, as a secondary-side operation, the secondary-side rectifier diode DO1 has a period DON according to the waveform shown in FIG.
Thus, an operation in which a rectified current flows can be obtained. This is
The operation is based on + M (polarity mode) described with reference to FIG. As a result of this rectification operation, the resonance voltage V2 obtained at both ends of the secondary side parallel resonance capacitor C2 is, as shown in FIG. 11B, a DC output during the period DOFF when the rectification diode DO1 is off. A period DON during which a sinusoidal voltage having a peak level of about 2 to 3.5 times the voltage EO (EO1, EO2) is applied and the rectifier diode DO1 is turned on
To the DC output voltage EO (EO1, EO2).

FIG. 8 shows another example of a voltage resonance type switching power supply circuit which can be constructed based on the invention proposed by the present applicant. In FIG. 8,
6 and 7 are denoted by the same reference numerals, and the description of the parts having the same configuration will be omitted. The switching power supply circuit shown in FIG. 1 is designed so that, for example, a commercial AC power supply in Europe or the like is a so-called AC 200 V system and the maximum load power is about 120 W or more.

In the power supply circuit shown in FIG. 1, a full-wave rectification circuit including a bridge rectification circuit Di and a smoothing capacitor Ci is provided as a rectification and smoothing circuit for receiving an AC input voltage VAC and obtaining a DC input voltage. A rectified smoothed voltage Ei corresponding to a level that is one time the AC input voltage VAC is generated. This is the commercial AC power supply AC
Since the AC input voltage VAC of the AC 200 V system is supplied, there is no need to provide a voltage doubler rectifier circuit as in the power supply circuits shown in FIGS. 6 and 7.

The switching power supply circuit shown in FIG. 1 includes a choke coil CH. The choke coil CH has a configuration in which the inductor Lc and the detection drive winding NB are transformer-coupled. Detection drive winding NB
Then, an alternating voltage corresponding to the switching period is excited by the switching output transmitted from the primary winding N1 of the orthogonal type insulating converter transformer PRT to the inductor Lc. The self-excited oscillation circuit (NB, CB, LB, RB) for driving the switching element Q1 generates a drive current for driving the switching element Q1 at a fixed switching frequency based on the alternating voltage. Generate.

The orthogonal type insulated converter transformer PRT is
It has a function of transmitting the switching output of the switching element Q1 to the secondary side and performing constant voltage control of the secondary side output. As a structure of the orthogonal type insulating converter transformer PRT, a three-dimensional core is formed by joining the ends of two magnetic legs of two double U-shaped cores having four magnetic legs. Then, the primary winding N1 and the secondary winding N2 are wound around predetermined two magnetic legs of the three-dimensional core in the same winding direction, and further, the control winding NC is connected to the primary windings N1 and N1. By being wound around predetermined two magnetic legs so as to be in a winding direction orthogonal to the secondary winding N2, it is configured as a saturable reactor. In this case, the opposing surfaces of the magnetic legs of the double U-shaped core are joined, and no gap is formed.

The above orthogonal type insulated converter transformer PRT
One end of the primary winding N1 is connected to the collector of the switching element Q1, and the other end is connected to the choke coil C as shown in the figure.
It is connected to the positive electrode (rectified smoothed voltage 2Ei) of the smoothing capacitor Ci via the series connection of the inductor Lc of H.

On the secondary side of the orthogonal insulation converter transformer PRT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. The alternating voltage excited by the secondary winding N2 by this parallel resonance circuit becomes a resonance voltage. That is, a voltage resonance operation is obtained on the secondary side. For the secondary parallel resonance circuit formed as described above, a center tap is provided for the secondary winding N2, and the rectifier diodes DO1, DO2, DO3, DO4 and the smoothing capacitors CO1, CO2 are provided. As shown in the figure, a set of [rectifier diodes DO1, DO2, smoothing capacitor CO1] and [rectifier diodes DO3, DO4, smoothing capacitor CO1]
2], two sets of full-wave rectifier circuits are provided.
A full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] receives a resonance voltage supplied from the secondary parallel resonance circuit to generate a DC output voltage EO1, and generates a [rectifier diode DO3, DO4, smoother. Similarly, the full-wave rectifier circuit composed of the capacitor CO2] receives the resonance voltage supplied from the secondary parallel resonance circuit and generates the DC output voltage EO2. Also in this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1.

The control circuit 1 supplies a DC current, the level of which is varied according to the level of the DC voltage output EO1 on the secondary side, to the control winding NC of the quadrature-type insulating converter transformer PRT. The constant voltage control is performed as described.

For example, when the DC output voltage EO1 on the secondary side fluctuates with the fluctuation of the AC input voltage VAC or the load power, the control circuit 1 changes the control current flowing through the control winding NC within a required range. . Since the control winding NC is wound around the orthogonal type insulated converter transformer PRT, the leakage inductance (L1, L2) in the orthogonal type insulated converter transformer PRT which is a saturable reactor.
Is obtained. As described above, the primary winding N
The leakage inductance L1 of 1 forms a parallel resonance circuit on the primary side and the leakage inductance L2 of the secondary winding N2 forms a parallel resonance circuit on the secondary side. , The inductances L1 and L2 are variably controlled. This operation changes the resonance impedance of the primary side and the secondary side, so that the switching output transmitted from the primary side to the secondary side also changes. As a result, the output of the secondary side DC voltage (EO1, EO2) changes. A constant voltage is achieved. Note that such a constant voltage control method is hereinafter referred to as a “parallel resonance frequency control method”.

[0027]

By the way, in the voltage resonance type converter having the configuration shown in FIGS. 6 and 7,
AC input voltage VAC is AC100V system and maximum load power is 1
In order to cope with the condition of 20 W or more, a DC input voltage of 2Ei level is obtained by a voltage doubler rectification method. Therefore, as shown in FIG. 10, the resonance voltage Vcr of 1800 V is provided across the switching element Q1 and the parallel resonance capacitor Cr when the switching Q1 is off.
Occurs. Therefore, for the switching element Q1 and the parallel resonance capacitor Cr, it is required to select a high withstand voltage product of 1800V. In this case, especially for the switching element Q1, the saturation voltage VCE (SAT) and the storage time t
Since the STG and the fall time tf are large and the current amplification factor hFE is small, the switching loss and the drive power increase, and the power loss as a power supply circuit increases accordingly.

In the configuration shown in FIG. 6 and FIG.
0 (b), the DC output voltage EO (EO1,
A secondary voltage V2 that is about twice to about 3.5 times EO2) is applied. For this reason, the rectifier diodes (DO1, DO2) used in the half-wave rectifier circuit have a DC output voltage of 135, for example.
If the voltage is about V, a high withstand voltage product of about 500 V is required, and accordingly, the forward voltage drop VF and the reverse recovery time trr
Becomes large, which causes an increase in power loss. For example, if the switching frequency is increased, it is possible to reduce the size of various component elements. However, it is difficult to set the switching frequency fs to a high value due to the above-described circumstances. It has been found that the conversion efficiency decreases. Further, as described above, a voltage doubler rectifier circuit is required to obtain a highly reliable DC input voltage, so that two relatively large smoothing capacitors are required and the substrate area is increased.

In the power supply circuits shown in FIGS. 6 and 7,
There is a limit to the improvement of the control sensitivity of the constant voltage control circuit system. To compensate for this, the controlled winding N of the orthogonal control transformer PRT is used.
It is necessary to secure the required constant voltage control range by expanding the variable range of the inductance of R. For this purpose, it is necessary to increase the number of turns of the controlled winding NR, so that the orthogonal control transformer PRT becomes large and expensive.

On the other hand, in the power supply circuit shown in FIG.
In order to obtain a DC input voltage by using the AC 200 V system, a normal full-wave rectifier circuit that performs equal-size voltage rectification, such as a bridge rectifier circuit, may be used.

However, also in the circuit shown in FIG. 8, under the condition that the AC input voltage is increased by about 20% from the nominal value,
It is inevitable that a resonance voltage of about 1800 V is applied to both ends of the switching element Q1 // parallel resonance capacitor Cr.

[0032]

SUMMARY OF THE INVENTION Accordingly, the present invention has been made in view of the above-mentioned problems, and has been developed in view of the promotion of reduction in size and weight and cost of a resonance-type power supply circuit, as well as sensitivity to constant voltage control and power conversion efficiency. It is an object of the present invention to improve various characteristics as a resonance type power supply circuit.

Therefore, as a power supply circuit of the present invention, a rectifying / smoothing means which receives a commercial AC power supply, generates a rectified smoothed voltage having a level corresponding to the same level as the commercial AC power supply level, and outputs it as a DC input voltage. A gap is formed so as to obtain a required coupling coefficient that is loosely coupled, and an insulating converter transformer provided for transmitting the primary output to the secondary side, and a DC input including two sets of switching elements. A switching unit configured to intermittently output a voltage by a push-pull operation and output the voltage to a primary winding of an isolated converter transformer, and at least a leakage inductance component including a primary winding of the isolated converter transformer and a capacitance of a parallel resonant capacitor. And a primary-side parallel resonance circuit formed to make the operation of the switching means a voltage resonance type. And the. Further, the secondary-side configuration includes a secondary-side resonance circuit formed by a leakage inductance component of the secondary winding of the insulating converter transformer and a capacitance inserted into the secondary-side, and a secondary-side resonance circuit. And a DC output voltage generating means configured to input an alternating voltage obtained in the secondary winding of the insulated converter transformer and perform a full-wave rectification operation to generate a secondary DC output voltage. . Further, as the constant voltage control means,
The constant frequency control is performed by variably controlling the switching frequency of the switching means according to the level of the secondary DC output voltage.

According to the above configuration, the primary side is provided with a resonance type converter for performing a push-pull operation by two sets of switching elements while providing a parallel resonance circuit for performing a voltage resonance type switching operation. Thus, the isolation converter transformer is loosely coupled. Then, on the secondary side, a secondary-side DC output voltage is generated by a full-wave rectifier circuit to supply power to a load. That is, in the present embodiment, the resonance circuit is provided on the primary side and the secondary side, and for a required load condition, the electromagnetic energy obtained by the series resonance operation on the primary side and the resonance operation on the secondary side is obtained. Utilizing this, a direct-current output voltage is generated by a full-wave rectifier circuit to cope with the situation. Accordingly, the primary side is not a voltage doubler rectifier circuit, but has an AC input voltage level of 1.
The configuration includes a full-wave rectifier circuit that generates a rectified smoothed voltage corresponding to twice.

[0035]

FIG. 1 is a circuit diagram showing a configuration of a power supply circuit according to a first embodiment of the present invention. The power supply circuit shown in this figure is provided with a voltage resonance type switching converter which includes two switching elements and performs a push-pull operation as described later. In this figure, the same parts as those in FIGS. 6 to 8 are denoted by the same reference numerals, and description thereof will be omitted.

In the power supply circuit according to the present embodiment shown in this figure, as a rectifying and smoothing circuit for inputting an AC input voltage VAC and obtaining a DC input voltage, a full-wave circuit comprising a bridge rectifier circuit Di and a smoothing capacitor Ci. A rectifier circuit is provided to generate a rectified smoothed voltage Ei corresponding to a level that is one-time the AC input voltage VAC. That is, in the present embodiment, a voltage doubler rectifier circuit is not provided as in the related art. In this specification, a rectifier circuit that generates a rectified smoothed voltage Ei corresponding to one time of the level of the AC input voltage VAC as shown in FIG. 1 is also referred to as an “equal voltage rectifier circuit”. .

In the power supply circuit shown in FIG.
An orthogonal drive transformer PRT is provided for switchingly driving two sets of switching elements Q1 and Q2 to be described later by a push-pull operation and variably controlling the switching frequency of the switching elements Q1 and Q2. The structure of the orthogonal drive transformer PRT is, for example, 4
A three-dimensional core is formed by joining the ends of the two magnetic legs of two double U-shaped cores having two magnetic legs.
Then, for two predetermined magnetic legs of the three-dimensional core,
The detection winding ND and the drive windings NB1 and NB2 are wound in the same winding direction, and the control winding NC is wound in a direction orthogonal to the detection winding ND and the drive windings NB1 and NB2. This constitutes a saturable reactor.

In this case, the detection winding ND is connected to the primary winding N1A.
And the collector of the switching element Q1 are inserted in series.

In this case, the driving windings NB1 and NB2 are formed by dividing one winding by center tapping the ground with respect to the ground. The end of the drive winding NB1 is connected to the base of the switching element Q1 via a series connection of the base current limiting resistor RB1 and the resonance capacitor CB1. The end of the drive winding NB2 is connected to the base of the switching element Q2 via the base current limiting resistor RB2 and the resonance capacitor CB2. That is, a self-excited oscillation drive circuit for the switching element Q1 is formed by the drive winding NB1-base current limiting resistor RB1-resonance capacitor CB1,
The drive winding NB2-base current limiting resistor RB2-resonant capacitor CB2 forms a self-excited oscillation drive circuit for the switching element Q2. In the detection winding ND, an alternating voltage corresponding to a switching output is detected by a switching operation described later. In the drive windings NB1 and NB2, alternating voltages of opposite polarities 180 ° out of phase with each other are obtained in accordance with the switching output detected by the resonance current detection winding ND.

In this embodiment, two switching elements Q1 and Q2 are provided for the push-pull operation. The switching element Q1 includes the drive circuit (drive winding NB1
-Base current limiting resistor RB1-Resonant capacitor CB1), clamp diode DD1, parallel resonant capacitor Cr1
Are connected as shown in the figure, and a drive circuit (drive winding NB2-base current limiting resistor RB2-resonance capacitor CB2), a clamp diode DD2, and a parallel resonance capacitor Cr2 are connected to the switching element Q2 as shown in the figure. .

In this case, the insulation converter transformer PIT
Is divided into primary windings N1A and N1B by a center tap, and an end of the primary winding N1A is connected to a collector of the switching element Q1. Primary winding N1B
Is connected to the collector of the switching element Q2 as described above. The taps of the primary windings N1A and N1B are connected to the positive electrode of the smoothing capacitor Ci via a series connection of the inductance winding Lc of the choke coil CH. The emitters of the switching elements Q1 and Q2 are connected to the primary side ground.

In this case, the above-described parallel resonance capacitor C
r1 is the leakage inductance component of the primary winding N1A (L1
A) and the combined inductance (L1A + Lc) of the inductance winding Lc form a parallel resonance circuit for causing the switching element Q1 to operate in a voltage resonance type. Similarly, the parallel resonance capacitor Cr2 is connected to the primary winding N1B.
By using the leakage inductance component (L1B) and the combined inductance (L1A + Lc) of the inductance winding Lc, a parallel resonance circuit for making the switching element Q2 operate in a voltage resonance type is formed.

In such a configuration on the primary side, the driving winding N
Since alternating voltages of opposite polarities are obtained in B1 and the driving winding NB2, the self-excited oscillation driving circuit including the driving winding NB1 and the self-excited oscillation driving circuit including the driving winding NB2 respectively have opposite polarities. , A drive current (base current) as an alternating current flows through each base of the switching elements Q1 and Q2. Thereby, the switching element Q1,
As for Q2, an operation of alternately turning on / off at a switching frequency determined by a constant of the self-excited oscillation driving circuit is obtained. That is, a switching operation by push-pull is obtained in a voltage resonance type. Switching element Q1
Is supplied to the primary winding N1A via the detection winding ND, and the switching output of the switching element Q2 is supplied to the primary winding N1B. Then, the current flows to the smoothing capacitor Ci via the inductance winding Lc of the choke coil CH. By providing the voltage resonance type converter that performs the push-pull operation in this manner, the present embodiment can cope with a maximum load power of, for example, 200 W or more.

The insulating converter transformer PIT provided in the power supply circuit shown in FIG.
For example, there is provided an EE-type core in which E-type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other. The EE-type core is wound around a central magnetic leg on a primary side and a secondary side. The primary winding N1 and the secondary winding N2 (N2A, N2B) are wound in a state of being divided using the divided bobbin B in which the mounting portion is divided. In the present embodiment, a gap G is formed for the center magnetic leg as shown in the figure. This gap G is the E type core CR
1. The center magnetic leg of CR2 can be formed by forming it shorter than the two outer magnetic legs. In the present embodiment, the gap G has a required gap of 1 mm or more. As a result, for example, a loose coupling is achieved by a coupling coefficient smaller than that of the insulating converter transformer PIT shown in FIG. 9 as a conventional example, so that a saturated state is hardly obtained. The coupling coefficient k in this case is, for example, k ≒ 0.85.

In the present embodiment, the same configuration as that described above with reference to FIG. 8 is employed on the secondary side of the insulating converter transformer PIT. In other words, a set of [rectifier diodes DO1, DO2, smoothing capacitor CO1]
There are provided two sets of center-tap full-wave rectifier circuits including a set of [rectifier diodes DO3, DO4, smoothing capacitor CO2]. The secondary parallel resonance circuit is formed by the leakage inductance component L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2.
Here, the primary side parallel resonance circuit (N1A + Lc // Cr,
N1B + Lc // Cr) is the series resonance frequency of fo1, and the parallel resonance frequency of the secondary side parallel resonance circuit is fo1.
Assuming that 2, fo1 ≒ fo2, the capacitance of the secondary-side series resonance capacitor Cs is selected.

When the orthogonal drive transformer PRT is provided as in the present embodiment, the following operation is obtained. Also in this case, the control circuit 1 varies the level of the control current (DC current) flowing through the control winding NC according to a change in the secondary DC output voltage level caused by a change in the load power or the primary DC input voltage. By doing so, the inductances LB1 and LB2 of the drive windings NB1 and NB2 wound on the orthogonal drive transformer PRT are variably controlled. Therefore, in this case, the driving winding N
The resonance condition of the series resonance circuit in the self-excited oscillation driving circuit for the switching element Q1 formed including the inductance LB1 of B1 and the switching condition for the switching element Q2 formed including the inductance LB2 of the driving winding NB2 are shown. The resonance conditions of the series resonance circuit in the self-excited oscillation drive circuit change together.
As a result, the switching element Q
An operation of varying the switching frequency of Q1 and Q2 is obtained, and this operation stabilizes the secondary DC output voltage.

As described above, in the power supply circuit of the present embodiment, the primary side is provided with the parallel resonance circuit for making the switching operation a voltage resonance type, and the secondary side is connected with the parallel resonance circuit. Provided full-wave rectifier circuit. And
The constant voltage control operation is configured to variably control the switching frequency of the primary-side voltage resonance type converter. In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

According to the configuration of the power supply circuit described so far, the full-wave rectifier circuit on the secondary side performs full-wave rectification utilizing the state where the mutual inductance is in the operation mode of + M and -M. You. This rectifying operation is performed by transmitting electromagnetic energy transmitted from the primary winding to the secondary winding of the isolated converter transformer PIT, which has been loosely coupled (coupling coefficient k = 0.85), to a primary-side parallel resonance circuit (voltage resonance). circuit)
And a parallel resonance circuit (voltage resonance circuit) on the secondary side, thereby reducing the resonance impedance of the resonance circuit between the primary side and the secondary side to reduce the secondary side. It also operates to amplify the resonance current. Then, the power supplied to the load side also increases due to the amplifying action of the secondary side resonance current, thereby increasing the maximum load power. That is, in the present embodiment, the push-pull operation of the voltage resonance type converter on the primary side and the operation as the above-described composite resonance type converter are used in combination to realize a large increase in the maximum load power.

Also, by increasing the maximum load power as described above, in the present embodiment, the rectifying and smoothing circuit for generating the DC input voltage needs to adopt the double voltage rectification method to cover the load power. There is no. For this reason, as described with reference to FIG. 1, for example, the configuration of a normal equal-voltage rectifier circuit using a bridge rectifier circuit can be employed. Thereby, for example, the rectified smoothed voltage Ei when the AC input voltage VAC = 144 V becomes lower than 200 V. For example, the resonance voltage Vcr obtained at both ends of the parallel connection circuit of the switching element Q1 // secondary-side parallel resonance capacitor Cr changes the switching element Q1 by the action of the primary-side parallel resonance circuit on the rectified smoothed voltage Ei. Occurs when off,
In this embodiment, the resonance voltage V
cr is about 1/1 of the resonance voltage Vcr (1800 V) generated in the power supply circuit of the conventional example shown in FIGS.
It will be suppressed to about 2. That is, the resonance voltage Vc
r is suppressed to about 700 V at the peak. Therefore, in the present embodiment, the switching element Q1 and the parallel resonance capacitor Cr are, for example, 9
What is necessary is to select a withstand voltage product of about 00V.

Also, on the secondary side, a full-wave rectifier circuit that performs a rectification operation during both the positive and negative periods of the excitation voltage of the secondary winding N2A is provided. An operation is obtained in which the voltage applied to both ends of the diode is clamped to the level of the secondary DC output voltage EO. As a result, as the rectifier diode forming the rectifier circuit on the secondary side, a withstand voltage product corresponding to the level of the secondary side DC output voltage EO may be selected.

Further, in the present embodiment, as described above, the switching element Q1 and the parallel resonance capacitor C
As for the rectifier diode on the secondary side, a low breakdown voltage product can be used as compared with a conventional rectifier diode, so that the element is less expensive. Therefore, for example, the switching element Q
1 and secondary side rectifier diodes having improved characteristics (the switching element Q1 can be used to obtain the saturation voltage VCE
(SAT), accumulation time tSTG, fall time tf, current amplification factor hFE
A rectifier diode having good characteristics such as a forward voltage drop VF and a reverse recovery time trr can be selected. Due to such improvement in characteristics, in the present embodiment, the switching frequency can be set higher than in the conventional case, and accordingly, reduction of power loss and reduction in size and weight of various components are promoted. That is, it is possible to improve various characteristics such as power conversion efficiency as compared with the related art, and to promote reduction in size, weight, and cost. In actuality, the range from the maximum load power to the minimum load power is 9
High efficiency of 0% or more is obtained.

Further, from the viewpoint of reducing the size and weight of the power supply circuit, the conventional configuration including the voltage doubler rectifier circuit for generating the DC input voltage requires two sets of rectifier diodes and a smoothing capacitor. However, in the present embodiment, for example, a full-wave rectifier circuit using a normal bridge rectifier circuit can be used, so that a set of a block-type smoothing capacitor and a bridge rectifier diode can be employed. Therefore, cost reduction and downsizing of parts can be achieved.

Further, since the primary side rectifier circuit is the same-size voltage rectifier circuit, the number of turns of the primary winding N1 is reduced as compared with the conventional case, and as a result, the orthogonal drive transformer PR
Even if the variable range of the inductance by T is narrower than in the past, the constant voltage control is performed stably. As a result, the number of windings of the control winding NC and the like wound around the orthogonal drive transformer PRT is reduced, and the orthogonal drive transformer PR is reduced.
T can be reduced in size and weight and cost can be reduced.

Further, as a result of variably controlling the switching frequency as a constant voltage control operation on the primary side, the resonance impedance of the parallel resonance circuit on the primary side and the resonance impedance of the two parallel resonance circuits on the secondary side are controlled simultaneously. Operation is obtained. Therefore, the control sensitivity for the constant voltage control is improved as compared with the conventional case, and for example, even under the condition of the same switching frequency variable width (control current variable width) as in the conventional case, the variation in the AC input voltage and the load power is suppressed. The control range is expanded. This also makes it possible to reduce the number of turns of the control winding NC and the like, thereby contributing to downsizing of the orthogonal drive transformer PRT.

FIG. 3 is a circuit diagram showing a configuration of a power supply circuit according to a third embodiment of the present invention. The same parts as those in FIG.

In the present embodiment, the structure of the primary side is the same as that of the circuit of FIG. 1, and the insulating converter transformer PIT has the same structure as that described above with reference to FIG. However, in this case, the secondary windings N2A and N2B are independently wound as secondary windings wound around the insulating converter transformer PIT.

One end of the secondary winding N2A is connected to the secondary side ground, and the other end is connected to the anode of the rectifier diode DO1 and the rectifier diode DO2 via a series connection of a series resonance capacitor Cs.
Is connected to the connection point of the cathode. The cathode of the rectifier diode DO1 is connected to the positive electrode of the smoothing capacitor CO1, and the anode of the rectifier diode DO2 is connected to the secondary side ground. The negative side of the smoothing capacitor CO1 is connected to the secondary side ground. In such a connection form,
A voltage doubler full-wave rectifier circuit comprising a set of [secondary winding N2A, series resonance capacitor Cs, rectifier diodes DO1, DO2, smoothing capacitor CO1] is provided. Here, the series resonance capacitor Cs forms a series resonance circuit corresponding to the on / off operation of the rectifier diodes DO1 and DO2 by its own capacitance and the leakage inductance component of the secondary winding N2A. Also here, assuming that each series resonance frequency of the primary-side parallel resonance circuit (N1A + Lc // Cr, N1B + Lc // Cr) is fo1, and that the parallel resonance frequency of the secondary-side series resonance circuit is fo2, fo1 ≒ The capacitance of the secondary-side series resonance capacitor Cs is selected so as to be fo2.

In the secondary winding N2B, a normal full-wave rectifier circuit composed of rectifier diodes D05 and D06 and a smoothing capacitor CO2 is formed with the center tap provided on the secondary winding N2B grounded. As a result, the secondary side DC output voltage EO2 is obtained.

As described above, the power supply circuit of the present embodiment
A composite resonance type switching converter in which a primary side is provided with a series resonance circuit for making a switching operation a current resonance type, and a secondary side is provided with a series resonance circuit for obtaining a voltage doubled full-wave rectification operation. Is configured as

[Second winding N2A, series resonance capacitor Cs1, rectifier diodes DO1, DO2, smoothing capacitor CO]
The double voltage full-wave rectification operation by the set [1] is as follows. When a switching output is obtained on the primary winding N1 by the switching operation on the primary side, this switching output is excited by the secondary winding N2A. During the period when the rectifier diode DO1 is turned off and the rectifier diode DO2 is turned on, the primary winding N1 and the secondary winding N2A operate in the depolarization mode in which the polarity is -M, and the secondary winding N2A is operated. The operation of charging the series resonance capacitor Cs1 with the rectified current IC2 rectified by the rectifier diode DO2 is obtained by the series resonance effect of the leakage inductance of N2A and the series resonance capacitor Cs1. During the period in which the rectifier diode DO2 is turned off and the rectifier diode DO1 is turned on to perform the rectification operation, the polarity of the primary winding N1 and the secondary winding N2 becomes + M, and the secondary polarity winding is performed. Line N2A
In the state where the series resonance occurs in which the potential of the series resonance capacitor Cs1 is added to the voltage induced in the smoothing capacitor C
This is an operation for charging O1. As described above, the rectifying operation is performed by using both the polarity adding mode (+ M; forward operation) and the depolarizing mode (-M; flyback operation), so that the smoothing capacitor CO1 is obtained.
In this case, a DC output voltage EO1 corresponding to almost twice the induced voltage of the secondary winding N2 is obtained.

As described above with reference to FIG. 2, the configuration for obtaining the above-mentioned voltage doubler full-wave rectifying operation is to form a gap G with respect to the insulating converter transformer PIT to achieve loose coupling by a required coupling coefficient. This is achieved by obtaining a state in which the isolated converter transformer PIT is more difficult to be saturated. For example, when the gap G is not provided for the insulating converter transformer PIT as in the related art, there is a high possibility that the insulating converter transformer PIT becomes saturated during flyback operation and the operation becomes abnormal. It is difficult to desirably perform the voltage doubling rectification operation as described in the first embodiment.

The waveform diagram of FIG. 4 shows the operation of the main part of the power supply circuit shown in FIG. 3 by the switching cycle.
4 (a) to 4 (g) show the operation waveforms of the respective components when the maximum load power is 240W.
(H) to FIG. 4 (n) show the minimum load power 60
4A to 4G show operation waveforms at the time of W.

In the case of the maximum load power of 240 W, the switching elements Q1 and Q2 perform the switching operation at the switching frequency substantially corresponding to the lower limit of the switching frequency control range. The switching operations of the switching elements Q1 and Q2 at this time are shown in FIGS. 4 (a) to 4 (d). According to this figure, the switching element Q1 is in the period T1
To turn on during the period T2, and the voltage Vc1 between the collector and the emitter of the switching element Q1 becomes a sinusoidal pulse in the period T1 as shown in FIG. And the primary winding N1A
As shown in FIG. 4 (b), the switching current I1 flowing from the switching element Q1 through the detection winding ND to the collector of the switching element Q1 reverses from the positive level to the negative level during the period T1 (off-period). , A smooth waveform close to a sine wave, which changes so as to reverse from the negative level to the positive level in response to the turn-on, is obtained.

The switching element Q2 operates so as to be turned off in the period T3 and turned on in the period T4.
The periods T3 and T4 correspond to the periods T1 and T2 corresponding to the on / off timing of the switching element Q1, respectively.
Is obtained at a timing having a phase difference of 180 °. Therefore, the collector-emitter voltage Vc2 of the switching element Q2 shown in FIG. 4C and the switching current I2 flowing from the primary winding N1B to the collector of the switching element Q2 shown in FIG.
4A and 4B have the same waveform shape as the waveforms shown in FIGS. 4A and 4B, and have a 180 ° phase difference with respect to the waveforms shown in FIGS. As can be seen from the waveform diagrams of FIGS. 4A to 4D, the switching element Q
1 and Q2 perform a switching operation at alternately on / off timings. That is, a push-pull operation is obtained.

When the minimum load power is 60 W, the switching elements Q1 and Q2 perform the switching operation at the switching frequency corresponding to substantially the upper limit of the switching frequency control range.
As can be seen from the collector-emitter voltages Vc1 and Vc2 of the switching elements Q1 and Q2 and the switching currents I1 and I2 shown in FIGS.
3) is fixed, and the switching frequency is raised so that the ON period (T2, T4) is shortened.

The switching output current I0 flowing from the primary windings N1A and N1B to the smoothing capacitor Ci has the waveform shown in FIG. 4 (e) at the maximum load power and shown in FIG. 4 (l) at the minimum load power. A waveform is obtained. FIG.
The switching output current I0 shown in FIG.
4D has a waveform obtained by synthesizing the switching currents I1 and I2. The switching output current I0 shown in FIG. 4L is the switching current I1 and I2 shown in FIGS. Is a synthesized waveform.

In this case, as the operation of the secondary side series resonance circuit, the series resonance current I3 flowing through the secondary side series resonance capacitor is as shown in FIG. 4 (f) at the maximum load power and FIG. As shown in m), a sinusoidal alternating current with a frequency twice the switching frequency is obtained. The series resonance current I3 is rectified by the rectifier diode DO2 in the period T5 and rectified by the rectifier diode DO1 in the period T6, whereby the above-described voltage-doubling operation is obtained. The rectifier diode D
The voltage V2 between both ends of O2 is shown in FIG.
At the time of the minimum load power, as shown in FIG.
During the period T5 when the rectifier diode DO2 is on, 0
At the level, a waveform is obtained in which a pulse is obtained in the off period T6. Note that the voltage V2 across the rectifier diode DO2 is actually the secondary-side DC output voltage EO (E0
The waveform is clamped at the level of O1). On the rectifier diode DO1 side, 1 in FIG.
Almost the same waveform having a phase difference of 80 ° is obtained.

Here, in the present embodiment, as in the case of the power supply circuit of FIG. 1, a DC input voltage on the primary side is obtained by the configuration of a normal unity voltage rectification circuit having a bridge rectification circuit. That is, for example, FIGS. 4 (a) and 4 (c)
The switching element Q1,
The collector-emitter voltages Vc1 and Vc2 of Q2 are 70
It is suppressed to the peak level of 0V. Then, at the time of the minimum load power, as shown in FIGS.
It is suppressed to a peak level of 500V. The switching is performed by the operation of the orthogonal drive transformer PRT under the control of the control circuit 1 as shown as the transition from FIG. 4 (a) → FIG. 4 (h) and FIG. 4 (c) → FIG. 4 (j). By controlling the frequency control, the switching element Q
This is due to the fact that a complex control operation of controlling the ON period of Q1 and the conduction angle is obtained. Thus, for the switching elements Q1, Q2 and the parallel resonance capacitor Cr, for example, a 900V withstand voltage product may be selected.

Further, as described above, the voltage across the secondary-side rectifier diode is suppressed to a level equivalent to the secondary-side DC output voltage EO. As the rectifier diode forming the doubler voltage rectifier circuit, a withstand voltage product corresponding to the level of the secondary side DC output voltage EO may be selected.

Further, in the case of the present embodiment, the secondary-side DC output voltage is obtained by the voltage doubler full-wave rectifier circuit, which is equivalent to, for example, the secondary-side DC output voltage obtained by the equal-voltage rectifier circuit. In order to obtain the above level, the secondary winding N2A of the present embodiment requires only half the number of turns of the conventional winding. This reduction in the number of turns leads to a reduction in the size and weight of the insulating converter transformer PIT and a reduction in cost.

Also in this case, since low-voltage products can be used for the switching elements Q1 and Q2 and the rectifier diode on the secondary side, as in the case of the first embodiment, reduction in size and weight and reduction in power consumption can be achieved. Along with cost reduction,
The switching frequency can be set higher. The actual switching frequency is, for example, 50 KHz conventionally.
However, in this embodiment, the frequency is about 100 KHz.
Can be as high as that.

Further, in the power supply circuit of the second embodiment, the switching frequency is variably controlled as the constant voltage control operation on the primary side, so that the resonance impedance of the primary side series resonance circuit and the secondary side series resonance circuit The operation of controlling both the resonance impedance and the resonance impedance at the same time is obtained. Therefore, also in this case, the control sensitivity for the constant voltage control is improved, and for example, the constant voltage control range with respect to the fluctuation of the load DC input voltage is expanded, and the minimum load power is also expanded.

FIG. 5 shows a configuration of a power supply circuit according to a third embodiment of the present invention. The power supply circuit shown in this figure is also provided with a push-pull voltage resonance type converter on the primary side. In this figure, the same parts as those in FIGS. 1 and 3 are denoted by the same reference numerals, and description thereof will be omitted.

In the power supply circuit shown in FIG.
As switching elements Q1 and Q2, bipolar transistors (BJT) Q11 and Q12, damper diodes (Zener diodes) DD1 and damper diodes D, respectively.
A Darlington circuit formed by connecting D2 and resistors R11 and R12 as shown in the figure is provided. In this case, as the connection mode of the Darlington circuit, the collector of the transistor Q11 is connected to the collector of the transistor Q12, the emitter of the transistor Q11 is connected to the emitter of the transistor Q12, and the emitter of the transistor Q12 is grounded. . Further, the anode of the damper diode DD1 is connected to the emitter of the transistor Q11, and the cathode of the damper diode DD1 is connected to the transistor Q11 via the resistor R11.
Connected to the base. The anode of the damper diode DD2 is connected to the emitter of the transistor Q12, and the cathode is connected to the collector of the transistor Q12. The resistor R12 is connected in parallel between the base and the emitter of the transistor Q12. In the Darlington circuit thus formed, the base of the transistor Q11 is connected to the switching element Q shown in each of the above embodiments.
1 and the collector contacts of the transistors Q11 and Q12 become equivalent to the collector of the switching element Q1. Further, the emitter of the transistor Q12 is equivalent to the emitter of the switching element Q1.

In this case, the self-excited oscillation circuit for driving the switching element in a self-excited manner is omitted.
Instead, a configuration in which the oscillation / drive circuit 2 is provided and switching driving by separately excited is performed is employed. For this reason, in the present embodiment, the isolated converter transformer PIT
Are provided with windings N4A and N4B. And winding N
A DC voltage of, for example, +12 V is generated by a half-wave rectifier circuit including 4A, rectifier diode D1, and capacitor CA, and a DC voltage of, for example, -12V is generated by a half-wave rectifier circuit including winding N4B, rectifier diode D2, and capacitor CB. To be. The DC voltage of +12 V and −12 V is supplied to the oscillation / drive circuit 22 as an operation power supply. The structure of the insulating converter transformer PIT according to the present embodiment is first described in FIG.
A structure similar to the core shown in FIG. The difference from the first embodiment is that the windings N4A and N4B are additionally wound on the primary side.

The oscillation / drive circuit 2 is activated by an activation resistor RS, and generates an oscillation signal having a cycle having a required switching frequency fs. And, by using the operating power supply of + 12V / -12V,
The oscillating signal is converted into a positive (on) / negative (off) switching drive current for each switching cycle and output to the base terminal of the switching element Q1. As a result, the switching element Q1 is driven to perform a switching operation at a required switching frequency. When a Darlington circuit is employed for the switching element Q1 as in the present embodiment, higher power conversion efficiency can be obtained than when the switching element Q1 is a single bipolar transistor, for example.

In the control circuit 1 shown in this figure, for example, a DC signal (detection signal) of a level varied according to the level of the secondary DC output voltage EO1 as a detection input is sent to the oscillation / drive circuit 2. I am trying to supply. here,
In the oscillation / drive circuit 2, if the off-period of the switching element Q1 is made constant and the on-period is variably controlled to vary the switching frequency in accordance with the detection signal input from the control circuit 1, The same operation as that shown in FIG. 4 is obtained, and the same effects as those of the power supply circuit of each embodiment described above are obtained.

Further, in the isolated converter transformer PIT of the present embodiment, secondary windings N2A and N2B are provided. The secondary winding N2A of the present embodiment is wound with a different number of turns from the conventional one as described later. One end of the secondary winding N2A is connected to a connection point between the anode of the rectifier diode DO1 and the cathode of the rectifier diode DO2 via a series connection of a series resonance capacitor Cs1, and
It is connected to the connection point between the anode of the rectifier diode DO3 and the cathode of the rectifier diode DO4 via the series connection of the series resonance capacitor Cs2. The other end of the secondary winding N2A is
It is connected to the connection point between the negative electrode of the smoothing capacitor CO10 and the positive electrode of the smoothing capacitor CO11. The anode of the rectifier diode DO2 and the cathode of the rectifier diode DO3 are connected to a connection point between the negative electrode of the smoothing capacitor CO10 and the positive electrode of the smoothing capacitor CO11. The smoothing capacitor CO10 and the smoothing capacitor CO11 are connected in series such that the negative electrode of the smoothing capacitor CO10 and the positive electrode of the smoothing capacitor CO11 are connected, and then the positive electrode of the smoothing capacitor CO10 is connected to the cathode of the rectifier diode DO1, And the negative electrode of CO11 are connected to the secondary side ground.

In such a connection form, as a result,
[Series resonant capacitor Cs1, rectifier diodes DO1, DO
2. A first voltage doubler full-wave rectifier circuit composed of a set of [smoothing capacitor CO10] and a second voltage doubler full-wave rectifier circuit composed of a set of [series resonance capacitor Cs2, rectifier diodes DO3, DO4, smoothing capacitor CO11]. Are formed, and these first
And the output of the second voltage doubler rectifier circuit (smoothing capacitor CO1
0, CO11) are connected in series and provided.
As a whole rectifier circuit combining the first and second voltage doubler rectifier circuits, both ends of a series-connected smoothing capacitor CO10-smoothing capacitor CO11 have an alternating voltage of the secondary winding N2A. A secondary output voltage corresponding to four times is obtained. That is, a quadruple voltage rectifier circuit is formed as an entire rectifier circuit combining the first and second voltage doubler rectifier circuits. The rectifying operation of the quadruple voltage rectifier circuit will be described later.

The series resonant capacitor Cs1 is connected in series with the on / off operation of the rectifier diodes DO1 and DO2 in the first voltage doubler rectifier circuit by its own capacitance and the leakage inductance component (L2) of the secondary winding N2A. Form a resonance circuit. Similarly, the series resonance capacitor C
s2 forms a series resonance circuit corresponding to the on / off operation of the rectifier diodes DO3 and DO4 in the second voltage doubler rectifier circuit by its own capacitance and the leakage inductance component (L2) of the secondary winding N2A. As the resonance frequency of these series resonance circuits, the parallel resonance frequency of the primary side parallel resonance circuit (N1, Cr) is fo1,
The series resonance frequency of the secondary side series resonance circuit (N2A, Cs1) is fo2, and the same secondary side series resonance circuit (N2A, Cs1).
Assuming that the series resonance frequency in 2) is fo3, fo1 ≒ fo
The capacitance of the secondary side series resonance capacitors Cs1 and Cs2 is selected so that 2 ≒ fo3.

As described above, the power supply circuit according to the present embodiment is provided with a parallel resonance circuit on the primary side for making the switching operation a voltage resonance type, and a quadruple voltage rectification circuit on the secondary side. And two sets of voltage doubler rectifier circuits forming the quadruple voltage rectifier circuit are provided with two sets of series resonance circuits for obtaining current resonance operation. That is, this embodiment also adopts a configuration as a composite resonance type switching converter.

Next, the operation of the aforementioned quadruple voltage rectifier circuit will be described. When a switching output is obtained on the primary winding N1 by the switching operation on the primary side, this switching output is excited by the secondary winding N2A. The quadruple voltage rectifier circuit performs the rectification operation by inputting the obtained alternating voltage to the secondary winding N2A, and comprises a [series resonance capacitor Cs1, rectifier diodes DO1, DO2, and a smoothing capacitor CO10]. The operation of the first voltage doubler rectifier circuit is described below. First, during a period T1 in which the rectifier diode DO1 is turned off and the rectifier diode DO2 is turned on, the secondary winding N1 operates in the depolarized mode in which the polarity of the primary winding N1 and the secondary winding N2A is -M. Due to the series resonance effect of the leakage inductance of the line N2A and the series resonance capacitor Cs1, an operation of charging the series resonance capacitor Cs1 with the rectified current I2 rectified by the rectifier diode DO2 is obtained. Then, the rectifier diode DO2 is turned off, the rectifier diode DO1 is turned on, and a period T2 for performing the rectification operation
, The polarity of the primary winding N1 and the secondary winding N2A is +
The operation becomes an additional polarity mode of M, and the smoothing capacitor CO10 is charged in a state where series resonance occurs in which the potential of the series resonance capacitor Cs1 is added to the voltage induced in the secondary winding N2A. As described above, the positive polarity mode (+ M; forward operation) and the negative polarity mode (−
M; flyback operation), the rectification operation is performed in the smoothing capacitor CO10, so that the DC voltage (rectified smoothing voltage) corresponding to almost twice the induced voltage of the secondary winding N2A is obtained in the smoothing capacitor CO10. ) Is obtained. The same operation is performed by the second voltage doubler full-wave rectifier circuit composed of a set of [series resonance capacitor Cs2, rectifier diodes DO3, DO4, and smoothing capacitor CO11].
At both ends of O11, a DC voltage corresponding to approximately twice the induced voltage of the secondary winding N2A is obtained.

The voltage doubler rectification operation is performed by each of the first and second voltage doubler rectifier circuits as described above. As a result, a secondary capacitor is connected across both ends of the series-connected smoothing capacitor CO10 and smoothing capacitor CO11. A secondary side DC output voltage EO1 corresponding to approximately four times the induced voltage of the winding N2A is obtained.

In the secondary winding N2B, after a center tap provided on the secondary winding N2B is grounded, an ordinary full-wave rectifier circuit composed of rectifier diodes D05 and D06 and a smoothing capacitor CO2 is formed. As a result, the secondary side DC output voltage EO2 is obtained.

As described above, also in the present embodiment, as the operation of the quadruple voltage rectifier circuit on the secondary side, the two sets of voltage doubler full-wave rectifier circuits are in the operation mode in which the mutual inductance is + M and -M. Double voltage full-wave rectification is performed using the state. This is because electromagnetic energy transmitted from the primary winding to the secondary winding of the isolated converter transformer PIT, which has been loosely coupled (coupling coefficient k = 0.85), is transmitted to the primary side parallel resonance circuit (voltage resonance circuit). It can be seen that the operation is coupled between the secondary-side series resonance circuits (current resonance circuits). Therefore, also in this embodiment, the operation is performed such that the resonance impedance of the resonance circuit on the primary side and the secondary side is reduced to amplify the secondary side resonance current, so that the maximum load power is greatly increased. The increase will be achieved.

In this embodiment, the quadruple voltage rectifier circuit is provided on the secondary side as described above. For example, the voltage applied to each secondary side rectifier diode is reduced by the secondary DC voltage. The output voltage EO1 is clamped to a level equivalent to 1/2. As a result, as the rectifier diode forming the quadruple voltage rectifier circuit on the secondary side, a withstand voltage product corresponding to a level equivalent to 1/2 of the secondary side DC output voltage EO1 may be selected.

Further, since the secondary side DC output voltage is obtained by the quadruple voltage rectifier circuit, for example, a level equivalent to the secondary side DC output voltage obtained by the unity voltage rectifier circuit can be obtained. For example, the secondary winding N of the present embodiment
For 2A, the number of turns is only 1/4 of the conventional number. This reduction in the number of turns leads to a reduction in the size and weight of the insulating converter transformer PIT and a reduction in cost.

Further, as a result of variably controlling the switching frequency as a constant voltage control operation on the primary side, in the present embodiment, the resonance impedance of the primary side parallel resonance circuit and the resonance impedance of the two sets of series resonance circuits on the secondary side are also obtained. An operation of controlling at the same time as the resonance impedance is obtained, control sensitivity for constant voltage control is improved as compared with the related art, and a control range for fluctuations in AC input voltage and load power is expanded.

In the power supply circuit of the present invention, the combination of the secondary side resonance circuit and the rectifier circuit with respect to the primary side parallel resonance circuit is limited to those shown in FIGS. 1, 3 and 5. Instead, they can be configured in any combination. As one specific example, for example, FIG.
May be combined with, for example, the secondary-side configuration including the series resonance circuit and the voltage doubler rectifier circuit shown in FIG.

FIG. 1 shows a power supply circuit according to the present invention.
In addition to the configuration shown in FIGS. 3 and 5, the configuration may be appropriately changed in accordance with actual use conditions.
As the switching element, other component elements other than the bipolar transistor and the MOS-FET may be employed.

[0091]

As described above, according to the present invention, as a switching power supply circuit, a voltage resonance type converter (primary side parallel resonance circuit) by a push-pull operation is used for a primary side.
Is provided. Further, by making the insulating converter transformer loosely coupled, an operation mode (+ M / -M) in which the mutual inductances of the primary winding and the secondary winding have polarities opposite to each other is obtained. On the secondary side, a secondary-side parallel resonance circuit or a secondary-side series resonance circuit is formed with respect to the secondary winding, and the secondary wave is provided by providing a full-wave rectifier circuit using this resonance circuit. A side DC output voltage is obtained.

As described above, the primary side voltage resonance type converter is configured to perform the push-pull operation, and the secondary side is configured to perform the full-wave rectification of the resonance output to supply power to the load. It is possible to greatly increase the maximum load power that can be handled as compared with the related art. Accordingly, for example, even in the case of accommodating a relatively heavy load condition in the AC 100 V system, the primary side is not a voltage doubler rectifier circuit but a rectifier of a level corresponding to the AC input voltage level by a normal full-wave rectifier circuit. It can be configured to input a smoothing voltage.

The following can be said from the above configuration. For example, conventionally, in order to meet the above condition, it is necessary to obtain a rectified smoothed voltage corresponding to twice the AC input voltage level by a voltage doubler rectifier circuit. As the capacitor, it is necessary to select a withstand voltage product corresponding to the switching voltage generated according to the rectified smoothing voltage level.

On the other hand, in the present invention, since the switching voltage depending on the rectified smoothing voltage level is about 1/2 of the conventional one, the withstand voltage of the switching element and the primary side resonance capacitor is about 1/2 of the conventional one. Goods can be used.

On the secondary side, the alternating voltage is positive /
As a result of the full-wave rectification operation performing the rectification operation in both negative periods,
The voltage applied to the rectifier diode is suppressed substantially equal to the rectified smoothed voltage level. When the secondary side rectifier circuit is a quadruple voltage rectifier circuit, the voltage applied to the rectifier diode is suppressed to about の of the rectified smoothed voltage level. As a result, a secondary rectifier diode having a lower withstand voltage than the conventional rectifier diode can be selected. As a result, it is possible to reduce costs for the switching element, the primary-side parallel resonance capacitor, the secondary-side rectifier diode, and the like. In addition, it is also possible to easily select a switching element and a secondary rectifier diode having improved characteristics and set a high switching frequency, thereby improving the power conversion efficiency. Also, it is possible to reduce the size and weight of circuit components around the switching element. Further, as described above, since the circuit for obtaining the rectified smoothed voltage from the commercial AC power supply is a normal equal-voltage rectifier circuit, for example, a normal set of a block-type smoothing capacitor and a bridge rectifier diode may be used. So you can
Also in this respect, the cost and the circuit scale can be reduced.

Further, according to the present invention, when a voltage doubler rectifier circuit is employed as the full-wave rectifier circuit provided on the secondary side,
For example, if a DC output voltage of the same level as that of the case where the equal-voltage rectifier circuit is provided is to be obtained, the number of turns of the secondary winding can be reduced to about １ ／ of the conventional case. When a quadruple voltage rectifier circuit is employed, the number of turns of the secondary winding can be reduced to about 1/4.

Further, in the circuit configuration of the present invention, the primary side operates so as to perform the constant voltage control by variably controlling the switching frequency. At this time, according to the switching frequency being varied, Since the operation of simultaneously controlling the resonance impedance of the primary-side series resonance circuit and the resonance impedance of the secondary-side resonance circuit (series resonance circuit or parallel resonance circuit) is obtained, the constant voltage control range is expanded. Further, thereby, for example, if the power supply circuit of the present invention is a self-excited type, it is possible to reduce the number of windings of the control winding and the like in the orthogonal transformer, thereby achieving reduction in size, weight, and cost. In addition, since the primary side rectifier circuit is a unity voltage rectifier circuit, the number of turns of the primary winding N1 can be reduced as compared with the conventional case, so that the number of turns of the winding can be reduced. Is achieved.

As described above, according to the present invention, the cost, the size and the weight of the resonance type power supply circuit, and the improvement of various characteristics such as the power conversion efficiency and the constant voltage control range are promoted.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration example of a power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a cross-sectional view illustrating a configuration of an insulating converter transformer of the power supply circuit according to the present embodiment.

FIG. 3 is a circuit diagram illustrating a configuration example of a power supply circuit according to a second embodiment;

FIG. 4 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

FIG. 5 is a circuit diagram illustrating a configuration example of a power supply circuit according to a third embodiment;

FIG. 6 is a circuit diagram showing a configuration of a power supply circuit as a conventional example.

FIG. 7 is a circuit diagram showing a configuration of a power supply circuit as a conventional example.

FIG. 8 is a circuit diagram showing a configuration of a power supply circuit as a conventional example.

FIG. 9 is a sectional view showing a configuration of an insulating converter transformer as a conventional example.

FIG. 10 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

FIG. 11 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIGS. 6 and 7.

[Explanation of symbols]

1 control circuit, 2 oscillation / drive circuit, Ci smoothing capacitor, C2 secondary parallel resonance capacitor, Cs1, C
s2 Secondary-side series resonance capacitor, Di bridge rectifier circuit, DO1, DO2, DO3, DO4 rectifier diode, PR
T orthogonal drive transformer, PIT isolated converter transformer, NC control winding, Q1, Q2 switching element

Continued on the front page F term (reference) 5G065 AA00 AA08 DA06 DA07 EA06 FA02 GA06 GA07 HA04 JA01 LA01 MA01 MA03 MA10 NA01 NA03 NA04 NA09 5H730 AA02 AA14 AA15 BB23 BB25 BB52 BB57 BB76 BB77 BB80 BB94 DD02 EE07 EE07 EE07

Claims (8)

[Claims] A rectifying / smoothing means for receiving a commercial AC power supply, generating a rectified smoothed voltage having a level corresponding to the same level as the commercial AC power supply level, and outputting the rectified smoothed voltage as a DC input voltage; A gap is formed so as to obtain the coupling coefficient of the above, an insulating converter transformer provided for transmitting the primary side output to the secondary side, and two sets of switching elements, and the DC input voltage is intermittently operated by a push-pull operation. Switching means configured to output to the primary winding of the insulated converter transformer, and at least a leakage inductance component including the primary winding of the insulated converter transformer and a capacitance of a parallel resonance capacitor, A primary-side parallel resonance circuit in which the operation of the switching means is a voltage resonance type; A secondary resonance circuit formed by a leakage inductance component of a secondary winding of the capacitor and a capacitance inserted on the secondary side; and a secondary resonance circuit formed by including the secondary resonance circuit. DC output voltage generating means configured to input an alternating voltage obtained to the secondary winding and perform a full-wave rectification operation to generate a secondary DC output voltage; and a level of the secondary DC output voltage. And a constant voltage control means for performing a constant voltage control by variably controlling a switching frequency of the switching means according to the above. 2. The secondary-side resonance circuit is a secondary-side parallel resonance circuit formed by connecting a secondary-side parallel resonance capacitor to the secondary winding in parallel. The means is configured to input an alternating voltage obtained by the resonance operation of the secondary parallel resonance circuit, and to generate a secondary DC output voltage corresponding to substantially equal to the input voltage. Claim 1.
3. The switching power supply circuit according to item 1. 3. The secondary-side resonance circuit is a secondary-side series resonance circuit formed by connecting a secondary-side series resonance capacitor to the secondary winding in series. The means includes inserting the secondary side series resonance capacitor into the secondary side rectified current path, and inputting an alternating voltage obtained in the secondary winding of the insulating converter transformer, and setting the input voltage level to approximately 2%. 2. The switching power supply circuit according to claim 1, wherein the switching power supply circuit is configured to obtain a secondary side DC output voltage corresponding to the doubled output voltage. 4. The secondary-side resonance circuit is a secondary-side series resonance circuit formed by connecting a secondary-side series resonance capacitor to the secondary winding in series. Means are such that the secondary side series resonance capacitor is inserted into the secondary side rectified current path, and the alternating voltage obtained in the secondary winding of the insulating converter transformer is inputted, so that the input voltage level is approximately 4%. 2. The switching power supply circuit according to claim 1, wherein the switching power supply circuit is configured to obtain a secondary side DC output voltage corresponding to the doubled output voltage. 5. The switching power supply circuit according to claim 1, wherein said switching means includes a self-excited oscillation drive circuit for performing a switching operation in a self-excited manner. 6. The constant voltage control means comprises: at least a driving winding forming the self-excited oscillation driving circuit; and a winding wound in a direction orthogonal to the driving winding, and 6. The switching power supply circuit according to claim 5, further comprising: a quadrature transformer as a saturable reactor wound with a control winding to which a control current of a level varied according to the fluctuation is supplied. 7. The switching power supply circuit according to claim 1, wherein said switching means includes a drive circuit for performing a switching operation by a separately excited system. 8. The switching power supply circuit according to claim 1, wherein said switching means is configured to use a Darlington circuit formed with a bipolar transistor as one switching element.