JP2002345236A - Switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To lower the breakdown voltage of a switching element. SOLUTION: To use this circuit as a compound resonance converter that is a self-excited and stabilized by switching frequency-control system, a DC-AC control transformer is omitted. Then, a soft-starter circuit is provided to operate based on the level detection of a primary DC voltage generated based on the alternating voltage of tertiary windings wound on the insulated converter transformer when starting a power supply. This structure enables a circuit to be protected even without providing an overcurrent detecting circuit and the like supposed to be used, for example.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a circuit employing a switching converter of a type such as a flyback converter or a forward converter is widely known. Since these switching converters have a rectangular switching operation waveform, there is a limit in suppressing switching noise. Also, due to its operating characteristics,
It has been found that there is a limit in improving the power conversion efficiency. Therefore, the present applicant has previously proposed various switching power supply circuits using various resonance type converters.
The resonance type converter can easily obtain high power conversion efficiency and realize low noise because the switching operation waveform is sinusoidal. It also has the advantage that it can be configured with a relatively small number of parts.

[0005] FIG. 5 is a circuit diagram showing an example of a switching power supply circuit as a prior art, which can be formed based on the invention proposed by the present applicant. As a basic configuration of the power supply circuit shown in this figure, a voltage resonance type converter is provided as a primary side switching converter.

In the power supply circuit shown in FIG. 1, a bridge rectifier circuit Di and a smoothing capacitor Ci convert one of the AC input voltage VAC from the commercial AC power supply AC (AC input voltage VAC).
A rectified smoothed voltage Ei corresponding to the double level is generated. A switch SW for turning on / off the power supply is inserted in the line of the commercial AC power supply AC. Also, a rush current limiting resistor Ri is inserted into the line of the commercial AC power supply AC, for example, to suppress a rush current flowing into the smoothing capacitor when the power is turned on.

[0005] A single-ended single-end system is adopted as a voltage resonance type converter that receives and inputs the rectified smoothed voltage Ei (DC input voltage) and is intermittent. The drive system employs a self-excited configuration. In this case, the switching element Q1 forming the voltage resonance type converter includes:
A high breakdown voltage bipolar transistor (BJT; junction transistor) is selected. A primary side parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. A series connection circuit of a clamp diode DD and a resistor RD is connected between the base and the emitter. Here, the parallel resonance capacitor Cr is connected to the primary winding N1 of the insulation converter transformer PIT.
A primary-side parallel resonance circuit is formed together with the leakage inductance L1 obtained as described above, whereby an operation as a voltage resonance type converter can be obtained. To the base of the switching element Q1, a self-excited oscillation drive circuit including a drive winding NB, a resonance capacitor CB, and a base current limiting resistor RB is connected. The switching element Q1 is switched by being supplied with a base current based on an oscillation signal generated by the self-excited oscillation driving circuit. In addition, at the time of startup, it is started by a startup current flowing from the line of the rectified smoothed voltage Ei to the base via the startup resistor Rs.

The orthogonal control transformer PRT is configured by winding a control winding Nc so that the winding direction of the drive winding NB and the current detection winding ND is orthogonal to the winding direction. It is provided for controlling the switching frequency of the primary side voltage resonance type converter as described later.

[0007] The insulating converter transformer PIT is provided for transmitting the switching output of the switching converter obtained on the primary side to the secondary side. This insulated converter transformer PIT has a primary winding N with respect to an EE type core.
1 and the secondary winding N2 are divided and wound, and a gap G is formed with respect to the center magnetic leg so that a loose coupling state with a required coupling coefficient can be obtained, and a saturated state can be obtained. I try to be hard to be.

The primary winding N1 of the insulating converter transformer PIT is connected between the line of the DC input voltage (rectified smoothed voltage Ei) and the collector of the switching element Q1. The switching element Q1 performs switching with respect to the DC input voltage. As a result, the switching output of the switching element Q1 is supplied to the primary winding N1, and an alternating voltage having a cycle corresponding to the switching frequency is generated. I do.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary side parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary side winding N2 and the capacitance of the secondary side parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. With this parallel resonance circuit,
The alternating voltage induced in the secondary winding N2 becomes a resonance voltage.
That is, a voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, the primary side is provided with a parallel resonance circuit for performing a voltage resonance type switching operation, and the secondary side is provided with a parallel resonance circuit for obtaining a voltage resonance operation. In the present specification, such a switching converter configured to operate with a resonance circuit provided for the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

The isolated converter transformer PIT in this case
First, the anode of the rectifier diode DO1 is connected to the winding end of the secondary winding N2, and the cathode is connected to the positive terminal of the smoothing capacitor CO1, thereby forming a half-wave rectifier circuit. Has formed. With this half-wave rectifier circuit, a secondary side DC output voltage EO1 is obtained across the smoothing capacitor CO1. In this case, a tap is provided for the secondary winding N2, and a half-wave rectifier circuit composed of a rectifier diode DO2 and a smoothing capacitor CO2 is formed as shown in FIG. Then, depending on the half-wave rectifier circuit, the secondary-side DC output voltage E O which is lower than the secondary-side DC output voltage EO1 is provided.
O2 is obtained. Note that, specifically, the secondary side DC output voltage EO1 = 135V and the secondary side DC output voltage EO2 = 15V.

These secondary side DC output voltages EO1, EO2 are:
Each is supplied to a required load circuit. The secondary side DC output voltage EO1 is used as a detection voltage of the control circuit 1, and the secondary side DC output voltage EO2 is used as the control circuit 1
Is branched and output as the operating power supply for the power supply.

The control circuit 1 allows a variable DC current to flow through the control winding Nc of the orthogonal control transformer PRT as a control current in accordance with the level of the secondary DC output voltage EO1. By varying the level of the control current flowing through the control winding Nc, the orthogonal control transformer PRT is controlled to vary the inductance of the drive winding NB. As a result, the resonance frequency of the resonance circuit including the drive winding NB and the resonance capacitor CB in the self-excited oscillation drive circuit changes, and the switching frequency of the switching element Q1 is variably controlled. By changing the switching frequency of the switching element Q1 in this way, the secondary DC output voltage is controlled to be constant. That is, the power supply is stabilized. In this specification, the constant voltage control by such an operation is described.
It is also referred to as “switching frequency control method”.

In the power supply circuit shown in FIG. 5, when the switch SW is turned on and the commercial AC power supply AC is turned on, the smoothing capacitor Ci from the rush current limiting resistor Ri via the diode of the bridge rectifier circuit Di. , A rectified smoothed voltage Ei, which is a voltage across the smoothing capacitor Ci, is raised to a level corresponding to the AC input voltage VAC. Then, the starting current flows from the line of the rectified smoothed voltage Ei to the base of the switching element Q1 via the starting resistance Rs, and the switching element Q1 is turned on to start oscillation and start the switching operation.

However, accompanying such an operation at the time of starting, an excessive charging current flows to the smoothing capacitors CO1 and CO2 on the secondary side of the insulating converter transformer PIT. Also, an excessive collector current flows from the smoothing capacitor Ci to the collector of the switching element Q1 via the current detection winding NA and the primary winding N1. Further, the levels of the secondary side DC output voltages EO1 and EO2 at this time are as follows.
This is a transient state in which the voltage rises to a required level. Therefore, at this time, switching frequency control according to the level of the secondary DC output voltage EO1 is not performed. At this time, the switching element Q1 operates at the lowest switching frequency determined by the inductance of the drive winding NB and the capacitance of the capacitor CB.
In the case of this power supply circuit, when the switching operation is performed at the lowest switching frequency, the ON period of the ON period and the OFF period of the switching device Q1 becomes longer. During the period, the peak level of the voltage resonance pulse as the parallel resonance voltage V1 generated at both ends of the parallel circuit of the switching element Q1 // parallel resonance capacitor Cr also becomes excessive.

Therefore, as a practical example of the circuit having the configuration shown in FIG. 5, an overcurrent limiting circuit for limiting an excessive collector current flowing through the switching element Q1 at the time of startup may be provided. FIG. 6 shows a configuration example of a power supply circuit provided with an overcurrent limiting circuit based on the circuit configuration shown in FIG. In this figure, the same parts as those in FIG. The overcurrent limiting circuit 10 shown in FIG. 6 detects a current flowing through the emitter of the switching element Q1 by a circuit including a current detecting resistor RE and voltage dividing resistors R11 and R12. And
By detecting the excessively high level of the emitter current and turning on the transistor Q10, the base current of the switching element Q1 flows from the diode D2 through the collector and emitter of the transistor Q10. As a result, the forward base current of the switching element Q1 is suppressed, and the collector current flowing to the collector of the switching element Q1 can be limited.

However, when the overcurrent limiting circuit 10 is provided as shown in FIG. 6, for example, a current detecting resistor RE is connected in series with the emitter of the switching element Q1. Because
A power loss occurs in the current detection resistor RE. The power loss in the current detection resistor RE increases particularly under a heavy load, and is a factor that lowers the power conversion efficiency.

FIG. 7 is a waveform chart showing the operation of the circuit shown in FIG.
1 and the collector current IQ1 of the switching element Q1. For example, an intermediate load is assumed, and the AC input voltage VAC =
At the time of steady operation at about 120 V, FIG.
As shown in (a) and (b), the voltage resonance pulse V1 is about 700 Vp, and the collector current IQ1 of the switching element Q1 is about 4.5A. On the other hand, under the condition that the maximum load power (Pomax = 150 W) is set and the AC input voltage VAC is increased to about 120 V, as shown in FIGS. 7C and 7D, the voltage resonance pulse V
1 rises to about 900 Vp, and the switching element Q1
Collector current IQ1 rises to about 6.5A. That is, the voltage resonance pulse V1 and the collector current IQ1 at the time of the maximum load power are increased by about 20 to 30% as compared with the steady state. Such an operation includes a margin corresponding to an increase in the AC input voltage VAC and a variation in components constituting the overcurrent limiting circuit 10 because the circuit shown in FIG. This is the result of circuit design in consideration of a malfunction margin during normal operation. The switching element Q1 is, for example, 1200 V corresponding to the maximum load power.
Must be selected. The higher the withstand voltage of the switching element, the larger and more expensive it becomes. In addition, the switching characteristics are inferior.

In the power supply circuits shown in FIGS. 5 and 6, since the rise time of the secondary DC output voltage at startup is fast, for example, a level change of the secondary DC output voltage is detected at startup. Therefore, a circuit cannot be configured to suppress such a sharp rise in level. That is, a so-called soft start operation cannot be obtained by detecting the level fluctuation of the secondary side DC output voltage.
For this reason, there is a problem that there is no malfunction margin at the time of start-up, and the reliability of the power supply circuit is accordingly reduced.

[0019]

In view of the above-mentioned problems, the present invention is configured as a switching power supply circuit as follows. That is, a switching means including a switching element for switching the DC input voltage, an insulating converter transformer for transmitting an output of the switching means obtained on the primary winding to a secondary winding, and a primary winding of the insulating converter transformer. And a primary-side parallel resonance circuit formed by the primary-side parallel resonance capacitor and provided to make the operation of the switching means a voltage resonance type. In addition, a secondary parallel resonance circuit formed by connecting a secondary parallel resonance capacitor in parallel to a secondary winding wound on an insulating converter transformer, and a secondary parallel resonance circuit obtained in this secondary parallel resonance circuit DC output voltage generating means is provided which is configured to obtain a DC output voltage by inputting an alternating voltage and performing a rectification operation. A primary winding connected in series to a primary winding of the insulating converter transformer and detecting an output of the switching means; and a secondary winding to which a switching voltage detected by the primary winding is transmitted. A switching unit configured to include a drive transformer and at least a series resonance circuit including a series resonance capacitor and a series resonance inductor connected in series with a secondary winding of the drive transformer, and to supply a switching drive signal to the switching element; Drive means. Further, a series circuit of a division capacitor connected to a connection point between the series resonance capacitor and the series resonance inductor and the conduction control element, and the conduction control element according to the level of the DC output voltage obtained by the DC output voltage generation means. By changing the amount of conduction of the element and variably controlling the amount of current flowing through the dividing capacitor, a constant voltage control means that controls the switching frequency of the switching element and performs constant voltage control on the DC output voltage is provided. Prepare. A primary-side DC voltage generating means provided with a tertiary winding wound on the primary side of the insulating converter transformer PIT, and rectifying an alternating voltage obtained in the tertiary winding to obtain a primary-side DC voltage; And soft start means for variably controlling the conduction amount of the conduction control element so that the constant voltage control means has an operation tendency to lower the DC output voltage based on the primary side DC voltage obtained at the time of startup. is there.

In the above configuration, in the compound resonance type converter in which the switching element is driven by self-excited switching, the amount of conduction in the conduction control element is varied according to the DC output voltage on the secondary side for constant voltage control. ing. Thus, the current flowing through the split capacitor connected in parallel with the series resonance capacitor in the switching driving means is variably controlled to control the switching frequency of the switching element. With such a configuration of constant voltage control, for example, in the case of the self-excited type, it is possible to omit the orthogonal control transformer used for the switching frequency variable control. Then, the present invention winds a tertiary winding on the primary side of the insulated converter transformer to generate a primary side DC voltage based on the alternating voltage excited in the tertiary winding. The primary side DC voltage rises from, for example, 0 level to a certain steady level at the time of startup, and therefore corresponds to the rise of the secondary side DC output voltage at the time of startup. And the soft start means of the present invention, at the time of startup,
The amount of conduction of the conduction control element is variably controlled such that the switching frequency is variably controlled with a control tendency to obtain an operation of lowering the secondary side DC output voltage in accordance with the level rise of the primary side DC voltage. A soft start circuit is provided. That is, the soft start circuit is configured to indirectly detect a rise in the level of the secondary-side DC output voltage on the primary side to obtain a soft start operation.

[0021]

FIG. 1 shows a configuration example of a switching power supply circuit according to an embodiment of the present invention. The power supply circuit shown in this figure employs a configuration as a complex resonance type switching converter including a voltage resonance type converter on the primary side and a parallel resonance circuit on the secondary side. In the power supply circuit shown in this figure, first, as a rectifying and smoothing circuit for receiving a commercial AC power supply (AC input voltage VAC) to obtain a DC input voltage, a full-wave rectifying circuit including a bridge rectifying circuit Di and a smoothing capacitor Ci. And generates a rectified smoothed voltage Ei corresponding to a level that is one time the AC input voltage VAC. Further, FIG. 3 shows a switch SW inserted into a line of a commercial AC power supply for turning on / off the operation of the power supply circuit.

As the switching converter that receives the rectified smoothed voltage Ei (DC input voltage) and is intermittent,
A voltage resonance type converter including a stone switching element Q1 and performing a so-called single-ended switching operation is provided. The voltage resonance type converter here has a self-excited type configuration, and a high breakdown voltage bipolar transistor (BJT: junction type transistor) is used as the switching element Q1. Switching element Q
The collector 1 is connected to the positive electrode of the smoothing capacitor Ci via the primary winding N1 of the insulating converter transformer PIT and the primary winding (detection winding) NA of the drive transformer CDT. In this case, the emitter is connected to the primary side ground via the primary current detection resistor R2. That is, the primary current detection resistor R2 is inserted between the emitter of the switching element Q1 and the primary side ground as a path for the switching current.

The collector of the switching element Q1
A parallel resonance capacitor Cr is connected in parallel between the emitters. The capacitance of the parallel resonance capacitor Cr and the primary winding N of the insulating converter transformer PIT
The primary side parallel resonance circuit is formed by the leakage inductance obtained in (1). And
The resonance operation by the parallel resonance circuit is obtained according to the switching operation of the switching element Q1, so that the switching operation of the switching element Q1 is a voltage resonance type.

A clamp diode DD is inserted between the base and the emitter of the switching element Q1 via an inductor LB. That is, the clamp diode DD
Is connected to a connection point between the time constant capacitor CB1 and the inductor LB, which will be described later, and the anode of the clamp diode DD is connected to the primary side ground. This clamp diode DD is a low-speed recovery type diode.

Further, in this embodiment, as a starting resistor for starting the switching element Q1, the starting resistor is divided into two starting resistors Rs1 and Rs2 and connected in series. Then, the starting resistors Rs1-Rs2 connected in series are connected.
Is inserted between the line of the rectified smoothed voltage Ei and the inductor LB. As a result, when the power supply is started, the starting current obtained through the starting resistors Rs1 to Rs2 flows through the inductor LB to the base of the switching element Q1, thereby turning on the switching element Q1. The switching operation is started.

The primary side of the drive transformer CDT is a detection winding NA. The detection winding NA is connected in series to the primary winding N1 of the insulation converter transformer PIT, and detects the switching output of the switching element Q1. The secondary winding to which the switching voltage is transmitted from the detection winding NA is a drive winding NB that forms a self-excited oscillation drive circuit for the switching element Q1.

As the drive transformer CDT, for example, one using an H-shaped ferrite core as shown in FIG. 4A or one using an EI-12 type ferrite core as shown in FIG. 4B can be adopted. In the case of FIG.
It is formed by winding the detection winding NA and the driving winding NB as a double insulated wire on the ferrite core 100 having the insulated shape while securing insulation. In the case of FIG. 4B, the I-type core 102 and the E-type core 103 are combined as shown.
A gap G is formed at the junction of the magnetic legs of the I-shaped core 102 and the E-shaped core 103. The split bobbin 103 is arranged on the center magnetic leg of the E-shaped core 103, and the detection winding NA and the drive winding NB are wound around the split bobbin 103, respectively. The drive transformer CDT as shown in FIG. 4A or FIG. 4B enables a significant reduction in size and weight as compared with, for example, the orthogonal control transformer PRT described with reference to FIG.

As shown, an LC series connection circuit of [drive winding NB-time constant capacitor CB-inductor LB] is connected to the base of the switching element Q1. This series connection circuit is a self-excited oscillation drive circuit for driving the switching element Q1 in a self-excited manner. The inductor LB has, for example, a structure as a ferrite bead inductor.

In this case, the drive winding NB of the drive transformer CDT has the detection winding NA connected in series to the primary winding of the insulating converter transformer PIT as described above.
An alternating voltage as a switching output voltage obtained in the primary winding N1 is excited. As the self-excited oscillation drive circuit, a series resonance circuit is formed by the capacitor CB1, the inductance of the drive winding NB, and the inductor LB. The resonance frequency of this series resonance circuit is determined by the inductance of the drive winding NB and the inductor LB and the resonance capacitor CB
And a capacitance of 1.

In the self-excited oscillation drive circuit, an alternating voltage as a drive voltage is generated in the drive winding NB excited by the detection winding NA connected in series with the primary winding N1 of the insulating converter transformer PIT. This drive voltage is
Through a series resonance circuit (LB-CB1-NB),
The drive current is output to the base of the switching element Q1. As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit. Then, the switching output obtained at the collector is transmitted to the primary winding N1 of the insulating converter transformer PIT.

In this embodiment, the capacitor CB2
, A series circuit of MOS-FET (Q2) is formed, and this series circuit is inserted between the connection point between the capacitor CB1 and the inductor LB and the primary side ground. Here, if the resistance of the MOS-FET (Q2) is omitted, the capacitor CB2 can be regarded as connecting a path that branches the alternating current flowing through the capacitor CB1 to flow to the primary side ground. Then, it can be seen that the MOS-FET (Q2) has a conduction control function of controlling the amount of current flowing through the capacitor CB2.

The drain of the MOS-FET (Q2) is connected to the capacitor CB2, and the source is connected to the primary side ground. Also, the clamp diode DD2 is a MOS-FE
The connection between the drain and source of T (Q2) is made in parallel in the direction shown. In this case, a so-called body diode incorporated in the MOS-FET (Q2) can be used as the clamp diode DD2.

In the case of the circuit shown in this figure, a tertiary winding N3 is wound around the primary side of the insulating converter transformer PIT. A half-wave rectifier circuit composed of a diode D1 and a capacitor C1 is connected to the tertiary winding N3. A low-level DC voltage E1 of a predetermined level is obtained at both ends of the capacitor C1. The low-voltage DC voltage E1 is connected to the gate of the MOS-FET (Q2) via the collector-emitter of the phototransistor forming the photocoupler PC and further via a resistor. Therefore, at the gate of the MOS-FET (Q2), a voltage across the resistor R1 is generated according to the level of the current flowing from the photodiode, and this is applied to the gate of the MOS-FET (Q2) as a gate voltage. .

A soft start circuit 2 is provided on the primary side of the power supply circuit according to the present embodiment. This soft start circuit 2 includes a PNP transistor Q4
And an NPN transistor Q3. The collector of the transistor Q4 is connected to the connection point of the divided starting resistors Rs1 and Rs2, and the emitter is connected to the gate of the MOS-FET (Q2) via the resistor R2. The base is connected to the collector of the transistor Q3 via the resistor R3.

The base of the transistor Q3 is connected to the connection point of the voltage dividing resistors R5 and R6. The voltage dividing resistor R5 is connected to the low voltage DC voltage E1 via the anode → cathode of the Zener diode ZD, and the voltage dividing resistor R6 is connected to the primary side ground. Also, the transistor Q3
Are also connected to the primary side ground. A time constant capacitor C3 is connected between a connection point between the base of the transistor Q4 and the resistor R3 and the emitter of the transistor Q3. The circuit configuration of the soft start circuit 2 is as described above, and this operation will be described later.

The insulating converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side. The insulating converter transformer PIT has an EE-type core in which two sets of E-type cores made of, for example, a ferrite material are combined so that their magnetic legs face each other. A primary winding N1 and a secondary winding N2 are wound around the legs in a divided state using a divided bobbin. Then, a gap is formed with respect to the center magnetic leg. As a result, loose coupling with a required coupling coefficient can be obtained. The gap consists of two central magnetic legs of two sets of E-shaped cores.
It can be formed by making it shorter than the outer magnetic leg of the book. The coupling coefficient k is set to a loosely coupled state of, for example, k ≒ 0.85, so that a saturated state is hardly obtained.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary side parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary side winding N2 and the capacitance of the secondary side parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. With this parallel resonance circuit,
The alternating voltage induced in the secondary winding N2 becomes a resonance voltage.
That is, a voltage resonance operation is obtained on the secondary side.

In other words, this power supply circuit has a parallel resonance circuit on the primary side for making the switching operation a voltage resonance type, and a parallel resonance circuit on the secondary side for obtaining the voltage resonance operation. It is referred to as a “resonant switching converter”.

For the secondary side of the power supply circuit formed as described above, for the secondary side parallel resonance circuit (N2 // C2), for the secondary side rectifier diode DO1 and the smoothing capacitor CO1
And a secondary DC output voltage EO1 corresponding to an almost equal level of the alternating voltage generated in the secondary parallel resonance circuit (N2 // C2). ing. Here, a tap output is provided for the secondary winding N2, and a secondary rectifier diode D02 is provided between the tap output and the secondary ground as shown in FIG.
And a half-wave rectifier circuit composed of a smoothing capacitor CO2 to obtain a low-voltage secondary-side DC output voltage EO2. In this case, the secondary side DC output voltage EO1 is input to the control circuit 1 as a detection voltage for constant voltage control.

The control circuit 1 functions as an error amplifier having the DC output voltage EO1 as a detection input, and outputs a current of a variable level according to the level of the DC output voltage EO1. Here, when the level of the DC output voltage EO1 is reduced, the control circuit 1 is configured to increase the amount of output current in accordance with the reduction, and the output current is supplied to the photodiode of the photocoupler PC. Let it flow. That is, the output of the control circuit 1 is fed back to the primary side, whereby the switching frequency of the switching element Q1 is controlled as described below to achieve constant voltage control. The constant voltage control system is provided to insulate the primary side and the secondary side of the insulating converter transformer PIT in a DC manner.

Here, it is assumed that the load power of the secondary-side DC output voltage EO1 has changed such that the level of the secondary-side DC output voltage EO1 becomes low under heavy load conditions. In this case,
MOS-FE from control circuit 1 via photocoupler PC
The current level flowing to the gate of T (Q2) will be increased. Thereby, the MOS-FE is set by the resistor R1.
The control voltage (gate voltage) applied to the gate of T (Q2) also increases.

The conduction state of the MOS-FET (Q2) is controlled by the gate voltage whose level changes in this manner, whereby the conduction angle of the capacitor CB2 connected in series with the MOS-FET (Q2) is controlled. Is done. The series circuit of the MOS-FET (Q2) and the capacitor CB and the low-speed recovery type clamp diode DD1 are connected in parallel to the connection point of the capacitor CB1 and the inductor LB, respectively, so that the switching frequency of the switching element Q1 is changed. And the resonance current (base current IB of the switching element Q1) of the series resonance circuit (LB-CB1-NB) is controlled, so that the conduction angle and the switching frequency of the switching element Q1 are simultaneously variable. A controlled operation is obtained.

The operation in which the switching frequency and the conduction angle are simultaneously variably controlled is an operation based on the above-described complex control method. The period during which the switching element Q1 is off is constant, and the period during which the switching element Q1 is on (conduction angle). Is controlled. By such an operation, the secondary side DC output voltage E
When the level of O1 decreases, an effect of increasing the level is obtained, and a constant voltage is achieved.
That is, the control is performed such that the TON period during which the switching element Q1 is on is prolonged at the time of heavy load, and is shortened at the time of light load.

Thus, the secondary side DC output voltage EO1
The switching frequency is variably controlled in accordance with the level of, for example, the energy transmitted from the primary side to the secondary side is varied, and as a result, the level of the secondary side DC output voltage is varied. Become. That is, the power supply is stabilized.

As can be seen from the above description, in the power supply circuit of the present embodiment, the MOS
A control voltage variably controlled in accordance with the level of the secondary-side DC output voltage is applied to the gate electrode of the FET (Q2). For example, depending on the conduction state of the MOS-FET (Q2), The switching frequency of the switching element Q1 is variably controlled. In other words, the switching element is controlled by a complex control method in which the conduction angle and the switching frequency are simultaneously varied, but in order to obtain such an operation, the configuration may be such that the orthogonal control transformer is omitted. Will be possible. This allows
The problem of the variation of the inductance value caused by the variation of the gap of the orthogonal control transformer is solved. In particular, since the variation of the inductance of the drive transformer in the present invention is about ± 5%, it is possible to set a small margin for the range of the AC input voltage, so that the circuit design can be simplified. . Further, the problem of difficulty in manufacturing the orthogonal control transformer is solved. Further, the AC / DC power conversion efficiency is improved.

Further, since the switching frequency is not controlled by supplying control power to the control winding of the orthogonal control transformer, the reactive power at light load can be reduced, and the power loss can be reduced. Further, a small and lightweight drive transformer can be selected, and a low breakdown voltage and small capacity MOS-FET may be used as the auxiliary switching element, which is preferable in terms of design and cost.

The resonance current as the switching drive signal has a sinusoidal waveform, and the base current of the switching element has a larger peak value in the reverse base current than in the forward base current, and the fall time of the switching element Q1 is shorter. Therefore, the switching loss when the switching element Q1 is turned off is reduced. Further, there is obtained an advantage that the heat generation of the switching element Q1 is small.

Next, the operation of the soft start circuit 2 will be described. According to the constant voltage control operation described above,
If the control is performed by increasing the gate voltage of the MOS-FET (Q2), the switching frequency is controlled to be increased, thereby lowering the increased secondary-side DC output voltage. Therefore, at the time of startup, which is a transitional period from when the switch SW is turned on until the steady operation is performed, if an operation is performed to increase the gate voltage of the MOS-FET (Q2) and increase the switching frequency, The steep rise of the secondary side DC output voltage E at the time of the start is suppressed. That is, a soft start operation at the time of startup is obtained. The soft start circuit 2 of the present embodiment is configured to perform the above operation.

Here, when the switch SW is turned on and the line of the commercial AC power supply AC is connected, the AC input voltage V
AC is supplied to a rectifying / smoothing circuit including a rectifying / smoothing circuit (Di, Ci). That is, a rectified current flows into the smoothing capacitor Ci via the bridge rectifier circuit Di, and raises the rectified smoothed voltage Ei, which is a voltage across the smoothing capacitor Ci, to a level corresponding to the AC input voltage VAC. When the rectified smoothed voltage Ei increases in this way, a starting current flows into the base of the switching element Q1 via the starting resistors Rs1 to Rs2, and when the switching element Q1 is turned on by the starting current, The oscillation operation of the self-excited oscillation drive circuit is started, and the switching operation is also started.

As the switching operation is started, an alternating voltage is generated in the primary winding N1 of the isolated converter transformer PIT, and an alternating voltage is also generated in the tertiary winding N3 excited by the primary winding N1. Begins. This alternating voltage is rectified by the diode D1 and charged in the capacitor C1, so that the low-voltage DC voltage E1, which is a voltage across the capacitor C1, appears to rise to, for example, a steady level. Will be. At this time, on the secondary side of the insulating converter transformer PIT, charging of the smoothing capacitors CO1 and CO2 is started by a rectified current obtained by rectifying the alternating voltage excited by the secondary winding N2. Secondary side DC output voltage E
O1 and EO2 are in the process of increasing their levels. In other words, it can be said that the level change of the low-voltage DC voltage E1 on the primary side at the time of startup responds to the level change of the secondary-side DC output voltages EO1 and EO2.

Therefore, the soft start circuit 2 detects the level change of the low-voltage DC voltage E1 as the level change of the secondary-side DC output voltages EO1 and EO2. That is, in the soft start circuit 2, the voltage dividing resistor R5-is connected between the low-voltage DC voltage E1 and the primary side ground via the cathode and the anode of the Zener diode ZD.
R6 is connected, and a circuit composed of these elements detects an increase in the level of the low-voltage DC voltage E1 and causes a base current to flow from the connection point of the voltage dividing resistors R5-R6 to the base of the transistor Q3. .

When the base current starts flowing through the transistor Q3 after the start as described above, the current flows through the collector of the transistor Q3. Therefore, the base current of the transistor Q4 also flows through the resistor R3. Will be. As a result, in the transistor Q4, the current obtained from the line of the rectified smoothed voltage Ei via the starting resistor Rs1 flows from the emitter-collector via the resistor R2. Then, the gate voltage of the MOS-FET (Q2), which is the potential of the resistor R1, is increased by this current. At this time, since the secondary DC output voltage is in the process of rising and the control circuit 1 has not started operation, the photocoupler PC is not conducting.

By increasing the gate voltage of the MOS-FET (Q2) in this manner, the switching frequency of the switching element Q1 is controlled so as to increase to, for example, about 200 KHz which is close to the upper limit of the control range. Will be done. As described above, the constant-voltage control system according to the present embodiment performs control with a tendency to decrease the secondary-side DC output voltage as the switching frequency increases. By increasing the frequency, an operation of suppressing a sharp rise in the secondary side DC output voltage is performed.

When the soft start circuit 2 starts the above-mentioned operation and the base is made to flow to the transistor Q4, this base current flows into the time constant capacitor C3 as a charging current, and the time constant The potential of the capacitor C3 will be raised. Then, the base potential of the transistor Q4 increases, so that the collector current of the transistor Q4 decreases with the lapse of time.
The gate voltage of the OS-FET (Q2) also decreases. Therefore, the switching frequency gradually decreases. Then, as the switching frequency decreases, the level of the secondary DC output voltage is gradually increased to a steady level. The potential of the time constant capacitor C3 is equal to the base of the transistor Q4.
When the potential becomes equal to or higher than the emitter-to-emitter voltage, the base current stops flowing through the transistor Q4, and the transistor Q4
The forcible control of the switching frequency based on the collector current is stopped thereafter. That is, the soft start operation ends. Then, at the time when the soft start operation is completed, the secondary side DC output voltage has been raised to a level close to a steady state, and thereafter,
The control is shifted to the normal constant voltage control under the control of the control circuit 1. Further, according to the above description, the operation time from the start to the start and the end of the operation of the soft start circuit 2 is determined by the time constant (capacitance) of the time constant capacitor C3.

FIG. 2 shows the primary-side voltage resonance pulse V1 and the level change of the collector current IQ1 of the switching element Q1 over time at the time of startup. The primary-side voltage resonance pulse V1 is a pulse voltage obtained when both ends of the parallel circuit of the switching element Q1 and the parallel resonance capacitor Cr are turned off. Further, the characteristic shown in this figure is that the AC input voltage VAC = 10
0 V and the maximum load power Po = 150 W.

When the soft start circuit 2 of the present embodiment is not provided, for example, as shown by a broken line in FIG. 2, the primary side voltage resonance pulse V1 and the collector current IQ1
Both rise sharply immediately after startup. In this case, the switching operation is started approximately 20 ms after the switch is turned on. However, the primary-side voltage resonance pulse V1 and the collector current IQ1 continue to increase, and decrease after approximately 40 ms. Although it goes,
The steady level is reached after about 100 ms.

Therefore, in the present embodiment, for example, the capacitance of the time constant capacitor C3 is, for example, 1
Selection is performed so as to obtain a time constant of 00 ms. That is, as the soft start circuit 2, 100
It is set so that the above-described soft start operation is performed over a period of ms. In this way, as shown by the solid line in FIG. 2, the primary-side voltage resonance pulse V1 and the collector current IQ1 can be set to levels below the steady state at the time of startup. In other words, overvoltage and overcurrent occurring on the primary side during a certain period after the start-up are suppressed, and it is possible to stably shift to the subsequent steady-state operation.

FIG. 3 shows a change in the level of the secondary side DC output voltage EO1 at the time of startup with the elapse of time. Also in the case shown in this figure, when the soft start circuit 2 is not provided, the secondary side DC output voltage EO1 exceeds the steady level about 20 ms after the start, as shown by the broken line. Peaks in the state where
Until 0 ms elapses, a level higher than the steady state appears. On the other hand, when the soft start circuit 2 selected so as to obtain a time constant of 100 ms for the capacitance of the time constant capacitor C3 is provided, as shown by the solid line, the primary side voltage resonance pulse V1 and the It can be seen that a soft start operation in which the collector current IQ1 is set to a level equal to or lower than the steady state is obtained.

As described above, in the power supply circuit according to the present embodiment, the overcurrent protection and the overvoltage protection by the soft start circuit 2 are performed. For example, the overcurrent protection circuit shown in FIG. It is no longer necessary to have 10. For this reason, in the present embodiment, it is possible to reduce the power loss caused by the current detection resistor connected to the emitter of the switching element Q1. In the circuit of the present embodiment, the primary current detecting resistor R2 is inserted into the emitter of the switching element Q1, but in the present embodiment, the switching element Q1 and the clamp diode DD are connected to the primary. Since the connection is made directly without passing through the current detection resistor R2, no power loss occurs due to the current detection resistor as in the circuit shown in FIG.

Further, in the power supply circuit according to the present embodiment, a margin corresponding to the rise of the AC input voltage VAC and a malfunction margin during a normal operation due to a variation in components constituting the overcurrent limiting circuit 10 are taken into consideration. There is no need to design a circuit. For this reason, in the present embodiment, it is possible to suppress the rise of the voltage resonance pulse V1 at the time of the maximum load power, so that it is possible to select a low withstand voltage product for the switching element Q1. Also,
A small current capacity can be selected. If a switching element having a low withstand voltage and a small capacity can be selected, its shape can be made smaller. For example, in the circuit shown in FIG. 6, the switching element Q1 is a large package of TO-3P with a withstand voltage of 1200 V. In the circuit of the present embodiment shown in FIG.
This makes it possible to form a medium-sized package of TO-220 with a withstand voltage. In addition, since the switching characteristics are improved by using a low breakdown voltage product, the power loss in the switching element Q1 is also reduced.

Further, for example, in the overcurrent limiting circuit 10 provided in the power supply circuit shown in FIG.
E is a winding resistance, and the transistor Q10 and the diode D2 have a current capacity of 1A. On the other hand, if the configuration of the soft start circuit 2 shown in FIG. 1 is used, a semiconductor having a lower withstand voltage and a smaller current capacity can be selected as a semiconductor provided in each circuit system. Circuit size and cost can be reduced.

As can be seen from the description of the time constant capacitor C3 with reference to FIGS. 2 and 3, the soft start circuit 2 of the present embodiment The operating time of the soft start circuit 2 is changed and set by changing the capacitance (time constant) of the time constant capacitor C3 in accordance with the rise time at the time of startup of the side voltage resonance pulse V1 and the collector current IQ1. . In this way, the soft start circuit 2 can be operated with an appropriate operation time in a power supply circuit of any specification. For example, in the power supply circuits shown in FIGS. 5 and 6, it is difficult to take a margin for a malfunction at the time of start-up, but as in the present embodiment, the soft start circuit 2 whose operation time can be variably set is provided. With such a configuration, it is possible to greatly increase the margin of malfunction at startup.

Although the embodiments have been described above, the present invention is not limited to the configurations shown in the drawings as the above embodiments. For example, in each of the above-described embodiments, the case of a single-end system including one set of switching elements is described. However, a so-called push-pull system including two sets of switching elements is used.
It may be a self-excited voltage resonance type converter. Also, the secondary side may be provided with a rectifier circuit having a circuit configuration other than that shown in each drawing.

[0064]

As described above, the present invention employs, as a basic configuration, a configuration in which a quadrature control transformer is omitted as a composite resonant converter which is stabilized by a self-excited switching frequency control method. In addition, for example, at the time of starting the power supply, a soft start circuit that operates based on the detection of the level of the primary side DC voltage generated based on the alternating voltage of the tertiary winding wound on the insulating converter transformer is provided. I have. As a result, in the present invention, it is possible to protect the circuit without providing an overcurrent detection circuit or the like which has been used so far. For this reason, since a low withstand voltage product can be selected as the switching element, the switching element can be made small and inexpensive. That is, the circuit can be reduced in size and cost. In addition, by selecting a low breakdown voltage product, switching characteristics are improved, and power conversion efficiency is also improved. In addition, since it is not necessary to insert an overcurrent detection resistor provided in the above-described overcurrent detection circuit, it is possible to reduce power loss in this regard.

[Brief description of the drawings]

FIG. 1 is a circuit diagram illustrating a configuration example of a switching power supply circuit according to an embodiment of the present invention.

FIG. 2 is an explanatory diagram illustrating a change in level of a primary-side voltage resonance pulse and a collector current of a switching element at the time of startup in the power supply circuit of the present embodiment.

FIG. 3 is an explanatory diagram showing a level change of a secondary-side DC output voltage at the time of startup in the power supply circuit of the present embodiment.

FIG. 4 is an explanatory diagram of a drive transformer of the power supply circuit according to the embodiment;

FIG. 5 is a circuit diagram showing a configuration example of a switching power supply circuit as a prior art.

FIG. 6 is a circuit diagram showing a configuration example when an overcurrent limiting circuit is provided as a switching power supply circuit of the prior art.

FIG. 7 is a waveform diagram showing a primary-side parallel resonance pulse and a switching current corresponding to a steady state and a maximum load power in the circuit shown in FIG. 6;

[Explanation of symbols]

1 control circuit, SW switch, PC photo coupler,
Q1 switching element, Q2 MOS-FET PIT
Insulation converter transformer, CDT drive transformer, N1 primary winding, N2 secondary winding, N3 tertiary winding,
DD clamp diode, Cr primary side parallel resonance capacitor, NB drive winding, CB1, CB2 capacitor, LB
Inductor, Q3, Q4 transistor, R5, R6 voltage dividing resistor

Claims (2)

[Claims] A switching means having a switching element for switching a DC input voltage; an insulating converter transformer for transmitting an output of the switching means obtained on a primary winding to a secondary winding; A primary-side parallel resonance circuit formed by a primary winding and a primary-side parallel resonance capacitor and provided so that the operation of the switching means is of a voltage resonance type; and a secondary winding wound around the insulating converter transformer. And a secondary parallel resonance circuit formed by connecting the secondary parallel resonance capacitor in parallel, and a DC output voltage by inputting an alternating voltage obtained in the secondary parallel resonance circuit and performing a rectification operation. DC output voltage generating means configured to obtain A drive transformer having a primary winding for detecting the output of the switching means and a secondary winding for transmitting a switching voltage detected by the primary winding; and a drive transformer connected in series with at least the secondary winding of the drive transformer. A switching drive means for supplying a switching drive signal to a switching element, the switching drive means being provided with a series resonance circuit including a series resonance capacitor and a series resonance inductor, and a connection point between the series resonance capacitor and the series resonance inductor. A series circuit comprising a divided capacitor and a conduction control element connected thereto; and a conduction amount of the conduction control element varied according to a level of the DC output voltage obtained by the DC output voltage generating means, thereby forming the divided capacitor. The switching frequency of the switching element is controlled by variably controlling the amount of current flowing through And a tertiary winding wound on the primary side of the insulating converter transformer PIT. Primary-side DC voltage generating means configured to obtain a primary-side DC voltage by rectifying an alternating voltage that is obtained, based on the primary-side DC voltage obtained at the time of startup,
A switching power supply circuit, comprising: soft start means for variably controlling the amount of conduction of the conduction control element so that the constant voltage control means tends to operate to reduce the DC output voltage. 2. The switching power supply circuit according to claim 1, wherein the soft start means includes a time constant circuit for setting an operation period at the time of starting.