JP2003092879A - Switching power supply - Google Patents

Abstract

PROBLEM TO BE SOLVED: To lower the breakdown voltage of a switching element. SOLUTION: As a self-excited composite resonance type converter being settled by a switching frequency control system, an arrangement where an orthogonal control transformer is eliminated is employed. When the load being imposed to a switching element is increased due to short circuit on the load side, for example, an overload protective circuit detects a current of undue level being generated in the power supply circuit and then starts operation. The circuit can thereby be protected without requiring a conventional overcurrent detection circuit, or the like.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply in various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a switching power supply circuit that employs a switching converter such as a flyback converter or a forward converter is widely known. Since the switching operation waveforms of these switching converters are rectangular waves, there is a limit in suppressing switching noise. Also, due to its operating characteristics,
It is known that there is a limit to the improvement of power conversion efficiency. Therefore, the applicant has previously proposed various switching power supply circuits using various resonant converters.
The resonant converter can easily obtain high power conversion efficiency, and the switching operation waveform has a sinusoidal waveform to realize low noise. It also has the advantage that it can be configured with a relatively small number of parts.

The circuit diagram of FIG. 7 shows an example of a switching power supply circuit as a prior art, which can be constructed based on the invention previously proposed by the present applicant. As a basic configuration of the power supply circuit shown in this figure, a voltage resonance type converter is provided as a primary side switching converter.

In the power supply circuit shown in this figure, the bridge rectifier circuit Di and the smoothing capacitor Ci are used to change the AC input voltage VAC from the commercial AC power supply AC (AC input voltage VAC) to 1
The rectified and smoothed voltage Ei corresponding to the doubled level is generated. Further, a switch SW for turning on / off the power source is inserted in the line of the commercial AC power source AC. In addition, an inrush current limiting resistor Ri is also inserted in the line of the commercial AC power supply AC so as to suppress the inrush current flowing into the smoothing capacitor when the power is turned on, for example.

As the voltage resonance type converter which receives and connects the rectified and smoothed voltage Ei (DC input voltage), a single-ended system using one stone is adopted. The drive system is a self-excited type. In this case, the switching element Q1 forming the voltage resonance type converter is
A high voltage bipolar transistor (BJT; junction type transistor) is selected. A primary side parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. A series connection circuit of a clamp diode DD and a resistor RD is connected between the base and the emitter. Here, the parallel resonance capacitor Cr is the primary winding N1 of the insulating converter transformer PIT.
The primary side parallel resonance circuit is formed together with the leakage inductance L1 obtained in FIG. A self-excited oscillation drive circuit composed of drive winding NB-resonance capacitor CB-base current limiting resistor RB is connected to the base of switching element Q1. The switching element Q1 is switching-driven by being supplied with a base current based on an oscillation signal generated by the self-excited oscillation drive circuit. It is to be noted that, at the time of starting, it is started by the starting current flowing from the line of the rectified and smoothed voltage Ei to the base via the starting resistor RS.

The orthogonal control transformer PRT is constructed by winding the control winding NC so that the winding directions of the drive winding NB and the current detection winding NA are orthogonal to each other. It is provided to control the switching frequency of the primary side voltage resonance type converter as described later.

The insulating converter transformer PIT is provided for transmitting the switching output of the switching converter obtained on the primary side to the secondary side. This isolation converter transformer PIT has a primary winding N with respect to the EE type core.
By winding 1 and the secondary winding N2 separately, and forming a gap G with respect to the central magnetic leg, a loose coupling state with a required coupling coefficient can be obtained and a saturation state can be obtained. I try not to be easily affected.

The primary winding N1 of the insulating converter transformer PIT is connected between the line of the DC input voltage (rectified and smoothed voltage Ei) and the collector of the switching element Q1. The switching element Q1 performs switching with respect to the DC input voltage, whereby the switching output of the switching element Q1 is supplied to the primary winding N1, and an alternating voltage having a cycle corresponding to the switching frequency is generated. To do.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary side parallel resonant capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonant capacitor C2 are used. A parallel resonant circuit is formed. With this parallel resonant circuit,
The alternating voltage induced in the secondary winding N2 becomes a resonance voltage.
That is, the voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, the parallel resonance circuit for making the switching operation a voltage resonance type is provided on the primary side, and the parallel resonance circuit for obtaining the voltage resonance operation is provided on the secondary side. In the present specification, the switching converter configured such that the resonance circuits are provided for the primary side and the secondary side as described above is also referred to as a “composite resonance type switching converter”.

Insulation converter transformer PIT in this case
On the secondary side, first, the half-wave rectification circuit is formed by connecting the anode of the rectifying diode DO1 to the end of the secondary winding N2 and connecting the cathode to the positive terminal of the smoothing capacitor CO1. Is forming. Depending on this half-wave rectifier circuit, the secondary side DC output voltage EO1 is obtained across the smoothing capacitor CO1. Further, in this case, a tap is provided on the secondary winding N2, and a half-wave rectifying circuit including a rectifying diode D02 and a smoothing capacitor C02 is formed for the tap output as shown in the figure. Then, depending on this half-wave rectification circuit, the secondary side DC output voltage E lower than the secondary side DC output voltage EO1 is obtained.
O2 is obtained. In addition, specifically, the secondary side DC output voltage EO1 = 135V and the secondary side DC output voltage EO2 = 15V.

These secondary side DC output voltages EO1 and EO2 are
It will be supplied to each required load circuit. The secondary side DC output voltage EO1 is used as a detection voltage for the control circuit 1, and the secondary side DC output voltage EO2 is used as the detection voltage.
It is branched and output as the operating power supply of.

The control circuit 1 causes a DC current variable according to the level of the secondary side DC output voltage EO1 to flow through the control winding NC of the orthogonal control transformer PRT as a control current. By varying the level of the control current flowing through the control winding NC, the quadrature control transformer PRT is controlled to vary the inductance of the drive winding NB. As a result, the resonance frequency of the resonance circuit composed of the drive winding NB and the resonance capacitor CB in the self-excited oscillation drive circuit changes, and the switching frequency of the switching element Q1 is variably controlled. By varying the switching frequency of the switching element Q1 in this manner, the secondary side DC output voltage is controlled to be constant. That is, the power supply is stabilized. In addition, in this specification, regarding constant voltage control by such an operation,
It is also called a "switching frequency control method".

By the way, in the power supply circuit shown in FIG. 7, when the switch SW is turned on and the commercial AC power supply AC is turned on, the smoothing capacitor Ci is passed from the inrush current limiting resistor Ri through the diode of the bridge rectifier circuit Di. A current flows in, and the rectified and smoothed voltage Ei, which is the voltage across the smoothing capacitor Ci, is raised to a level corresponding to the AC input voltage VAC. Then, a starting current flows into the base of the switching element Q1 from the line of the rectified and smoothed voltage Ei through the starting resistor RS, the switching element Q1 is turned on, oscillation starts, and a switching operation is started.

However, with such a start-up operation, an excessive charging current flows to the smoothing capacitors CO1 and CO2 on the secondary side of the insulating converter transformer PIT. Further, an excessive collector current also flows from the smoothing capacitor Ci to the collector of the switching element Q1 via the current detection winding NA-primary winding N1. Further, the levels of the secondary side DC output voltages EO1 and EO2 at this time are in a transient state where they rise to a required level. Therefore, at this time, switching according to the level of the secondary side DC output voltage EO1 is performed. No frequency control is performed. At this time, the switching element Q1 operates at the lowest switching frequency determined by the inductance of the drive winding NB and the capacitance of the capacitor CB. In the case of this power supply circuit, when the switching operation is performed at the lowest switching frequency, the ON period becomes longer between the ON period and the OFF period of the switching element Q1. Therefore, in the OFF period of the switching element Q1, The peak level of the voltage resonance pulse as the parallel resonance voltage V1 generated at both ends of the parallel circuit of the switching element Q1 // parallel resonance capacitor Cr also becomes excessive.

In addition to the start-up, the switching element Q may also be activated by short-circuiting the load side, for example.
The burden on 1 will increase. That is, when the load is short-circuited, the level of the current flowing through the secondary side DC output voltage line is higher than in the normal state, and an excessive current flows through this line. Further, depending on the operation of the control circuit 1 at this time, the switching frequency of the switching element Q1 is lowered to almost the lower limit. Therefore, an excessive level of collector current also flows through the switching element Q1 on the primary side, and an excessive level of voltage is generated as the primary side parallel resonance voltage V1. As a result, the load on the switching element Q1 increases. Of.

Therefore, as a practical matter of the circuit having the configuration shown in FIG. 7, when an excessive collector current is generated in the switching element Q1 at the time of such an abnormality at the time of start-up and load short-circuit, An overcurrent limiting circuit may be provided to limit the current. FIG. 8 shows a configuration example of a power supply circuit provided with an overcurrent limiting circuit based on the circuit configuration shown in FIG. In this figure, the same parts as those in FIG. 7 are designated by the same reference numerals and the description thereof will be omitted.

The overcurrent limiting circuit 10 shown in FIG. 8 detects the current flowing through the emitter of the switching element Q1 by a circuit composed of a current detecting resistor RE and voltage dividing resistors R11 and R12. Then, by detecting an excessively high level emitter current and turning on the transistor Q10, the base current of the switching element Q1 is caused to flow from the diode D2 through the collector-emitter of the transistor Q10. As a result, the forward base current of the switching element Q1 is suppressed, and it becomes possible to limit the excessive collector current flowing in the collector of the switching element Q1.

[0019]

By the way, the power supply circuits shown in FIGS. 7 and 8 are orthogonal control transformers PRT.
This orthogonal control transformer PRT is equipped with
In order to reduce the amount of control current flowing through the control winding, the gap G must be made as small as about 10 μm. For this reason, an accuracy error of the gap thickness inevitably occurs at the time of manufacture, but this causes variations in the inductance value of the drive winding NB wound around the orthogonal control transformer PRT. Further, variations in the magnetic permeability of the ferrite core, deviations when the magnetic legs are joined, and the like also cause variations in the inductance value of the drive winding NB. Therefore, the allowable value of the inductance LB is
For example, it must fluctuate ± 10%. Therefore, the amplification factor hFE of the switching element Q1 and the accumulation time t
Although the stg varies, the constant voltage guarantee range of the composite resonance type converter against this variation is, for example, the AC input voltage VAC = 10 when the commercial AC power supply is a 100V system.
To achieve 0V ± 10%, the orthogonal control transformer PR
The variable inductance range of T must be designed with a sufficient margin. In other words, it becomes difficult to design the margin for practical use.

The winding specifications of the orthogonal control transformer PRT are as described above, and it is not necessary to wind the control winding NC, the detection winding NA and the drive winding NB in directions orthogonal to each other. Moreover, the winding process becomes very complicated. That is, the orthogonal control transformer PRT is difficult to manufacture, and cost reduction is also difficult.

The DC control current flowing through the control winding NC of the orthogonal control transformer PRT is supplied from the DC output voltage EO2 line (15V line) on the secondary side of the insulating converter transformer PIT, and its supply power is 0, for example. .9W ~
Although it fluctuates within the range of 0.15 W, this supplied power is reactive power, and the power loss at light load increases.

In addition, the waveform diagram of FIG. 9 shows that the operation of the circuit shown in FIG.
1 and the collector current IQ1 of the switching element Q1 are shown. For example, when the load is intermediate, the AC input voltage VAC =
At the time of steady operation of about 120 V,
As shown in (a) and (b), the voltage resonance pulse V1 is about 700 Vp, and the collector current IQ1 of the switching element Q1 is about 4.5 A. On the other hand, under the condition that the maximum load power (Pomax = 150 W) is set and the AC input voltage VAC is increased to about 120 V, as shown in FIGS. 9C and 9D, the voltage resonance pulse V
1 rises to about 900 Vp and switching element Q1
Collector current IQ1 rises to about 6.5A. That is, the voltage resonance pulse V1 and the collector current IQ1 at the maximum load power are increased by about 20 to 30% as compared with the steady state. Since such an operation includes the overcurrent limiting circuit 10 in the circuit shown in FIG. 8, the margin corresponding to the rise in the AC input voltage VAC and the variation in the components forming the overcurrent limiting circuit 10 are caused. This is due to the result of the circuit design considering the malfunction margin at the time of steady operation. The switching element Q1 is, for example, 1200 V, corresponding to the maximum load power.
It is necessary to select a high withstand voltage product. The higher the breakdown voltage of the switching element, the larger and more expensive it becomes. Also, the switching characteristics are inferior.

[0023]

Therefore, in view of the above problems, the present invention is configured as a switching power supply circuit as follows. That is, a switching means including a switching element for inputting a DC input voltage to perform switching, a primary winding and a secondary winding, and the output of the switching means obtained in the primary winding is used as the secondary output. A primary side parallel resonance formed by an insulating converter transformer transmitting to the winding, a primary winding of the insulating converter transformer and a primary side parallel resonance capacitor, and provided so as to make the operation of the switching means a voltage resonance type. Input the circuit and the secondary side resonance circuit formed by connecting the secondary side resonance capacitor to the secondary winding of the isolation converter transformer, and the alternating voltage obtained in the secondary side resonance circuit. And a DC output voltage generating means configured to obtain a DC output voltage by performing a rectifying operation. Further, at least the detection winding and the drive winding are wound, and the detection winding is connected in series to the primary winding of the insulating converter transformer so that the output of the switching means is transmitted to the drive winding. Drive transformer, a series resonance circuit formed by the drive winding and a resonance capacitor, and switching drive means for switching the switching element based on the output of the series resonance circuit. A control element, a time constant circuit provided so that the waveform of the control input terminal potential of the transistor element as the conduction control element is substantially sawtooth, and the control input terminal potential of the control input terminal potential in accordance with the level of the DC output voltage. By varying the rising period to variably control the amount of current conduction in the conduction control element, the switching element By varying the level of the drive signal and variably controlling the switching frequency of the switching element according to the varied level of the drive signal, constant voltage control of the DC output voltage is performed. And constant voltage control means. Further, by detecting an excessive level of current generated in the power supply circuit, the switching element operates so that the switching frequency is maintained within a predetermined range, and the excessive level generated in the switching element is detected. A protective means is provided to control such that a large current and voltage are suppressed.

The power supply circuit having the above-mentioned structure has a basic structure as a composite resonance type converter which is provided with a drive transformer and drives the switching element by self-excitation. Further, for constant voltage control, a conduction control element connected to the switching element is provided. When the conduction control element is caused to have a waveform of the control input terminal potential (base-emitter voltage) of a substantially sawtooth wave, the sawtooth wave starts rising and reaches a peak. By varying the rising period, the current conduction amount in the conduction control element can be varied. Thus, the base current of the switching element is controlled by the amount of current conduction in the conduction control element, and the switching frequency of the switching element is variably controlled. With such a configuration of constant voltage control, it becomes possible to omit the orthogonal control transformer used for variable switching frequency control in the case of the self-excited type, for example.

Further, depending on the protection means,
It becomes possible to suppress an excessive resonance voltage applied to the switching element and an excessive switching output current (collector current) flowing through the switching element in the event of an abnormality such as a short-circuit on the secondary side load. As a result, it is possible to reduce the load on the switching element when the load is short-circuited, and it is possible to ensure the reliability of the circuit against an abnormality such as when the load is short-circuited.

[0026]

BEST MODE FOR CARRYING OUT THE INVENTION <First Embodiment> FIG. 1 shows the configuration of a power supply circuit according to a first embodiment of the present invention. The power supply circuit shown in FIG. 1 has a configuration as a composite resonance type switching converter having a voltage resonance type converter on the primary side and a parallel resonance circuit on the secondary side. In the power supply circuit shown in this figure, first, a bridge rectification circuit D is used as a rectification smoothing circuit for inputting a commercial AC power supply (AC input voltage VAC) to obtain a DC input voltage.
A full-wave rectifier circuit including i and a smoothing capacitor Ci is provided to generate a rectified and smoothed voltage Ei corresponding to a level of the AC input voltage VAC. Further, a switch SW for turning on / off the power source is inserted in the line of the commercial AC power source AC.

As a switching converter which receives the rectified and smoothed voltage Ei (DC input voltage) and is intermittent,
A voltage resonance type converter including a stone switching element Q1 and performing a switching operation by a so-called single-ended system is provided. The voltage resonance type converter here has a self-excited configuration, and a high breakdown voltage bipolar transistor (BJT; junction type transistor) is used as the switching element Q1.

The collector of the switching element Q1 is connected to the positive electrode of the smoothing capacitor Ci via the primary winding N1 of the insulating converter transformer PIT and the detection winding NA of the drive transformer CDT. Further, the emitter is connected to the primary side ground via the resistor RE. This resistor RE is, for example, the base-emitter voltage VB of the switching element Q1.
A resistor having a resistance value such that E1 is 1.4 V or more is selected. As a result, the base-emitter voltage VBE1 of the switching element Q1 is 1.4 V, which was 0.7 V in the past.
It will be raised to V and the control range for stabilization will be expanded.

Further, as shown in the figure, the line from the connection point of the resistor RE and the emitter of the switching element Q1 is inputted to the overload protection circuit 2 described later, and the resistor RE is short-circuited on the load side, for example. In the event of
The excessive current generated in the switching element Q1 is detected.

Further, the collector of the switching element Q1
A parallel resonance capacitor Cr is connected in parallel between the emitters. The capacitance of the parallel resonance capacitor Cr and the primary winding N of the insulating converter transformer PIT
The leakage inductance obtained in 1 forms a primary side parallel resonant circuit. Then, according to the switching operation of the switching element Q1, the resonance operation by the parallel resonant circuit is obtained, so that the switching element Q1
The switching operation of is a voltage resonance type.

A clamp diode DD is connected between the base and emitter of the switching element Q1. Here, the anode of the clamp diode DD is connected to the emitter (primary side ground), and the cathode is connected to the base. In the case of the first embodiment, a slow recovery type diode is selected as the clamp diode DD.

The drive transformer CDT is provided to drive the switching element Q1 by self-excitation. In this case, the primary side of the drive transformer CDT is the detection winding N
A, the detection winding NA is an insulating converter transformer P
By being connected in series to the primary winding N1 of IT, the switching output of the switching element Q1 transmitted to the primary winding N1 of the insulating converter transformer PIT is detected. The drive winding NB is connected to the secondary side where the alternating voltage obtained in the detection winding NA is induced.
Is wound. The drive winding NB forms a self-excited oscillation drive circuit for switching-driving the switching element Q1.

The drive transformer CDT around which the above windings (NA, NB) are wound is, for example, an H-shaped ferrite magnetic core as shown in FIG. 4 or EI in FIG.
A type ferrite core can be used. In the case of FIG. 4, it is formed by winding the detection winding NA and the drive winding NB on the H-shaped ferrite magnetic core 100.

In the case of FIG. 5, the I type core 102 and the E type core 103 are combined as shown to form an EI type core. Then, the split bobbin 10 is attached to the central magnetic leg of the E-shaped core 103
3 is arranged and the detection winding NA and the drive winding NB are respectively wound on the split bobbin 103 as shown in the drawing.

The drive transformer CDT having the structure shown in FIGS. 4 and 5 is considerably small and lightweight when compared with the orthogonal control transformer PRT described with reference to FIG. 8, for example. Further, with such a configuration, the drive transformer CDT has characteristics as a supersaturation reactor, so that the drive transformer CDT
The inductance characteristics of the detection winding NA and the drive winding NB wound around T have an inductance value as an element when the current flowing through each winding is at "0" level, and the absolute value level of this current is large. As it becomes, the inductance will decrease proportionally.

Since the detection winding NA of the drive transformer CDT is connected in series with the primary winding N1 of the insulating converter transformer PIT, the drive winding NB of the drive transformer CDT in the self-excited oscillation drive circuit is connected to the primary winding N1. An alternating voltage is excited by the transmitted switching output voltage. In the self-excited oscillation drive circuit described above, the capacitor CB and the inductance of the drive winding NB
Form a series resonant circuit.

As described above, the drive winding NB of the self-excited oscillation drive circuit is excited by the detection winding NA to generate an alternating voltage as a drive voltage. The drive voltage causes the series resonance circuit (NB-CB) to oscillate in a self-excited manner to obtain a resonance output. The resonance output causes the base current IB as a switching drive signal to flow through the base of the switching element Q1. As a result, the switching element Q1 performs the switching operation at the switching frequency determined by the resonance frequency of the series resonance circuit. The switching output obtained at the collector is used as the primary winding N1 of the insulation converter transformer PIT.
Communicate to.

Further, the starting resistor RS is inserted between the line of the rectified and smoothed voltage Ei and the base of the switching element. For example, when the power supply is activated, a base current flows from the rectified and smoothed voltage Ei through the activation resistor RS to the base of the switching element Q1 to start the switching operation.

Further, the circuit shown in this figure is provided with a conduction control element Q2 which is an NPN type bipolar transistor for small signals. The collector of the conduction control element Q2 is connected to the base of the switching element Q1. The emitter of the conduction control element Q2 is grounded to the primary side ground. Therefore, the collector-emitter of the conduction control element Q2 is connected in parallel to the clamp diode DD which is a Schottky diode, for example.

A tertiary winding N3 is wound around the primary side of the insulating converter transformer PIT, and a half wave composed of a diode D1 and a capacitor CO3 is provided to the tertiary winding N3 as shown in the figure. By connecting a rectifier circuit, a low voltage DC voltage is obtained. The low-voltage DC voltage is connected to the base (control input terminal) of the conduction control element Q2 via the phototransistor of the photocoupler PC and the resistor R1. A resistor R2 is inserted between the base and emitter of the conduction control element Q2. A base-emitter voltage VBE2 (control input terminal potential) is obtained across the resistor R2. Since the circuit on the base side of the conduction control element Q2 is formed as described above, the conduction control element Q2 functions as the collector current IQ2 in accordance with the amount of current conduction that changes in the phototransistor of the photocoupler PC. The conduction amount is controlled so as to be variable. In addition, photo coupler P
The current conduction amount of the C phototransistor is controlled by
This is the operation of the control circuit 1 provided on the secondary side, which will be described later.

Further, in this embodiment, a small-capacity film capacitor C3 is inserted between the base and the emitter of the conduction control element Q2. That is, the capacitor C3 is a resistor R2 that generates the base-emitter voltage VBE2.
It can be considered that a time constant circuit is formed by being connected in parallel with respect to. The capacitor C3 is charged and discharged with a current which tends to flow to the base of the conduction control element Q2 via the phototransistor of the photocoupler PC and the resistor R1.
As will be described later, the base-emitter voltage VBE is almost corresponding to the period during which the switching element Q1 is turned on.
The action of increasing the level of 2 proportionally is obtained.
In other words, the base-emitter voltage VBE2 during the period when the switching element Q1 is turned on has a substantially sawtooth wave.

The insulating converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side. This insulating converter transformer PIT is provided with an EE type core in which, for example, two sets of E type cores made of ferrite material are combined so that their magnetic legs face each other. Is used to wind the primary winding N1 and the secondary winding N2 separately. A gap is formed with respect to the central magnetic leg. As a result, a loosely coupled state with a required coupling coefficient is obtained. The gap can be formed by making the center magnetic legs of the two sets of E-shaped cores shorter than the two outer magnetic legs. Further, as the coupling coefficient k, for example, a loose coupling state of k≈0.85 is obtained, and accordingly, it is difficult to obtain a saturated state.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In the circuit shown in FIG. 1, the secondary side parallel resonance capacitor C2 is connected to the secondary winding N2.
Are connected in parallel. Therefore, in this case, the parallel resonant circuit is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonant capacitor C2. With this parallel resonant circuit, the secondary winding N2
The alternating voltage induced in the coil and the alternating voltage obtained in the detection winding NA become a resonance voltage. That is, the voltage resonance operation is obtained on the secondary side.

That is, this power supply circuit is provided with a parallel resonance circuit for providing a voltage resonance type switching operation on the primary side and a parallel resonance circuit for obtaining a voltage resonance operation on the secondary side. It is a "resonant switching converter".

For the secondary side of the power supply circuit formed as described above, there is provided a half-wave rectifying circuit consisting of a secondary side rectifying diode DO1 connected to the secondary winding N2 and a smoothing capacitor CO1. As a result, the secondary side DC output voltage EO1 corresponding to the almost equal level of the alternating voltage induced in the secondary winding N2 is obtained. Further, here, a tap output is provided for the secondary winding N2, and as shown in the drawing, between the tap output and the secondary side ground, there is provided a half side composed of a secondary side rectifying diode DO2 and a smoothing capacitor CO2. By connecting the wave rectification circuit, a low-voltage secondary side DC output voltage EO2 is obtained.

The control circuit 1 functions as an error amplifier which receives the DC output voltage EO1 as a detection input. That is, for the constant voltage control, it functions so that a current of a level corresponding to the DC output voltage EO1 flows through the photodiode of the photocoupler PC. A current corresponding to the amount of current flowing through the photodiode arranged on the secondary side flows through the phototransistor arranged on the primary side, which is caused by the conduction control element Q2.
It appears as a change in the base-emitter voltage VBE2.
A base current corresponding to the base-emitter voltage VBE2 is supplied to the base of the conduction control element Q2.
As a result, the level of the collector current IQ2 flowing through the collector of the conduction control element Q2 is also changed according to the level of the secondary side DC voltage EO2. Depending on this operation, the switching frequency of the switching element Q1 is changed, and thereby the secondary side DC output voltage is stabilized so as to be constant. The constant voltage control operation for the secondary side DC output voltage in the power supply circuit will be described later.

Further, as described above, the conduction control element Q
By inserting the capacitor C4 between the base 2 and the emitter 2, the base-emitter voltage VBE2 becomes a sawtooth wave substantially corresponding to the period when the switching element Q1 is turned on. As a result, the switching element Q1 The effect of reducing the switching loss is obtained.

The power supply circuit of this embodiment is provided with an overload protection circuit 2 as shown in addition to the configuration described above. This overload protection circuit 2 is shown in FIG.
As shown in the broken line, the resistor R3, R4, R5, the transistor Q3, the small signal thyristor Q4, the capacitor C4, and the Zener diode D2 are provided. First, in the overload protection circuit 2, the resistor R3 is connected in series with the positive terminal of the smoothing capacitor Ci as shown in the figure.
The cathode of the Zener diode D2 is connected to the resistor R3, and the anode of the Zener diode D2 is grounded to the primary side ground. Further, the emitter of a PNP transistor Q3 is connected to the connection point between the resistor R3 and the Zener diode D2. With such a configuration, the rectified and smoothed voltage Ei is input via the resistor R3, and the voltage thus obtained is stably obtained by the Zener diode D2 as a DC voltage of, for example, about 5V. Then, a current corresponding to this DC voltage of about 5 V is supplied to the emitter of the transistor Q3, and this is used as the operating power supply of the overload protection circuit 2.

The collector of the transistor Q3 is connected to the base of the above-mentioned conduction control element Q2 via a resistor. The base of the transistor Q3 is connected to the anode of the small signal thyristor Q4 via a resistor. The cathode of the small signal thyristor Q4 is grounded to the primary side ground. A resistor R5 is connected to the gate, and the resistor R5 is connected to the connection point between the emitter of the switching element Q1 and the resistor RE as described above. As shown in this figure, small signal thyristor Q
If a parallel circuit composed of a capacitor C4 and a resistor R4 is interposed between the gate of 4 and the primary side ground, an excessive level of current detected by the resistor RE will first flow into this parallel circuit. Then, a voltage corresponding to this current level is applied to the gate of the small signal thyristor Q4 after that, so that circuit malfunction can be prevented.

According to the overload protection circuit 2 configured as described above, when an abnormal condition such as a short-circuit on the load side occurs, an excessive current and voltage are generated in the switching element Q1 and the current flows through the resistor RE. By detecting an excessive level of current, the amount of base current of the conduction control element Q2 is increased, and the switching element Q2 is increased.
Although the operation of controlling by increasing the switching frequency of 1 can be obtained, this operation will be described later.

The above is the description of the structure of the power supply circuit in the present embodiment. With such a structure, the operation shown in FIG. 2 can be obtained in the main part of the power supply circuit. FIG. 2 is a waveform diagram showing the operation of the main part of the power supply circuit of this example. The operation waveforms of each part shown in this figure are obtained at the rated maximum load within the steady operation range of this circuit.

First, in the drive winding NB wound around the drive transformer CDT, an alternating voltage due to the excitation action is generated as described above. Then, the self-excited oscillation drive circuit (N
B-CB) performs self-excited oscillation operation based on the alternating voltage generated in the drive winding NB. That is, the series resonance circuit formed by the time constant capacitor CB and the drive winding NB performs a resonance operation, and the resonance output is caused to flow as the base current to the base of the switching element Q1.

Depending on the resonance operation of this series resonance circuit (CB-NB), a sinusoidal resonance voltage corresponding to the switching period is generated in the time constant capacitor CB, which causes
Base current IB flowing through the base of switching element Q1
Becomes as shown in FIG. Further, this resonance voltage causes the damper current ID flowing in the clamp diode DD1 as shown in FIG. 2 (f) and the collector current IQ2 of the conduction control element Q2 as shown in FIG. 2 (g) to branch and flow. It

Period T during which the switching element Q1 is turned on
When ON, the clamp diode DD first conducts, and the damper current ID flows as shown in FIG. 2 (f). The damper current ID in this period flows into the primary winding N1 via the base-collector PN junction of the switching element Q1. Accordingly, the collector current IQ1 of the switching element Q1 in this period is as shown in FIG.
As shown in (b), a waveform flowing in the negative direction is obtained. Further, as shown in FIG. 2 (c), the base current IB rises to the positive polarity at the start of the period TON and immediately falls to the 0 level immediately thereafter.

After that, the clamp diode DD is turned off. At this time, the base current IB (Fig. 2 (c))
As a result, the forward current IB1 due to the positive polarity flows. This forward current IB1 is the drive winding NB → time constant capacitor CB
It is obtained by causing a positive resonance current to flow in the path of. Then, after that, when the resonance current reverses to the negative polarity, the base current IB also reverses to the negative polarity due to the base accumulated carrier disappearance time tstg. As a result, in this period, as shown in FIG. 2C, the reverse current IB2
Will flow. In this way, the base current IB
As the current flows, the switching element Q1 becomes conductive, and as shown in FIG. 2B, the collector of the switching element Q1 has a positive collector current IQ1.
Flows.

Then, the base current IB (FIG. 2 (c))
Becomes zero level when the base accumulated carrier extinction time tstg in which the reverse current IB2 flows is completed, whereby the switching element Q1 shifts to a period TOFF in which it is turned off.

The resonance voltage V1 obtained across the primary side parallel resonance capacitor Cr by the switching operation of the switching element Q1 as described above is shown in FIG.
As shown in FIG. 5, a waveform having a 0 level during the period TON when the switching element Q1 is on and a sinusoidal pulse during the period TOFF when the switching element Q1 is off is obtained. This indicates that the primary side switching converter is a voltage resonance type operation.

Since the collector of the conduction control element Q2 is connected to the base of the switching element Q1, a collector current IQ2 flows through the collector of the conduction control element Q2 as shown in FIG. 2 (g). Here, the base-emitter voltage VBE2 of the conduction control element Q2 has a negative polarity during the period in which the clamp diode DD is conductive at the start of the period TON and then has a positive polarity, as shown in FIG. 2 (h).
In this case, the base-emitter voltage VBE2
Shows a waveform as a sawtooth wave that gradually increases by reversing from a certain level due to the negative polarity to the positive polarity. In this way, the base-emitter voltage VBE2 in the period TON does not maintain a constant level but has a sawtooth wave as shown in FIG.
As indicated by the collector current IQ2 in (g), the collector current IQ2 having a positive polarity is turned on near the end of the TON period.
It is possible to operate so as to flow.

The base-emitter voltage VBE2 becomes a sawtooth wave, as described above.
This is because the capacitor C3 is inserted in parallel between the base-emitter of No. 2 and the timing at which the conduction control element Q2 transitions from the off state to the on state within the TON period is, for example, the capacitance of the capacitor C3 ( Time constant) and the resistance value (time constant) of the resistor R2 connected in parallel with the capacitor C3.

Next, referring to FIG. 2 also, the constant voltage control operation of the power supply circuit shown in FIG. 1 will be described. First, assuming that the level of the secondary side DC output voltage EO1 rises due to, for example, an increase in the AC input voltage VAC or a decrease in load power, the control circuit 1 raises this level of increase as described above. It operates so as to increase the current of the photodiode (primary side phototransistor) of the photocoupler PC according to the level.

As the phototransistor current increases,
Since the amount of charging current to the capacitor C3 increases correspondingly, the slope of the sawtooth waveform as the base-emitter voltage VBE2 shown in FIG. 2 (h) increases. Depending on this, the switching element Q1 becomes O
The operation is performed so as to accelerate the conduction timing of the conduction control element Q2 within the TON period when N is set. As a result, the collector current IQ2 flowing through the conduction control element Q2 (see FIG.
(G)), for example, as compared with the case of heavy load conditions,
A large amplitude will be obtained.

The amplitude of the collector current IQ2 of the conduction control element Q2 becomes large, and the collector current I in the period TON is increased.
When the current amount of Q2 increases, the base current I
An operation that changes so that the amount of B current is reduced can be obtained. As a result, the base accumulated carrier disappearance time (tstg) of the switching element Q1 shown in FIG. 2C is shortened. Along with this, the length of the period TON during which the switching element Q1 is turned on is shortened and varied.

As the period length within the period TON becomes shorter, the switching frequency of the switching element Q1 becomes higher. By variably controlling the switching frequency, for example, the resonance impedance of the primary side parallel resonance circuit is changed, and the power transmitted from the primary side to the secondary side of the insulating converter transformer PIT is also changed. It will be variable. Finally, the level of the secondary side DC output voltage EO1 is also variably controlled, and as a result, the control circuit 1 outputs a current corresponding to the level of the secondary side DC output voltage EO1 to the photodiode of the photocoupler PC. The constant voltage control operation of maintaining the level of the secondary side DC output voltage at a constant level is realized by supplying the voltage to the.

In the present embodiment, when the switching frequency is variably controlled, the period TOFF during which the switching element Q1 is off is constant, and the period TON during which it is on is variable. That is, also in this case, the constant voltage control operation by the composite control method is obtained.

Next, the operation of the overload protection circuit 2 provided in the power supply circuit of the present embodiment when the load side is shorted by exceeding the rated maximum load will be described.

First, when the load side is short-circuited, the level of the current flowing through the secondary side DC output voltage line is higher than in the normal state, and an excessive current flows through this line.
Further, the switching frequency of the switching element Q1 at this time is lowered to almost the lower limit. Therefore, an excessive level of collector current also flows in the switching element Q1. Thus, an excessive current flowing through the switching element Q1 is detected by the resistor RE connected to the emitter of the switching element Q1. In other words, the resistor RE can obtain a voltage VE which is equal to or higher than a predetermined value according to the excessively large emitter current.

As described above, depending on the generation of the voltage VE across the resistor RE in response to the excessive current, the resistor R5
Therefore, the required voltage between both ends can be obtained. The voltage thus obtained is applied to the gate of the small signal thyristor Q4 connected to the resistor R5, and the small signal thyristor Q4 becomes conductive.

The conduction of the small signal thyristor Q4 causes a base current to flow to the base of the transistor Q3 connected to the anode of the small signal thyristor Q4 via a resistor, which causes the transistor Q3.
Conducts. By thus conducting the small signal thyristor Q4 and the transistor Q3, the overload protection circuit 2 generates a control current for overload protection. Then, this protection control current is applied to the transistor Q3.
Control element Q2 connected to the collector of the resistor via a resistor
Flow into the base of the switching element Q2, and as a result, the base current of the conduction control element Q2 increases and the switching frequency of the switching element Q1 is controlled to increase. The operation of the overload protection circuit 2 can be stopped by turning off the switch SW once. Then, since the thyristor Q4 is reset by this, the power supply circuit can start the steady operation by turning on the switch SW again.

FIG. 3 shows operation waveforms in each part of the power supply circuit when the overload protection circuit 2 operates. First, the overload protection circuit 2 starts to operate, and a protection control current is supplied to the base of the conduction control element Q2, so that the conduction control element Q2
The waveform of the base-emitter voltage VBE2 of is shown in Fig. 2 (g).
From the waveform of the power supply circuit shown in Fig. 3 during normal operation
It is shown that the transition is made to (g) and the amplitude thereof is increased. Further, along with this, the waveform of the collector current IQ2 also becomes as shown in FIG. 2 (h) to FIG. 3 (h), and it can be seen that the amplitude in the period TON in which the switching element Q1 is ON becomes large.

As described above, when the amplitude of the collector current IQ2 of the conduction control element Q2 is increased and the amount of the collector current IQ2 in the period TON is increased, the amount of the resonant current is changed so as to be reduced. The action to do is obtained. Base current IB obtained based on this resonance current
2 changes from the waveform shown in FIG. 2 (c) to the waveform shown in FIG. 3 (c), for example, which shortens the base accumulated carrier disappearance time (tstg) of the switching element Q1. . Along with this, the length of the period within the period TON in which the switching element Q1 is turned on is shortened and varied.

As the period TON becomes shorter, the switching frequency of the switching element Q1 is controlled so as to become higher, as described above in the constant voltage operation. That is, comparing the waveform of the resonance voltage V1 during the steady operation shown in FIG. 2A with the waveform of the resonance voltage V1 shown in FIG. 3A, the waveform shown in FIG. The period is shortened and the switching frequency of the switching element Q1 is increased.

Then, as the switching frequency becomes higher, the collector current IQ1 of the switching element Q1 becomes as shown in FIG. 3B, and the resistor RE connected to the emitter of the switching element Q1. The voltage VE between both ends of V and the base-emitter voltage VBE1 of the switching element Q1 are as shown in FIG.
The waveform as shown in FIG. 3D is obtained, and it can be seen that the current and voltage generated in each of the base, emitter, and collector of these switching elements Q1 are smaller than those in the case of FIG. Further, even if the damper current ID of the clamp diode DD is reduced, the amount of the current is reduced as shown in FIG.

As described above, the current and the voltage generated in each of the base, the emitter and the collector of the switching element Q1 are reduced, so that the load on the switching element Q1 is reduced.

As described above, the overload protection circuit 2 allows the switching element Q1 to be operated when an abnormality such as a load short circuit occurs.
Since it is possible to reduce the load on the switching element Q1 in the present embodiment, it is possible to select a switching element Q1 having a higher breakdown voltage and a smaller current capacity, which is low cost and small in size. You will be able to use one.

Although the first embodiment has been described above, if the constant voltage control circuit system according to the present embodiment is configured as described above, the orthogonal type circuit provided in the power supply circuit of FIG. 7 as the prior art is provided. The control transformer PRT will be omitted. As a result, in the present embodiment, the problem of the variation in the inductance value of the drive winding NB due to the variation in the gap at the time of manufacturing the orthogonal control transformer PRT is solved. Therefore, the margin for the range of the AC input voltage VAC can be set to be small, and the circuit design can be facilitated. Further, the problem of difficulty in the manufacturing process of the orthogonal control transformer PRT is solved. AC / DC
The power conversion efficiency can also be improved.

Further, unlike the example of FIG. 7, the control power is not supplied to the control winding NC of the orthogonal control transformer PRT to control the switching frequency, so that the reactive power at the light load is reduced and the power is reduced. The loss can be reduced.

Further, the drive transformer CDT can be constituted by an H-shaped core or an ultra-small EI-12 type ferrite core, and when the orthogonal control transformer PRT is provided as shown in the prior art of FIG. The size and weight can be significantly reduced compared to.

Further, according to the structure of the present embodiment, the current flowing through the conduction control element Q2 is very small, and the voltage applied to the conduction control element Q2 is also low.
Therefore, for the bipolar transistor as the conduction control element Q2, a low withstand voltage small capacity product having a withstand voltage of 30 V and a rated current of 0.1 A or less may be selected. For example, the present applicant previously provided an active clamp circuit that clamps a primary side parallel resonance voltage or a secondary side resonance voltage for a composite resonance type switching converter, and stabilizes the power supply by controlling the conduction angle of the active clamp circuit. Although various configurations have been proposed for achieving this, in this case, the switching element (transistor) that forms the active clamp circuit has a high voltage corresponding to the primary side parallel resonance voltage level or the secondary side resonance voltage level. It is necessary to select a pressure resistant product, which is disadvantageous in terms of cost and size. On the other hand, in the present embodiment, a low withstand voltage small capacity product is selected as the bipolar transistor as the conduction control element Q2, so that it is possible to realize cost reduction, size reduction and weight reduction. is there.

Further, as described above with reference to the waveform diagram of FIG. 2, in the present embodiment, by providing the capacitor C3 between the base and the emitter of the conduction control element Q2, the voltage between the base and the emitter thereof can be reduced. VBE2 is the period TO
It is designed to have a sawtooth wave in N. As a result, the conduction control element Q2 is not made conductive for the entire period TON, but is made conductive only in the terminal period within the period TON. This conduction period is the base accumulated carrier disappearance time (tst
Corresponding to g), the reverse current IB2 is made to flow through the base of the switching element Q1 during this period. When the conduction control element Q2 conducts during this period and the collector current IQ2 flows, the reverse current IB2 of the base current IB also increases.

The drive transformer CDT employs an H-shaped core or an EI type core as shown in FIGS. 4 and 5, but the drive transformer CDT using these cores can obtain a saturated state. Since it is easy, the inductance of the drive winding NB hardly changes during the period IB1 in which the base current IB has a positive polarity, and the inductance of the drive winding NB decreases rapidly due to saturation of the magnetic core in the period IB2 in which the base current IB has a negative polarity. become.

As a result, the peak level of the base current IB flowing through the base of the switching element Q1 is as shown in FIG.
As shown in (c), the reverse current IB2 becomes much larger than the forward current IB1. As a result, the fall time of the switching element Q1 is shortened and the switching loss when the switching element Q1 is turned off is reduced. At the same time, the accumulation time tstg is also shortened,
The power loss in the switching element will be less. Moreover, the control range of the constant voltage can be expanded.

Further, in this embodiment, the resistor RE is inserted between the emitter of the switching element Q1 and the primary side ground. The voltage VE across the resistor RE becomes as shown in FIG. 2 (e), whereby the base-emitter voltage VBE1 of the switching element Q1 is raised as shown in FIG. 2 (d). For example, the base-emitter voltage VBE1
Is raised from 0.7V to 1.4V in this example. Therefore, the constant voltage control range is also expanded by providing the resistor RE as in the present embodiment.

Further, as described above, the overload protection circuit 2 is provided in the present embodiment. By providing the overload protection circuit 2, the switching element Q1 for the abnormal condition such as a load short circuit is provided. Since the overcurrent protection and the overvoltage protection are performed, it is not necessary to provide the overcurrent limiting circuit 10 as shown in FIG. 8, for example. Therefore, in the power supply circuit according to the present embodiment, the circuit design is performed in consideration of the margin corresponding to the rise of the AC input voltage VAC and the malfunction margin during the steady operation due to the variation of the components forming the overcurrent limiting circuit 10. It will not be necessary to do.

Therefore, in the present embodiment, it is possible to select a low breakdown voltage product as a component of each required part such as the switching element Q1. Also, as for the current capacity, a small capacity can be selected. If a switching element having a low breakdown voltage and a small capacity can be selected, its shape can be made smaller. For example, in the circuit shown in FIG. 8, the switching element Q1 is a large package of TO-3P with a withstand voltage of 1200V, but the circuit of the present embodiment shown in FIG. 1 is a medium-sized package of TO-220 with a withstand voltage of 700V. Is something that can be done. In addition, since the switching characteristics are improved by using the low withstand voltage product, the power loss in the switching element Q1 is also reduced.

Furthermore, in the overcurrent limiting circuit 10 provided in, for example, the power supply circuit shown in FIG. 8, an operation of directly inputting an excessive current generated in the switching element Q1 and directly limiting the base current of the switching element Q1. Therefore, there is a limit to the reduction in withstand voltage and the reduction in current capacity of the semiconductor constituting the overcurrent limiting circuit 10. On the other hand, in the present embodiment, in controlling the switching frequency of the switching element Q1 in order to reduce the load on the switching element Q1 in the event of an abnormality such as a short-circuit on the load side, the overload protection circuit 2 conducts conduction control. Since it suffices to control the amount of current that should flow through the element Q2, it is possible to select a semiconductor with a lower withstand voltage and a smaller current capacity as the semiconductor provided for these elements. Also in this respect, in the case of the present embodiment, downsizing and cost reduction of the circuit can be achieved. <Second Embodiment>

FIG. 6 shows the configuration of a power supply circuit according to the second embodiment of the present invention. In this figure, the same parts as in FIG.
Only parts different from the example of FIG. 1 will be described.

In this case, the resistor RE is not inserted between the emitter of the switching element Q1 and the primary side ground,
A clamp diode DD is connected between the base of the switching element Q1 and the primary side ground. In this example, the clamp diode DD is a fast recovery type diode.

As shown in the figure, the base of the switching element Q1 is [base current limiting resistor RB-
A series connection circuit of a time constant (for resonance) capacitor CB-driving winding NB] is connected. This series connection circuit is a self-excited oscillation drive circuit for switching-driving the switching element Q1 by self-excitation. In this case, for example, at the time of starting the power source, the rectified and smoothed voltage Ei flows through the starting resistor RS and the base current through the drive winding NB-base current limiting resistor RB to the base of the switching element Q1 to perform the switching operation. Is designed to start.

The drive transformer CDT in the power supply circuit shown in FIG. 6 includes a detection winding NA and a drive winding NB.
In addition, the control winding NC is wound. As shown in the drawing, the winding direction of each of these windings is such that the drive winding NB and the control winding NC are in opposite phase, and the detection winding NA is in phase with this winding NB. ing.

As the drive transformer CDT around which the above windings (NA, NB, NC) are wound, for example, an open magnetic circuit type with an H-shaped ferrite core or a closed magnetic circuit type with an EI type ferrite core can be adopted. . In the case of the H-shaped ferrite core, the detection winding NA, the drive winding NB, and the control winding NC are wound around the H-shaped ferrite core in the same manner as described in FIG. It is formed.

Also in the case of the EI type ferrite core, the I type core and the E type core are combined to form the EI type core as in the case of FIG. Then, a split bobbin is arranged on the central magnetic leg of the E-shaped core, and the detection winding NA, the drive winding NB, and the control winding NC are wound around the split bobbin, respectively. Also in this case, the drive transformer CDT having a closed magnetic circuit structure using these H-shaped ferrite cores or EI type ferrite cores is considerably smaller and lighter than the orthogonal control transformer PRT described in FIG. 7, for example. There is.

By the way, in the drive transformer CDT, the detection winding NA and the drive winding NB are arranged on the primary side of the insulating converter transformer PIT. On the other hand, the control winding NC is arranged on the secondary side of the insulating converter transformer PIT. Therefore, for example, in order to secure the direct current insulation between the primary side and the secondary side, a photo coupler or the like is generally provided. On the other hand, in the present embodiment, by selecting a triple insulated wire for the control winding NC, the insulation property with respect to the detection winding NA and the drive winding NB is ensured. A sufficient insulation state is obtained without any interposition.

Further, the drive transformer C of the present embodiment
The control winding NC wound around the DT has its winding start end side grounded to the secondary side ground and its winding end end connected to the collector of the conduction control element Q2. The emitter of the conduction control element Q2 is grounded to the secondary side ground. That is, the control winding NC and the conduction control element Q2 (collector-emitter)
Can be regarded as forming a series connection circuit connected in series on the secondary side of the insulating converter transformer PIT. In this case, the conduction control element Q2 is NP
N-type bipolar transistors have been selected.

The base (control input terminal) of the conduction control element Q2 is connected to the collector of a PNP type transistor Q3 provided in the control circuit 5 described later via the resistor R1. A resistor R2 is inserted between the base and emitter of the conduction control element Q2. A base-emitter voltage VBE2 (control input terminal potential) is obtained across the resistor R2.

Then, similarly to the first embodiment, the capacitor C3 is inserted between the base and the emitter of the conduction control element Q2. In other words, also in this case, the capacitor C3
Is connected in parallel to the resistor R2 which generates the base-emitter voltage VBE2, thereby forming a time constant circuit. The capacitor C3 is charged and discharged with a current that tends to flow from the collector of the transistor Q3 on the control circuit 5 side to the base of the conduction control element Q2. The operation of proportionally increasing the level of the base-emitter voltage VBE2 can be obtained almost corresponding to the period when Q1 is turned on. That is, the switching element Q1
Base-emitter voltage VBE during the period when the transistor turns on
2 is a sawtooth wave.

As shown in the figure, on the secondary side of the insulating converter transformer PIT, a tap is provided on the secondary winding N2, and then the rectifying diodes DO1, DO2, DO3, DO4 are provided.
By connecting the smoothing capacitors CO1 and CO2 as shown in the figure, two sets of [rectifier diodes DO1, DO2, smoothing capacitor CO1] and [rectifier diodes DO3, DO4, smoothing capacitor CO2] are provided. A full wave rectifier circuit is provided. A full-wave rectifier circuit consisting of [rectifier diodes DO1, DO2, smoothing capacitor CO1] generates a DC output voltage EO1, and [rectifier diodes DO3, DO4, smoothing capacitor C1].
The full-wave rectifier circuit consisting of O2] produces a DC output voltage EO2. That is, in the circuit shown in this figure, a full-wave rectifier circuit is provided to obtain the DC output voltage on the secondary side. In this case, the secondary side DC output voltage EO1 is
Is input as a detection voltage for constant voltage control. The secondary side DC voltage EO2 is used as an operating power source of the control circuit 5.

In the case of this embodiment, the secondary winding N2
Is further provided with a center tap, and one of the lines has a small signal thyristor Q of the overload protection circuit 3 described later.
5 cathodes are connected. Also, as shown in the figure, an overcurrent protection resistor RO is inserted in the other side of this line,
Grounded to secondary earth. By thus inserting the overcurrent protection resistor R0 between the secondary winding N2 and the secondary side ground, for example, in the event of an abnormality such as a short-circuit on the secondary side load, the load side passes through the secondary side ground through the secondary side ground. A voltage corresponding to an excessive level of current flowing into the overcurrent protection resistor RO is generated, so that a voltage is applied to the cathode of the small signal thyristor Q5, which will be described later. It serves as a detection resistor for starting the operation of the overload protection circuit 3. In this case, it is assumed that a low resistance is selected as the overcurrent protection resistor R0 so that a voltage across both ends is hardly generated at the rated load.

In the control circuit 5, resistors R3 to R4 are connected in series between the DC output voltage EO1 and the secondary side ground, and the control terminal of the shunt regulator Q9 is connected to this connection point (voltage dividing point). . For the resistor R4,
The capacitor C5 is connected in parallel. The anode of the shunt regulator Q9 is grounded to the secondary side ground, and the cathode is connected to the base of the PNP type transistor Q3 via the resistor R7. The transistor Q3 is provided to amplify the current flowing through the anode of the shunt regulator Q9. The emitter of the transistor Q3 is connected to the line of the secondary side DC voltage EO2, and the collector is a resistor R1.
Is connected to the base of the conduction control element Q2 described above via. The output of this collector becomes the detection output of the control circuit 5. A resistor R6 is placed between the base and the emitter.
Have been inserted.

The control circuit 5 formed by the above-mentioned connection form functions as an error amplifier having the DC output voltage EO1 as a detection input. That is, the DC output voltage EO1 is applied to the resistor R
By inputting the voltage divided by 3 and R4 to the control terminal of the shunt regulator Q9 as a control voltage, a current having a level corresponding to the DC output voltage EO1 flows through the shunt regulator Q9. Then, this current is amplified by the transistor Q3 and output from the collector.

According to the level change of the collector current of the transistor Q3, the base-emitter voltage VBE2 of the conduction control element Q2 changes, and the base current corresponding to the base-emitter voltage VBE2 is conduction controlled. Although it is made to flow to the base of the element Q2, the level of the collector current IQ2 flowing to the collector of the conduction control element Q2 is also changed according to the level of the secondary side DC voltage EO1. This means that the level (amplitude) of the current flowing through the control winding NC connected between the collector of the conduction control element Q2 and the secondary side ground is variably controlled. According to this operation, the switching frequency of the switching element Q1 is changed, and thereby the secondary side DC output voltage is stabilized so as to be constant. The control circuit 5 according to the present embodiment,
The constant voltage control operation by the conduction control circuit system will be described later.

Further, as described above, in this embodiment, the capacitor C3 is inserted between the base and the emitter of the conduction control element Q2 as in the first embodiment. The -emitter voltage VBE2 becomes a sawtooth wave substantially corresponding to the period when the switching element Q1 is turned on, and as a result, in the case of this example as well, the switching loss in the switching element Q1 is reduced as in the first embodiment. It is designed to have the effect of causing it.

Further, also in the second embodiment, an overload protection circuit 3 is provided as shown in addition to the configuration of the power supply circuit described so far. This overload protection circuit 3 includes a small signal thyristor Q as shown.
5, transistor Q6, Q7, capacitor C4, resistor R8. First, in the overload protection circuit 3, as described above, the cathode of the small signal thyristor Q5 is connected to the center tap of the secondary winding N2. The gate of the small signal thyristor Q5 is grounded to the secondary side ground via the parallel circuit of the resistor R8 // capacitor C4, and when the load on the secondary side is short-circuited, the The voltage corresponding to the excessive level of current flowing from the ground on the secondary side through the resistor R8 is small signal thyristor Q.
It is intended to be applied to the gate of 5. In this way, when the load on the secondary side is short-circuited, a voltage is applied to the gate of the small signal thyristor Q5, and as described above, the small signal thyristor Q5 is activated by the overcurrent protection resistor RO.
By applying a voltage to the cathode of the small signal thyristor Q5, the small signal thyristor Q5 is made conductive, and the operation of the overload protection circuit 3 is started. As described above, by connecting a parallel circuit composed of the capacitor C4 and the resistor R8 between the gate of the small signal thyristor Q5 and the secondary side ground, it is possible to prevent the circuit malfunction.

The anode of the small signal thyristor Q5 is connected to the base of a PNP type transistor Q6 via a resistor. And the emitter of this transistor Q6
For example, a DC voltage of about 5V is input from a standby power supply circuit (not shown) that is provided in the power supply circuit,
This is the power supply for the operation of the overload protection circuit 3. The collector of the transistor Q6 is connected to the base of an NPN type transistor Q7 via a resistor. Further, as shown in the figure, a resistor is inserted between the base of the transistor Q7 and the secondary side ground. The emitter of the transistor Q7 is grounded to the secondary side ground, and the collector is connected to the winding start end of the control winding NC. Thus, the transistor Q7
Since the collector of is connected to the control winding NC, the overload protection circuit 3 can control the amount of current flowing through the control winding NC. The operation will be described later.

The above is the description of the configuration of the power supply circuit according to the second embodiment. The steady-state operation of the main part of the power supply circuit having such a configuration is the same as that of the first embodiment. The same operation as that in the steady state by the power supply circuit can be obtained. That is, also in the power supply circuit of this embodiment, the operation waveforms of the respective parts shown in FIGS. 2A to 2H can be obtained. Therefore,
The description of the operation of the main part of the power supply circuit will be omitted.

Here, by the control circuit 5 and the conduction control circuit system, for example, when the level of the secondary side DC output voltage EO1 rises because the AC input voltage VAC rises or the load power decreases. The constant voltage control operation will be described.

First, when the level of the secondary side DC voltage EO1 rises, the control circuit 5 operates so as to increase the collector current of the transistor Q3 which is the detection output as described above. . Transistor Q3
When the collector current of increases, the capacitor C3
Since the amount of charging current to the conduction control element Q2 increases, the slope of the sawtooth waveform as the base-emitter voltage VBE2 of the conduction control element Q2 increases. As a result, the conduction control element Q2 operates to accelerate the conduction timing within the TON period in which the switching element Q1 is turned on. As a result, as the collector current IQ2 flowing through the conduction control element Q2, a larger amplitude can be obtained than in the case of heavy load or steady operation, for example.

Here, as described above, since the drive winding NB and the control winding NC are in a tightly coupled state in the drive transformer CDT, they are equalized to the drive winding NB. It can be seen that the control winding NC is connected in parallel. Therefore, in terms of an equalizing circuit, for example, in the self-excited oscillation circuit (CB-NB-RB), the winding end of the control winding NC is connected to the connection point between the winding end end of the drive winding NB and the resonance capacitor CB. A circuit with the ends connected will be formed. According to this equalization circuit, the control winding N
The collector current IQ2 of the conduction control element Q2 flowing in C is the resonance current which becomes the output of the self-excited oscillation circuit (CB-NB-RB). It can be considered that the current is obtained by branching the connection point with NC as a branch point.

Therefore, when the amplitude of the collector current IQ2 of the conduction control element Q2 becomes large and the amount of the collector current IQ2 in the period TON increases, that much,
It is possible to obtain an operation that changes so that the amount of the resonance current decreases. Then, as a waveform of the base current IB obtained based on this resonance current, the base accumulated carrier disappearance time (tstg) of the switching element Q1 shown in FIG. The length of the period TON during which is ON is variable as the length becomes shorter.

When the period TON is shortened in this manner, the switching frequency of the switching element Q1 is controlled to be high as described in the first embodiment. By variably controlling the switching frequency, for example, the resonance impedance of the primary side parallel resonance circuit is changed, and the power transmitted from the primary side to the secondary side of the insulating converter transformer PIT is also changed. It will be variable. As a result, the level of the secondary side DC output voltage is finally variably controlled, and the power source is stabilized.

As described above, also in the power supply circuit of the present embodiment, the same effect as the effect of the constant voltage control operation in the first embodiment is obtained.

In the present embodiment as well, when the switching frequency is variably controlled, the period TOFF during which the switching element Q1 is off is constant, and the period TON during which it is on is variable. That is,
Also in this case, the constant voltage control operation by the composite control method is obtained.

Next, the operation of the overload protection circuit 2 provided in the power supply circuit of the present embodiment when the load on the secondary side exceeds the rated maximum load and is short-circuited will be described.

First, when the load on the secondary side is short-circuited, an excessive current flows into the overcurrent protection resistor RO from the load side via the secondary side ground as described above, and the overcurrent protection resistor RO shown in FIG. The potential of the voltage VO at both ends rises. In this way, the excessive current generated by the load short circuit is detected by the overcurrent protection resistor RO. Then, by the voltage VO thus detected, a negative voltage is applied to the cathode of the small signal thyristor Q5. Similarly, an excessive current from the load side flows into the parallel circuit of the capacitor C4 // resistor R8 via the secondary side ground, and the positive voltage corresponding to the level of this current is for small signals. It is applied to the gate of thyristor Q5. As a result, the small signal thyristor Q5 becomes conductive.

When the small signal thyristor Q5 conducts in this way, a base current is generated at the base of the transistor Q6 connected to the anode of the small signal thyristor Q5 via a resistor, and the transistor Q6 conducts.
When the transistor Q6 becomes conductive, a base current is generated in the base of the transistor Q7 which is connected to the collector of the transistor Q6 via a resistor, and the transistor Q7 becomes conductive.

If the transistor Q7 becomes conductive in this manner, the collector current of the transistor Q7 will flow, but since the collector of the transistor Q7 is connected to the control winding NC, as a result, the current flowing through the control winding NC. The quantity will increase as it is pulled up by this transistor Q7.

By controlling so as to increase the amount of current flowing through the control winding NC in this manner, the switching element Q1 is operated in the same manner as in the constant voltage control operation described above.
Will be controlled so that the switching frequency becomes higher. Then, in this way, the switching element Q1
By controlling the switching frequency to be high, it is possible to suppress an excessive current and voltage generated in the switching element Q1 in the same manner as described in the overload protection circuit 2 of the first embodiment. Becomes possible,
As a result, it is possible to protect the switching element Q1 from an overload state in the event of an abnormality such as a load short circuit.

The above is the description of the power supply circuit as the second embodiment. However, as described above, the operation of this circuit is almost the same as that of the circuit described in the first embodiment.
And the effect can be obtained. However, in the case of the circuit of the second embodiment, the following effects are particularly obtained. In other words, in this circuit, the series connection circuit including the control winding NC and the conduction control element Q2 is connected to the primary side self-excited oscillation drive circuit in terms of equalization, but in reality, the secondary side ground. The circuit is provided between the insulating converter transformer PIT and the secondary side of the insulating converter transformer PIT. Then, as described above, the control winding NC
For example, select a triple insulated wire to secure DC insulation from the primary side, and then drive winding NB on the primary side.
I am trying to get a state that is magnetically coupled with. As a result, in the present embodiment, in order to secure the direct current insulation state between the secondary side control circuit system and the primary side switching converter, it is not necessary to add a component element such as a photo coupler. Is what you get. For example, if
If a series connection circuit consisting of the control winding NC and the conduction control element Q2 is provided on the primary side, it must be galvanically isolated from the control circuit system on the secondary side by interposing a photocoupler, for example. become.

Further, unlike the overload protection circuit 2 of the first embodiment, the overload protection circuit 3 of the present embodiment is also provided in the event of an abnormality such as a short circuit of the load of the secondary side DC output voltage. In order to reduce the load on the switching element Q1, by detecting the current when the load is short-circuited on the secondary side and controlling the amount of current that should flow in the control winding NC also arranged on the secondary side, switching is performed. The operation of controlling the switching frequency of the element Q1 is obtained. That is, in the case of the present embodiment, the control operation by the overload protection circuit 3 is completed on the secondary side. Therefore, in the present embodiment, a photocoupler for insulating the primary side and the secondary side from each other is not necessary for obtaining the control operation by the overload protection circuit 3. As a result, the number of parts can be reduced, and the circuit production cost can be reduced.

Although each of the above-described embodiments adopts the single-end system having one set of switching elements, it is a self-excited voltage resonance converter of the so-called push-pull system having two sets of switching elements. It doesn't matter. Also on the secondary side,
A rectifying circuit having a circuit configuration other than that shown may be provided.

[0120]

As described above, the present invention adopts a self-excited configuration including a drive transformer as a composite resonance type converter. At the same time, the switching frequency control for constant voltage control is performed by controlling conduction of the conduction control element to variably control the amount of current conduction in the self-excited oscillation drive circuit (switching drive means) that drives the switching element. I am trying to do it. Then, with such a configuration, it becomes possible to omit the orthogonal control transformer which has been required for the switching frequency control up to now.

As a result, the problem of the variation in the inductance value due to the variation in the gap of the orthogonal control transformer is solved. In particular, the variation of the inductance of the drive transformer in the present invention is about ± 5%. For this reason, it is possible to set a small margin for the range of the AC input voltage, so that the circuit design can be facilitated. Further, the problem of difficulty in the manufacturing process of the orthogonal control transformer can be solved. Since it is not configured to supply the control power to the control winding of the quadrature control transformer to control the switching frequency, it is possible to reduce the reactive power at the light load and reduce the power loss. That is, the AC / DC power conversion efficiency can be improved. Further, since the drive transformer in the present invention is much smaller than the orthogonal control transformer, it is possible to promote the reduction in size and weight of the power supply circuit.

Since a low voltage and a small level current are applied to the conduction control element (bipolar transistor), the power required for constant voltage control is also reduced.
That is, the reactive power in the power supply circuit is reduced, which also promotes the reduction of the AC / DC power conversion efficiency of the power supply circuit. Further, as the conduction control element,
Since a low withstand voltage and small capacity product is also selected, reduction in size and weight and cost reduction are promoted also in this respect.

Further, in the present invention, the base-emitter voltage (control input terminal potential) of the conduction control element is made to have a sawtooth wave within the ON period of the switching element. As a result, an operation that makes the peak of the reverse current flowing through the base of the switching element larger in the ON period can be obtained, so that the fall time and the accumulation time of the switching element are shortened and the switching loss is reduced accordingly. To be done. In other words, this also promotes the improvement of the power conversion efficiency.

Further, by inserting a resistor between the emitter of the switching element and the primary side ground, the control range of the conduction control element is expanded, and accordingly, constant voltage control is performed against fluctuations in the AC input voltage. The range will also be expanded.

Further, according to the present invention, an overload protection circuit is provided to increase the switching frequency of the switching element at the time of abnormality such as short-circuiting of the load on the secondary side. I try to suppress the voltage. As a result, it is possible to reduce the load on the switching element in the event of an abnormality such as a load short circuit, and to ensure the reliability of the circuit against an abnormality such as a load short circuit. Further, as the switching element, it is possible to select a switching element having a lower withstand voltage and a small current capacity, which makes it possible to reduce the cost and size of the circuit.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing a configuration example of a switching power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a waveform diagram showing the operation of the main part of the power supply circuit during steady operation (at rated maximum load).

FIG. 3 is a waveform chart showing an operation of a main part when the overload protection circuit included in the power supply circuit is operating.

FIG. 4 is a perspective view showing a structural example of a drive transformer having an H-shaped core.

FIG. 5 is a sectional view showing a structural example of a drive transformer having an EI type core.

FIG. 6 is a circuit diagram showing a configuration example of a switching power supply circuit according to a second embodiment of the present invention.

FIG. 7 is a circuit diagram showing a configuration of a power supply circuit as prior art.

FIG. 8 is a circuit diagram showing a configuration of a power supply circuit as a prior art including an overcurrent limiting circuit.

9 is a waveform chart showing the operation of the circuit shown in FIG.

[Explanation of symbols]

1, 5 Control circuit, 2, 3 Overload protection circuit, Di bridge rectifier circuit, Ci smoothing capacitor, Cr primary side parallel resonance capacitor, C2 secondary side parallel resonance capacitor,
DO1, DO2, DO3, DO4 Rectifier diode, PIT insulation converter transformer, N1 primary winding, N2 secondary winding, N3 tertiary winding, CDT drive transformer, NA
Detection winding, NB drive winding, NC control winding, SW AC
Switch, CO1, CO2, CO3, CO4 smoothing capacitor,
Q1 switching element, Q2 conduction control element, Q3, Q
6, Q7 transistor, Q4, Q5 small signal thyristor, Q9 shunt regulator, C3 time constant capacitor, C4 capacitor, D2 Zener diode, RB
Base current limiting resistance, RS starting resistance, RO overcurrent protection resistance

Claims (3)

[Claims] 1. A switching means comprising a switching element for inputting a DC input voltage to perform switching, a primary winding and a secondary winding, and an output of the switching means obtained from the primary winding. A primary formed by an insulating converter transformer transmitting to the secondary winding, a primary winding of the insulating converter transformer and a primary side parallel resonance capacitor, and provided so that the operation of the switching means is a voltage resonance type. Side parallel resonance circuit, a secondary side resonance circuit formed by connecting a secondary side resonance capacitor to the secondary winding of the insulating converter transformer, and an alternating voltage obtained in the secondary side resonance circuit. Is input to perform a rectification operation to obtain a DC output voltage, and a DC output voltage generating unit, at least the detection winding, and the drive winding are A drive transformer for transmitting the output of the switching means to the drive winding by connecting the detection winding in series with the primary winding of the insulating converter transformer, and the drive winding and the resonance capacitor. A switching drive means that has a series resonance circuit formed, and that drives the switching element based on the output of the series resonance circuit, a conduction control element, and a control input terminal potential of the transistor element as the conduction control element. And a time constant circuit provided so as to have a substantially sawtooth waveform, and the rising period of the control input terminal potential is varied according to the level of the DC output voltage to variably control the amount of current conduction in the conduction control element. By changing the level of the drive signal of the switching element, By such switching frequency of the switching element is variably controlled according to the level of No.,
The switching frequency of the switching element is within a predetermined range by detecting a constant voltage control means for performing constant voltage control on the DC output voltage and an excessive level of current generated in the power supply circuit. A switching power supply circuit comprising: a protection unit that operates so as to be maintained inside and that controls an excessive current and voltage generated in the switching element to be suppressed. 2. The constant voltage means connects an output terminal (collector) of the conduction control element and an input terminal of the switching element, and a drive signal of the switching element branches to an output terminal of the conduction control element. The level of the drive signal of the switching element is changed according to the change of the current conduction amount in the conduction control element, and the protection means controls the primary side DC input voltage. While inputting as operating power supply,
In response to detecting the excessive current level,
The switching frequency of the switching element is configured to be maintained within a predetermined range by supplying a control current to the control input terminal of the conduction control element. Switching power supply circuit. 3. The constant voltage control means winds the control winding, which is supposed to be on the secondary side of the insulating converter transformer, on the drive transformer so as to be tightly coupled to the drive winding. In addition, by providing at least a conduction control circuit formed by connecting the control winding and the transistor element as the conduction control element in series, the conduction control element according to the level of the DC output voltage. Is configured to variably control the level of the drive signal of the switching element by variably controlling the amount of current conduction in the control winding and varying the control current flowing in the control winding. It is characterized in that the control frequency to be supplied to the control element is branched so that the switching frequency of the switching element is maintained within a predetermined range. Switching power supply circuit according to claim 1,.